KR100745770B1 - Complex Coefficient Transversal Filter and Wireless Communication System - Google Patents

Abstract

The present invention relates to a method and apparatus for a wireless communication system employing a complex coefficient transversal filter for removing an image. According to the present invention, a phase delay of a real part signal and an imaginary part signal is used in a multiband and broadband portable wireless communication system. By applying a complex coefficient transversal filter, it is possible to improve the image suppression ratio in a wide band. And, the circuit size can be simplified and the power consumption can be reduced.

Complex coefficient transversal filter, real part signal, imaginary part signal, surface acoustic wave

Description

Complex coefficient transversal filter and radio communication system employing the same

1 is a block diagram of a complex coefficient transversal filter that removes an image of the present invention;

2 (a) is a graph showing the arrangement of the variable tap coefficients according to the first embodiment of the present invention;

2 (b) is a graph showing frequency characteristics according to the first embodiment of the present invention;

3 (a) is a graph showing the arrangement of variable tap coefficients according to a second embodiment of the present invention;

3 (b) is a graph showing a frequency characteristic according to a second embodiment of the present invention;

4 (a) is a graph showing the arrangement of variable tap coefficients according to a third embodiment of the present invention;

4 (b) is a graph showing a frequency characteristic according to a third embodiment of the present invention;

5 is a graph showing a relationship between an imaginary part variable tap coefficient and a frequency characteristic of the present invention;

6 is a graph showing an arrangement of variable tap coefficients according to a fourth embodiment of the present invention;

7 is a graph showing frequency characteristics according to a fourth embodiment of the present invention;

8 illustrates a complex coefficient transversal surface acoustic wave filter according to a fifth embodiment of the present invention;

9 is a view showing a real part electrode unit according to a fifth embodiment of the present invention;

10 is a graph illustrating frequency conversion in a low IF receiver;

11 is a block diagram illustrating a receiver using a complex coefficient transversal filter to remove an image of the present invention;

12 is a graph showing frequency conversion in a receiver using a complex coefficient transversal filter to remove an image of the present invention;

13 is a graph showing a receiver block diagram of a super heterodyne scheme;

14 is a graph showing frequency conversion in a super heterodyne receiver;

15 is a block diagram showing a receiver of a low IF method using a complex coefficient transversal filter to remove an image of the present invention;

16 is a block diagram illustrating a transmitter using a complex coefficient transversal filter to remove an image of the present invention;

17 is a graph showing frequency conversion in a direct conversion type transmitter;

18 is a graph showing frequency conversion in a direct conversion transmitter using a complex coefficient transversal filter for removing an image of the present invention.

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a method and apparatus for a wireless communication system employing a complex coefficient transversal filter for removing an image, and more particularly, to a complex coefficient transformer for removing an image using phase delays of a real part signal and an imaginary part signal. A wireless communication system method and apparatus to which a vertex filter is applied.

In a wireless communication system, there are a heterodyne method, a low IF method, and a direct conversion method for transmitting and receiving a signal wirelessly.

First, in a heterodyne receiver, a received radio frequency (RF) signal is converted into an intermediate frequency (IF) and then a baseband signal. In this case, a bandpass filter (BPF) for selecting a desired radio frequency (RF) and a bandpass filter having high rejection characteristics in an intermediate frequency (IF) band are used. The image is removed by double filtering. Up to now, the communication method has been easy to remove the image because the signal frequency band is relatively narrow band. However, due to the rapid expansion of portable wireless devices, there is a demand for multiband and wideband wireless communication systems. In order to remove an image in such a wireless communication system, a poly-phase filter, which is a complex coefficient filter composed of a resistor and a capacitor, may be used. If the filter is multistage to realize high attenuation characteristics and wideband satisfying the system requirements, insertion loss increases, and it is difficult to remove images satisfying the system requirements.

Meanwhile, in order to implement a simple system in a multiband and wideband communication system, the use of a direct conversion method has been studied. In a direct conversion receiver, a radio frequency (RF) received signal is directly converted into a baseband signal, and a low frequency radio frequency (RF) is used. The received signal is converted into an intermediate frequency (IF) signal slightly away from the baseband signal. In this case, there is a difficulty in image removal due to the low intermediate frequency.

In order to solve this problem, the present invention provides a circuit scale by applying a complex coefficient transversal filter that removes an image by using a phase delay of a real part signal and an imaginary part signal to improve an image suppression ratio in a wide band. It is an object of the present invention to provide a wireless communication system apparatus and method that can simplify and reduce the power consumption.

In order to solve the above technical problem, a complex coefficient transversal filter which removes an image by using a phase delay of a real part signal and an imaginary part signal delays an input signal by an integer multiple of one period and at least two real part delay signals. And multiplying at least one of the input signal and the real part delayed signal by an absolute value of a real part variable tap coefficient to generate at least two real part multiplication signals, and the at least two real part A real part signal generation unit generating the real part signal by sequentially adding a multiplication signal in pairs; And delaying the input signal by a half cycle to generate a phase delay signal, delaying the phase delay signal by an integer multiple of one cycle to generate at least two imaginary delay signals, and generating the phase delay signal and the imaginary delay. Multiply at least one of the signals by an absolute value of an imaginary part variable tap coefficient to generate at least two or more imaginary part multiplication signals, and sequentially add and add the at least two imaginary part multiply signals to the imaginary part And an imaginary signal generator for generating a signal.

In addition, the real part signal generation unit of the complex coefficient transversal filter for removing the image of the present invention may include at least one of the input signal and the real part delay signal by an absolute value of the real part variable tap coefficient. Multiply to generate the real part multiplication signal, where M is an integer of at least 5 or greater.

On the other hand, the imaginary part signal generator of the complex coefficient transversal filter for removing the image of the present invention, at least one signal of the phase delay signal and the imaginary part delay signal by the absolute value of the imaginary part variable tap coefficient It may include (M-1) imaginary part signal coefficient multipliers for multiplying to generate the imaginary part multiplication signal.

In addition, the real part signal generation unit of the complex coefficient transversal filter for removing the image of the present invention may receive at least two signals of the real part multiplication signal through a polarization terminal and a signal input through the polarization terminal. (M-1), wherein M is an integer of at least 5, wherein the polarizing terminal alternately adds a positive polarity or a negative polarity to the input signal. It is desirable to.

On the other hand, the imaginary part signal generation unit of the complex coefficient transversal filter for removing the image of the present invention, the at least two signals of the imaginary part multiplication signal is input through the polarization terminal, the signal input through the polarization terminal It is preferable to include (M-2) adders for generating the imaginary part signal by adding s, and the polarity providing terminal alternately gives a positive polarity or a negative polarity to the input signal.

In addition, the M real part signal coefficient multipliers of the complex coefficient transversal filter for removing the image of the present invention may include first, second, M-1, and first absolute values of the real part variable tap coefficients smaller than one. The M real part signal coefficient multiplier and the third to M-2 real part signal coefficient multipliers whose absolute value of the real part variable tap coefficient is 1 may be included.

On the other hand, in the (M-1) imaginary part signal coefficient multipliers of the complex coefficient transversal filter for removing the image of the present invention, the absolute value of the imaginary part variable tap coefficients is the first and second real part signal coefficient multipliers. The first and second M-1 imaginary signal coefficient multipliers and the second through M-th absolute values of the imaginary part variable tap coefficients are in a range of 0.4 to 0.6 multiplied by the sum of the absolute values of the real part variable tap coefficients of And an imaginary part signal coefficient multiplier.

In addition, the real part signal generator of the complex coefficient transversal filter which removes the image of the present invention includes (M-1) real part signal delayers for receiving a signal and delaying one period to output the real part delay signal. It is preferable that the (M-1) delay units are arranged in series with each other.

On the other hand, the imaginary part signal generation unit of the complex coefficient transversal filter to remove the image of the present invention, by delaying the input signal by half a period to generate a phase delay signal and a signal delayed by one period (M-2) imaginary part signal delayers output as the imaginary part delay signal, wherein a first imaginary part signal delayer of the (M-2) imaginary part signal delayers is connected in series with the phase delay unit Preferably, the (M-2) imaginary part signal retarders are connected in series with each other.

In addition, the complex coefficient transversal filter for removing the image of the present invention may further include a transmission electrode unit for converting the input signal into a surface acoustic wave form to output to the real part signal generator and the imaginary part signal generator.

On the other hand, the transmission electrode portion of the complex coefficient transversal filter to remove the image of the present invention, the transmission positive electrode and the transmission secondary electrode consisting of a comb-like fingers arranged to face each other on a substrate containing a piezoelectric material And receiving electrical signals through a transmitter input terminal respectively connected to the transmitter positive electrode and the transmitter sub-electrode, and receiving a surface acoustic wave signal through a transmitter output terminal respectively connected to the transmitter positive electrode and the transmitter sub-electrode. It is desirable to transmit.

In addition, the real part signal generating part of the complex coefficient transversal filter for removing the image of the present invention is a real part positive electrode and a real part composed of comb-shaped fingers arranged to face each other on a substrate including a piezoelectric material. A real part electrode having a sub part electrode, receiving a surface acoustic wave signal through a real part input terminal connected to the real part positive electrode and the real part negative electrode, and receiving an electrical signal through a real part output terminal. It may include wealth.

On the other hand, the imaginary part signal generation part of the complex coefficient transversal filter for removing the image of the present invention, an imaginary part positive electrode and an imaginary part composed of comb-shaped fingers arranged to face each other on a substrate including a piezoelectric material. An imaginary electrode having a secondary sub-electrode, receiving a surface acoustic wave signal through an imaginary part input terminal connected to the imaginary part positive electrode and the imaginary part negative electrode, and applying an electrical signal through an imaginary part output terminal. It may include wealth.

In addition, the real part electrode part of the complex coefficient transversal filter for removing the image of the present invention is a real part according to the degree of overlap of the finger of the real part positive electrode and the finger of the real part negative electrode. It is desirable to adjust the absolute value of the variable tap coefficient.

On the other hand, the imaginary part electrode part of the complex coefficient transversal filter for removing the image of the present invention, the imaginary number according to the degree of overlap of the finger of the imaginary part positive electrode and the finger of the imaginary part negative electrode It is desirable to adjust the absolute value of the negative variable tap coefficient.

In addition, the real part electrode part of the complex coefficient transversal filter that removes the image of the present invention may include alternating repeating of the finger of the real part positive electrode and the finger of the real part negative electrode. have.

On the other hand, the imaginary electrode portion of the complex coefficient transversal filter to remove the image of the present invention, it may include that the fingers of the imaginary positive electrode and the finger of the imaginary sub-electrode alternately repeated. have.

In addition, the real electrode portion of the complex coefficient transversal filter for removing the image of the present invention, it is preferable to adjust the delay time according to a constant pitch of the real portion finger (finger).

On the other hand, the imaginary electrode portion of the complex coefficient transversal filter for removing the image of the present invention, it is preferable to adjust the delay time according to a constant pitch of the imaginary finger (finger).

In order to solve the other technical problem, the filtering method for removing the image by using the phase delay of the real part signal and the imaginary part signal (a) delays the input signal by a first value and multiplied by a second value, Generating a real part signal by sequentially adding the multiplied result; And (b) delaying the input signal by a third value and multiplying by a fourth value, and sequentially adding the multiplied result to generate an imaginary part signal.

In addition, the step (a) of the filtering method for removing the image of the present invention, in the wireless communication method applying a complex coefficient transversal filter to remove the image using the phase delay of the real part signal and the imaginary part signal, at least two Delaying an input signal by an integer multiple of one period to generate the real part delay signal, and realizing at least one of the input signal and the real part delay signal to generate at least two real part multiplication signals Multiplying by an absolute value of a variable tap coefficient; and sequentially adding and pairing the at least two real part multiplication signals to generate the real part signal.

In addition, step (b) of the filtering method for removing an image of the present invention may include: delaying the input signal by half a period to generate a phase delay signal, and generating the at least two imaginary delay signals. Delaying the delay signal by an integer multiple of one period, and generating at least one of the phase delay signal and the imaginary delay signal by an absolute value of an imaginary variable tap coefficient to generate at least two imaginary multiplication signals; And multiplying the at least two imaginary part multiplication signals sequentially in order to generate the imaginary part signal.

In order to solve the another technical problem, a receiver using a complex coefficient transversal filter that removes an image by using a phase delay of a real part signal and an imaginary part signal delays an input signal by a first value and then by a second value. Multiply and sequentially add the multiplied result to generate a real part signal, delay the input signal by a third value, multiply by a fourth value, and sequentially add the multiplied result to add an imaginary part signal. Generating a complex coefficient transverse filter; A frequency converter for converting a frequency of the result generated by the complex coefficient transversal filter; And a demodulator for demodulating the result converted by the frequency converter.

In order to solve the another technical problem, a transmitter using a complex coefficient transversal filter for removing an image using the phase delay of the real part signal and the imaginary part signal includes a modulator for modulating the baseband signal; A frequency converter for converting a frequency of the result modulated by the modulator; And delaying the input signal by the first value and multiplying the result converted by the frequency converter by a second value, sequentially adding the multiplied result to generate a real part signal, and generating the converted result. And a complex coefficient transverse filter for delaying by three values and multiplying by a fourth value, and sequentially adding the multiplied results to produce an imaginary signal.

Hereinafter, with reference to the accompanying drawings will be described in detail the operating principle of the preferred embodiment of the present invention. In the following description of the present invention, detailed descriptions of well-known functions or configurations will be omitted if it is determined that the detailed description of the present invention may unnecessarily obscure the subject matter of the present invention. Terms to be described later are terms defined in consideration of functions in the present invention, and may be changed according to intentions or customs of users or operators. Therefore, the definition should be made based on the contents throughout the specification.

1 is a block diagram of a complex coefficient transversal filter that removes the image of the present invention.

The complex coefficient transverse filter 10 includes a real part signal generator 12 and an imaginary part signal generator 14, and a signal input from the common input terminal A is connected to the real part signal generator 12. Branched to the imaginary part signal generator 14. The real part signal from the real part signal generator 12 is obtained from its output terminal B, and the imaginary part signal from the imaginary part signal generator 14 is obtained from the output terminal C thereof. Each complex coefficient transversal filter includes a delay 16, an adder 22, and a coefficient multiplier 20.

The delay unit 16 plays a role of generating a delay time of one clock, that is, one cycle (T seconds), and it is possible to use, for example, a D flip-flop. The adder 22 and the coefficient multiplier 20 can be configured with a logic circuit. And a coefficient multiplier 20 arranged between the input terminal A of the complex coefficient transverse filter and the adder 22, and a coefficient multiplier 20 for connecting the output of each delay unit and the adder 22 to each other. Let's call it a tap. The coefficient multiplier whose absolute value of the coefficient is 1 is equivalent to the connection line connecting the delay output terminal and one input terminal of the adder to negative or positive polarity, but this is also included in the tap.

For example, in the real part signal generator 12 illustrated in FIG. 1, the first tap is a coefficient multiplier 20 having a coefficient a1, and the second tap is a coefficient multiplier having a coefficient a2. (20). A first delay 16 having a delay time T seconds is inserted between the input terminals of the coefficient multiplier 20. The output from the coefficient multiplier is then added by the first adder 22. In the present embodiment, the coefficient after the third is set to one. The second coefficient from the output terminal B becomes a2 again, and the coefficient connected to the output terminal B becomes a1 again.

In the imaginary part signal generator 14 illustrated in FIG. 1, the output side of the first delay unit 16 having a delay time of half a period (T / 2 seconds) connected to the input terminal A has a coefficient b1. It is connected to the coefficient multiplier 20 and the second delay unit 16 having. In this case, the first tap is the first coefficient multiplier 20. The second tap is the direct connection of the adder 22 with the output of the delay 16 having a delay time T seconds. After the second tap, the coefficient is 1, and the output from the last delayer 16 is output to the output terminal C through the coefficient multiplier 22 having the coefficient b1 like the first tap. do.

In the real part signal generator, the number of real part signal coefficient multipliers 20, i.e., the number of taps, M-M is an integer of 5 or more-, and the real part signal delay unit 16 is (M-1). The number of real part signal adders 22 is (M-1). In the imaginary part signal generator, the number of imaginary part signal coefficient multipliers 20, that is, the number of taps (M-1), the imaginary part signal delay unit 16 (M-1), the imaginary part There are (M-2) signal adders 22. In this embodiment, the absolute values of the coefficients of the first two real part signal coefficient multipliers 20 on the input side of the real part signal generator and the two real part signal coefficient multipliers 20 on the output side are less than one. Each of them is designated, and a total of four real part signal coefficient multipliers 20 corresponding thereto are disposed. The absolute value of the coefficient of the other real part signal coefficient multiplier can be set.

In the imaginary part signal generator 14, the first and last coefficients are designated b1, and a total of two imaginary part signal coefficient multipliers 20 corresponding thereto are arranged. The absolute value of the other coefficients can be set to one tap.

Next, the simulation result of the transversal complex coefficient filter 10 constructed by the digital circuit designed with | a1 | = 0.5, | a2 | = 0.5, and | b1 | = 0.48 will be described.

2 (a) is a graph showing the arrangement of the variable tap coefficients according to the first embodiment of the present invention.

2 (b) is a graph showing frequency characteristics according to the first embodiment of the present invention.

In the simulation, MathWorks' MATLAB was used. And the characteristic of the complex coefficient filter 10 which concerns on a present Example is shown by the thick line. The thin line is a characteristic of the complex coefficient transverse surface acoustic wave filter, which will be described in detail below.

First, in the real part signal generator, a1 = -0.5 at the first tap, a2 = 0.5 at the second tap, and an absolute value until the ninth tap after the third tap. The sign is repeated alternately with a minus and a plus. The tenth tap is a2 = 0.5, and the eleventh tap is a1 = -0.5. On the other hand, in the imaginary part signal generation unit, b1 = -0.48, the sign of which the absolute value is positive plus one and the negative is repeated alternately from the second tap to the ninth tap. Finally, b1 = 0.48. In this case, it is preferable to select with | b1 | = (| a1 | + | a2 |) x (an arbitrary value around 0.4 to 0.6). The symbol represents the polarity of the signal input to the adder, plus represents addition, and minus represents subtraction. As illustrated in Figs. 1 and 2 (a), the polarities of the signals added from the taps are alternately repeated with plus and minus.

In FIG. 2 (b), the horizontal axis represents a normalization frequency and the vertical axis represents the amplitude of the signal passing through the filter in decibels (dB: decibel). The center frequency of the passband is represented by the point of normalization frequency 0.5 in this figure, and the amplitude of this point indicates that the attenuation is almost zero. On the other hand, at the normalized frequency of -0.5 representing the image frequency, the amplitude is about -53 dB and can be sufficiently attenuated compared to the desired frequency. That is, image suppression is possible by the simple filter configuration illustrated in Fig. 2A.

Next, the frequency characteristics of the amplitude when the numerical values of the variable tap coefficients a1, a2, and b1 are changed will be described below.

3 (a) is a graph showing the arrangement of the variable tap coefficients according to the second embodiment of the present invention.

 As illustrated in Fig. 3A, the numerical values of the real part variable tap coefficients a1 and a2 are set to 0.5 as in the first embodiment. The imaginary part variable tap coefficient b1 is 0.5.

3 (b) is a graph showing frequency characteristics according to the second embodiment of the present invention.

The passband characteristic near the desired frequency indicated by the normalization frequency of 0.5 can be said to be a low insertion loss which is almost the same as in the first embodiment. On the other hand, at the image frequency represented by -0.5, the amplitude can be -100 dB or less, and the attenuation amount of the image signal can be increased. However, the band having a high attenuation amount becomes narrower as compared with the first embodiment.

4 (a) is a graph showing the arrangement of the variable tap coefficients according to the third embodiment of the present invention. In the third embodiment, as illustrated in FIG. 4A, a1 = -0.2, a2 = 0.8, and | b1 | = 0.49.

4B is a graph showing frequency characteristics according to the third embodiment of the present invention.

The passband characteristics near the desired frequency allow a low insertion loss which is almost the same as in the first embodiment. On the other hand, at the image frequency, the amplitude is about -58 dB and can be made slightly larger than in the first embodiment. In addition, the band having a high attenuation amount can be made slightly wider than in the first embodiment.

5 is a graph showing a relationship between an imaginary part variable tap coefficient and a frequency characteristic of the present invention. In other words, b1 dependence of the amount of attenuation in the vicinity of the image.

Here, the solid line is a1 = -0.5 and a2 = 0.5, and represents the amount of attenuation when b1 is changed. The dotted line indicates a1 = -0.2, a2 = 0.8, and indicates the amount of attenuation when b1 is changed. At | b1 | = 0.5, the amount of attenuation becomes infinity, and the amount of attenuation falls even if | b1 | is larger or smaller than this.

As described in the first, second, and third embodiments, by changing a1, a2, and b1, it is possible to control a sudden change in amplitude and its frequency characteristic mainly in the vicinity of the image frequency. As a result, by selecting these three variables, it is possible to realize a filter that meets the specifications of the target communication system.

Next, the complex coefficient transversal filter according to the fourth embodiment will be described. In general, for high desired frequencies and for multiple modulation schemes, complex coefficient transversal filters that increase the number of taps and change the coefficients slightly from tap to tap to remove the image more completely. Design.

6 is a graph showing the arrangement of the variable tap coefficients according to the fourth embodiment of the present invention, wherein the number of variable tap coefficients is 41 and the configuration of the fourth embodiment in which the variable tap coefficients gradually change is shown in FIG. Indicates. The upper graph shows the arrangement of the real part tap coefficients and the lower graph shows the arrangement of the imaginary part tap coefficients.

7 is a graph showing frequency characteristics according to the fourth embodiment of the present invention.

At the point of the image frequency of -0.5, the amplitude can be removed by -70 dB or less. A large number of variable taps is 41, but a tap having a coefficient having an absolute value of 1 is disposed near the center portion, whereby the circuit scale can be simplified and power consumption can be reduced. In addition, the dotted line in FIG. 7 shows the amplitude characteristic which consists of fixed coefficients, and has 11 taps. In this case, the real part tap coefficient is (-1, 1, -1, 1, -1, 1, -1, 1, -1,1, -1), and the imaginary part tap coefficient is (-1, 1, -1,1, -1,1, -1, 1, -1,1). The amplitude at the image frequency is about -30 dB.

8 is a diagram illustrating a complex coefficient transversal surface acoustic wave filter 50 according to a fifth embodiment of the present invention.

As a medium for transmitting a surface acoustic wave (SAW), piezoelectric materials such as lithium tantarate (LiTaO 3), lithium nitride (LiNbO 3), and quartz can be used. Advances in microfabrication techniques can achieve sufficient performance even in frequencies above 3 GHz.

The input terminal D corresponding to the input terminal A of the first embodiment is connected to the positive electrode 561 of the transmission electrode portion 56. In addition, the negative electrode 562 is connected to the ground. A comb-like finger 58 arranged on the substrate including the piezoelectric material to be opposed to each other, wherein the finger of the positive electrode and the finger of the negative electrode are alternately repeated, and the input signal This causes a piezoelectric effect and excites surface acoustic waves (SAW). Finger 58 may be multiple.

The transmitted surface acoustic wave SAW serves as the real part positive electrode 521 constituting the real part signal generator 52 and is returned to the electric signal, thereby output terminal corresponding to the output terminal B of the first embodiment. It is output as (E). The sub-electrode 522 constituting the real part signal generator 52 is connected to ground. In this way, a real part signal generator 52 including a positive electrode 521 and a negative electrode 522 having a comb-like finger 58 arranged to face each other is configured.

The finger 58 illustrated in FIG. 8 has a tap coefficient as illustrated in FIG. 1. If the fingers overlap 100% of each other, the tap coefficient is 1; otherwise, the tap coefficient is zero. The real signal generator 52 has a configuration in which a plurality of fingers 58 having a delay time interval of one period (T seconds) are arranged in the surface acoustic wave (SAW) transfer direction.

Moreover, in the finger part which comprises each tap of each receiving electrode, the finger part which connects with the positive electrode is arrange | positioned to the side near the transmission electrode 56, and the next finger part. In 58, a finger connected to the sub-electrode portion is arranged closer to the transmission electrode. Such alternating arrangements correspond to alternating positive and negative signs of a1 and a2.

Similarly, the imaginary part positive electrode 541 constituting the imaginary part signal generator 54 is transmitted to the output terminal F corresponding to the output terminal C of the first embodiment. The imaginary part negative electrode 542 constituting the imaginary part signal generator 54 is connected to the ground. Also in this case, the constant pitch of the finger 58 is set so that the delay time is one period (T seconds). The imaginary positive electrode finger and the imaginary negative electrode finger are one pair less than the real positive electrode finger and the real negative electrode finger, and the delay time is half a period (T / 2 seconds) is disposed so as not to be inside. This has the same function as the digital circuit configuration illustrated in FIG. 1.

The frequency characteristic of the amplitude of the fifth embodiment, which is a complex coefficient surface acoustic wave filter having a tap coefficient of | a1 | = | a2 | = 0.5 and | b1 | = 0.48, represented in FIG. represented by a thin line in b). As in the first embodiment, the insertion loss can be zero at the desired frequency, and the amplitude can be -55 dB near the image frequency. In other bands, noise other than the image can be removed. Similarly, also in FIG. 4 and FIG. 5, it is exemplified that the complex coefficient transversal surface acoustic wave filter has similar characteristics to the complex coefficient transversal filter having the digital circuit configuration.

9 is a diagram illustrating a real part electrode part according to a fifth exemplary embodiment of the present invention.

9 is a schematic plan view of the real part receiving electrode therein. Higher frequencies reduce the width and constant pitch of the finger—corresponding to delay time. In this example, two fingers are constituted by one pair in order to prevent deterioration in characteristics due to unnecessary reflection of surface acoustic wave SAW between the fingers.

In the fifth embodiment, by determining a1, a2, b1 and three types of coefficients, frequency characteristics of amplitude sufficient for image suppression can be obtained as illustrated in Fig. 2B. The coefficient of b1 can be selected to an appropriate size as described in the relation of | b1 | = (| a1 | + | a2 |) x (near 0.4 to 0.6). Moreover, the area | region whose coefficient is not 1 is limited to both ends. Thus, a small part of the acoustic excitation effect of the electrode is possible to the minimum and the insertion loss of the filter is small. In addition, the size can be reduced.

In the first and fifth embodiments, the filter design becomes very simple with a simpler circuit configuration than the various filter circuits conventionally applied. Moreover, since a transversal filter can be applied to either a digital circuit or an analog circuit, the application range is very wide. In addition, by designing at least three absolute values of the tap coefficients, it is possible to select the attenuation characteristics of the filter according to the application, such as to prioritize the image suppression ratio or to remove the image signal at two frequencies. have.

Next, an operation of suppressing an image by using such a simplified complex coefficient filter will be described in detail.

FIG. 10 is a graph illustrating frequency conversion in a low IF receiver.

The desired frequency with the center frequency fc represents its system signal 1000 constituting a multiband or wideband system. An image signal 1002 for this own system signal exists around -fc. In addition, other system signals and noise 1004 exist near the own system signal. For example, in the upper graph of FIG. 10, it exists in a frequency band slightly lower than its system signal, and this overlaps with another system signal or noise in the vicinity of the image signal (right side of the image signal in this figure). 1006). It is difficult to remove other system signals or noise in the RF Bandpass Filter 1008 having a passband characteristic illustrated in the dotted line in FIG. 10.

The intermediate frequency (IF = fc-fL) of the Low IF method is selected lower than the superheterodyne method, for example, in the range of several MHz or less. In the lower graph of Fig. 10, it is easy to extract only (fc-fL) because the frequency difference between the (fc-fL) signal and the (fc + fL) signal is large. In addition, the desired frequency (fc-fL) converted from its system signal overlaps the image signal of another system signal or noise (-fc + fL), and also the image of its system signal. This image disturbance greatly reduces the communication quality.

Fig. 11 is a block diagram showing one example of the configuration of a receiver using the transversal complex coefficient filter 10 according to the present invention.

The signal is received by the antenna 60, amplified by a low noise amplifier (LNA) 62, and the digital circuit configuration of the first embodiment or the transverse complex coefficient of the SAW of the second embodiment. Filtering is performed by the filter 10. Thereafter, demodulation is completed via the frequency converter 64 and the demodulator 66 as the down converter. The demodulator 66 demodulates the modulated signal into base bands. Further, when the complex coefficient filter 10 is constituted by a digital circuit, an A / D converter 68 illustrated in a broken line is inserted between the LNA 62 and the transversal complex coefficient filter 10.

In addition, the operation of suppressing the image using the complex coefficient filter according to the present invention will be described.

FIG. 12 is a graph illustrating a radio frequency (RF) signal before filtering in a low IF receiver using a complex coefficient transversal filter to remove an image of the present invention.

The desired wave having the center frequency fc is represented by its system signal 1000 constituting a multiband or wideband system. An image signal 1002 for this system signal exists around -fc. In addition, other system signals and noise 1004 exist near the own system signal. For example, in the first graph of FIG. 12, the signal is present at a slightly lower frequency band than its own system signal, and the superimposed one is located near the image signal (the right side of the image in this figure). This exists. The transversal complex coefficient filter according to the present invention has a band characteristic 1010 for passing its system signal centered on fc and blocking an image signal centered on -fc.

The second graph of FIG. 12 is a graph showing a radio frequency (RF) signal after filtering in a low IF receiver using a complex coefficient transversal filter to remove an image of the present invention.

Most of their system signals enter the frequency converter without attenuation. On the other hand, the image signal is blocked from passing through large attenuation by the complex coefficient transversal filter 10. The other system signal or noise is located at the end of the passband, passes slightly while attenuating, and is in the vicinity of its system signal. On the other hand, an image of another system signal or noise is blocked for large attenuation like an image signal. That is, in the step of passing through the complex coefficient transversal filter 10, most of the image signal can be removed.

The third graph of FIG. 12 is a graph illustrating a signal after frequency conversion in a low IF receiver using a complex coefficient transversal filter to remove an image of the present invention.

The intermediate frequency (IF = fc-fL) of the IF stage of the low IF method is lower than that of the superheterodyne method and is selected, for example, in a range of several MHz or less. In Fig. 12C, the (fc-fL) signal and the (fc + fL) signal can be made to extract only (fc-fL) because the difference in frequency is large. In addition, the desired frequency (fc-fL) converted from its system signal may overlap another image signal or noise image signal (-fc + fL), but this image signal is sufficiently suppressed by a complex coefficient transversal filter. The reception performance does not decrease. In addition, the direct conversion method can be described in almost similar operation.

In addition, the low IF method using the complex coefficient transversal filter 10 will be described in comparison with the conventional super heterodyne method.

13 is a graph showing a receiver block diagram of a super heterodyne system.

The signal from the antenna 60 is first input to an RF bandpass filter to select a desired frequency. The signal amplified by the low noise amplifier (LNA) 62 and the local signal from the local oscillator 65 are then multiplied by the mixer 66 and converted to the intermediate frequency IF.

FIG. 14 is a schematic diagram showing signals at each step of the block diagram illustrated in FIG.

14 is a graph showing a radio frequency (RF) signal in a super heterodyne receiver.

Here, it is a real signal at the radio frequency (RF) stage, its system signal 1000 exists near the center frequency fc, and other system signal or noise 1004 exists in a slightly lower frequency band. In the RF bandpass filter 1008 having a pass characteristic represented by a dotted line, other system signals or noise cannot be sufficiently removed. Also, near -fc, there is a repeated image of them.

The second graph of FIG. 14 is a graph showing an intermediate frequency (IF) signal in a super heterodyne receiver.

Here, it is a real signal at the intermediate frequency IF stage, and the local oscillation signal fL from the local oscillator 65 and the center frequency fc are multiplied by the mixer 66 to generate an IF signal f _IF . . In this case, another system signal or noise 1004 is also frequency-converted and generated near the IF signal f _IF (1014). The repeated images are generated near the negative IF signal (-f _IF ) (1016). The narrowband may also be filtered by an IF bandpass filter having steep attenuation characteristics (1012) to remove other frequency-converted system signals or noise components disclosed in the second graph of FIG. Moreover, the intermediate frequency IF in the super heterodyne system is generally high, ranging from tens to hundreds of MHz. As described above, image suppression is performed by dual filtering between the RF bandpass filter 70 and the IF bandpass filter 72.

The third graph of FIG. 14 is a graph showing a final signal in a super heterodyne receiver.

In contrast, in the direct conversion method illustrated in FIG. 11, an RF bandpass filter is unnecessary, and the signal is substantially transverse without passing through the RF bandpass filter. It is input to the (transversal) type complex coefficient filter (10). In addition, since the frequency of the IF signal is low or substantially absent in comparison with the super heterodyne scheme, an intermediate frequency band pass filter having a wideband or steep attenuation characteristic is unnecessary. As described above, in the direct conversion communication system using the transversal complex coefficient filter 10 of the present embodiment, image signal suppression is possible without requiring double filtering, and the system can be simplified. Do.

15 is a block diagram illustrating a receiver of a low IF system using a complex coefficient transversal filter to remove an image of the present invention.

The real part signal is input from the terminal A, and the real part signal of the complex signal is output to the real part signal generator 12 in the complex coefficient transversal filter 10 as in the first and fifth embodiments. . On the other hand, the real signal is input to the imaginary part signal generator 14 from the terminal A, and the imaginary part signal of the complex signal is output. The real part signal and the imaginary part signal have a phase difference of 90 degrees.

The local oscillator 80 has a frequency of a difference between a radio frequency (RF) signal and an intermediate frequency (IF) signal, and outputs a complex local signal in which a real part is a cosine wave and a imaginary part is a sine wave. . In the direct conversion method, a local signal having the same frequency as a radio frequency (RF) signal is used. The mixer 86 constituting the complex mixer 82 multiplies the real part of the complex side signal and the real part of the complex local signal and inputs it to the positive input terminal of the subtractor 90. The mixer 94 multiplies the output of the real-side side transverse filter 12 by the imaginary part of the complex local signal and inputs the result to one input terminal of the adder 96.

The mixer 88 multiplies the output of the imaginary part signal generator 14 and the real part of the complex local signal and outputs it to the other input terminal of the adder 96. The mixer 92 further comprises an imaginary part signal generator 14. Multiply the output of the multiplier by the imaginary part of the complex local signal and input the result to the sub-input terminal of the subtractor 90.

The subtractor 90 subtracts the output signal of the mixer 92 from the output signal of the mixer 86 and outputs the result to the terminal G as a real part of the complex signal. The adder 96 adds the output signal of the mixer 88 and the output signal of the mixer 94, and outputs the result to the terminal H as an imaginary part of the complex signal.

The baseband generator 108 includes the BPFs 100 and 101, the AGC amplifiers 102 and 103, the A / D converters 104 and 105, the local transmitter 122, and the precomplex mixer 130. And an imbalance corrector 120 and LPFs 116 and 117.

The band pass filters (BPFs) 100 and 101 pass complex bands with a predetermined band centered on negative and positive intermediate frequency (IF) signal frequencies and output complex signals. The automatic gain control amplifiers (AGC Amplifiers) 102 and 103 control the gain in accordance with the voltage input to the terminal L.

The analog / digital converters 104 and 105 convert the analog output of the automatic gain control amplifier (AGC Amplifier) into a digital signal and input the complex signal output to the imbalance corrector 120. By analog-to-digital conversion (A / D conversion), digital signal processing is possible in a demodulator at a later stage.

The imbalance corrector 120 is composed of a compensation value memory 106 and a multiplier 118, and digitally corrects an output signal amplitude difference and a phase difference (unbalance) in two analog digital converters (A / D converters). do. As a result, image suppression can be better performed in the target band. In the compensation value memory 106, the amplitude ratio and the phase difference (compensation value) of the analog signal processing section before the two analog digital converters (A / D converters) are stored in advance. The multiplier 118 multiplies the output signal of the analog-to-digital converter (A / D converter) 105 with the compensation value input from the compensation value memory 106, and outputs the result.

The local oscillator 122 outputs a complex local signal having the same frequency as the intermediate frequency IF signal. Since the complex mixer 130 has the same configuration as that of the complex mixer 82, a baseband signal including the frequency zero component at the terminal J and the terminal K can be obtained as a complex signal by almost the same operation. In the portable wireless communication receiver multibanded and widened by such a down converter, image removal is possible.

The present invention is not limited to the receiver, and the image suppression can be similarly performed at the transmitter. A transmitter using a complex coefficient transversal filter to remove an image is described as follows.

16 is a block diagram illustrating a transmitter using a complex coefficient transversal filter to remove the image of the present invention.

The baseband signal is modulated by the modulator 130 and then upconverted by the complex local signal by the frequency converter 132, which is an up-converter, and input to the complex coefficient transverse filter 134. It is then amplified by a power amplifier (PA) 138 and a signal is transmitted from the transmitting antenna 140. Furthermore, in the case where the complex coefficient transversal filter 134 is a digital filter composed of digital circuits, the digital-to-analog converter 136 is provided between the amplifier 138 and the complex coefficient transversal filter 134. Insert it.

17 is a graph illustrating a baseband signal and a radio frequency (RF) in a direct conversion transmitter.

After multiplying the real signal, which is a real signal, by the frequency converter 132 in the mixer, the desired frequency signal 1018 and the image disturbance signal 1020 exist near the center frequency fc as shown in Fig. 17 (b). Images 1022 and 1024 where they are repeated exist near -fc.

18 is a schematic diagram illustrating image suppression using the complex coefficient transversal filter 134 of the present invention.

18 is a graph showing a baseband signal in a direct conversion transmitter using a complex coefficient transversal filter to remove an image of the present invention.

The second graph of FIG. 18 is a graph showing a radio frequency (RF) signal before filtering 1026 in a direct conversion transmitter using a complex coefficient transversal filter to remove an image of the present invention.

18 is a graph showing a radio frequency (RF) signal after filtering in a direct conversion transmitter using a complex coefficient transversal filter that removes an image of the present invention.

18 is a graph showing a radio frequency (RF) signal in a direct conversion transmitter using a complex coefficient transversal filter to remove an image of the present invention.

The complex signal at the baseband stage is up converted. In the radio frequency (RF) stage before filtering, the desired frequency 1028 and the image jammer 1030 are near the center frequency fc, and the image signals 1032 and 1034 are near -fc.

The complex coefficient transversal filter 134 according to the present invention has a passing characteristic such as a dotted line, and the image signal near -fc in the radio frequency (RF) stage has a large attenuation amount, so that the image signal can be made very small. As a result, image suppression becomes possible, and transmission output which is a real signal with little image influence can be obtained. This up-converter and a transmitter using the same are suitable for multiband and broadband portable wireless communication systems and the like.

The above description is merely illustrative of the technical idea of the present invention, and those skilled in the art to which the present invention pertains may make various modifications and changes without departing from the essential characteristics of the present invention.

Therefore, the embodiments disclosed in the present invention are not intended to limit the technical idea of the present invention but to describe the present invention, and the scope of the technical idea of the present invention is not limited by these embodiments.

Even if a person skilled in the art makes various design changes regarding the shape, size, arrangement, etc. of the circuit element, SAW material, electrode, etc. constituting the complex coefficient transversal filter, they are included in the present invention as long as they do not depart from the gist.

The protection scope of the present invention should be interpreted by the following claims, and all technical ideas within the equivalent scope should be interpreted as being included in the scope of the present invention.

As described above, according to the present invention, an image suppression ratio can be improved in a wide band by applying a complex coefficient transversal filter that removes an image by using a phase delay of a real part signal and an imaginary part signal. In addition, there is an effect that can simplify the circuit scale and reduce power consumption.

Claims (24)

Delaying the input signal by an integer multiple of one period to generate at least two real part delayed signals, and multiplying the input signal and at least one of the real part delayed signals by an absolute value of the real part variable tap coefficient A real part signal generation unit configured to generate at least two real part multiplication signals and to sequentially add pairs of the at least two real part multiplication signals to generate the real part signal; And Delaying the input signal by a half cycle to generate a phase delay signal, delaying the phase delay signal by an integer multiple of one cycle to generate at least two imaginary delay signals, and generating the phase delay signal and the imaginary delay signal. Multiply at least one signal by an absolute value of an imaginary variable tap coefficient to generate at least two imaginary part multiplication signals, and sequentially add and add the at least two imaginary part multiplication signals in pairs to form the imaginary part signal Complex coefficient transversal filter comprising an imaginary part signal generator for generating a. The method of claim 1, The real part signal generator, M multiplying at least one of the input signal and the real part delay signal by an absolute value of the real part variable tap coefficient to generate the real part multiplication signal, where M is an integer of at least 5 A complex coefficient transverse filter comprising: a real part signal coefficient multiplier of a; The method of claim 2, The imaginary part signal generator, (M-1) imaginary part signal coefficient multipliers for multiplying at least one of the phase delay signal and the imaginary part delay signal by an absolute value of the imaginary part variable tap coefficient to generate the imaginary part multiplication signal Complex coefficient transversal filter, characterized in that it comprises. The method of claim 1, The real part signal generator, (M-1) receiving at least two signals of the real part multiplication signal through a polarization terminal and adding the signal input through the polarization terminal to generate the real part signal, wherein M is at least Is an integer of 5 or greater, including an adder of, The polarization terminal is a complex coefficient transversal filter, characterized in that for giving the input signal alternating the plus polarity or minus polarity. The method of claim 1, The imaginary part signal generator, And (M-2) adders for receiving at least two signals of the imaginary part multiplication signal through a polarization terminal and adding the signals input through the polarization terminal to generate the imaginary part signal. The polarization terminal is a complex coefficient transversal filter, characterized in that for giving the input signal alternating the plus polarity or minus polarity. The method of claim 3, wherein The M real part signal coefficient multipliers are First, second, M-1, and Mth real part signal coefficient multipliers having an absolute value of the real part variable tap coefficient less than one; And And a third to M-th real part signal coefficient multiplier having an absolute value of the real part variable tap coefficient of one. The method of claim 6, The (M-1) imaginary part signal coefficient multipliers, The first and M-1 imaginary part signals in which the absolute value of the imaginary part variable tap coefficient is in a range of 0.4 to 0.6 multiplied by the sum of the absolute values of the real part variable tap coefficients of the first and second real part signal coefficient multipliers. Coefficient multiplier; And And a second to M-th imaginary part signal coefficient multiplier whose absolute value of the imaginary part variable tap coefficient is one. The method of claim 1, The real part signal generator, (M-1) real part signal delayers which receive a signal and delay one period and output the signal as the real part delay signal; And said (M-1) delay units are arranged in series with each other. The method of claim 1, The imaginary part signal generator, A phase delayer delaying the input signal by a half cycle to generate a phase delay signal; And (M-2) imaginary part signal delayers receiving a signal and delaying one period to output the imaginary part delay signal; Including, A first imaginary signal delay unit of the (M-2) imaginary signal delay units is connected in series with the phase delay unit, and the (M-2) imaginary signal delay units are connected in series with each other. Complex coefficient transversal filter. The method of claim 1, And converting the input signal into a surface acoustic wave shape and transmitting the signal to the real part signal generator and the imaginary part signal generator. The method of claim 10, The transmission electrode unit, A transmitter positive electrode and a transmitter negative electrode composed of comb-shaped fingers arranged to face each other on a substrate including piezoelectric material, and through a transmitter input terminal connected to the transmitter positive electrode and the transmitter negative electrode, respectively. And a surface coefficient wave type signal transmitted through a transmitter output terminal respectively connected to the transmitter positive electrode and the transmitter negative electrode. The method of claim 10, The real part signal generator, A real part positive electrode and a real part negative electrode composed of comb-shaped fingers arranged to face each other on a substrate including a piezoelectric material, and a real part connected to the real part positive electrode and the real part negative electrode, respectively. And a real part electrode part configured to receive a surface acoustic wave type signal through a negative input terminal and to receive an electrical signal through the real part output terminal. The method of claim 10, The imaginary part signal generator, An imaginary part positive electrode and an imaginary part negative electrode composed of comb-shaped fingers arranged to face each other on a substrate including a piezoelectric material, and an imaginary part connected to the imaginary part positive electrode and the imaginary part negative electrode, respectively. And a imaginary part electrode part receiving a surface acoustic wave type signal through a negative input terminal and an electric signal applied through an imaginary part output terminal. The method of claim 12, The real part electrode part, Complex coefficient transversal filter, characterized in that the absolute value of the real variable variable tap coefficient is adjusted according to the degree of overlap of the finger of the real part positive electrode and the finger of the real part negative electrode. . The method of claim 13, The imaginary part electrode part, Complex coefficient transversal filter, characterized in that the absolute value of the imaginary variable tap coefficient is adjusted according to the degree of overlap of the finger of the imaginary part positive electrode and the finger of the imaginary part negative electrode. . The method of claim 12, The real part electrode part, And a finger of the real positive electrode and a finger of the real negative electrode are alternately repeated. The method of claim 13, The imaginary part electrode part, And a finger of the imaginary positive electrode and a finger of the imaginary negative electrode are alternately repeated. The method of claim 12, The real part electrode part, And a delay coefficient according to a constant pitch of the real part finger. The method of claim 13, The imaginary part electrode part, And a delay coefficient is adjusted according to a constant pitch of the imaginary fingers. (a) delaying an input signal by a first value and multiplying by a second value, and sequentially adding the multiplied result to generate a real part signal; And (b) delaying the input signal by a third value and multiplying by a fourth value, and sequentially adding the multiplied results to generate an imaginary part signal. The method of claim 20, Step (a) is Delaying the input signal by an integer multiple of one period to generate at least two real part delay signals; Multiplying at least one of the input signal and the real part delay signal by an absolute value of a real part variable tap coefficient to generate at least two real part multiplication signals; And And sequentially pairing and adding the at least two real part multiplication signals to generate the real part signal. The method of claim 20, Step (b) is Delaying the input signal by a half period to generate a phase delay signal; Delaying the phase delay signal by an integer multiple of one period to produce at least two imaginary delay signals; Multiplying at least one of the phase delay signal and the imaginary delay signal by an absolute value of an imaginary variable tap coefficient to produce at least two imaginary part multiplication signals; And And sequentially pairing and adding the at least two imaginary part multiplication signals to generate the imaginary part signal. Delaying the input signal by a first value and multiplying by a second value, sequentially adding the multiplied results to generate a real part signal, delaying the input signal by a third value, and multiplying by a fourth value A complex coefficient transverse filter configured to sequentially add the multiplied result to generate an imaginary part signal; A frequency converter for converting a frequency of the result generated by the complex coefficient transversal filter; And And a demodulator for demodulating the result converted by the frequency converter. A modulator for modulating the baseband signal; A frequency converter for converting a frequency of the result modulated by the modulator; And Delaying the input signal by the first value by a first value and multiplying the result by the frequency converter by a second value, sequentially adding the multiplied result to generate a real part signal, and converting the converted result by a third And a complex coefficient transverse filter for delaying by a value and multiplying by a fourth value and sequentially adding the multiplied result to generate an imaginary part signal.

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