JPS6123758B2 - - Google Patents

Description

[Detailed description of the invention]

The present invention relates to an electric motor control device that reduces torque pulsation in a self-limiting synchronous motor such as a transistor motor. In permanent magnet rotating field type audio and video motors, if torque ripple occurs and speed fluctuations occur, the sound quality and image quality will deteriorate, so it is necessary to take measures to reduce the ripple. Motor torque is defined as being proportional to the product of armature current and field magnetic flux density. Controlling the motor armature current to a constant value is relatively easy, but in this case, the field magnetic flux density changes sinusoidally with changes in the rotor position, so the torque cannot be made constant regardless of the rotor position. Therefore, as an example of the torque ripple reduction method, a method has been proposed in which the field magnetic flux density or the corresponding speed electromotive force is detected and fed back, and a current value is calculated to keep the torque constant, thereby controlling the current. However, the means for detecting the field magnetic flux density requires a speed electromotive force detection winding, a magnetic flux detection element, or a rotation detector that can obtain a signal equivalent to the field magnetic flux density and can be used as a substitute. Furthermore, a current calculator is required to calculate the desired current. As described above, the conventional control device requires some kind of calculation means to reduce the torque ripple of the motor and has a complicated configuration. The present invention provides a torque ripple reduction method with a simple configuration by controlling the input DC voltage of an inverter that switches the armature winding of a motor to a constant value. FIG. 1a shows the basic configuration of the motor of the present invention. In the figure, 1 is a motor, 2 is an inverter, 3 is a voltage controller, and 4 is an inverter energization controller. The motor 1 is, for example, a rotating field type in which a permanent magnet is used as a field rotor, and has a stator having multiphase AC windings. The inverter 2 is a bridge type inverter having a number of phases equal to the number of phases of the motor 1, and an inverter energization controller 4 controls opening and closing of each of the constituent transistors according to the position of the rotor. The voltage controller 3 receives the voltage reference signal Vs as an input signal and controls the input DC voltage to the inverter to a desired value. It is assumed that the field magnetic flux interlinking with the AC winding changes approximately sinusoidally as the motor rotates. Further, the inverter 2 applies an AC voltage having a predetermined phase difference to each phase AC winding of the motor 1. Next, to explain the operation and effects of the present invention, a three-phase motor in which each winding is star-connected will be described as an example. The configuration is shown in Figure 1b. Now, when the DC input voltage Vdc of the inverter is constant, the inverter DC input current idc is limited to the speed electromotive force eu~ew of each phase winding and the winding resistance R _M selected by the energization control of the inverter 2. It becomes an electric current. It goes without saying that the inverter DC input current idc is equal to each phase AC winding current selected by the inverter. On the other hand, the field magnetic flux density interlinking with each phase AC winding is in phase with the velocity electromotive force of each phase. It is also assumed that the speed electromotive force has a value proportional to the speed, and the peak value of the field magnetic flux density is Bm. The individual torques of each phase winding are τu, τv,
τw is each phase current (iu ~ iw) and each phase field magnetic flux density
Proportional to the product of Bu, Bv, and Bw. The motor generated torque τm (the sum of τu to τw) is the sum of the torques generated by the two-phase windings that are selected and fed by the 120° energization control of the inverter. As a result, the motor generated torque is equivalently expressed as the product of the magnetic flux density Bτ, which is the sum of the interlinkage magnetic flux densities of each phase, and the inverter DC current idc. In FIG. 1b, 3a is a battery, 5 is a rotor position detector, 8 is a detection means for inverter input voltage Vdc, and this output Vf is added with a different sign to the setting signal Vs, and transistor 6 is added to amplifier 7. As a result of the control, it is assumed that the Vdc voltage is controlled to a constant value. FIG. 2 is a state transition diagram for explaining the above process. In the figure, 1 is the velocity electromotive force waveform of each phase eu~ew
2 is the inverter DC input voltage Vdc and each phase electromotive force eu
~Waveform where ew is rectified by inverter energization control
edc, and 3 is the inverter DC input current idc where the difference voltage between the inverter DC input voltage Vdc and the electromotive force rectified waveform edc in 2 is controlled by the winding resistance _RM and becomes a waveform including ripples. show. 4 to 6 show the field magnetic flux density Bu to Bw of each phase and the current iu to iw of each phase, and 7 shows the inverter DC input current IDC and each phase of the winding selected by the inverter energization control. In the composite magnetic flux density Bτ of the field magnetic flux densities Bu to Bw, 8 represents the motor generated torque τθ. In other words, the electromotive force shown in FIG. Since the voltage obtained by subtracting the power rectified waveform e _dc from the inverter DC input voltage V _dc is applied to the windings for the two energized phases, the current flowing to these windings is determined by the winding resistance of each phase winding. R _M
(Actually, since it is for two phases, the current is limited by 2R _M ). Therefore, the inverter DC input current i _dc
The waveform in FIG. 2 shows the shape obtained by subtracting e _dc from V _dc in FIG. 2, and it is shown in _FIG.
The ripple state of is opposite to that of e _dc . Here, the motor generated torque τθ will be explained. If the input DC voltage V _dc of this inverter is controlled to a constant value, π6/6<θ<π/6 from Fig. 2 7 and 8.
, the ripple of τθ is one period, and the magnetic flux density Bτ and inverter DC current idc are respectively
It is expressed by equations (1) and (2). Bτ=√3Bm cosθ (1) However, Bm is the field magnetic flux density However, e _n is the peak value of the speed electromotive force of each phase winding, so the torque τθ at the rotation angle θ is τθ = Kτ・Bτ・idc = Kv cosθ{1−K′v cosθ} (3) However, Kτ: Torque constant. By the way, the torque τθ(0) at θ=0 and the torque τθ(π/6) at θ=±π/6 are respectively τθ(0)=Kv(1−K′v) (6) becomes. When these are equal, K′v=4−2√3 is obtained as a condition for K′v. Inverter DC input voltage at this time
Vdc is from equation (5) becomes. That is, the DC input voltage Vdc of the inverter is the voltage √3 induced in the armature windings connected in series.
It is almost twice that of em. By the way, the inverter DC current idc and magnetic flux density Bτ when satisfying this equation (9) are as shown in FIG. The magnetic flux density Bτ is the maximum value √3Bm at θ-0, and its maximum value is √3Bm at θ=±π/6.

[Formula] Doubles. On the other hand, the inverter DC current idc can be expressed separately into a DC current I0 and a current increment Δi. This current increment △i changes with the rotation angle, is zero at θ=0, and θ=
The maximum value is △1 at ±π/6. That is, from equation (2), the inverter DC input current i _dc exhibits a waveform change that repeats in one cycle with −π/6≦θ≦π/6, and when θ=0, i _dc takes the minimum value.
My suggestion is that the value of i _dc when θ=0 is the direct current I ₀
is giving. Therefore, On the other hand, since i _dc shows the maximum value when θ=±π/6, the increment Δ1 from the minimum value I ₀ is expressed by the following equation. In addition, when the inverter DC input voltage V _dc satisfies equation (9) in relation to the peak value e _n of the speed electromotive force generated in each phase winding, the torque τθ
The difference between the maximum value and the minimum value of , that is, the torque ripple becomes minimum in the range of -π/6≦θ≦π/6. In other words (3
) The torque τθ shown in equation ( ) is the torque ripple when Kv′ is given when satisfying τθ(0)=τθ(±π/6), that is, Kv′=4−2√3 in equation (8). becomes the minimum. Torque ripple increases whether Kv' is larger or smaller than 4-2√3. Kv′=4−2√
3, the minimum value of torque τθ is τmin=τθ
(0) = τθ (±π/6) = Kv (2√3-3), and the maximum value τmin is θ = arccos (1/2 Kv') ≒ 21.1 (degrees)
At the time of It is. Therefore Therefore, the torque cripple is 0.5%. This torque τθ is normalized and shown in FIG. 3. Here, we will add an explanation about the relationship between Kv′ and torque τθ. Torque τθ is −π/6≦θ≦π/
It shows a shape change that repeats with 6 as one cycle, and also shows a symmetrical shape when -π/6≦θ≦π/6 with respect to θ=0, so the change in torque τθ over O≦θ≦π/6. Just look at it. The results of analyzing the torque τθ in this range are summarized as follows. especially At this θ = ±21.1 (degrees), the peak value of the speed starting force induced in the two-phase windings when connected in series is √
3e _n cos (±21.1°) ≒ 1.616e _n . On the other hand, in the case of Kv′=4−2√3, from equation (5), V _dc
= √3e _n /Kv'≒3.232e _n , so the voltage drop due to the resistance of the winding is 3.232e _n -1.616e _n = 1.616e _n
This is equal to the peak value of the speed electromotive force. Rewriting equation (2) gives the following equation. however here Therefore, i _dc =Jo+△J(1-cosθ) =Jo{1+△J/Jo(1-cosθ)}=Jo{
1+ KJ(1−cosθ)} Therefore, equation (3) can be written as the following equation. τθ=Kτ・Bτ・i _dc =√3KτBm・cosθ・
Jo・{1+KJ(1−cosθ)} =√3KτBmJo{1+KJ(1−cosθ)}
cosθ……From equation (12) and equation (13) In other words, since KJ has a relationship with Kv' as shown in equation (13), Kv' = 4, which minimizes the torque ripple, as described above.
- In the case of 2√3

[Formula] becomes. At this time, the difference between τ _nio and τ _nio is the minimum, and τ _ni
o /τ _nio ≒1.005, so torque ripple is approximately 0.5%
It is. From the above equation (15), the above equation A becomes as follows.

[Formula] That is, when 1.16≦KJ≦1.37, τ _nio = T (θ ₁ ) where cosθ ₁ = 1/2Kv'
= 1+KJ/2KJ τ _nio =T(O) Also

[Formula] That is, when 1≦KJ≦1.16, τmin=T(θ ₁ ) However, cosθ ₁ = 1/2Kv' = 1+KJ/2KJ τmin=T(π/6) Here, torque ripple is defined as τ _nio /τ _nio . There is a diagram. As can be seen from this figure, when KI≒1.16, the torque ripple, that is, the difference between τmax and τmin, becomes the minimum,
This corresponds to about 0.5% in the above explanation. When the rotational speed of the motor is almost constant, if the resistance of the armature winding circuit is constant, △I is also constant,
KI decreases and increases as the load increases and decreases, but
Corresponding to this ratio, the difference between τmax and τmin, that is, the torque ripple, both increases. However, the ratio is small; for example, the torque ripple only increases by about 3% for a ±20% variation in Io, indicating that a sufficiently small torque ripple can be achieved. Furthermore, if
When the load torque is constant, even if the rotational speed fluctuates by ±20%, it can be seen that the torque ripple only increases by about 3% because it can be assumed that only ΔI changes by about ±20%. It can be seen that by performing constant voltage control in this manner, the torque ripple can be controlled within a small value range that has little effect on the magnitude of the load and the increase/decrease in rotational speed. As described above, the control device of the present invention has a simple configuration that does not use a field magnetic flux detection means or a current calculator.
If formula (9) or formula (10) is satisfied, torque ripple can be drastically reduced. FIG. 5a shows a first embodiment in which an additional resistor Ro is inserted in series with the motor winding in order to equivalently vary the winding resistance R _M in order to satisfy the relationship of equation (10) of the present invention. This is an example. By adjusting this resistance value, the desired minute torque ripple can be achieved. This method is easy and simple to adjust when performing constant load and constant rotation speed control, when the motor characteristics that satisfy equation (10) are unknown, or when the desired motor characteristics cannot be obtained. be. Figure 5b shows the additional resistance shown above in order to speed up the start-up.
This is an example in which the first embodiment is improved by inserting a switch SW in parallel with Ro. This switch
The SW is short-circuited at startup and opened when the rated rotation speed is reached. It goes without saying that this increases the starting current, shortens the starting time, and thereafter provides the desired minute torque ripple. Note that this switch (SW) may be a semiconductor switch element such as a transistor thyristor. FIG. 6 is an example in which an additional resistor Ro is inserted in series with the inverter DC input current path in FIG. 5a of the first embodiment. This embodiment has almost the same effect as the first embodiment, but this case has the advantage that the number of additional resistors Ro can be reduced to one. Furthermore, by providing a switch in parallel with the additional resistor Ro, it is possible to eliminate the start-up problem as in the embodiment of FIG. 5b. In addition, in Figure 1b, 9
Placing the position Ro in is consistent with this method. Although a detailed explanation will be omitted here, in the case of two-phase, for example, the torque ripple can be reduced to about 3% as shown in FIG. As described above, according to the present invention, it is possible to perform constant voltage control of the inverter DC input voltage and reduce torque ripple without using a field magnetic flux detection means or a current calculator. In particular, if the relationship between the motor speed electromotive force em and the inverter DC voltage Vdc satisfies the above equation (9), the torque ripple can be approximately 0.5%. Similarly, if inverter DC voltage constant control is performed for a motor with multiple phases other than three phases, a current increment that contradicts the increase and decrease of the field magnetic flux can be obtained.
It can be seen that the maximum torque can be made almost equal and that torque ripple can be reduced.

[Brief explanation of the drawing]

Fig. 1 is a basic circuit diagram of the transistor motor of the present invention, Figs. 2 to 4 are operation explanatory diagrams, and Figs.
FIG. 7 is a configuration diagram showing an embodiment of the transistor motor of the present invention. In the figure, 1 is a motor, 2 is an inverter, and 3 is a motor.
is a voltage controller, 4 is an inverter energization controller, 5 is a position detector, and Ro is an additional resistor. Note that the same reference numerals in each figure indicate the same or corresponding parts.

Claims (1)

[Scope of Claims] 1. A multi-phase synchronous motor having a multi-phase armature winding and a field consisting of a plurality of poles, a rotor position detection means for detecting the position of the rotor of this multi-phase synchronous motor,
an inverter that connects a plurality of the armature windings in series and switches and controls the current flowing through them according to the output of the rotor position detection means; and a voltage induced in the series-connected armature windings. A control device for an electric motor, comprising: voltage control means for maintaining the input DC voltage of the inverter at a voltage value approximately twice that of the DC voltage input to the inverter. 2. The motor control device according to claim 1, wherein the armature winding circuit has an impedance other than that of the armature winding.