JPS5821518B2 - Museiliyushidendoukinoseigiyosouchi - Google Patents

Description

[Detailed description of the invention] The present invention relates to a control device for a commutatorless motor.

In general, in non-commutator motors in which the armature current of a synchronous motor is controlled in a sinusoidal manner, it is possible to control the current flowing in the U phase, ■ phase, and W phase in a sinusoidal manner by synchronizing the electromotive force of each phase. Performs speed change control.

At this time, each phase current is controlled according to a current reference value by a converter connected in a cross connection or in anti-parallel connection.

FIG. 1 shows a conventional U-phase current control system of a control device using cross-wired converters.

The armature current for the V phase and the W phase is also controlled by the same configuration.

In this figure, a positive side converter 2 that supplies a positive direction current and a reverse side converter 3 that supplies a negative direction current are connected to the U phase of an armature winding 1 of a synchronous motor.

A DC reactor 4 is provided between the positive converter 2 and the reverse converter 3 to suppress circulating current between them.

A secondary winding separately provided in a transformer 6 is connected to the positive side converter 2 and the reverse side converter 3, respectively.
A three-phase alternating current power supply 5 is connected to the primary winding.

A current transformer 7A is inserted between the positive converter 2 and the secondary winding, and a current transformer 7B is inserted between the reverse converter 3 and another secondary winding. The output current of each converter is detected.

These current transformers 7A and 7B each have a current detection circuit 8A,
8B is connected to output a voltage proportional to the output current of the current transformer.

These are added by a combining circuit 9 and applied to an adder 10.

On the other hand, the current reference value ST is added to the adder 10.

This current reference value ST is generally the product of the output voltage of the speed control device, the armature voltage of the synchronous motor, and the in-phase voltage.

The output of the adder 10 is applied to a switching circuit 13 via a current control circuit 11 and a phase control circuit 12.

The switching circuit 13 switches and applies the output of the phase control circuit 12 to either the positive side converter 2 or the reverse side converter 3.

On the other hand, the current reference value ST is also applied to the switching logic circuit 14, and the switching logic circuit 14 causes the switching circuit 1 to
3 switching is controlled.

In the above configuration, when the polarity of the current reference value ST is positive, the switching logic circuit 14 selects the positive converter 2 and controls its firing phase.

As a result, the armature current is controlled to be proportional to the current reference value ST.

Next, when the polarity of the current reference value ST is reversed to negative, the switching logic circuit 14 selects the reverse converter 3.

As a result, the negative half cycle of the armature current is controlled.

By the way, in such a cross connection method, the positive side converter 2,
Since each AC power source of the reverse side converter 3 is composed of separate transformer windings and is insulated from each other, and is provided with a DC reactor 4 for suppressing circulating current, the forward side conversion is performed during forward/reverse switching. A state in which the converter 2 and the opposite converter 3 are electrically connected at the same time is allowed.

Therefore, since current control is performed continuously, the response of the current control system can be improved to a limit determined by the power supply frequency and the number of rectification phases.

This has the advantage of having less distortion in the output current waveform and being able to operate up to a higher frequency. However, it is economically disadvantageous because the power transformer is expensive and a DC reactor is required.

FIG. 2 shows the configuration of a U-phase current control system using anti-parallel connected converters, and the same parts as in FIG. 1 are represented by the same symbols.

In this figure, the secondary winding of a transformer 6 is commonly connected to the positive converter 2 and the reverse converter 3.

As a result, one current transformer 7 and one current detection circuit 8 are provided in common.

With this configuration, since the power source is common to the positive converter 2 and the reverse converter 3, all the thyristors of the positive converter 2 are completely turned off when switching from the positive side to the reverse side, for example. If a gate pulse is not applied to the reverse converter 3 after that, a path will be created that directly shorts the power supply lines by the positive and reverse thyristors, and a large short-circuit current will flow.

For this reason, during forward/reverse switching, it is necessary to allow a period (dead time) in which all the thyristors of the forward-side converter and reverse-side converter are off and the current is zero.

However, during that period, the feedback loop of the current control system is cut off, so the current control circuit 11 swings in a direction that increases the current more than the current reference ST.

For this reason, after the switching is completed, a gate pulse with an ignition phase that is too advanced is applied to the sirisk constituting the converter, causing an overcurrent to flow.

In order to avoid such inconveniences, the current control circuit 11 has conventionally been configured as shown in FIG. 3.

In this figure, the current reference ST is applied to the analog switch 21 via a resistor 20, while the current feedback signal fed back via the current detection circuit 8 is applied to the analog switch 21 via a resistor 22.

A DC operational amplifier 23 is connected to the other end of this switch, and a filter capacitor 24 and a resistor 25 are connected in parallel to this.
and a delay compensation circuit consisting of a capacitor 26 connected in series.

In the above configuration, if the analog switch 21 is turned off during the dead time of forward/reverse switching, the input current to the DC operational amplifier 23 is cut off, and the charges in the capacitors 24 and 26 are held almost unchanged.

As a result, a gate pulse is applied to the cylindrical converter in the same phase as before switching, so no overcurrent will flow immediately after switching.

However, in reality, the bias current of the operational amplifier 23, the leakage current flowing through the analog switch 21, and the capacitor 24
In order to maintain the electric charge of the capacitor 24 well during the dead time of forward and reverse switching by discharging from the capacitor 25 to the resistor 25 and the capacitor 26, the capacitance of the capacitor 24 itself had to be increased.

As a result, the delay in current control increases, and as the output frequency increases, the phase delay and amplitude attenuation of the output current with respect to the current reference ST increase, and the limit of the output frequency is limited to a relatively low value.

As mentioned above, the cross wiring method has a good current control response, but has the disadvantage that the main circuit configuration is complicated and expensive, while the anti-parallel wiring method has a simple main circuit configuration, but the current control response is good. There was a drawback that it worsened.

The present invention aims to solve the above-mentioned problems with forward/reverse switching in the anti-parallel connection system and to provide a control device for a commutatorless motor that improves the responsiveness of current control.

Embodiments of the present invention will be described below with reference to the drawings.

Figure 4 shows only the U-phase current control system, but
The same is true for phases.

In this figure, one U-phase armature winding of a synchronous motor is connected to positive and reverse converters (hereinafter referred to as converters) 30 that are connected in antiparallel.

Here, it is assumed that the resistance value of one U-phase armature winding is Ru, the effective inductance is Lu, and the induced voltage is e. The converter 30 has the same configuration as 2.3 in FIG.

An AC power supply 31 is added to this converter 30, and a current transformer 7 for detecting the U-phase current i1 is installed in this AC power supply 31.

A current detection circuit 8 is connected to the current transformer 7 and outputs a voltage proportional to the output current of the current transformer 7.

This voltage is applied to a summer 10 consisting of a resistor 20.22.

On the other hand, a current reference value ST having an arbitrary waveform, for example a sinusoidal waveform, is applied to the adder 10 and the switching logic circuit 14.

The output of adder 10 is applied to current control circuit 11A.

This current control circuit 11A includes a DC operational amplifier 23, a filter capacitor 24, a resistor 25, and a capacitor 26.
The analog switch 21 is composed of a delay compensation circuit connected in series and a parallel circuit with the analog switch 21.

The output of the current control circuit 11A is added to the adder 32.

On the other hand, a back electromotive force compensation circuit 33 and an impedance drop compensation circuit 34 are newly provided, and their outputs are also added to the adder 32.

FIG. 5 shows a back electromotive force compensation circuit 33 in the embodiment shown in FIG.
3 shows the configuration of a portion including the impedance drop compensation circuit 34 in more detail.

The back electromotive force compensation circuit 33 is a sine wave having the same phase as the back electromotive force eu generated in the U-phase armature winding of the synchronous motor, and a signal Vu whose peak value is constant regardless of the rotational speed of the synchronous motor and the speedometer generator. 36 is multiplied by the multiplier 35 to obtain the signal e.
form u'.

The impedance drop compensation circuit 34 compensates for the delay caused by the effective inductance of the motor and the voltage drop caused by the resistance.
, the capacitor C and the amplifier 37, the resistor R2 connected between the output terminals, the anti-series Zener diode ZD
Consisting of

By appropriately selecting the resistors R1 and R2 and the capacitor C, the above formula (9), ie, e.

Control that satisfies -0 can be performed.

The output of the adder 32 is applied via the phase control circuit 12 to the switching circuit 13 for the positive side converter and the reverse side converter, and this switching circuit 13 is controlled by the switching logic circuit 14 to convert either one of the converters. It is designed to apply a gate pulse to the device.

In the above configuration, the U-phase armature current is transferred to the 1u converter 3.
If the output voltage of 0 is set as e5, then the following equation is obtained.

In addition, the input voltage of the phase control circuit 12 is determined by K.
Then, e5-KepH・・・・・・・・・・・・・・・・・・
・・・・・・・・・・・・・・・・・・(2)

Next, by eliminating e5 from equations (1) and (2), ePH
Solving for , it can be expressed as .

Here, the output of the back electromotive force compensation circuit 33 is e/u, and the output of the impedance drop compensation circuit 34 is e'B+e'□.
If the output of the current control circuit 11A is er, then the following equation is obtained.

When formula 4 is rearranged, it is expressed as.

e'Rj e'L j e so that this \ always holds true
If 'U is given to the adder 32, eo=0 from equations 5 to 8.

This e. - The relationship between 〇 is current reference ST and armature current ■.

This only holds true if there is an equilibrium between the two. The armature current (iu in the U phase) and induced voltage (eu in the V phase) of the motor are sinusoidal waves, and therefore the control device shown in FIG. 4 needs to produce an output that causes sinusoidal energization.

This control device has an output e of the current control circuit 11A.

and the signal eu′ for its compensation, (eR′+eL′
).

Of these, e.

is formed by the deviation between the current reference ST and the current feedback signal luF, and a delay circuit consisting of capacitors 24 and 26 and a resistor 25 is provided to always generate this deviation.

This is e.

Considering the case of controlling only by e. This is because the electric motor cannot be energized unless this always occurs, so it is necessary to create a deviation using this delay circuit.

However, this delay circuit causes a phase delay and amplitude attenuation of the armature current 1u as the output frequency increases.

A back electromotive force compensation circuit 33 and an impedance drop compensation circuit 34 are provided to compensate for this phase delay and amplitude attenuation, and compensate for the back electromotive force and impedance drop while the motor is being driven.

Due to this compensation, the current control circuit 11A does not need to produce an output e in a steady state.

So e. -O, 1 uF=ST, and control without steady-state errors is possible.

e, even as \0. If is small enough, i up ”i
ST, which allows for highly accurate current control.

Here, the setting conditions of the compensation circuit elements for ensuring that equations (6), (scheme), and equation (8) always hold will be explained in more detail using relational equations.

1UF in FIG. 5 is a value detected by the converter 7 and the current detection circuit 8 on the AC power supply side of the converter 30 for the U-phase current iU in FIG. 4. If the conversion coefficient is 2, then jUF=2jU・・・・・・・・・・・・・・・
・・・・・・・・・・・・・・・・・・・・・00) In Figure 5, if compensation is performed correctly so that ec becomes 0, iup will be the standard value ST. Since it converges to , 1UF=sT ・・・・・・・・・・・・・・・・・・
・・・・・・・・・・・・・・・・・・ aυ.

From α0) formula and 01 formula, ST-K21U ・・・・・・・・・・・・・・・・・・
・・・・・・・・・・・・・・・・・・・・・(12) The amplification rate of the impedance drop compensation circuit 34 is from FIG. In order to ensure that formulas 6) and (7) always hold, it is sufficient to substitute formulas (6) and (formula 09) into formula 09. In this way, the impedance drop compensation circuit 34 in FIG. The constants R1, R2, and C may be selected such that the steady-state gain is proportional to the armature winding resistance RU and the differential time constant CR2 is proportional to the armature effective inductance LU.

In addition, in order to ensure that equation (8) always holds true, if the voltage of the tachometer generator is v'ro in Figure 5, then VTG = R3e()... ...
・・・・・・・・・・・・・・・・・・・・・・・・ Therefore eu'-VUX VTG = VUK3 eU"""l
By substituting the equation lH+++l+++++++Hα9) into the equation (8), 3 can be selected as the voltage coefficient of the tachometer generator using the equation (20).

Therefore, current control is performed with the output voltage of the current control circuit 11A hardly swinging.

As a result, by turning on the analog switch 21 during the dead time of forward/reverse switching to lock the output voltage of the current control circuit 11A at zero, and then turning off the analog switch 21 after the switching is completed, the optimum phase is set. Current control will begin.

The current control signal input to the open current control circuit 11A when the analog switch 21 is on has no effect on the output.

Therefore, there is no overcurrent due to too much phase advance after switching is completed, and no current rise delay due to too much phase delay.

In addition, since the input to the operational amplifier 23 is not cut off during the dead time to maintain the state, the capacitance of the filter capacitor 24 and the capacitor 26 of the delay compensation circuit can be minimized, which speeds up the response of the current control system. be able to.

Further, since the current control circuit 11A can perform current control with almost zero swing, the apparent gain of current control becomes very high, and the responsiveness of the armature current can be significantly improved.

As shown above, according to the present invention, the responsiveness of current control can be improved even when forward/reverse converters connected in antiparallel are used.

[Brief explanation of the drawing]

1 and 2 are block diagrams showing a conventional control device for a non-commutated motor, FIG. 3 is a circuit diagram of a current control circuit used in the device shown in FIG. 2, and FIG. 4 is a non-commutated motor according to the present invention. FIG. 5 is a block diagram showing a control device for a sub-motor, and is a detailed circuit diagram including the back electromotive force compensation circuit 33 and impedance drop compensation circuit 34 of FIG. 4. 1A... U-phase armature winding, 7... Current transformer, 8... Current detection circuit, 10, 32...
...Adder, 11A...Current control circuit, 12.
... Phase control circuit, 13 ... Switching circuit,
14...Switching logic circuit, 20, 22. 25...Resistor, 21...Analog switch, 23...DC calculation amplifier, 24, 26
... Capacitor, 33 ... Back electromotive force compensation circuit, 34 ... Impedance drop compensation circuit.

Claims (1)

[Claims] 1. A cycloconverter-type non-converter in which the current of each phase of the synchronous motor is controlled by a positive side converter and a reverse side converter connected in antiparallel, respectively, according to the relative speed between the armature of the synchronous motor and the field. In a commutator motor, a device that outputs a signal proportional to the back electromotive force induced in the armature and a device that outputs a signal proportional to the impedance drop of the armature are provided. The firing phase of the converter is controlled by the sum of the signal proportional to the impedance drop, the signal proportional to the impedance drop, and the output signal of the current control circuit that controls the armature current. A control device for a commutatorless motor, characterized in that the output voltage of the current control circuit is locked to zero.