JPH10224105A - Bandpass filter device - Google Patents

Abstract

(57) [Problem] To provide a band-pass filter device having a large attenuation in a stop band as compared with a conventional example. An input / output separating means and a signal reflecting means are provided, and a signal reflected by the signal reflecting means among signals input from the first terminal of the input / output separating means is converted to a third signal.
Wherein the signal reflecting means comprises: a first resonance circuit including three transmission lines constituting first and second directional couplers; and a third and fourth bandpass filter device.
And a second resonance circuit including three transmission lines constituting the directional coupler of the first embodiment, and the second section of the input / output separation means at the connection between the first and second resonance circuits of the signal reflection means. Connected to terminal.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

[0001] The present invention relates to a band-pass filter device.

[0002]

2. Description of the Related Art FIG. 20 is a plan view showing the structure of a conventional band-pass filter device using a microstrip line. The conventional filter device of FIG. 20 includes microstrip line resonators MR101, MR102, MR
103 and an input microstrip line M100
And a microstrip line M200 for output, and a dielectric substrate 1 having a ground conductor formed on the entire lower surface.
10, strip conductors 100, 101, 102,
103 and 200 are formed and configured as follows.

A strip conductor 100 including a strip conductor 100a and a strip conductor 100b is formed on an upper surface of a dielectric substrate 110. As a result, a microstrip line M100 is formed. Next, the strip conductor 101 including the strip conductor 101a and the strip conductor 101b is formed such that the strip conductor 101a faces the strip conductor 100b at a predetermined interval. Thereby, microstrip line resonator MR101 is formed so as to be electromagnetically coupled to microstrip line M100.

A strip conductor 102 composed of a strip conductor 102a and a strip conductor 102b is formed so that the strip conductor 102a faces the strip conductor 101b at a predetermined interval. Thereby, the microstrip line resonator MR102
Are formed so as to be electromagnetically coupled to the microstrip line resonator MR101. Further, the strip conductor 103 including the strip conductor 103a and the strip conductor 103b is formed such that the strip conductor 103a faces the strip conductor 102b at a predetermined interval. Thereby, the microstrip line resonator MR103 is formed so as to be electromagnetically coupled to the microstrip line resonator MR102.

The strip conductor 200 composed of the strip conductor 200a and the strip conductor 200b is formed such that the strip conductor 200a faces the strip conductor 103b at a predetermined interval. Thus, microstrip line M200 is formed so as to be electromagnetically coupled to microstrip line resonator MR103. The conventional bandpass filter device configured as described above has the strip conductors 100b and 10b.
1a, 101b, 102a, 102b, 103a, 10
3b and 200a, the width w and the length L10 are set to predetermined values, and the microstrip line resonator M
R101 and microstrip line resonator MR102
And the microstrip line resonator MR10
By setting the distance between the microstrip line resonator 2 and the microstrip line resonator MR103 to predetermined values, it is possible to pass signals in a desired frequency range.

FIGS. 21 to 26 are graphs showing simulation results of the conventional band-pass filter device formed as described above. 21, 22, and 23 respectively show the fractional band Δf / f ₀ defined by the ratio of the frequency range Δf of the pass band to the center frequency f ₀ of the pass band as 0.1, 0.007, ₀ . 9 is a graph showing a simulation result when each parameter is set to be 0056. Here, in FIGS. 21, 22, and 23, the center frequency f ₀ of the pass band is set to be approximately 18 GHz. 24, 25, and 26 respectively show the fractional bandwidth Δf / f ₀ as 0.033, 0.02, 0.01.
9 is a graph showing a simulation result when each parameter is set so as to be as follows. Here, in FIGS. 24, 25, and 26, the center frequency f ₀ of the pass band is set to be approximately 3 GHz. This conventional bandpass filter device is designed, for example, by G. Matthaei, L. Youn.
g, EMTJone, “Microwave Filters, Impedance-Matchin
g Networks, And Coupling Structures ”, ARTECH HOUSE,
INC, pp 472-476, 1964.

[0007]

However, in the conventional band pass filter device, in order to reduce the loss in the pass band, the microstrip line resonator MR101 and the microstrip line resonator MR1 are required.
02, the microstrip line resonator MR102 and the microstrip line resonator MR103 are strongly coupled to each other, and the microstrip line M100
It is necessary to strongly couple the microstrip line resonator MR101 and the microstrip line M200 with the microstrip line resonator MR103.
As a result, there is a problem that the fractional band is widened and the amount of attenuation in the stop band cannot be increased. To increase the attenuation in the stop band, the microstrip line resonator MR101 and the microstrip line resonator MR102 and the microstrip line resonator MR102 and the microstrip line resonator M
It is necessary to weakly couple R103 to each other and weakly couple microstrip line M100 to microstrip line resonator MR101 and microstrip line M100 to microstrip line resonator MR101. As a result, there is a problem that the fractional band becomes narrow and the loss in the pass band increases.

An object of the present invention is to solve the above problems and to provide a bandpass filter device capable of increasing the amount of attenuation in a stop band as compared with the conventional example.

[0009]

SUMMARY OF THE INVENTION The present invention has been made to achieve the above-mentioned object, and the present invention relates to claim 1 of the present invention.
The band-pass filter device has first, second, and third terminals, outputs a signal input from the first terminal from the second terminal, and receives a signal input from the second terminal. Input / output separating means for outputting a signal from the third terminal, and signal reflecting means connected to the second terminal of the input / output separating means, wherein the first terminal of the input / output separating means is provided. A band-pass filter device that outputs a signal of a predetermined pass band by outputting, from the third terminal, a signal reflected by the signal reflection unit among the signals input from the signal reflection unit, A first and a second transmission line provided facing each other at a predetermined interval such that each end faces each other and each other end faces each other, and one end corresponds to the first and second transmission lines. Of the first transmission line so as to face each end of the transmission line. Provided between the second transmission line,
A third directional coupler distributedly coupled to a part of the first transmission line to form a first directional coupler and a third directional coupler distributedly coupled to a part of the second transmission line to form a second directional coupler; A first resonance circuit including a transmission line and a fourth transmission line connected to one end of the third transmission line and having a length of about の of the guide wavelength λg at the center frequency of the pass band; ,
Fifth, provided at a predetermined distance from each other, such that each end faces each other and each other end faces each other.
And a sixth transmission line, and the fifth transmission line is provided between the fifth and sixth transmission lines such that one end thereof is opposed to one end of each of the fifth and sixth transmission lines. A third transmission line which is distributedly coupled to a portion of the sixth transmission line, and a seventh transmission line which is distributedly coupled to a portion of the sixth transmission line to constitute a fourth directional coupler; Is connected to the other end of the fourth transmission line, and a second resonance circuit including a resistor whose other end is connected to one end of the seventh transmission line is provided. It is characterized in that it is connected to the second terminal of the input / output separating means at a connection portion with a resistor.

According to a second aspect of the present invention, in the band-pass filter device according to the first aspect of the present invention, the first resonance circuit further includes a first and second transmission lines. The other end of the third transmission line is provided so as to face the other end of the third transmission line, and a portion of the first transmission line other than the portion which is distributedly coupled to the third transmission line is the first transmission line. Distributed coupling with the transmission line, and
And an eighth transmission line that is distributedly coupled to the second transmission line at a portion other than the portion that is distributedly coupled to the third transmission line in the transmission line of the second resonance circuit. A portion provided between the fifth transmission line and the sixth transmission line such that one end thereof is opposed to the other end of the seventh transmission line, and the portion of the fifth transmission line which is distributedly coupled to the seventh transmission line; Is distributedly coupled to the fifth transmission line at a portion other than the sixth transmission line, and is distributedly coupled to the sixth transmission line at a portion other than the portion of the sixth transmission line that is distributedly coupled to the seventh transmission line. A ninth transmission line is provided.

Further, according to a third aspect of the present invention, in the band-pass filter device according to the second aspect, the band-pass filter device according to the second aspect further includes a portion between the other end of the eighth transmission line and the ground terminal, and 9 is characterized in that a negative resistance circuit is provided between the other end of the transmission line and the ground end.

According to a fourth aspect of the present invention, there is provided a band-pass filter device according to one of the first to third aspects, wherein the first and second transmission lines and A variable capacitor capable of changing the capacitance is provided between one end and the other end of the fifth and sixth transmission lines and the ground end.

According to a fifth aspect of the present invention, in the band-pass filter device according to any one of the first to fourth aspects, the input / output separating means is a circulator. It is characterized by.

According to a sixth aspect of the present invention, there is provided the band-pass filter according to any one of the first to fourth aspects, wherein the input / output separating means includes a 3 dB directional coupling. It is characterized by being a vessel.

[0015]

BEST MODE FOR CARRYING OUT THE INVENTION

<First Embodiment> FIG. 1 is a block diagram showing a configuration of a band-pass filter device F1 according to a first embodiment of the present invention. Bandpass filter device F1 of the first embodiment
Has terminals T6a, T6b, and T6c, and has terminals T6a
Output from the terminal T6b and the signal inputted from the terminal T6b.
A circulator 6 that outputs a signal input from the terminal T6c from a terminal T6c, and a reflection circuit 71 whose input / output terminal T71 is connected to a terminal T6b of the circulator 6 via a transmission line 2c, and a signal input from a terminal T6a. Among them, the signal of the predetermined band reflected by the reflection circuit 71 is output from the terminal T6c, thereby passing the signal of the predetermined band.

Hereinafter, the configuration of the bandpass filter device F1 according to the first embodiment will be described in detail with reference to the drawings. In the reflection circuit 71 of the filter device F1, the transmission line 1 and the transmission line 11 having both ends released and having the same length are provided so as to face each other at a predetermined interval. The transmission line 2 includes a transmission line 2a,
2b and 2c, a transmission line 2a having a predetermined length from one end of the transmission line 2 is connected to the transmission line 1 and the transmission line 1
1 and distributedly coupled to a portion (hereinafter referred to as a transmission line 1a) having a predetermined length from one end of the transmission line 1 and having a predetermined length from one end of the transmission line 11. A portion (hereinafter, referred to as a transmission line 11a) is distributedly coupled to each other. In this manner, the transmission lines 1a and 2 provided to face each other so as to be distributedly coupled to each other.
a constitutes the directional coupler 10, and the transmission lines 2a and 11a provided to face each other so as to be distributedly coupled to each other.
Thus, the directional coupler 10a is configured. Here, in FIG. 1, the end of the transmission line 2a denoted by reference numeral T11 is an input / output terminal of the directional coupler 10, 10a.

One end of the resistor 5 is connected to the input / output terminal T11 of the transmission line 2 at a position of a length L1, and one end of the transmission line 4 is connected to the other end of the resistor 5. Here, the position of the length L1 from the input / output terminal T11 of the transmission line 2 is the input / output terminal T71 of the reflection circuit 71.
The input / output terminal T6b of the circulator 6 is connected to 71 via the transmission line 2c. Here, the circulator 6 is an input / output separation element that outputs a signal input from the terminal T6a to the input / output terminal T71 via the terminal T6b and outputs a signal input from the terminal T6b via the terminal T6c. Also, the input / output terminal T11 of the transmission line 2
The part of the length L1 is the transmission line 2b. Further, the transmission line 3 includes transmission lines 3a and 3b.
The transmission line 3a, which is a portion having a predetermined length from one end of the transmission line 3, is provided so as to be distributed-coupled with the transmission line 4 at a predetermined interval. In this manner, the directional coupler 20 is configured by the transmission line 4 and the transmission line 3a that are distributedly coupled to each other. Further, the transmission line 12 includes transmission lines 12a, 1
2b, a transmission line 12a which is a predetermined length from one end of the transmission line 12 is provided so as to be distributed-coupled with the transmission line 4 at a predetermined interval. In this way, the directional coupler 20a is formed by the transmission line 4 and the transmission line 12a that are distributedly coupled to each other.

Here, the input / output terminal T21 of the directional couplers 20 and 20a is one end of the transmission line 4 and the other end of the resistor 5 is connected. The input / output terminal T22 of the directional coupler 20, which is one end of the transmission line 3, is open, and the input / output terminal T24 of the directional coupler 20, which is the other end of the transmission line 4, is open. The input / output terminal T23 of the directional coupler 20 is connected to an open transmission line 3b having a length L3. The input / output terminal T23a of the directional coupler 20a has a length L3.
The transmission line 12b having the open end is connected. Transmission line 1
2, the input / output terminal T2 of the directional coupler 20a.
2a is opened.

In the bandpass filter F1 of the first embodiment configured as described above, the two directional couplers 10 and 10a sharing the transmission line 2a and the directional coupler 1
0, the open transmission line 1b connected to the input / output terminal T13, the open transmission line 11b connected to the input / output terminal T13a of the directional coupler 10a, and the directional coupler 1b.
A first resonance circuit 810 is configured by the transmission line 2b connected to the common input / output terminal T11 of 0 and 10a. Also, two directional couplers 2 sharing the transmission line 4
0, 20a, an open-ended transmission line 3b connected to the input / output terminal T23 of the directional coupler 20, and a directional coupler 2
The second resonance circuit 820 is configured by the open transmission line 12b connected to the input / output terminal T23a of Oa and the resistor 5 connected to the common input / output terminal T21 of the directional couplers 20 and 20a. . Further, the first resonance circuit 81
One end of the zero transmission line 2b and one end of the resistor 5 of the second resonance circuit 820 are connected to form the reflection circuit 71. Here, a connection point between the resistor 5 and the transmission line 2b is an input / output terminal T71 of the reflection circuit 71.

In the band-pass filter F1 of the first embodiment configured as described above, the input high-frequency signal input through the terminal T6a of the circulator 6 is transmitted to the reflection circuit 71 through the terminal T6b of the circulator 6. The input high-frequency signal input and reflected by the reflection circuit 71 is input to the circulator 6 via the terminal T6b,
6c. That is, the reflection circuit 71
The parameters of the respective components are set such that the reflection coefficient increases in the pass band of the band-pass filter F1.

Hereinafter, the reflection circuit 7 in the first embodiment will be described.
The reflection characteristic of No. 1 will be described in detail. Here,
For the sake of simplicity, the operation of the present invention will be described based on the band-pass filter device shown in FIG. Here, the reflection circuit 71d shown in FIG.
1 using one directional coupler 10 and the directional coupler 1
The input / output terminal T13 of the directional coupler 10 is configured by grounding the input / output terminal T13 of the directional coupler 10 via the transmission line 1b of the length L2, and the second resonance circuit 82 uses one directional coupler 20. T
The configuration is the same as that of the reflection circuit 71 of the first embodiment, except that the reflection circuit 23 is grounded via a transmission line 3b having a length L3. First, an operation related to the directional coupler 10 in the reflection circuit 71 will be described with reference to FIG. FIG.
Resonant circuit is configured and terminates at the input and output terminal T13 of the directional coupler 10 the impedance Z _L of the load 7a in FIG. 13 8
1 is a circuit diagram of FIG. Here, the load 7a corresponds to the transmission line 1b in FIG. 13, and the input / output terminals T12 and T14 are
The directional coupler 10 is configured so as to have an open end similarly to the directional coupler 10. In the resonance circuit 81 of FIG.
Directional coupler 1 which is a coupled transmission line between the transmission line 2a and the transmission line 2a
Each element of the zero Z matrix is given by the following equations 1 to 4 using the even mode impedance Z _0e and the odd mode impedance Z _0o of the coupled transmission line. Here, each element of the Z matrix of the coupled transmission line is E.
MTJones and JTBolljahn, “Coupled-strip-transm
ission-line filters and directional couplers ”, IR
E Transactions on Microwave Theory and Technique
s, MTT-4 No4, pp75-81, determined using the method shown in April 1956. Also, it is assumed that there is no loss in the transmission lines 1a and 2a.

[0022]

## EQU1 ## Z ₁₁ = Z ₂₂ = Z ₃₃ = Z ₄₄ = −j (1/2) (Z _0e + Z _0o ) cotθ

## EQU2 ## Z ₁₂ = Z ₂₁ = Z ₃₄ = Z ₄₃ = −j (1/2) (Z _0e −Z _0o ) cotθ

## EQU3 ## Z ₁₃ = Z ₃₁ = Z ₂₄ = Z ₄₂ = −j (1/2) (Z _0e −Z
_0o ) cosecθ

## EQU4 ## Z ₁₄ = Z ₄₁ = Z ₂₃ = Z ₃₂ = -j (1/2) (Z _0e + Z
_0o ) cosecθ

Here, θ in Equations 1 to 4 is an electrical length given by the following Equation 5. Also, l in Equation 5
Is the physical length of the directional coupler 10, and k is the phase constant.

[0024]

## EQU5 ## θ = kl

Also, as shown in FIG.
12 and T14 are open ends, and the input / output terminal T13 is terminated by the load 7a, so that the input / output terminals T11 and T1
2, T13, respectively in the T14 of voltages V _1, V _2,
V ₃ and V ₄ are the currents I ₁ and I _{3 at} the input / output terminals T11 and T13 and the elements Z ₁₁ , Z ₃₃ , Z ₂₁ , Z ₄₃ ,
Using Z ₁₃ , Z ₃₁ , Z ₄₁ , and Z ₂₃ , the following equations 6 to 9 are used.
Can be represented by

[0026]

V ₁ = Z ₁₁ I ₁ + Z ₁₃ I ₃

V ₂ = Z ₂₁ I ₁ + Z ₂₃ I ₃

V ₃ = Z ₃₁ I ₁ + Z ₃₃ I ₃ = −Z _L I ₃

V ₄ = Z ₄₁ I ₁ + Z ₄₃ I ₃

From the above, the impedance Z ₁ when the directional coupler 10 is viewed from the input / output terminal T11 can be expressed by the following equation (10).

[0028]

Z ₁ = V ₁ / I ₁ = −j (Z _0e + Z _0o ) (cotθ) / 2 + {(1/4) (Z _0e −Z _0o ) ² co
sec ² θ} / {− j (1/2) (Z _0e + Z _0o ) cot θ + Z _L }

Here, in the band-pass filter device of FIG. 13, since the load 7a is constituted by the transmission line 1b having one end grounded, the impedance Z _{L of the} load 7a can be expressed by the following equation (11). . The impedance Z ₁ represented by the number 10 can be represented by the number 12. Here, Z ₀ is the characteristic impedance of the transmission line 1b, and L2 is the length of the transmission line 1b. β is transmission line 1
b is the phase constant.

[0030]

## EQU11 ## Z _L = jZ ₀ tan (βL2)

Z ₁ = −j {(Z _0e + Z _0o ) cot θ} / 2 + j {(1/4) (Z _0e −Z _0o ) ²
cosec ² θ} / {(1/2) (Z _0e + Z _0o ) cot θ + Z ₀ cot (βL2)}

Accordingly, the resonance circuit 81 has a resonance frequency f ₀ at which the denominator of the second term on the right-hand side of Expression 12 becomes ₀ , that is, Expression 1
Resonate at a resonance frequency f ₀ that satisfies the condition 3. In other words, the resonance frequency f ₀ of the resonance circuit 81 can be set to a predetermined value by setting the length L2 of the transmission line 1b to a predetermined value.

[0032]

(1/2) (Z _0e + Z ₀₀ ) cotθ = Z ₀ tan (βL2)

[0033] FIG. 4 is a Smith chart showing the impedance Z _1. Here, the impedance Z ₁ sets the length L2 to a predetermined value, and was determined by calculation by sweeping the frequency from 17GHz to 17.88GHz. As a result of the calculation, the resonance frequency f ₀ was 17.4 GHz. In Equation 12, the impedance Z ₁ is, transmission lines 1a, but 2a is assumed to be lossless, in FIG. 4 was calculated as the transmission lines 1a, 2a has a loss.
As a result, the impedance Z ₁ is, as shown in FIG. 4,
A locus is drawn inside the circle of R = 0 on the Smith chart. When one end of the transmission line 2b having a length L1 set to １ ／ times the guide wavelength λg at the resonance frequency f ₀ is connected to the input / output terminal T11 of the directional coupler 10, the other ends of the transmission line 2b are connected. impedance Z _1a of the resonant circuit 81 when viewed from the end, as shown in FIG. 6, a locus that the locus of the impedance Z ₁ of Figure 4 is rotated 180 degrees on the Smith chart. That is, by connecting the transmission line 2b in the input-output terminal T11 of the directional coupler 10, the impedance Z _1a of the resonant circuit 81 when viewed from the other end of the transmission line 2b, becomes minimum at the resonance frequency f ₀ In this way, it is set to increase at frequencies f ₁ and f ₂ apart from the resonance frequency f ₀ .

Next, the resonance circuit 82 shown in FIG. 3 will be described. Resonant circuit 82 includes a resonant circuit 82a consisting of a load 7b of the directional coupler 20 and the impedance Z _L,
Resistance 5 having a resistance equal to characteristic impedance Z ₀
It is constituted by and. Here, the resonance circuit 82a is configured similarly to the resonance circuit 81, and the resistor 5 is connected to the directional coupler 2
0 is connected to the input / output terminal T21. The resistance value of the resistor 5, that is, the value of the characteristic impedance Z ₀ is set to 50 ohms. The load 7b corresponds to the transmission line 3b in FIG. In the resonance circuit 82 configured as described above, as described above, the impedance of the load
Since set equal to a impedance Z _L, the resonant circuit 82 resonates at the resonance frequency f _0, the resistance 5 and 50
Since the set ohms, the impedance Z ₂ looking into the directional coupler 20 from the terminal T5 of the resistor 5, the frequency away from the resonance frequency f _0, becomes approximately 50 ohms.

[0035] FIG. 5 is a Smith chart showing the impedance Z _2. Here, the impedance Z _2, like the impedance Z _1, set the length L3 to a predetermined value, and was determined by calculation by sweeping the frequency from 17GHz to 17.88GHz. As shown in FIG. 5, the frequency f ₁ = 17.00 GHz and the frequency f ₂ = 17.88 G
Each of the impedances Z ₂ in Hz is located substantially at the center of the Smith chart and has a frequency of 17.00 GHz.
From the trajectory of the impedance Z ₂ at the time of the sweep until 17.88GHz draws a circle. Figure 8 is a graph showing frequency characteristics of the phase φ of the input end reflection coefficient S ₁₁ and the reflection signal at terminal T5 of the resistor 5.

Next, a description will be given impedance Z ₇₁ when viewed reflected circuit 71d from the terminal T71d. Here, the reflection circuit 71d is for the other end of the transmission line 2b in the resonant circuit 81 and the terminal T5 of the resistor 5 of the resonant circuit 82 is configured by connecting the impedance Z ₇₁ is above the impedance Z ₂ It can be described as follows using the impedance _Z1a and First, the resonance frequency f
In the vicinity of ₀ = 17.40GHz, as apparent from FIGS. 5 and 6, the resistance component and reactance component of the impedance Z ₂ of the resonant circuit 82, is very high compared to the impedance Z _1a. Further, since the reflection circuit 71d can be regarded as a circuit in which the resonance circuit 81 and the resonance circuit 82 are connected in parallel between the terminal T71d and the ground terminal, the impedance Z of the reflection circuit 71d near the resonance frequency f ₀ is considered. ₇₁ is impedance Z because Z ₂ ≫Z _1a
It is almost equal to _1a . On the other hand, at the frequencies f ₁ and f ₂ apart from the resonance frequency f ₀ , the resistance component and the reactance component of the impedance Z _1a of the resonance circuit 81 are:
It becomes very high compared to the impedance Z _2. Therefore, the frequency f ₁ and the frequency f ₂ distant from the resonance frequency f ₀
Impedance Z ₇₁ in the approximately equal to the impedance Z ₂ of the resonant circuit 82 as viewed from the terminal T5,
That is, it is approximately 50 ohms. In this case, the terminal T71
The impedance Z ₇₁ of the reflection circuit 71d when viewed from the point d.
Draws a locus on the Smith chart as shown in FIG. Impedance Z ₇₁ of the reflection circuit 71d obtained by synthesizing the impedance Z ₂ shown in the impedance Z _1a and 5 shown in FIG. 6, in the vicinity of the resonance frequency f _0, a value substantially equal to the impedance Z _1a, i.e., on the Smith chart, It takes a value distant from the center, and at a frequency distant from the resonance frequency f ₀ , takes a value of approximately 50 ohms, that is, a value located approximately at the center on the Smith chart. Thus, the reflection circuit 71d, as shown in FIG. 1, since the end of the transmission line 2b in the resonant circuit 81 and the terminal T5 of the resistor 5 of the resonant circuit 82 is configured by connecting, in the vicinity of the resonance frequency f ₀ , possible to increase the input end reflection coefficient S ₁₁ when viewed reflected circuit 71d from the terminal T71d, in frequency away from the resonance frequency f _0, the reflection circuit 71d from the terminal T71d
An input end reflection coefficient S ₁₁ when viewed can be reduced.

The inventor has set the length L1 of the transmission line 2b to the resonance frequency f ₀ in order to confirm the operation of the reflection circuit 71d.
Set to be substantially 1/4 of the guide wavelength λg in, was determined by simulating the frequency characteristic of the phase φ of the input end reflection coefficient S ₁₁ and the reflection signal at terminal T71d reflective circuit 71d. FIG. 9 shows the result. As is clear from FIG. 9, the resonance frequency f ₀ and the frequency band in the vicinity thereof are 17.4 GHz to 17.52 GHz.
The input end reflection coefficient S ₁₁ in the frequency band between
Larger than the input end reflection coefficient S ₁₁ at other frequencies, and is flat. In FIG. 9, the length L1 of the transmission line 2b is 1650 μm, 1700 μm,
The respective simulation values when set to 40 μm are shown. Therefore, the length L1 of the transmission line 2b is set to λ
g / 4 be only ± 3% is varied around the resonant frequency f ₀ and the frequency band width which the input end reflection coefficient S ₁₁ in the vicinity thereof becomes flat, i.e. band-pass filter apparatus F
It can be seen that the width of the passband at 1 hardly changes.

Next, the length L1 of the transmission line 2b is set to λg / 4
As an example of the case where the value is set to a value greatly shifted from the above, a case where the length L1 of the transmission line 2b is set to 0 will be considered. In this case, the impedance Z of the reflection circuit 71d
_71, the resonant circuit 8 when viewed from the impedance Z ₁ and the terminal T5 when viewed directional coupler 10 from the terminal T11
2 and the impedance Z ₂ . That is, at the frequency f ₁ and the frequency f ₂ apart from the resonance frequency f ₀ , the impedance Z ₁ has a sufficiently small value compared to the impedance Z ₂ ,
Impedance Z ₇₁ will be approximately equal to impedance Z ₁ , well below 50 ohms. Therefore, when the length L1 of the transmission line 2b is set to 0, the resonance frequency f ₀
Even at frequencies f ₁ and f ₂ distant from, the impedance Z ₇₁ becomes substantially equal to the impedance Z ₁ sufficiently smaller than 50 ohms, so that the reflection coefficient S ₁₁ at the input end of the reflection circuit 71d as viewed from the terminal T71d is growing.

In the above description, the same resonance
Frequency f₀A resonance circuit 81 configured to resonate with
Although the description has been given using the resonance circuit 82, the present invention is not limited to this.
The resonance frequency f of the resonance circuit 81₀And the resonance circuit 82
Resonance frequency f₈₂May be set differently from each other
No. That is, in the present invention, the resonance circuit 81
Wave number f₀And the resonance frequency f of the resonance circuit 82₈₂Is the band pass
In a predetermined frequency range including the pass band of the over-filter device,
Compared to frequencies outside the frequency range, the terminal T71d
Input end reflection coefficient S of the reflection circuit 71d as viewed from₁₁But
It is set to be large and flat. Where bandwidth
In the pass filter device, the resonance frequency of the resonance circuit 81
Number f₀And the resonance frequency f of the resonance circuit 82₈₂To the same value
By setting, the pass band of the band pass filter device
Can be set narrow, while the resonance of the resonance circuit 81
Frequency f₀And the resonance frequency f of the resonance circuit 82₈₂And each other
By setting different values,
A wide pass band can be set. Also, the resonance of the resonance circuit 81
Frequency f₀And the resonance frequency f of the resonance circuit 82₈₂Different from
If set to a value, the difference | f₀−f₈₂│
The pass band of the band-pass filter device F1 becomes wider.
Wear. Here, in the present invention, the resonance circuit 81
Wave number f₀And the resonance frequency f of the resonance circuit 82₈₂And is preferred
Alternatively, they are set to be different and close to each other. this
The input terminal of the reflection circuit 71d in the pass band.
Firing coefficient S₁₁To the resonance frequency f₀And the resonance frequency f₈₂Difference with
│f₀−f ₈₂Larger than when │ is increased
And more flattened, so that
The loss in the passband can be reduced and made flatter
You.

As described above, in the reflection circuit 71d, the length L2 of the transmission line 1b is set so that the resonance circuit 81 including the directional coupler 10 and the transmission line 1b resonates at a predetermined resonance frequency f _0. , The length L3 of the transmission line 3b is
Resonant circuit 8 composed of directional coupler 20 and transmission line 3b
2a is set to resonate at substantially the same resonance frequency f ₈₂ and the resonance frequency f _0. The length L1 of the transmission line 2b, the input end reflection coefficient S ₁₁ when viewed reflected circuit 71d from the terminal T71d is, to take a maximum value at a frequency near the resonance frequency f _0, the resonance frequency f ₀ Is set to a length that is approximately １ ／ times the guide wavelength λg at Thus, the input end reflection coefficient S ₁₁ of the reflection circuit 71d when viewed from the terminal T71d, greater than other frequency in the frequency band near the resonance frequency f ₀ of the resonant circuit 81, and can be flat.

Therefore, as shown in FIG. 1, a band-pass filter device can be constructed by connecting the terminal T6b of the circulator 6 to the terminal T71d via the transmission line 2c for leading out. That is, the high-frequency signal input to the terminal T6a of the circulator 6 is input to the reflection circuit 71d via the terminal T6b and the transmission line 2c. The reflection circuit 71d reflects a high-frequency signal in a predetermined frequency band of the input high-frequency signal, that is, a frequency band including the resonance frequency f ₀ and a frequency band near the resonance frequency f _0, and transmits the signal via the transmission line 2c and the terminal T6b. The high-frequency signal output to the circulator 6 and input to the terminal T6b is output from the terminal T6c of the circulator 6 via the circulator 6.

The above description relates to the case where the first resonance circuit 81 is configured using one directional coupler 10 and the second resonance circuit 82 is configured using one directional coupler 20. As described above, the same operation is performed when the first resonance circuit 810 and the second resonance circuit 820 each include two directional couplers in parallel as shown in FIG. In this case, the first resonance circuit 810 and the second resonance circuit 820 each have characteristics in which the resonance characteristics of two directional couplers are superimposed, and compare the no-load Q with the no-load Q of each resonance circuit. Can be higher. As a result, the reflection circuit 71 of the first embodiment including the first resonance circuit 810 and the second resonance circuit 820 has the resonance frequency of the first resonance circuit 810 and the second resonance circuit 820. In the vicinity, it has a very large reflection coefficient, and as it moves away from the resonance frequency,
The reflection coefficient suddenly decreases. Therefore, by connecting the circulator 6 to the input / output terminal T71 of the reflection circuit 71 having such reflection characteristics, a band-pass filter having excellent out-of-band attenuation characteristics can be configured.

[0043] Figure 10 is a graph showing a frequency characteristic of the input reflection coefficient S ₁₁ of the reflection circuit 71 in the input-output terminal T71, Fig. 11, the reflection in the band-pass filter device of the comparative example shown in FIG. 13 circuit 71d Input end reflection coefficient S
₁₁ is a graph showing frequency characteristics of No. ₁₁ . Comparing the graph of FIG. 10 with the graph of FIG. 11, the reflection circuit 71 of the first embodiment has a larger size than the reflection circuit 71d of the comparative example.
It can be seen that the reflection coefficient outside the pass band can be reduced.

FIG. 12 is a graph showing the frequency characteristic (pass characteristic) of the attenuation when the degree of coupling of the directional couplers 10 and 20 is changed in the bandpass filter of the comparative example of FIG. . Here, in FIG.
Transmission lines 1a, 2a constituting directional couplers 10, 20
And the distance S between the transmission lines 4 and 3a. As is clear from the graph of FIG. 12, the passing characteristics change when the degree of coupling is changed, but it is not possible to realize the passing characteristics higher than the first embodiment. That is, in the band-pass filter according to the first embodiment, since the first resonance circuit and the second resonance circuit are each configured by two resonance circuits, the attenuation outside the pass band can be increased.

<Second Embodiment> FIG. 14 is a block diagram showing a configuration of a bandpass filter device F2 according to a second embodiment of the present invention. The band-pass filter device F2 of the second embodiment includes two reflection circuits 71a, 71b and 90
And a reflection circuit 71a.
The reflection circuit 71b is connected to the input / output terminal T402 of the 90-degree hybrid circuit 40.
0 is connected to the input / output terminal T403. Here, the reflection circuits 71a and 71b have the same configuration as the reflection circuit 71 of the first embodiment. The 90-degree hybrid circuit 40 is an input / output separation element that operates as a 3 dB directional coupler in a predetermined frequency range. Further, the two reflection circuits 71a and 71b are configured to reflect signals having the same frequency.

In the band-pass filter device F2 of the second embodiment configured as described above, the input signal Sg1 input to the input / output terminal T401 of the 90-degree hybrid circuit 40 is distributed to the signal Sg1a and the signal Sg1b. The signal Sg1a is output to the reflection circuit 71a via the input / output terminal T402, and the signal Sg1b is output to the input / output terminal T40.
3 to the reflection circuit 71b. Here, the phase of the signal Sg1a is different from that of the signal Sg1b by π / 2. Therefore, the signal Sg1a and the signal Sg1b are expressed by the following Expressions 14 and 15, respectively.

[0047]

Sg1a = √ (1/2) · Sg1 · e ^{−i (π / 2)}

Sg1b = √ (1/2) · Sg1 · e− ^iπ

The signal Sg1a input to the reflection circuit 71a
Of the signal Sg2 is reflected by the reflection circuit 71a and output from the reflection circuit 71a, and the terminal T40
2 to the 90-degree hybrid circuit 40.
Since the signal Sg2 is reflected by the reflection circuit 71a, it has an amplitude obtained by multiplying the amplitude of the signal Sg1a by a coefficient r, and has a phase shifted by φ compared to the signal Sg1a. The coefficient r is a real number which is determined by the input end reflection coefficient S ₁₁ shown in FIG. 7, the phase φ 7
Are the values shown in the graph. Therefore, the signal Sg2 is expressed by the following equation (16). The second expression on the right side in Expression 16 is obtained by adding Sg1a in the first expression on the right side in Expression 16 to Expression 14
Is modified by substituting the expression on the right side of.

[0049]

Sg2 = r · Sg1a · e− ^iφ = r · ^{( 1/2) Sg1 · e− ^{i {(π / 2) + φ}}

The signal Sg2 input to the 90-degree hybrid circuit 40 via the terminal T402 is divided into a signal Sg2a and a signal Sg2b, and the signals Sg2a and Sg2
b is an input / output terminal T401 and an input / output terminal T40, respectively.
4 and are output. Here, the signal Sg2a is equal to the signal Sg.
The phase is different by π / 2 from g2b. Therefore, the signal Sg2a and the signal Sg2b are expressed by the following equations (17) and (1), respectively.
It is represented by 8. Note that the second expression on the right side in Expression 17 is a modification of the first expression on the right side in Expression 17 substituted for Sg2, and the second expression on the right side in Expression 16 is modified. Is Sg2 of the first expression on the right side in Expression 18.
Is modified by substituting the second expression on the right side of Expression 16 into

[0051]

Sg2a = √ (1/2) · Sg2 · e− ^{i (π / 2)} = r · (1/2) · Sg1 · e− ^{i (π + φ)}

Sg2b = {(1/2) · Sg2 · e− ^iπ = r · (1/2) · Sg1 · e− ^{i {(3π / 2) + φ}}

The signal Sg1b input to the reflection circuit 71b
Is reflected by the reflection circuit 71b and output from the reflection circuit 71b, and the signal Sg3
Is input to the 90-degree hybrid circuit 40 via the terminal T403. Since the signal Sg3 is reflected by the reflection circuit 71b, it has an amplitude obtained by multiplying the amplitude of the signal Sg1b by a coefficient r, and has a phase shifted by φ compared to the signal Sg1b. The coefficient r is a real number which is determined by the input end reflection coefficient S ₁₁ shown in FIG. 7, the phase φ is a value shown in the graph of FIG. Therefore, the signal S
g3 is expressed by the following equation 19 using the signal Sg1b.
The second expression on the right side of Expression 19 is a modification of the expression on the right side of Expression 15 substituted for Sg1b of the first expression on the right side of Expression 19.

[0053]

Sg3 = r · Sg1b · e− ^iφ = r · √ (1/2) · Sg1 · e− ^{i (π + φ)}

The signal Sg3 input to the 90-degree hybrid circuit 40 via the terminal T403 is divided into a signal Sg3a and a signal Sg3b, and the signal Sg3a and the signal Sg3
b is an input / output terminal T404 and an input / output terminal T40, respectively.
1 and are output. Here, the signal Sg3a is the signal Sg3.
The phase is different from g3b by π / 2. Therefore, the signal Sg3a and the signal Sg3b are expressed by the following equations (20) and (2), respectively.
It is represented by 1. Note that the second expression on the right side in Expression 20 is modified by substituting the second expression on the right expression in Expression 19 into Sg3 of the first expression on the right expression in Expression 20, and the second expression on the right expression in Expression 21 Is Sg3 of the first expression on the right side in Equation 21.
Into which the second expression on the right side of Expression 19 is substituted.

[0055]

Sg3a = {(1/2) · Sg3 · e− ^{i (π / 2)} = r · (1/2) · Sg1 · e− ^{i {(3π / 2) + φ}}

Sg3b = {(1/2) · Sg3 · e− ^iπ = r · (1/2) · Sg1 · e− ^{i {2π + φ}}

Here, as is apparent from Equations 17 and 21, the signal Sg2a output from the input / output terminal T401
And the signal Sg3b has a phase difference of π. That is,
Since the signals Sg2a and Sg3b are synthesized in opposite phases, the sum of the signals output from the input / output terminal T401 becomes 0,
No signal is output from the input / output terminal T401. On the other hand, as is apparent from Equations 18 and 20, the signals Sg2b and Sg3a output from the input / output terminal T404 are:
They have the same phase. Therefore, the signal Sg2b and the signal S
g3a is combined in phase, and the combined signal is output from input / output terminal T404 as output signal Sg4. As described above, the signal having a predetermined frequency among the signals input from the input / output terminal T401 of the 90-degree hybrid circuit 40 is output from the input / output terminal T404.

<Modification> In the present invention, the above-described first and second
Various modifications can be made to the embodiment. Less than,
A modification according to the present invention will be described with reference to the drawings.
FIG. 15 is a block diagram illustrating a configuration of a bandpass filter device F3 according to a first modification of the present invention. The band-pass filter device F3 according to the first modified example is different from the band-pass filter device F1 according to the first embodiment in that the transmission lines 1b and 11b are provided between the transmission lines 1b and 11b.
b is provided with a transmission line 31 distributed-coupled to the transmission line 3b and the transmission line 12b between the transmission line 3b and the transmission line 12b.
b, except that a transmission line 32 distributed-coupled to b is provided.
The configuration is the same as that of the first embodiment.

That is, in the first modification, the transmission line 1
b and the transmission line 31 form a directional coupler,
The transmission line 11b and the transmission line 31 form a directional coupler, the transmission line 3b and the transmission line 32 form a directional coupler, and the transmission line 12b and the transmission line 32
And a directional coupler are formed. Figure 16 is a graph showing a frequency characteristic of the input reflection coefficient S ₁₁ of the input-output terminal T71. Here, Lx is the length of the transmission line 2b. As described above, the input / output terminals T13, T13
Even if the loads connected to a, T23 and T23a are configured using directional couplers, they operate in the same manner as in the first embodiment and have the same effects.

FIG. 17 is a block diagram showing the configuration of a bandpass filter F4 according to a second modification of the present invention. The band-pass filter F4 of the second modified example is different from the band-pass filter F1 of the first embodiment in the following points, and is otherwise configured in the same manner. (1) The input / output terminal T13 of the directional coupler 10, the input / output terminal T13a of the directional coupler 10a, the input / output terminal T23 of the directional coupler 20 and the input / output terminal T23a of the directional coupler 20a are directly grounded. . (2) The input / output terminal T14 is grounded via the transmission line 33. (3) The input / output terminal T24 is grounded via the transmission line 34. Even with the configuration described above, the same operation and the same effect as in the first embodiment are obtained. In the modification of FIG. 17, as shown in FIG. 27, a transmission line 1ax distributed-coupled to the transmission line 1a and a transmission line 1ax distributed-coupled to the transmission line 11a.
1ax and a transmission line 3ax distributed-coupled to the transmission line 3a
And a transmission line 12ax that is distributedly coupled to the transmission line 12a. With this configuration, the out-of-band attenuation characteristics can be further improved.

FIG. 18 is a block diagram showing a configuration of a bandpass filter F5 according to a third modification of the present invention. The band-pass filter F5 of the third modification is different from the band-pass filter F3 of the first modification in that the transmission line 3
Negative resistance circuit 50 connected between one end of ground 1 and a ground end
a, and a negative resistance circuit 5 connected to one end of the transmission line 32.
0b, and the other points are configured similarly to the first modification. Here, the negative resistance circuit 50a includes a transistor 51a having a base terminal connected to one end of the transmission line 31, and a transmission line 52a connected between the emitter terminal and the ground terminal of the transistor 51a. The negative resistance circuit 50b includes a transistor 51b having a base terminal connected to the other end of the transmission line 32, and a transmission line 52b connected between the emitter terminal and the ground terminal of the transistor 51b.
Consists of The characteristic impedance and the electrical length of the transmission lines 52a and 52b are respectively set to the negative resistance circuit 5
0a and 50b are set to have negative resistance at a predetermined frequency. Here, the frequency range in which the negative resistance circuits 50a and 50b have negative resistance is set so as to include at least the pass band of the band-pass filter device F5. The band-pass filter F of the third modification configured as described above
5 has negative resistance circuits 50a and 50b, and can compensate for losses in the reflection circuit such as radiation loss at the open ends of the directional couplers 10, 10a, 20, and 20a and transmission loss in each transmission line. The transmission loss in the pass band can be reduced as compared with the first and second embodiments and other modifications. In addition, similarly to the present modification, the second
In the modified example, a negative resistance circuit 50a is provided at the tip of the transmission line 33, and the negative resistance circuit 50a is provided at the tip of the transmission line 34.
b may be provided. Even with the above configuration, the transmission loss in the pass band can be reduced as compared with the second modification of FIG.

FIG. 19 is a block diagram showing a configuration of a band-pass filter F6 according to a fourth modification of the present invention shown in FIG. The band-pass filter F6 according to the fourth modification is similar to the filter shown in FIG.
7, the input / output terminal T12 of the directional coupler 10, the input / output terminal T12a of the directional coupler 10a, the input / output terminal T22 of the directional coupler 20, and the directivity. Input / output terminal T of coupler 20a
The configuration is the same as that of the second modified example except that each of the capacitors 22a is grounded via a variable capacitor C1, a variable capacitor C2, a variable capacitor C3, and a variable capacitor C4. The band-pass filter F6 according to the fourth modification configured as described above has a variable capacitance capacitor C
1, variable capacitor C2, variable capacitor C3
By changing the capacitance value of the variable capacitor C4, the center frequency of the pass band can be changed.

<Other Modifications> In the above first and second embodiments and the first to fourth modifications, the circulator 6 or the 90-degree hybrid circuit 4 is used as the input / output separating means.
0 is used, but the present invention is not limited to this.
Other input / output separation means such as a wavelength distribution coupling type 3 dB directional coupler and a Lange coupler may be used. Even with the above configuration, the same operations as those of the first and second embodiments and the first to fourth modified examples are performed, and the same effects are obtained.

In the first and second embodiments described above, the respective ends of the transmission lines 1b and 11b and the respective ends of the transmission lines 3b and 12b are opened, but the present invention is not limited to this. Alternatively, one end of the transmission line 1b and one end of the transmission line 3b may be grounded. Even with such a configuration, it operates in the same manner as the first and second embodiments and has the same effect.

[0064]

As is apparent from the above description, the band-pass filter device of the present invention includes the input / output separating means and the signal reflecting means, and is connected to the first terminal of the input / output separating means. A band-pass filter device that outputs a signal of a predetermined pass band by outputting, from the third terminal, a signal reflected by the signal reflection unit among the input signals, wherein the signal reflection unit is A first resonant circuit including a first directional coupler and a second directional coupler, a second resonant circuit including a third directional coupler and a fourth directional coupler, At the connection between the fourth transmission line and the resistor,
, The amount of attenuation in the stop band can be increased as compared with the conventional example.

Also, the band pass filter device of the present invention
Between the other end of the eighth transmission line and the ground end,
By providing a negative resistance circuit between the other end of the transmission line and the ground end, the loss in the pass band can be extremely reduced.

Further, the band-pass filter device according to the present invention may be arranged such that one of the first and second transmission lines and one and the other ends of the fifth and sixth transmission lines are connected to a ground terminal. By providing a variable capacitance capacitor whose capacitance can be changed between the first and second components, the frequency of the pass band can be changed.

[Brief description of the drawings]

FIG. 1 is a block diagram of a bandpass filter device F1 according to a first embodiment of the present invention.

FIG. 2 is a band-pass filter device F1 according to the first embodiment;
FIG. 7 is a block diagram of a resonance circuit 81 for explaining the operation of FIG.

FIG. 3 is a band-pass filter device F1 according to the first embodiment;
FIG. 7 is a block diagram of a resonance circuit 82 for explaining the operation of FIG.

FIG. 4 shows an input / output terminal T in the resonance circuit 81 of FIG.
₆ is a Smith chart showing the impedance Z ₁ when viewing the directional coupler 10 from 11.

FIG. 5 is a Smith chart showing impedance Z ₂ when the directional coupler 20 is viewed from a terminal T5 in the resonance circuit 82 of FIG. 3;

6 is a Smith chart showing the impedance Z _1a of the resonant circuit 81 when viewed from the other end of the transmission line 2b.

7 is a Smith chart showing the impedance Z ₇₁ of the reflection circuit 71 as viewed from the terminal T71.

8 is a graph showing a frequency characteristic of the phase φ of the input reflection coefficient S ₁₁ and the reflection signal at terminal T5 of the resonant circuit 82 of FIG. 3.

9 is a directional coupler 1 at a terminal T71 in FIG.
Is a graph showing the frequency characteristics of the phase φ of the input reflection coefficient S ₁₁ and the reflection signal when viewed 0,20 side.

FIG. 10 shows an input / output terminal T71 of the reflection circuit 71 of FIG.
Is a graph showing a frequency characteristic of the input reflection coefficient S ₁₁ at.

11 is a graph showing a frequency characteristic of the input reflection coefficient S ₁₁ of the input and output terminals T71d reflective circuit 71d of the band-pass filter of the comparative example of FIG. 13.

FIG. 12 is a graph showing pass characteristics when the degree of coupling of the directional couplers 10 and 20 is changed in the bandpass filter of the comparative example of FIG.

FIG. 13 is a block diagram illustrating a configuration of a bandpass filter of a comparative example.

FIG. 14 is a block diagram of a bandpass filter device F2 according to a second embodiment of the present invention.

FIG. 15 is a block diagram illustrating a configuration of a bandpass filter device according to a first modification of the present invention.

16 is a graph showing a frequency characteristic of the input reflection coefficient S ₁₁ of the reflection circuit of Figure 15.

FIG. 17 is a block diagram illustrating a configuration of a bandpass filter device according to a second modification of the present invention.

FIG. 18 is a block diagram illustrating a configuration of a bandpass filter device according to a third modification of the present invention.

FIG. 19 is a block diagram illustrating a configuration of a bandpass filter device according to a fourth modification of the present invention.

FIG. 20 is a plan view of a conventional band-pass filter device.

FIG. 21 shows a conventional band-pass filter device.
Center frequency f ₀ is 18 GHz, fractional band Δf / f ₀ is 0.1
Is a graph showing a frequency characteristic of an output end transmission coefficient S ₂₁ and input end reflection coefficient S ₁₁ when set to.

FIG. 22 shows a conventional bandpass filter device.
Center frequency f ₀ is 18 GHz, fractional band Δf / f ₀ is 0.0
When set to 07 is a graph showing a frequency characteristic of an output end transmission coefficient S ₂₁ and input end reflection coefficient S ₁₁ of the.

FIG. 23 shows a conventional band-pass filter device;
Center frequency f ₀ is 18 GHz, fractional band Δf / f ₀ is 0.0
When set to 056 is a graph showing a frequency characteristic of an output end transmission coefficient S ₂₁ and input end reflection coefficient S ₁₁ of the.

FIG. 24 shows a conventional band-pass filter device.
Center frequency f ₀ is 3 GHz, fractional band Δf / f ₀ is 0.03
When 3 is a graph showing a frequency characteristic of an output end transmission coefficient S ₂₁ and input end reflection coefficient S ₁₁ of the.

FIG. 25 shows a conventional band-pass filter device;
Center frequency f ₀ is 3 GHz, fractional band Δf / f ₀ is 0.02
Is a graph showing a frequency characteristic of an output end transmission coefficient S ₂₁ and input end reflection coefficient S ₁₁ when set to.

FIG. 26 shows a conventional band-pass filter device.
Center frequency f ₀ is 3 GHz, fractional band Δf / f ₀ is 0.01
Is a graph showing a frequency characteristic of an output end transmission coefficient S ₂₁ and input end reflection coefficient S ₁₁ when set to.

FIG. 27 is a block diagram showing an example in which the second modification of FIG. 17 is further modified.

FIG. 28 is a block diagram showing a modification of the second modification of FIG. 17 which is different from that of FIG. 27;

[Explanation of symbols]

1, 1a, 1b, 1d, 1ad, 2, 2a, 2b, 2
c, 3, 3a, 3b, 3d, 3ad, 4, 11, 11
a, 11b, 12, 12a, 12b, 31, 32: transmission line, F1, F2, F3: band-pass filter device, 5: resistor, 7a, 7b: load, 10, 10a, 20, 20a: directional coupler , T11, T12, T12a, T13, T13a, T1
4, T21, T22, T23, T22a, T23a, T
24, T401, T402, T403, T404: input / output terminal, 6: circulator, T6a, T6b, T6c: terminal, 40: 90-degree hybrid circuit, 50a, 50b: negative resistance circuit, 51a, 51b: transistor, 71, 71a, 71b, 71d,..., A reflection circuit, 81, 82.

Claims (6)

[Claims] A first terminal having a first terminal, a second terminal, and a third terminal, a signal input from the first terminal being output from the second terminal, and a signal input from the second terminal being output from the second terminal. Input / output separating means for outputting from the third terminal, and signal reflecting means connected to the second terminal of the input / output separating means;
A band which outputs a signal of a predetermined pass band by outputting, from the third terminal, a signal reflected by the signal reflecting means among signals input from the first terminal of the input / output separating means. A first pass filter device, wherein the signal reflecting means is provided to face each other at a predetermined distance from each other such that each end faces each other and each other end faces each other.
And a second transmission line, and one end of the first transmission line is provided between the first and second transmission lines such that one end faces each end of the first and second transmission lines. A third transmission line that is distributedly coupled to a portion to form a first directional coupler and that is distributedly coupled to a portion of the second transmission line to form a second directional coupler; Λ at the center frequency of the pass band.
a first resonance circuit including a fourth transmission line having a length of about 1/4 of g, and a predetermined distance from each other such that each end faces each other and each other end faces each other. Fifth provided opposite
And a sixth transmission line, and the fifth transmission line is provided between the fifth and sixth transmission lines such that one end thereof is opposed to one end of each of the fifth and sixth transmission lines. A third transmission line which is distributedly coupled to a portion of the sixth transmission line, and a seventh transmission line which is distributedly coupled to a portion of the sixth transmission line to constitute a fourth directional coupler; Is connected to the other end of the fourth transmission line, and a second resonance circuit including a resistor whose other end is connected to one end of the seventh transmission line is provided. A band-pass filter device connected to the second terminal of the input / output separating means at a connection portion with a resistor. 2. The first resonance circuit further comprising: an end provided between the first and second transmission lines so as to face the other end of the third transmission line. Of the transmission line excluding the portion that is distributedly coupled with the third transmission line, is distributedly coupled with the first transmission line,
And an eighth transmission line that is distributedly coupled to the second transmission line in a portion of the second transmission line other than the portion that is distributedly coupled to the third transmission line. Further, one end is provided between the fifth and sixth transmission lines so as to face the other end of the seventh transmission line, and the fifth transmission line is distributed with the seventh transmission line. The sixth transmission line is distributed coupling-coupled with the fifth transmission line at a portion other than the portion coupled with the sixth transmission line, and the sixth transmission line is distributed coupling-coupled with the seventh transmission line of the sixth transmission line. 2. The band-pass filter device according to claim 1, further comprising a ninth transmission line that is distributedly coupled to the transmission line. 3. A negative resistance circuit is provided between the other end of the eighth transmission line and the ground terminal and between the other end of the ninth transmission line and the ground terminal. Claim 2
The band pass filter device according to claim 1. 4. The first and second transmission lines and the fifth transmission line.
And a variable capacitor capable of changing a capacitance between any one of the one end and the other end of the sixth transmission line and the ground end. The bandpass filter device according to one of the three. 5. The band pass filter device according to claim 1, wherein said input / output separating means is a circulator. 6. The band pass filter device according to claim 1, wherein said input / output separating means is a 3 dB directional coupler.