JPH09307306A - Microwave filter and its designing method - Google Patents

Abstract

PROBLEM TO BE SOLVED: To reduce a size, a cost and a loss in a microwave filter. SOLUTION: Flat or film-like dielectric resonators R3 and R4 are laminated in the direction of the thickness of a substrate and electromagnetically connected at a combining window 30 provided for an external conductive layer 24-2 positioned at the boundary of the both. Otherwise, the resonators are connected through a distribution constant element in the state of a flat surface of the substrate or a measuring combination line. Otherwise an input/output conductive layer and a strip conductive layer are combined through the distribution constant element. In addition, a loss is reduced by providing a groove for a part confronting an inner conductive layer among outer conductive layers.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a microwave filter using a dielectric resonator having a flat plate or film transmission line structure and a method for designing the same.

[0002]

2. Description of the Related Art Impedance when a finite-length transmission line is viewed from one end changes depending on what impedance the transmission line is terminated with, and what wavelength signal is transmitted through the transmission line. It also changes depending on what you do. For example, consider a case where one end of a transmission line is short-circuited and the short-circuited end side is viewed from the other end of the transmission line. In this case, the impedance of the transmission line looks infinite for a signal having a guide wavelength equal to ¼ of the line length of the transmission line. This phenomenon is called 1/4 wavelength resonance because it has the same effect as parallel resonance in a lumped constant circuit, and an element using this phenomenon is called a 1/4 wavelength resonator. Also, 1/2 of the line length of the transmission line
For a signal having a guided wavelength equal to, the impedance of its transmission line appears to be zero. This phenomenon is called 1/2 wavelength resonance because it has the same effect as series resonance.
An element using this phenomenon is called a 1/2 wavelength resonator.

The quarter-wave resonator and the half-wave resonator can be realized by using any dielectric as the dielectric layer of the transmission line. However, the wavelength of the signal in the dielectric body (wavelength in the tube) becomes shorter as the dielectric constant of the dielectric body becomes higher. Therefore, using a high dielectric constant material for the dielectric layer is advantageous in terms of downsizing. 1 using such material
The quarter-wave resonator and the half-wave resonator are usually called dielectric resonators, and their resonance frequencies generally belong to the microwave region. Further, by configuring a filter of a coupled resonance type or the like using a plurality of dielectric resonators, BPFs and Bs suitable for a high frequency (RF) circuit of a communication device in the microwave band.
RF etc. can be realized. This type of filter is generally called a dielectric filter.

A dielectric filter is disclosed in Japanese Patent Laid-Open No. 7-26391.
It can also be realized by selectively making the surfaces of a plurality of dielectric substrates into conductors and stacking the dielectric substrates as shown in Japanese Patent Laid-Open No. 0. When this method or structure is used, a plurality of dielectric resonators forming a single dielectric filter can be simultaneously manufactured (that is, using a common dielectric substrate), which simplifies the manufacturing process. It is possible to improve the degree of integration. Furthermore, since this method is a combination of general-purpose methods such as selective conductorization of the substrate surface and lamination of the substrates, it is possible to reduce the manufacturing cost and integrate it with other components or assemblies in the target RF circuit. Is suitable. The structure of the dielectric filter realized according to this method is partially simplified in FIG. The dielectric filter shown in this figure is a coupled resonance type filter composed of two quarter-wave dielectric resonators R1 and R2, and has five dielectric substrates 10 whose surfaces are selectively made conductive. -1 to 10
It is realized by stacking -5.

Of the five dielectric substrates 10-1 to 10-5, the first dielectric substrate 10-1 from the top in the figure has the upper surface thereof, and the first dielectric substrate 10 from the bottom in the figure. The lower surface of -5 is made into a conductor, whereby the outer conductor layers 12-1 and 12-2 shared by the resonators R1 and R2 are realized. Furthermore, on the upper surface of the third dielectric substrate 10-3 from the top in the figure, two inner conductor layers 16-1 and 16 are arranged along the depth direction in the figure and parallel to each other.
-2 is formed. Outer conductor layer 12-1, inner conductor layer 1
6-1 and the outer conductor layer 12-2 form a transmission line, that is, a triplate line having a structure in which an inner conductor layer is arranged between two outer conductor layers via a dielectric layer. Outer conductor layer 12-
Similarly, the inner conductor layer 16-2, the outer conductor layer 12-2, and the outer conductor layer 12-2 also form triplate lines. Furthermore, the inner conductor layer 1
6-1 and 16-2 are dielectric substrates 10-1 to 10-5.
Side short-circuit conductor layer 14-formed on the end face on the front side in FIG.
It is short-circuited to the outer conductor layers 12-1 and 12-2 via 1 to 14-5. In this way, the resonators R1 and R2 obtained by short-circuiting one end of each triplate line are both quarter wavelength resonators.

When the dielectric substrates 10-1 to 10-5 are laminated on the upper surface of the second dielectric substrate 10-2 from the top in the figure, both ends of the dielectric substrates 10-1 to 10-5 are placed inside the dielectric substrate 10-2. An inter-resonator coupling conductor layer 18 is formed so as to face the conductor layers 16-1 and 16-2. That is, the inter-resonator coupling conductor layer 18 is capacitively coupled to the inner conductor layers 16-1 and 16-2, respectively, and therefore, the resonators R1 and R2 are the inter-resonator coupling conductor layer 18 and the inner conductor layer 16-. 1 and 16-2 are coupled to each other via the capacitance (and the inductance of the inter-resonator coupling conductor layer 18 itself) associated with coupling with 1 and 16-2.
The degree of coupling can be adjusted by the position, shape and size of the inter-cavity coupling conductor layer 18. Further, on the upper surface of the fourth dielectric substrate 10-4 from the top in the figure, the dielectric substrates 10-1 to 1
When 0-5 are laminated, they are arranged so as to face the inner conductor layers 16-1 and 16-2 via the dielectric substrate 10-3, respectively.
Input / output conductor layers 20-1 and 20-2 are formed. That is, the input / output conductor layer 20-1 is the inner conductor layer 16-1.
And the input / output conductor layer 20-2 are capacitively coupled to the inner conductor layer 16-2, respectively, and therefore, the resonators R1 and R2.
Is the input / output conductor layers 20-1 and 20-2 and the inner conductor layer 16
-1 and 16-2 are connected to an external circuit (not shown) via an electrostatic capacitance associated with the connection. The degree of coupling can be adjusted by the positions, shapes and dimensions of the input / output conductor layers 20-1 and 20-2.

Here, in the frequency band in which both the resonators R1 and R2 function as a parallel resonator, all signals from the input / output conductor layers 20-1 or 20-2 are transmitted through the inter-resonator coupling conductor layer 18. Propagate to the other input / output conductor layer. In the upper and lower frequency bands, conversely, the input / output conductor layer 20-1
Alternatively, the signal from 20-2 is reflected by resonator R1 or R2. Therefore, the filter according to this conventional technique is a BPF whose pass band includes the resonance frequencies of the resonators R1 and R2. Further, this filter has the same advantages as the ordinary coupled resonance type filter and the dielectric filter,
Since it uses the substrate technology, it has the advantage of being excellent in miniaturization and high integration.

[0008]

However, the above structure has some problems. First, in order to prevent direct electromagnetic coupling between the two input / output conductor layers 20-1 and 20-2, the input / output conductor layer 20-1 and the input / output conductor layer 20-2 are The distance between must be widened to some extent. Therefore, the distance between the inner conductor layer 16-1 and the inner conductor layer 16-2 must be widened, and the dimension in the left-right direction in the drawing becomes large. Second, in order to freely adjust the nature and degree of inter-resonator coupling and input / output coupling, as shown in the figure, inter-resonator coupling conductor layer 18 is provided.
And the input / output conductor layers 20-1 and 20-2 are preferably formed on different dielectric substrates 10-2 and 10-4.
However, this increases the number of dielectric substrates.

One of the objects of the present invention is to improve the method or structure of inter-resonator coupling so as to prevent the direct electromagnetic coupling between at least two input / output conductor layers, and to secure the space therebetween. It is to realize a more compact and low-priced microwave filter. One of the objects of the present invention is to improve the method or structure of inter-resonator coupling so that at least two of the inner conductor layer, the input / output conductor layer and the inter-resonator coupling conductor layer can be arranged on the same plane. Therefore, it is to realize a smaller, thinner, and lower-priced microwave filter. One of the objects of the present invention is to realize a smaller, thinner, and lower-priced microwave filter by deforming the structure so that one resonator becomes a part of the other resonator. One of the objects of the present invention is to simplify the structure and increase the degree of design freedom by utilizing side coupling.

One of the objects of the present invention is to realize a microwave filter of lower cost by realizing a microwave filter capable of achieving the above-mentioned objects by a simple process. One of the objects of the present invention is at least 2
It is to realize a microwave filter having higher integration by setting the positional relationship between the input / output conductor layers provided individually. One of the objects of the present invention is to realize a microwave filter whose characteristics, in particular, the steepness of out-of-band attenuation and the arrangement of poles or zeros can be arbitrarily set by devising the arrangement of the inner conductor layer and its coupling means. . One of the objects of the present invention is
It is to realize a microwave filter with less loss by devising the conductor layer shape.

[0011]

In order to achieve such an object, a first structure of the present invention is a coupled resonance type microwave filter in which at least two inner conductors are formed on different surfaces. Layer, at least two input / output conductor layers coupled to any one of the inner conductor layers, and an outer conductor layer that electromagnetically shields the inner conductor layers and the input / output conductor layers from each other through a dielectric layer. Alternatively, a coupling window having a film-shaped transmission line structure and electromagnetically coupling the inner conductor layers by disturbing the electromagnetic shield is formed in the outer conductor layer.

In this structure, the inner conductor layers are electromagnetically coupled to each other through the coupling window (for example, the "small hole" of the outer conductor layer). Therefore, it is not necessary to provide a conductor layer such as an inter-resonator coupling conductor layer in the above publication, that is, a special conductor layer for coupling between dielectric resonators, and the structure is simpler and the processing is simplified as compared with the conventional one. Therefore, the cost of the microwave filter can be reduced. Furthermore, since this coupling window is adopted as the inner conductor layer coupling means, the positional relationship between the inner conductor layers is at least one outer conductor layer and the dielectric layer separated from the conventional relationship in which they are on the same plane. It has changed to a new relationship in which the transmission line structure is laminated in the thickness direction. Therefore, even when the number of inner conductor layers and thus the number of dielectric resonators is increased, the thickness is increased and the area is hardly changed.
As a result, a smaller microwave filter is realized.

In this configuration, since the input / output conductor layers are electromagnetically shielded by the outer conductor layer, it is not necessary to prevent direct electromagnetic coupling between the input / output conductor layers by securing a space as in the conventional case. Since it is possible to prevent the size enlargement caused by securing the space, a microwave filter which is smaller and less expensive than the conventional one can be realized. In this configuration, the inner conductor layer and the input / output conductor layer are electromagnetically shielded from the other inner conductor layer and the input / output conductor layer by the outer conductor layer (however, the electromagnetic coupling between the inner conductor layers via the coupling window is excluded. ) It is in a state. Therefore, the position of the input / output conductor layer can be set without considering the positional relationship with other input / output conductor layers, and a microwave filter having a high degree of freedom in design can be realized.

In the first structure described above, it is possible to arrange the inner conductor layer and the input / output conductor layer on different planes. However, arranging the inner conductor layer and the input / output conductor layer on the same plane can contribute to reduction in volume of the dielectric layer (for example, reduction in the number of dielectric substrates) and cost reduction due to simplification of processing. A microwave filter according to a second configuration of the present invention is characterized in that, in the first configuration, the input / output conductor layer and an inner conductor layer to be coupled by the input / output conductor layer are coupled via end-to-end capacitive coupling. To do. In this configuration, in addition to the same operation as in the first configuration, since the capacitive coupling between the end faces is utilized, the input / output conductor layer and the inner conductor layer can be arranged on the same plane. A compact, thin, low-cost microwave filter is realized. The same plane can also be obtained by connecting the inner conductor layer and the input / output conductor layer via a conductive member (or an inductance).

Further, in the prior art, since a plurality of input / output conductor layers were formed on the same plane, it was necessary to keep the distance between them, and it was difficult to draw them out to the positions close to each other. In the first and second configurations,
Since the plurality of input / output conductor layers are located on mutually different planes, it is possible to draw them out in close proximity. A microwave filter according to a third configuration of the present invention is
In the first configuration, the inner conductor layer and the input / output conductor layer are arranged such that the input / output conductor layers are arranged close to each other via the outer conductor layer. In this configuration, the simple method of setting the positional relationship between the input / output conductor layers and the connection / arrangement relationship between the inner conductor layers,
A microwave filter capable of performing input and output from the same end face, that is, a microwave filter with high integration is realized, and the configuration of the external circuit in the vicinity of the input / output conductor layer is downsized.

In the above-mentioned first to third structures, the input / output conductor layer, the inner conductor layer, the coupling window, the other inner conductor layers,
The electromagnetic fields are propagating in the order of ..., other input / output conductor layers. The path (electromagnetic field propagation path between the input and output conductor layers)
The electrical length and the equivalent circuit configuration of the are determined by the line length of the inner conductor layer that constitutes the electromagnetic field propagation path, the property of the coupling window (capacitive or inductive), and the like. Further, depending on the setting of the relative positional relationship between the inner conductor layer and the coupling window, a plurality of the electromagnetic field propagation paths can coexist at the same time. A design method according to a fourth configuration of the present invention is that, when designing the microwave filter according to the first configuration, a relative positional relationship between the inner conductor layer and the coupling window is determined from among the input / output conductor layers. It is characterized in that a plurality of electromagnetic field propagation paths from one to another are set to coexist at or near the resonance frequency. In such a method, therefore, by making a plurality of electromagnetic field propagation paths coexist, it becomes possible to design or set the characteristics of the microwave filter according to the first configuration in more detail, and the degree of freedom in design is improved. To do.

For example, the designing method according to the fifth aspect of the present invention is the same as the fourth aspect, except that in the fourth aspect, a plurality of the above-mentioned plurality of types are provided so as to exhibit a phase difference of π or −π [rad] at the frequency of the required attenuation pole. It is characterized in that a difference in electrical length between the electromagnetic field propagation paths is set. In a microwave filter designed according to this method, it is attenuated without adding an external circuit to a frequency where a pole or zero is not usually provided without adding an external circuit, such as in the pass band or in the vicinity of the pass band. It is possible to provide poles. In this way, the attenuation pole arrangement can be freely set depending on the positions of the inner conductor layer and the coupling window, etc., which improves the degree of freedom in the characteristic design of the microwave filter
In addition, the miniaturization and cost reduction can be realized by eliminating the external circuit.

In the first to third structures described above, the strength and nature of the coupling between the dielectric resonators (inner conductor layers) can be set by setting the position, shape and size of the coupling window. The designing method according to the sixth configuration of the present invention, in designing the microwave filter according to the first configuration, changes the coupling strength by the coupling window relatively steeply with respect to the change in frequency or wavelength. The position of the coupling window is set so that In the microwave filter obtained by this method, the coupling strength due to the coupling window changes relatively sharply with respect to the change in frequency or wavelength,
The cutoff characteristic (attenuation characteristic outside the pass band in the case of BPF) becomes steep.

Further, the designing method according to the seventh structure of the present invention, in designing the microwave filter according to the first structure, comprises: On the other hand, it is characterized in that the positions, shapes and dimensions of the first and second coupling windows, which are the coupling windows, are set so that they are electromagnetically coupled via the second coupling window. In this method, the characteristics of the microwave filter can be set or determined by a simple method of designing coupling between inner conductor layers by designing coupling windows.

For example, in the design method according to the eighth configuration of the present invention, in the seventh configuration, the electromagnetic coupling by the first coupling window and the electromagnetic coupling by the second coupling window emphasize each other. To match, the positions of the first and second coupling windows,
It is characterized by setting the shape and dimensions. In this configuration, stronger coupling can be realized as compared with the case where the inner conductor layers are coupled by a single coupling window, and the strength thereof can be set by the positions, shapes and dimensions of the first and second coupling windows.

In the designing method according to the ninth structure of the present invention, in the seventh structure, the lines of electric force from one of the inner conductor layers of the at least two dielectric resonators to the other are linked. The first coupling window is arranged at a position, the second coupling window is arranged at a position where the magnetic field lines near the inner conductor layer intersect, and the capacitive coupling by the first coupling window and the second coupling window are arranged. It is characterized in that the positions, shapes and dimensions of the first and second coupling windows are set so that a pole or a zero point is generated at a required frequency in combination with the inductive coupling by the coupling window. In this configuration, since the first coupling window is capacitive and the second coupling window is inductive, microwaves can be generated by a simple method of setting the positions, shapes and dimensions of the first and second coupling windows. It is possible to arbitrarily set the characteristics of the filter, particularly the arrangement of the poles or zeros.

A tenth structure of the present invention is, in a coupled resonance type microwave filter, at least two, which are formed on the same plane and whose end faces are capacitively coupled or inductively coupled to each other.
Inner conductor layers, at least two input / output conductor layers formed on the same plane as the inner conductor layers and capacitively coupled or inductively coupled to any one of the inner conductor layers, and an outer conductor formed on a plane different from the inner conductor layers. It is characterized by having a flat plate-shaped or film-shaped transmission line structure in which layers are laminated with a dielectric layer interposed therebetween.

In this structure, at least two inner conductor layers are arranged in the area direction of the transmission line structure (for example, in parallel) as in the prior art. However, since the coupling between these inner conductor layers is realized by the coupling between the end faces of these inner conductor layers, the inter-resonator coupling element is arranged separately from the dielectric layer for disposing the inner conductor layers. There is no need to provide a dielectric layer for installation, and a compact, simple, low-cost structure can be obtained. Further, since the inner conductor layer coupling is a lumped constant type such as capacitive coupling (eg end face capacitance) or inductive coupling (eg conductor connection), inner conductor layer coupling and adjustment of the strength thereof are simply a modification of the inner conductor layer shape. Can be realized by any means.

Although the input / output conductor layer in the tenth structure may be provided on a plane different from that of the inner conductor layer, it is preferable to provide the input / output conductor layer on the same plane in terms of downsizing, simplification and cost reduction of the structure. . However, in the configuration in which the inner conductor layers are coupled to each other on a single plane like the tenth configuration, the inner conductor layers are likely to be close to each other, so that the input / output conductor layers are arranged on the same plane as the inner conductor layers. In some cases, the input / output conductor layers may be close to each other. Sufficient spacing must be provided between the input and output conductor layers to eliminate the resulting unwanted direct coupling between the input and output conductors. In order to prevent the increase in size due to the securing of the space, the microwave filter according to an eleventh configuration of the present invention is the microwave filter according to the tenth configuration, wherein an input / output coupling element for capacitively coupling or inductively coupling the input / output conductor layers is provided. It is characterized in that it is formed on the same plane as the writing output conductor layer. In this configuration, the coupling is positively generated between the input / output conductor layers. As a result, the degree of freedom in designing the filter characteristic is improved. Further, since the input / output conductor layer and the inner conductor layer can be formed on the same plane, a dielectric layer for disposing the input / output conductor layer is provided separately from the dielectric layer for disposing the inner conductor layer. No need, small, simple,
A low cost structure can be obtained.

When viewed between two input / output conductor layers,
In the eleventh configuration, an electromagnetic wave propagation path (first path) from the first input / output conductor layer to the second input / output conductor layer via the first and second inner conductor layers and a first input / output conductor layer are provided. An electromagnetic wave propagation path (second path) from the output conductor layer to the second input / output conductor layer through the input / output coupling element is generated. Therefore,
By designing both paths, especially the phase shift amount, it is possible to add an attenuation pole and set its arrangement without adding an external circuit.
That is, the twelfth configuration of the present invention is the same as the eleventh configuration, in which the input / output conductor layers are coupled in any one mode of the capacitive coupling and the inductive coupling, and the input / output conductor layers are associated with each other. It is characterized in that the conductor layer and the conductor layer are coupled in the other mode. In this configuration, for example,
Since the coupling mode between the first inner conductor layer and the first input / output conductor layer is the same as the coupling mode between the second inner conductor layer and the second input / output conductor layer, dielectric resonance In the frequency band in which the device acts as an inductive or capacitive element (that is, a band separated from the resonance frequency to some extent), the signal passing through the first path and the second path
A phase difference of .pi. Or -.pi. [Rad] is generated between the two signals and the signal that has passed through the path of 1. Therefore, the two signals cancel each other, and the phase difference becomes 0 at the resonance frequency or in the vicinity thereof, and both signals strengthen each other. Due to such an action, a pole (and a zero point) is generated. Further, the arrangement can be easily determined by the design of each coupling element and the input / output conductor layer.

In the coupled resonance type microwave filter according to the thirteenth aspect of the present invention, at least two inner conductor layers are formed on the same plane so that inter-end-face side coupling occurs between the inner conductor layers. A dielectric, at least two input / output conductor layers formed on the same plane as the inner conductor layer and capacitively or inductively coupled to any one of them, and an outer conductor layer formed on a plane different from the inner conductor layer. Having a flat plate-shaped or film-shaped transmission line structure laminated via layers, and in the extension direction thereof so that signal propagation from one inner conductor layer to the other inner conductor layer related to the end-to-end side coupling can occur. Along the above, the mode of coupling between the end faces is changed.

In this configuration, instead of the input / output coupling element in the eleventh configuration, the input and output are coupled by a transmission line related to the coupling between the end faces of the inner conductor layers, thereby providing a simple structure. Furthermore, by changing the mode of the coupling between the end faces along the extension direction of the inner conductor layer, a step impedance can be generated in the transmission line related to the coupling between the end faces, and, by extension, the inner conductor layer and the outer layer. A difference in resonance frequency can be provided between the resonator formed of the conductor layer, that is, the resonator to be coupled, and the transmission line related to the end-face side coupling, that is, the resonator as the coupling means. As a result, the resonance frequency of the resonators to be coupled (the zero point or the passing frequency when the quarter wavelength resonators are coupled to each other) and the resonance frequency of the resonators that are the coupling means (the poles or the blocking frequency in the same case). Can be set to different values, and the pole and zero can be individually set by designing the change pattern of the coupling mode between the end faces along the extension direction of the inner conductor layer. Further, since the input / output coupling element is not necessary, the structure is simpler than that of the eleventh configuration. In addition, even if direct capacitive coupling or the like occurs between the input / output conductor layers, no problem will occur because the resonators are coupled.

Furthermore, the change of the coupling mode between the end faces in the thirteenth structure can be realized by a plane circuit. First of the present invention
The structure of No. 4 is such that in the thirteenth structure, an edge distance between adjacent inner conductor layers is relatively narrow at a portion close to the open end and relatively wide at a portion separated from the open end. The conductor width of the inner conductor layer is set. A fifteenth configuration of the present invention is characterized in that, in the thirteenth or fourteenth configuration, the inner conductor layers are connected to each other by a bridge conductor layer at a portion separated from the open end. Thus, by partially narrowing the space between the inner conductor layers near the open end side (fourteenth configuration), the electric field governs the mode of the end face side coupling between the inner conductor layers at that portion. Mode, and also,
By connecting the inner conductor layers with the bridge conductor layer at a portion separated from the portion related to the constriction (fifteenth configuration), the magnetic field governs the mode of end-face side coupling between the inner conductor layers at that portion. Mode can be set. Since all of these have a planar structure, they can contribute to miniaturization and thinning of the microwave filter.

In addition, when both the inner conductor layer spacing narrowing and the bridge conductor layer are adopted, it is possible to realize a high degree of freedom in design which cannot be obtained by a configuration in which either one is adopted alone. First, by designing the inner conductor layer spacing narrowing (for example, setting the length of the narrowed portion and the narrowed inner conductor layer spacing), from the resonator relating to one inner conductor layer to the resonator relating to the other inner conductor layer. The degree of signal propagation (equivalent coupling capacitance between resonators) can be set, and thus the zero point or the passing frequency can be set. At the same time, the inner conductor layer space narrowing has an effect of shifting the resonance frequency of the transmission line related to the end-to-end side coupling from the resonance frequency of the resonator to be coupled. When both inner conductor layer spacing narrowing and bridge conductor layers are used,
Further, by shifting the resonance frequency of the transmission line related to the end-to-end side coupling by designing the bridge conductor layer (for example, setting the position and shape / dimension), that is, designing the transmission line formed by the bridge conductor layer and the outer conductor layer. You can Therefore, in the design of the inner conductor layer spacing narrowing and the design of the bridge conductor layer, the resonance frequency of the transmission line related to the coupling between resonators, that is, the passband and the coupling between the end faces, that is, the stopband is designed independently. High design method can be adopted.

The sixteenth structure of the present invention is, in a coupled resonance type microwave filter, at least two inner conductor layers formed on the same plane, and at least two inner conductor layers formed on a plane different from the inner conductor layers. Individual input / output conductor layers, an input / output coupling element formed on the same plane as the input / output conductor layer and coupling these, and the input / output conductor layer and the input / output conductor layer formed on a plane different from the inner conductor layer. An outer conductor layer that electromagnetically shields between the output coupling element and the inner conductor layer has a flat plate or film transmission line structure in which a dielectric layer is laminated, and the electromagnetic shield is disturbed. Thus, a coupling window for electromagnetically coupling the input / output conductor layer and the inner conductor layer is formed in the outer conductor layer.

In this structure, the input / output conductor layers are coupled by the input / output coupling element, while the input / output conductor layers / inner conductor layers are coupled via the coupling window. Therefore, it is possible to obtain a coupled resonance type filter in which the inner conductor layers are coupled to each other through the coupling window and the input / output coupling element. Since this input / output conductor layer and the input / output coupling element are arranged on the same plane, this filter is small, thin, and low in cost. Furthermore, since the input / output conductor layers are positively connected to each other and used for connecting the inner conductor layers, it is not necessary to secure a space between the input / output conductor layers, and the size and cost are further reduced. In addition, all of these can be realized by the technique of selective conductorization and lamination of the substrate surface,
Since it can be carried out easily, it becomes even cheaper.

The designing method according to the seventeenth structure of the present invention is
In designing a microwave filter according to a sixteenth structure, a first structure including the inner conductor layer and the outer conductor layer is provided.
Compared with the dielectric constant of the dielectric layer used in the transmission line of
The dielectric constant of the dielectric layer used in the second transmission line composed of the input / output conductor layer, the input / output coupling element and the outer conductor layer is lowered. In this configuration, the conductor width in the second transmission line can be widened and the forming process thereof can be facilitated. Furthermore, since the impedance design of the first and second transmission lines can be made independent, it becomes easy to design the second transmission line according to the impedance of the external circuit, for example.

A microwave filter according to an eighteenth structure of the present invention is the microwave filter according to the sixteenth structure, wherein the chip-shaped dielectric or magnetic material is electromagnetically coupled between the input / output conductor layer and the inner conductor layer. It is characterized in that it is arranged in the vicinity of the coupling window and the input / output conductor layer so as to be interposed. In this configuration, the strength of the coupling can be adjusted by adjusting the arrangement of the chip-shaped dielectric or magnetic material.

A microwave filter according to a nineteenth structure of the present invention is an inner conductor layer, an input / output conductor layer formed on the same plane as the inner conductor layer, and the same plane as the inner conductor layer and the input / output conductor layer. An input / output inner conductor inter-layer coupling element formed on the above and inductively coupling or capacitively coupling between them, and an outer conductor layer formed on a plane different from the inner conductor layer are laminated through a dielectric layer into a flat plate shape or A _first resonator having a film-shaped transmission line structure, which is composed of the inner conductor layer and the outer conductor layer and resonates at a frequency f ₁ in the transmission line structure, the first resonator, and the input / output. Frequency f ₂ (where f ₁ ≠ f ₂ , | f ₁ −f ₂ | <
and a second resonator that resonates at ε, ε: a predetermined small value.

In this structure, the inner conductor layer and the outer conductor layer form a first resonator, and the first resonator and the input / output inner conductor interlayer coupling element form a second resonator. That is, since the first resonator is equivalent to the capacitance or the inductance except at the resonance frequency f ₁ , the input / output inner conductor interlayer coupling element (or the inductance or the capacitance related to the inductive coupling or the capacitive coupling) is used. The second resonator can be configured by using the second resonator. Since the difference between the resonance frequencies f ₁ and f ₂ of the _first and _second resonators is very small, in the present configuration, the second resonator is provided near the zero point or pole determined by the first resonator. The poles or zeros determined by Further, as compared with the conventional technique using coupling between dielectric resonators, since the first resonator is used as a part of the second resonator, the number of inner conductor layers is small and the inner conductor layers are small. Since the input / output conductor layer and the input / output inner conductor interlayer coupling element can be formed on the same plane, the number of dielectric layers is small, so that the microwave filter is smaller, thinner and less expensive. Further, since the input / output conductor layers may be directly connected by a conductor, it is not necessary to secure the space between them, which is also small and low in cost. In addition, since this structure can be realized only by selectively converting the surface of the substrate into a conductor and does not require substrate lamination, it can be easily manufactured, is small, thin and inexpensive as compared with the above-mentioned respective structures.

The design method according to the twentieth configuration of the present invention is:
In designing the microwave filter according to the nineteenth configuration, first, the resonance frequencies f1 and f2 of the _first and _second resonators are determined according to the frequencies of the poles or zeros to be realized, and then The inductance L related to the inductive coupling due to the conductor interlayer coupling element in the input output and the characteristic impedance Z ₁ of the first resonator are given by the following equation.

[Expression 3] Z ₁ <π ² L | f ₁ −f ₂ | Further, in the design method according to the twenty-first configuration of the present invention, in designing the microwave filter according to the nineteenth configuration, first, the first and second resonators are selected according to the frequency of the pole or zero to be realized. Of the resonance frequency f ₁ and f ₂ of the first resonator and the capacitance C related to capacitive coupling by the input / output inner conductor interlayer coupling element and the characteristic impedance Z ₁ of the first resonator are expressed by the following equation.

## EQU4 ## The feature is that Z ₁ > | f ₁ −f ₂ | / 4Cf ₂ ² is satisfied.

Here, the inductance L or the electrostatic capacitance C of the input / output inner conductor interlayer coupling element in the nineteenth structure usually has an upper limit in terms of the area of the dielectric substrate and the fineness of the conductor pattern formation. , Therefore characteristic impedance Z ₁
Is also constrained. The contents of this constraint are determined by the phase conditions in the first and second resonators, and can be approximately expressed by the above-mentioned equations. Therefore, in the configuration of the first 20 or 21, first, according to the frequency of the poles or zeros to be realized determining the resonance frequencies f ₁ and f _2. Next, the frequencies f ₁ and f ₂ are substituted into the above equations, and L or C is determined so that the equations hold. For example, when the first resonator at a frequency f ₂ are capacitive defines a L and Z ₁ in accordance with the equation according to the configuration of the 20 input and output in the conductor layers as the L and Z ₁ are realized Design the shape and size of the coupling element and the length of the inner conductor layer. Further, when the first resonator is inductive at the frequency f ₂ , C and Z ₁ are determined according to the equation relating to the 21st configuration, and the input and output inner conductor layers are set so that these C and Z ₁ are realized. Design the shape and size of the coupling element and the length of the inner conductor layer. The equations relating to the twentieth and twenty-first configurations are both constraint equations derived from the phase conditions in the first and second resonators, and an input / output inner conductor interlayer coupling element having a value satisfying this constraint equation is realized in principle. It is possible.
With such a simple means, in this configuration, the arrangement of the poles and zeros can be liberated, and the attenuation in the pass band can be reduced and the attenuation outside the band can be sharpened.

A microwave filter according to a twenty-second structure of the present invention is the microwave filter according to the first, tenth, thirteenth, sixteenth or nineteenth structure, wherein one portion of the outer conductor layer facing the inner conductor layer is It is characterized in that the portion is recessed so that the gap between the inner conductor layer and the inner conductor layer is narrower than the other portions. In this configuration,
Since the distance between the inner conductor layer and the outer conductor layer in the recessed portion is smaller than that in other portions, the current is likely to be concentrated in the portion of the inner conductor layer facing the recess of the outer conductor layer.
As a result, the current distribution in the width direction of the inner conductor layer becomes close to a flat distribution, and therefore the loss in the line becomes small.

[0039]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Preferred embodiments of the present invention will be described below with reference to the drawings. Note that, for simplification of description, the same or corresponding members used in different embodiments are denoted by common reference numerals, and description thereof will be omitted.

(1) Group of Embodiments Having Coupling Window for Resonator Coupling First Embodiment. FIG. 1 shows the structure of the microwave filter according to the first embodiment of the present invention. In this embodiment, four dielectric substrates 22-1 whose surfaces are selectively made conductive are provided.
22 to 22-4 are laminated. Of these four dielectric substrates 22-1 to 22-4, the top surfaces of the first and third dielectric substrates 22-1 and 22-3 from the top in the figure and the fourth dielectric substrate 22- The lower surface of 4 is made into a conductor over substantially the entire surface except for the portion of the coupling window 30, and thereby the outer conductor layers 24-1 to 24-3 are respectively realized. The outer conductor layers 24-1 to 24-3 are short-circuited to each other by the side surface short-circuit conductor layers 26-1 to 26-4. Although the side surface short-circuit conductor layers 26-1 to 26-4 are provided on the right end surfaces of the dielectric substrates 22-1 to 22-4 in this figure, in reality, the input / output conductor layers 32-1 and 32 -It suffices if it is provided so as to avoid contact with the -2. Further, the inner conductor layer 28-1 and the input / output conductor layer 32-1 are provided on the upper surface of the second dielectric substrate 22-2 from the top in the figure.
The inner conductor layer 2 is formed on the upper surface of the fourth dielectric substrate 22-4.
8-2 and the input / output conductor layer 32-2 are formed respectively. The inner conductor layers 28-1 and 28-2 are strip-shaped conductors, and are located substantially at the center of the corresponding dielectric substrate in the figure. Also, the input / output conductor layers 32-1 and 32
-2 is adjacent to the corresponding inner conductor layer via a minute gap (that is, so that the capacitive elements 33-1 and 33-2 relating to the capacitive coupling between the end faces are generated). Further, the input / output conductor layer 32-1
Are on the back side in the figure, and the input / output conductor layer 32-2 is on the front side.

Therefore, in this embodiment, the outer conductor layer 24-
First triplate line R3 having ground conductors 1 and 24-2 above and below, and outer conductor layers 24-2 and 24-3
A second triplate line R4, whose upper and lower ground conductors are connected to each other, is laminated in the thickness direction.
Further, the outer conductor layer 24-2 that electromagnetically shields between the resonators R3 and R4 has a coupling window 30 (for example, a "small hole opened in the outer conductor layer 24-2" that is a portion having no electromagnetic shielding property. ". There may be more than one). Due to the presence of this coupling window 30, the electromagnetic shield between the lines R3 and R4 is partially disturbed, and the lines R3 and R4 (more specifically, the inner conductor layers 28-1 and 28-2 thereof) are electromagnetically disturbed. Join. In particular, as shown in FIG. 1, the inner conductor layer 28-1 and the inner conductor layer 28-2
When the coupling window 30 is provided on the line connecting the two, the lines of electric force mainly interlink with the coupling window 30, and thus the coupling is generally a mode in which electric field coupling is dominant, that is, capacitive coupling. On the other hand, of the two ends of each of the inner conductor layers 28-1 and 28-2, the end opposite to the input / output conductor layers 32-1 and 32-2 is open.

As described above, in this embodiment, the two triplate lines R3 and R4 are capacitively coupled through the coupling window 30, and these are further coupled to the input / output conductor layer 32 by the capacitive elements 33-1 and 33-2. -1 and 32-2 to form a coupled resonant filter. Further, since the passband is formed at the frequency at which the triplate lines R3 and R4 resonate at 1/2 wavelength and in the vicinity thereof, this coupled resonance filter is a BPF that utilizes 1/2 wavelength resonance. Contains the half wavelength resonant frequency of each of the resonators R3 and R4. The dimensions of the inner conductor layers 28-1 and 28-2 may be slightly different depending on the required filter characteristics.

Since this embodiment is realized by using the technique of selectively conducting the surface of the substrate and the technique of laminating the substrate as in the above-mentioned conventional technique, the manufacturing is easy. Furthermore, the coupling window 3
At 0, the electromagnetic shield between the resonators R3 and R4 is disturbed and both are electromagnetically coupled, so that it is not necessary to use the inter-resonator coupling conductor layer 18 and the dielectric substrate 10-2 therefor as in the conventional case. The structure is simple and thin. Also, the coupling window 30
The resonators R3 and R4 by setting the position, shape, dimensions, etc.
Since the coupling between the two can be changed and the filter characteristic can be changed, various filter characteristics can be realized by a simple method of setting the position of the coupling window 30 and the like. Further, the input / output conductor layers 32-1 and 32-
Since the outer conductor layer 24-2 is electromagnetically shielded between the two, it is not necessary to set the interval to be wide, and the input / output conductor layers 32-1 and 32-2 are laminated in the thickness direction. . Therefore, the dimension (width) in the left-right direction in the drawing can be narrowed as compared with the conventional technique in which the space between the input / output conductor layers 20-1 and 20-2 needs to be maintained. Further, since the input / output conductor layers 32-1 and 32-2 and the inner conductor layers 28-1 and 28-2 are capacitively coupled between the end faces, they can be arranged on the same plane. In addition, the formation of the coupling window 30 is relatively easy. Therefore, according to the present embodiment, it is possible to realize a compact, simple and low-priced microwave filter, and further increase the degree of freedom in its design.

The input / output conductor layers 32-1 and 32-2 in this embodiment may be pulled out to the same side (for example, the front side in the drawing). In this way, the peripheral circuits (not shown) connected to the input / output conductor layers 32-1 and 32-2 can be concentrated in one place, so that the peripheral circuits can be downsized and integrated. As described above, the present embodiment has an advantage that the position of each input / output conductor layer can be set without considering the positional relationship with other input / output conductor layers, and thus the degree of freedom in design is high. Furthermore,
In the present embodiment, the number of resonators is two, but when the number of resonators is increased to three or more, unlike the prior art, only the thickness is increased and the area is not increased. Also in this respect, the present embodiment is suitable for miniaturization and high integration. Although the arrangement of the input / output conductor layers and the modification of the number of resonators are not shown, those skilled in the art will be able to easily carry out the invention with reference to the embodiments described later.

Second Embodiment. FIG. 2 shows the structure of the microwave filter according to the second embodiment of the present invention. This embodiment has a structure in which eight dielectric substrates 22-1 to 22-8 whose surfaces are selectively made conductive are stacked.
Of these eight dielectric substrates 22-1 to 22-8, odd-numbered dielectric substrates 22-1, 22-3, 22 from the top in the figure
-5 and 22-7 upper surface and eighth dielectric substrate 22-8
The lower surfaces of all are made conductive over the entire surface except for the coupling windows 30-1 to 30-3, whereby the outer conductor layers 24-1 to 24-5 are respectively realized. These outer conductor layers 24-1 to 24-5 correspond to the dielectric substrates 22-1.
22-8, side surface short-circuit conductor layers 26-1 to 26- provided on at least one surface (right surface in the figure) of the end surfaces
8 are short-circuited to each other, whereby each of the resonators R3 to R3
Six outer conductor layers have been realized. Further, even-numbered dielectric substrates 22-2, 22-4, 22-6 and 2 from the top in the figure
Each of the strip-shaped inner conductor layers 2 is provided on the upper surface of 2-8.
8-1 to 28-4 are formed. Each inner conductor layer 28-
At least one end of each of 1-28-4 is in an open state.
With such a structure, four triplate lines R3
A structure in which ~ R6 are laminated in the thickness direction is obtained.

Further, on the upper surfaces of the third, fifth and seventh dielectric substrates 22-3, 22-5 and 22-7 from the top in the figure, similar to the coupling window 30 in the first embodiment. And then
Coupling windows 30-1 to 30-3 are formed. Coupling window 3
0-1 is between the inner conductor layer 28-1 and the inner conductor layer 28-2,
The coupling window 30-2 includes the inner conductor layer 28-2 and the inner conductor layer 28-3.
The coupling window 30-3 electromagnetically couples the inner conductor layer 28-3 and the inner conductor layer 28-4. In the case of this embodiment, the coupling windows 30-1 to 30-3 are located on the line connecting the inner conductor layers to each other, so that they exhibit capacitive coupling. In this way, the inner conductor layers 2 located at both ends of the four inner conductor layers 28-1 to 28-4 which are chained via the coupling window
8-1 and 28-4 are provided with input / output conductor layers 32-1 and 32-2 in close proximity to each other so that capacitive elements 33-1 and 33-2 are formed by capacitive coupling between the end faces and on the same plane. There is. Therefore, the microwave filter realized in the present embodiment is a coupled resonance type BPF in which the frequency at which the triplate lines R3 to R6 resonate at 1/2 wavelength and the vicinity thereof are the passbands, and the resonators R3 to R6. Are electromagnetically coupled in sequence through the capacitive coupling windows 30-1 to 30-3, and these are coupled via the capacitive elements 33-1 and 33-2.
It has the structure combined with 2-1 and 32-2.

According to the above configuration, the same effect as that of the first embodiment can be obtained. Further, also in this embodiment, the same modifications as those shown in the first embodiment can be made. In the present embodiment, further, the input / output conductor layer 32-1
And 32-2 are located close to each other (in the figure, the front left corner of the dielectric substrate), the peripheral circuits (not shown) connected to the input / output conductor layers 32-1 and 32-2 can be integrated in one place. Therefore, miniaturization and integration of the peripheral circuit can be realized. Compared to the first embodiment, in the present embodiment in which a large number of resonators are stacked, the effect of making the input / output conductor layers close to each other can be realized is that the inner conductor layer 2 is extended in the extending direction of the inner conductor layer 28-1.
The extension direction of 8-2 is inclined by approximately π / 2 [rad], the extension direction of the inner conductor layer 28-3 is inclined by approximately π / 2 [rad] with respect to the extension direction of the inner conductor layer 28-2, and the inner conductor layer is further inclined. 28-3
The extension direction of the inner conductor layer 28-4 with respect to the extension direction of
The inner conductor layer 28-
This is because the electromagnetic field propagation direction was gradually rotated once by setting the direction or posture of 1-28-4. In addition, the coupling accompanied by such rotation is made possible by the coupling window 30-1.
˜30-3 is introduced and further coupling windows 30-1 to 30-3
This is because the 0-3 position is set to enable the above arrangement. In order to obtain the same effect while increasing the number of resonators, the angle formed by the inner conductor layers should be π / 2 [rad].
It may be made smaller, or the electromagnetic field propagation direction may be rotated twice or more.

Third Embodiment FIG. 3 shows the structure of the microwave filter according to the third embodiment of the present invention. This embodiment further includes three coupling windows 30-4 to 30-4 in addition to the second embodiment.
It has a configuration in which 0-6 is added. Coupling window 30-4 ~
30-6 is formed on the upper surfaces of the third, fifth, and seventh dielectric substrates 22-3, 22-5, and 22-7 in this order from the top, and the position thereof is the inner conductor layer 28-. 1 on the line connecting the portion of the inner conductor layer 32-1 near the input / output conductor layer 32-1 and the portion of the inner conductor layer 28-4 near the input / output conductor layer 32-2.
It is in a position where it does not intersect with 8-2 and 28-3.

Therefore, in this embodiment, at the resonance frequency or in the vicinity thereof, an electromagnetic wave propagation path similar to that of the second embodiment is generated on the one hand, and a new electromagnetic wave propagation path is generated on the other hand. The former is from the input / output conductor layer 32-1 to the capacitive element 33-
1, inner conductor layers 28-1 to 28-4, coupling windows 30-1 to 30-3
0-3 and the capacitive element 33-2, and the input / output conductor layer 32.
-2 (first path), and the latter is from the input / output conductor layer 32-1 to the capacitive element 33-1 and the coupling window 30-4.
The input / output conductor layer 32 through 30-6 and the capacitive element 33-2
It is a route (second route) to -2. In this way, when a plurality of electromagnetic field propagation paths that are different from each other, that is, generally have different responses to inputs, are made to coexist, the output is the additive synthesis of the responses to both electromagnetic field propagation paths. Therefore, by applying this, the characteristics of the filter can be designed or set in more detail. In this embodiment, two electromagnetic field propagation paths from the input / output conductor layer 32-1 to the input / output conductor layer 32-2 are made to coexist by setting the relative positional relationship between the inner conductor layer and the coupling window. Therefore, it is possible to set the characteristics based on such a principle relatively easily.

For example, pay attention to each capacitive coupling in the figure. The total number of capacitive couplings existing on the first path is five (two capacitive elements 33-1 and 33-2 and the coupling windows 30-1 to 30-3.
In contrast to the three capacitive couplings according to 1), the total number of capacitive couplings existing on the second path is three (two capacitive elements 33-1 and 33-2 and the coupling window 30-4. 30-6 is one capacitive coupling). Since the amount of phase shift for one capacitive coupling is −π / 2 [rad], the amount of phase shift caused by all capacitive coupling on the route is π between the first and second routes.
[Rad] Different. On the other hand, although the phase shift also occurs depending on the line length of each of the resonators R3 to R6 (the length of the inner conductor layers 28-1 to 28-4), these line lengths are appropriately set (for example, the mutual difference between them is adjusted). by setting to) be at the desired frequency f _r of the pass-band or near the resonator R3~R6
The total line length can be set to 0 [rad].
At such frequency f _r , the phase shift amount is determined only by capacitive coupling, so that the signal passing through the first path and the signal passing through the second path have opposite phases and thus cancel each other out. As a result, the attenuation pole appears at the frequency f _r.

According to this embodiment, the same effects as those of the first and second embodiments can be obtained. Furthermore, by setting the difference in the electrical length between the electromagnetic field propagation paths so as to exhibit a phase difference of π [rad] at the required attenuation pole frequency, it is normally possible to use external signals such as in the pass band or in the immediate vicinity of the pass band. It becomes possible to provide an attenuation pole without adding an external circuit at a frequency where an attenuation pole cannot be provided without adding a circuit. Further, this attenuation pole can be freely set depending on the positions of the inner conductor layer and the coupling window. The electromagnetic field propagation paths may be three or more.
Further, in general, a path corresponding to the above-mentioned second path can be provided other than between the first-stage and last-stage resonators.
Therefore, the input / output conductor layers 32-1 and 32-2 are not always required.
It is not necessary to arrange the inner conductor layer and the coupling window so that they are close to each other.

Fourth Embodiment. FIG. 4 shows the structure of the microwave filter according to the fourth embodiment of the present invention. This embodiment is an example in which a modification for shorting one end of the inner conductor layers 28-1 and 28-2 to the outer conductor layer is applied to the first embodiment. That is, the resonators R7 and R8 in this embodiment are both quarter-wave dielectric resonators. Further, as means for short-circuiting one end of the inner conductor layers 28-1 and 28-2 to the outer conductor layer, the ground portions 36-1 and 3 formed of a conductor are used.
6-2 is used. Grounding parts 36-1 and 36-2
Are respectively one end of the inner conductor layers 28-1 and 28-2 and the side short-circuit conductor layer 26 of the dielectric substrates 22-2 and 22-4.
-2 and 26-4 are short-circuited. in this way,
The microwave filter of the present embodiment couples two quarter-wave dielectric resonators R7 and R8 via a capacitive coupling window 30 and further inputs / outputs them via capacitive elements 33-1 and 33-2. This is a coupled resonance type BPF connected to the conductor layers 32-1 and 32-2, and the pass band thereof includes the resonance frequencies of the resonators R7 and R8.

Therefore, according to this embodiment, it is possible to realize a BPF having the same effect as that of the first embodiment. In addition, the resonators R3 and R4 in the first embodiment are 1/2
In contrast to the wavelength resonator, the resonators R7 and R8 in the present embodiment are quarter wavelength resonators, so the resonator R
The line length dimension of 7 and R8 is half that of the resonators R3 and R4. As a result, the size of the entire filter in the depth direction in the figure is also reduced to nearly 1/2.

Fifth Embodiment. FIG. 5 shows the structure of the microwave filter according to the fifth embodiment of the present invention. This embodiment is a modification of the fourth embodiment in which the number and position of the coupling windows are changed. That is, in the fourth embodiment, one coupling window 30 is arranged on a line connecting the centers of the resonators R7 and R8 (positions having a distance corresponding to 1/8 of the resonance wavelength from the end on the input / output conductor layer side). In contrast to this, in this embodiment, two coupling windows 30 are formed on the line connecting the positions of the resonators R7 and R8, which are slightly closer to the ends when viewed from the center.
-1 and 30-2 are arranged.

According to this embodiment, the same effect as that of the fourth embodiment can be obtained. Furthermore, the coupling windows 30-1 and 30-
2 is provided at a position closer to the end, that is, at a position where the change in the coupling strength with respect to the frequency change is larger than that in the center, so that a signal outside the pass band is transmitted from one resonator to the other resonator as compared with the fourth embodiment. Hard to propagate to. As a result, the cutoff characteristic (out-of-band attenuation characteristic due to the BPF in this embodiment) becomes steep. Further, since the two coupling windows are provided, the strength of the coupling can be secured. The number of coupling windows is 3
It may be more than one. The binding through the binding window may be inductive.

Sixth Embodiment. FIG. 6 shows the structure of the microwave filter according to the sixth embodiment of the present invention. This embodiment differs from the fourth embodiment in that the input / output conductor layers 32-1 and 32-2 are both on the front side and the ground portions 36-1 and 36 are.
-2 is located on the back side, a modification for setting the arrangement of each conductor layer, and the input / output conductor layers 32-1 and 32-2 of the inner conductor layers 28-1 and 28-2. A coupling window 30-1 is provided on the line connecting the parts closer to each other,
The coupling window 30 is located on the two lines connecting the sides of the portions near the grounding portions 36-1 and 36-2, that is, at a position where it is easy to interlink with the magnetic field lines.
-2 and 30-3 are modified.

According to this embodiment, the same effect as that of the fourth embodiment can be obtained. Further, as a result of the above-described modification, the input / output conductor layers 32-1 and 32-2 are located close to each other, so that the external circuit in the periphery thereof becomes compact. That is, by a simple method of setting the coupling / arrangement relationship of the inner conductor layers 28-1 and 28-2 and the like, it is possible to inexpensively and easily realize a microwave filter having a high degree of integration capable of executing input and output from the same end face. . The coupling mode realized by the coupling window 30-1 is capacitive coupling in which the electric field is dominant, and the coupling mode realized by the coupling windows 30-2 and 30-3 is a magnetic field dominant induction. As a result of the above-mentioned deformation, the capacitive coupling and the coupling window 3 related to the coupling window 30-1 are generated.
An attenuation pole appears at a frequency determined by the strengths of both inductive couplings 0-2 and 30-3. Therefore, the coupling window 30-1
By appropriately designing the position and shape of ~ 30-3,
Alternatively, the attenuation pole can be arranged at an arbitrary frequency by changing the number of coupling windows.

Seventh Embodiment. FIG. 7 shows the structure of the microwave filter according to the seventh embodiment of the present invention. This embodiment differs from the sixth embodiment in that it has coupling windows 30-2 and 30-3.
Is integrated at the window connecting portion (opening of the outer conductor layer here) 38. Even with this,
The same effect as the sixth embodiment can be obtained. Further, as is apparent from this, there are various modes of modification in the sixth embodiment.

Eighth Embodiment. FIG. 8 shows the structure of the microwave filter according to the eighth embodiment of the present invention. This embodiment is different from the fourth embodiment in that the number of quarter-wave dielectric resonators is increased from two (R7 and R8) to three (R7 to R9), and the number of resonators is increased. Accordingly, one coupling window is added (30-1 and 30-2 are provided), and the inner conductor layers 28-1 and 28- of the first and third resonators R7 and R9 from the top in the figure are changed. Input / output conductor layer 3
The coupling window 30 is formed on the line connecting the portions near 2-1 and 32-2.
-3 and 30-4 are provided. In order to realize the modification, the inner conductor layer 28-2 of the resonator (R8 in the figure) added by the modification is displaced to the right in the figure with respect to the inner conductor layers of the other resonators. Has been formed.

According to this embodiment, in addition to the same effects as in the fourth embodiment, the effects relating to the attenuation pole arrangement in the third embodiment can be realized by the three-resonator-coupled microwave filter. That is, in this embodiment, as in the third embodiment, two types of electromagnetic field propagation paths occur near the resonance frequencies of the resonators R7 to R9. One of them is a route (first route) passing through the coupling windows 30-1 and 30-2, and the other is a route (second route) passing through the coupling windows 30-3 and 30-4. In the first path, the signal input from the input / output conductor layer 32-1 is a total of −2π due to the four capacitive couplings.
It receives the phase shift of [rad] and the phase shift due to the line lengths of the inner conductor layers 28-1 and 28-3. On the other hand, in the second path, the signal input from the input / output conductor layer 32-1 undergoes a phase shift of −3π / 2 [rad] in total due to the three capacitive couplings. The phase shift amount due to the line lengths of the inner conductor layers 28-1 and 28-3 becomes a total of −π / 2 [rad] at a certain frequency near the resonance frequency of the resonators R7 to R9. Therefore, at this frequency, the phase shift difference between both paths is π [ra
d], and as a result, the attenuation pole appears as in the third embodiment. The frequency of the attenuation pole can be set appropriately according to the positions, shapes and dimensions of the coupling windows and the inner conductor layers. An attenuation pole can be added in the vicinity.

Two of the three capacitive couplings appearing on the second path are the capacitive elements 33-1 and 33-2. The remaining one is one capacitive coupling by the coupling windows 30-3 and 30-4, which appears near the resonance frequency of the resonators R7 to R9. Two of the four capacitive couplings appearing on the first path are capacitive elements 33-1 and 3-3.
3-2. The remaining two are combination windows 30-1 and 30.
Two capacitive couplings by -2, which are resonators R7
Appears in the vicinity of the resonance frequency of R9. Furthermore, the total phase shift due to the line lengths of the inner conductor layers 28-1 and 28-3 is -π.
/ 2 [rad] is the distance from the front end of the inner conductor layer 28-1 in the figure to the center (that is, the position facing the coupling window 30-1) and the center of the inner conductor layer 28-2 ( That is, the frequency is such that the sum of the distance from the position facing the coupling window 30-2) to the end on the near side in the figure is equal to ¼ of the wavelength.

In the inductive coupling window, π / 2 [ra
Since the phase shift component of d] can be provided, it is also possible to provide the same attenuation pole as in the present embodiment by utilizing this. in addition,
In the present embodiment, from the same viewpoint as in the sixth embodiment, the arrangement of the respective conductor layers is set so that both the input / output conductor layers 32-1 and 32-2 are located on the front side. If an input / output conductor layer is also provided in the second resonator R8 from the top in the figure, it is preferable to modify the arrangement of the inner conductor layer and the ground portion according to the sixth embodiment.

(2) Embodiment Group Having Lumped Element for Forming Attenuation Pole Ninth Embodiment FIG. 9 shows the structure of the microwave filter according to the ninth embodiment of the present invention. This embodiment has a structure in which two dielectric substrates 22-1 and 22-2 whose surfaces are selectively made conductive are stacked. Dielectric substrate 2
The upper surface of 2-1 and the lower surface of the dielectric substrate 22-2 are entirely made into a conductor, whereby outer conductor layers 24-1 and 24-2 are formed. Outer conductor layers 24-1 and 2
Between 4-2, short-circuit is performed by the side surface short-circuit conductor layers 26-1 and 26-2 formed on the end faces of the dielectric substrates 22-1 and 22-2. On the other hand, two inner conductor layers are formed substantially parallel to each other on the upper surface of the dielectric substrate 22-2. Shorted to 2. Therefore, in the structure of this embodiment, a plurality of inner conductor layers are formed on the same dielectric substrate 22-2, and each of these inner conductor layers is 1 /
Comprising four-wavelength dielectric resonators R7 and R8,
These resonators R7 and R8 are the outer conductor layers 24-1 and 2
4-2 is shared with the prior art structure described above.

However, there are some differences between this embodiment and the above-mentioned conventional technique. First, in the present embodiment, the function of coupling between the resonators R7 and R8 is realized by capacitive coupling between the end faces of the inner conductor layer. That is, the resonator R
The inner conductor layers 7 and R8 are respectively composed of front side (input / output side) portions 28-1a and 28-2a and rear side (short circuit end side) portions 28-1b and 28-2b in the figure. . Furthermore, the gap 39a between the former is the gap 3 between the latter.
Since it is set to be wider than 9b, the resonator R
Capacitive coupling occurs between 7 and R8. Therefore, in this embodiment, the inter-cavity coupling conductor layer 18 in the prior art is used.
And since the dielectric substrate 10-2 for it becomes unnecessary,
It is possible to simplify the structure, reduce the price, and reduce the size and thickness.

Further, the inner conductor layers 28-1a and 28-2a
By making the width of the resonators smaller than the width of the inner conductor layers 28-1b and 28-2b, the relationship of the gap 39a <the gap 39b is realized. Therefore, the lengths of the resonators R7 and R8 are equal to 1 of the resonance wavelength. It becomes shorter than the length corresponding to / 4, which makes it even smaller. That is, the inner conductor layers 28-1a and 28-
Since the stepwise change of the characteristic impedance is intentionally generated at the joint between the 2a and the inner conductor layers 28-1b and 28-2b, the line length is shortened by the so-called step impedance effect. At that time, the width of the inner conductor layer and its difference are set within a range in which an increase in reflection loss due to the step impedance can be allowed. Note that the gap 39a
Other structures may be used as long as capacitive coupling due to the above occurs.

Second, in the present embodiment, the input / output conductor layer 3
2-1 and 32-2 are provided on the same plane as the inner conductor layer, and through the capacitive elements 33-1 and 33-2 related to the end-to-end capacitive coupling between the input / output conductor layer and the inner conductor layer, the resonators R7 and R8
Is connected to the outside. Therefore, in this embodiment, since the dielectric substrate 10-4 for the input / output conductor layer in the prior art is not required, the structure can be simplified, the cost can be reduced, and the size and the thickness can be reduced. Note that this effect and the following effect can be realized by using an inductive element instead of the capacitive elements 33-1 and 33-2.

Thirdly, in this embodiment, the input / output conductor layer 3
A planar and lumped constant type inductive element 42 is provided between 2-1 and 32-2. The inductive element 42 includes an input / output conductor layer 32-1, a capacitive element 33-1 and a resonator R.
7. The electromagnetic field propagation path (second path) independent of the electromagnetic field propagation path (first path) reaching the input / output conductor layer 32-2 via the gap 7, the gap 39a, the resonator R8, and the capacitive element 33-2. Are offered. In this way, the input / output conductor layers 32-1 and 32
-2 and the input / output circuit having the inductive element 42 are formed on a single plane, thereby eliminating the need to maintain a space between the input / output conductor layers. It becomes small. Furthermore, a phase difference occurs between the signals that have passed through the above two electromagnetic field propagation paths due to the difference between the two paths, so in the present embodiment, an attenuation pole is formed in the immediate vicinity of the pass band without the addition of an external circuit. it can. The arrangement of the attenuation pole can be designed by selecting the gap 39a and the inductive element 42.

For example, at a frequency slightly lower than the resonance frequency of the resonators R7 and R8, the resonators R7 and R8 function as inductive elements, so that the phase shift generated in the first path is three. Capacitive elements (capacitive elements 33-1, 33-
2 and the gap 39a), which is a sum of −3π / 2 [rad] phase shift and the total of π [rad] phase shift caused by two inductive elements (resonators R7 and R8), −π. / 2
It becomes [rad]. On the contrary, at a frequency slightly higher than the resonance frequency of the resonators R7 and R8, since the resonators R7 and R8 also function as capacitive elements, the phase shift generated in the first path is five capacitive elements. Total of −5π / 2 [r
ad] = − π / 2 [rad]. On the other hand, the phase shift generated in the second path is π / 2 [ra by the inductive element 42.
d]. Therefore, at a frequency slightly lower or higher than the resonance frequency of the resonators R7 and R8, a difference of π [rad] occurs in the phase shift between both paths. As a result, the signal that has passed through the first path and the signal that has passed through the second path cancel each other out, so that these frequencies become attenuation poles. On the other hand, resonator R
At the resonance frequencies of 7 and R8 or in the vicinity thereof, the phase shift due to the resonators R7 and R8 can be ignored, so the phase shift generated in the first path is three capacitive elements (capacitive element 33).
-1, 33-2 and the gap 39a) total -3π / 2
[Rad], and the difference in phase shift between both paths becomes zero.
Therefore, a pass band with a small amount of attenuation can be realized. In addition,
The number of resonators is not limited to two.

Tenth Embodiment. FIG. 10 shows the first embodiment of the present invention.
The structure of the microwave filter which concerns on 0 embodiment is shown. This embodiment has a configuration in which the gap 39a in the ninth embodiment is replaced with a bridge conductor layer 44, and the inductive element 42 is replaced with a planar distributed constant type capacitive element 46. The bridge conductor layer 44 inductively couples between the resonators R7 and R8. Therefore, in the present embodiment, the input / output conductor layer 32
Since two electromagnetic wave propagation paths complementary to those in the ninth embodiment are formed between -1 and 32-2, it is possible to realize a BPF having the same effect.

(3) Embodiment group having distributed constant element for forming attenuation pole Tenth embodiment. FIG. 11 shows the structure of the microwave filter according to the eleventh embodiment of the present invention. In this embodiment, as in the ninth and tenth embodiments described above, quarter-wavelength dielectric resonators R7 and R8 sharing the outer conductor layers 24-1 and 24-2 are formed, and The inner conductor layers are located on the same plane. Further, the inner conductor layer of the resonator R7 has a wide conductor portion 28-1a and a narrow conductor portion 28-.
This embodiment is also similar to the ninth and tenth embodiments in that the inner conductor layer of the resonator R8 is formed of the wide conductor portion 28-2a and the narrow conductor portion 28-2b. There is. However, the present embodiment is different from the ninth and tenth embodiments in that the inductive element 42 and the capacitive element 46 are not used, and both the narrowed gap 39a and the bridge conductor layer 44 are adopted. There is. That is, in the above-described ninth and tenth embodiments, the input / output circuit 4
The inductive element 42 or the capacitive element 46 (more generally expressed as a lumped element) provided at 0 and the inter-resonator coupling realized by any one of the narrowed gap 39a and the bridge conductor layer 44 are used. Although the attenuation pole is formed in a lumped constant circuit manner, in the present embodiment, the input / output conductor layer 32-1 is used.
Attenuation pole is provided by a resonator (more specifically, a mode change along its extension direction or line length direction) related to inter-end surface side coupling between inner conductor layers by stopping lumped constant circuit coupling between 32-2. And forming a zero.

In order to make this point clearer, a comparative example shown in FIGS. 12 and 13 is assumed here. The comparative example of FIG. 12 has a configuration in which neither the narrowed gap 39a nor the bridge conductor layer 44 is adopted, while the comparative example of FIG. 13 uses the narrowed gap 39a, but the bridge conductor layer 44 is adopted. It is a configuration not done. So to speak, Fig. 11
The comparative example of FIG. 12 is obtained by adding the narrowed gap 39a to the comparative example of FIG. 12, and the embodiment of FIG. 11 is obtained by further adding the bridge conductor layer 44 to the comparative example of FIG. Therefore, by studying in the order of FIG. 12, FIG. 13, and FIG. It will become clear step by step.

The configurations of FIGS. 12, 13 and 11 are the same as those of the circuits shown in FIGS. 14, 15 and 16, respectively.
This can be equivalently expressed by the circuits shown in FIG. In these figures, the input / output conductor layer 32-
1 and 32-2 are terminal symbols, and capacitive elements 33-1 and 3
3-2 is a symbol of a capacitor, which is shown.
Further, the resonators R7 and R8 are the same as those shown in FIGS.
In FIG. 6, the shape of the inner conductor is shown, and in FIGS. 17, 18 and 19, it is shown by the symbol of the distributed constant line. Further, the side coupling generated between the end faces of the inner conductor layer whose shape is shown in FIGS. 14, 15 and 16 is
In FIGS. 17, 18 and 19, the distributed constant line R1
It is represented as 0. Since the transmission line R10 related to the end-to-end side coupling is a line having one end open and the other end shorted in the same manner as the resonators R7 and R8 are the lines having one end open and the other end short circuited, the transmission line R10 has the same value as that of the resonators R7 and R8. Functions as a quarter-wave dielectric resonator.

First, in the comparative example of FIG. 12, since the conductor widths of the inner conductor layers 28-1 and 28-2 are constant and the distance between them is also constant, the resonators R7, R8, and R10 are not affected. No step impedance is generated. Also,
The dielectric layers of the resonator R10 are the dielectric substrates 22-1 and 22-2, like the resonators R7 and R8. Therefore, since the line length (electrical length) of the resonator R10 is equal to that of the resonators R7 and R8, the resonators R7, R8, and R10 all resonate at the same frequency. That is, when the resonators R7 and R8 are resonating, the resonator R10
Also resonates, so that signal propagation from one input / output conductor layer to the other input / output conductor layer is blocked. Since the even mode propagation is dominant in the resonators R7 and R8 excited by the current source between the inner conductor layer and the outer conductor layer, its characteristic impedance can be regarded as equal to the even mode impedance.
Further, the characteristic impedance of the resonator R10 that can be regarded as being equivalently excited by the current source between the inner conductors is a phase θ = π / 2 corresponding to the line length from the short-circuited end (that is, ¼ wavelength).
2ZeZo / (Ze-Z) obtained by substituting into the formula of the image impedance between the side-coupled strip conductors.
o) (where Ze and Zo are even-mode and odd-mode impedances in side coupling, respectively).

Next, also in the comparative example of FIG. 13, the characteristic impedance of the resonators R7 and R8 is given as an even mode impedance. However, since the conductor widths of the inner conductor layers of the resonators R7 and R8 are changed stepwise, both of the portions (28-1a and 28-1a) having a wide conductor width of the inner conductor layer are provided.
-2a) and the narrow portion (28-1b and 28-2b) have different values. Also, the inner conductor layers 28-1a and 28-
Since the distance between 1b and the inner conductor layers 28-2a and 28-2b also changes stepwise, the characteristic impedance of the resonator R10 is 2ZaeZao / (Z at the gap 39a.
ae-Zao), 2ZbeZbo at the gap 39b.
/ (Zbe-Zbo) (where Zae and Zbe are even mode impedances in side coupling of each part, Za
o and Zbo are odd mode impedances in side coupling of each part). When the step impedance is generated in the resonators R7, R8, and R10 in this way, the actual size of the inner conductor becomes shorter than that of the comparative example of FIG. 12, and the longitudinal size of the entire microwave filter is also reduced. Furthermore, it is known that the even-mode and odd-mode impedances associated with side coupling decrease and increase with increasing edge spacing, respectively. Considering that the expression (above) giving the characteristic impedance of the resonator R10 is modified to be 2 / (1 / Zo-1 / Ze), the odd mode becomes more dominant in the gap 39a, It can be considered that the even mode becomes more dominant in the portion 39b. In other words, the gap 39
The coupling between the resonators R7 and R8 in the portion a is the gap 39.
The characteristic of capacitive coupling is stronger than that in part b.

As described above, the balance of the electromagnetic field between the resonators R7 and R8 is disturbed along the extension direction thereof, so that in the comparative example of FIG. 13 (and thus also in the embodiment of FIG. 11 described later), The resonators R7 and R8 can be coupled to each other through the capacitive coupling realized by the narrowed gap 39a. In other words, the effect of the above step impedance is not limited to the reduction of the substantial length of the line, and has the effect of shifting the substantial electrical length of the resonator R10, that is, the resonant frequency from the actual electrical length of the resonators R7 and R8. There is. Therefore, the comparative example of FIG. 13 (and FIG.
1) is a bandpass filter in which the resonance frequencies of the resonators R7 and R8 are the pass frequencies (or zero points) and the resonance frequency of the resonator R10 is the stop frequency (or the poles). Furthermore, the degree of capacitive coupling in the gap 39a is determined by the dielectric constants of the dielectric substrates 22-1 and 22-2 as well as the ratio of the size of the gap 39a to the size of the gap 39b (both including length and width). Can be set. Therefore, the pass frequency and the stop frequency can be determined by the shape and mutual spacing of the inner conductor layers of the resonators R7 and R8. Also, the ninth and first
It is not necessary to use the input-output coupling element as in the 0 embodiment, and therefore the structure is simpler.

However, in the comparative example of FIG. 12, two types of characteristics, the pass frequency and the stop frequency, have to be set by a single element, that is, the ratio of the size of the gap 39a and the size of the gap 39b. Has a relatively low degree of freedom. In the embodiment of FIG. 11, in order to improve this problem, the bridge conductor layer 44 is provided in the gap 39b, that is, at a position near the short-circuit end. The transmission line composed of the bridge conductor layer 44 and the outer conductor layers 24-1 and 24-2 is a resonator R7.
Since the mirror image symmetry plane of R8, that is, the plane at the same distance from the inner conductor center line of the resonator R7 and the inner conductor center line of the resonator R8, is crossed, a transmission line whose one end is open from the resonators R7 and R8 is formed. appear. In FIG. 19, this transmission line is
It is expressed as 11. Further, since the bridge conductor layer 44 short-circuits the inner conductor layers 28-1b and 28-2b, the bridge conductor layer 44 looks like a short-circuited transmission line from the resonator R10. FIG.
In the above, this transmission line is represented by R12. The connection position between the transmission lines R11 and R12 and the resonators R7, R8 and R10, the conductor width of the transmission lines R11 and R12,
When the line lengths of the transmission lines R11 and R12 change, the resonance frequency of the resonator R10 especially changes accordingly. Therefore, according to the present embodiment, the pass frequency and the stop frequency can be variably set also by designing the position and size of the bridge conductor layer 44.

As described above, according to this embodiment, the passing frequency and the blocking frequency are set to the gap 39a and the bridge conductor layer 44.
Can be changed in the design. Therefore, it is possible to design such that the stop frequency is provided in the vicinity of the pass frequency, and the width of the pass band can be designed relatively freely. Further, strictly speaking, when the position of the bridge conductor layer 44 is changed, the resonance frequencies of the resonators R7 and R8 also change, but since the change is small and can be practically ignored, the resonance frequency of the resonator R10 is designed. At this time, the change in the passing frequency due to the change in the position of the bridge conductor layer 44 and the like can be ignored, and therefore the design is easy. It should be noted that in the present embodiment as well, it is possible to obtain advantages such as facilitation of manufacturing by forming a planar circuit. Modifications such as application to coupling between three or more resonators and replacement of side short-circuit conductors with through holes are also possible.

Twelfth embodiment. FIG. 20 shows the first embodiment of the present invention.
The structure of the microwave filter which concerns on 2nd Embodiment is shown. In this embodiment, two half-wave dielectric resonators R3 and R are used.
It has a structure in which the same improvement as in the eleventh embodiment is applied to the microwave filter in which the four filters are combined. First, the inner conductor layer of the resonator R3 has a shape in which a wide conductor width portion 28-1a is added to both ends of a narrow conductor width portion 28-1b, and the inner conductor layer of the resonator R4 has a conductor width of Narrow part 28-2
It has a shape in which a wide conductor width portion 28-2a is added to both ends of b. As a result, a narrowed gap 39a is formed. Further, a portion or gap 39 isolated from the open end.
Two bridge conductor layers 44 are provided in b. Therefore, also in the present embodiment, a bandpass filter having the same advantages as the eleventh embodiment can be obtained. In addition, since the resonators R3 and R4 in the present embodiment are 1/2 wavelength resonators, it is possible to realize a filter having a sharp cutoff characteristic as compared with the eleventh embodiment. However, in terms of size, the eleventh embodiment is smaller. Note that various modifications can be made to this embodiment as in the eleventh embodiment.

Thirteenth Embodiment. FIG. 21 shows the first embodiment of the present invention.
The structure of the microwave filter which concerns on 3rd Embodiment is shown. This embodiment also has two half-wavelength dielectric resonators R3 and R.
It has a structure in which the same improvement as in the eleventh embodiment is applied to the microwave filter in which the four filters are combined. However, unlike the twelfth embodiment, the narrowed gap 39a is formed only at one end, and the number of the bridge conductor layers 44 is one. Even in this embodiment, there is a difference in degree,
The same effect as the eleventh embodiment can be obtained.

(4) Embodiment group having coupling windows for coupling input / output / inner conductor layers Fourteenth embodiment. FIG. 22 shows the structure of the microwave filter according to the fourteenth embodiment of the present invention. In this embodiment,
Four dielectric substrates 22 whose surfaces are selectively made conductive
It has a structure in which -1 to 22-4 are laminated. The upper surface of the third dielectric substrate 22-3 and the lower surface of the fourth dielectric substrate 22-4 from the top in the figure are almost entirely made into a conductor, whereby the outer conductor layer 24-2 and 24-3 is formed. Furthermore, the dielectric substrates 22-3 and 22-4
Side surface short-circuit conductor layers 26-3 and 26-4 that short-circuit the outer conductor layers 24-2 and 24-3 are formed on the end surface of the.
On the other hand, two inner conductor layers 28-1 and 28-2 are formed side by side on the upper surface of the dielectric substrate 22-4, and one end of the inner conductor layers 28-1 and 28-2 is laterally short-circuited via the ground portions 36-1 and 36-2. It is short-circuited to the conductor layer 26-4. As described above, in this embodiment, the two triplate lines R3 and R4, which are grounded at one end and share the outer conductor layers 24-2 and 24-3, are connected to the pair of dielectric substrates 22-3 and 22-4. It has a configuration realized by using it.

An input / output circuit 40 is formed above the outer conductor layer 24-2. First, a pair of input / output conductor layers 32-1 and 32-2 and a strip conductor layer 48 for short-circuiting these are formed on the upper surface of the second dielectric substrate 22-2 from the top in the figure, These together form a crank shape. The bent portion of this crank is located at a position just opposite to the inner conductor layers 28-1 and 28-2. Coupling windows 30-1 and 30-2 are formed on the upper surface of the outer conductor layer 24-2 at the portions on the line connecting the bent portion and the inner conductor layers 28-1 and 28-2, respectively. . Furthermore, the first dielectric substrate 2 from the top in the figure
The entire upper surface of 2-1 is made into a conductor, and thereby the outer conductor layer 24-1 is formed. In addition, the input / output conductor layer 3 of the end faces of the dielectric substrates 22-1 and 22-2 is used.
The outer conductor layer 2 is provided at a position sufficiently distant from 2-1 and 32-2.
Side short-circuit conductor layer 26-1 for short-circuiting 4-1 and 24-2
And 26-2 are formed. As described above, in this embodiment, the input / output circuit 40 that is coupled to the triplate lines R3 and R4 through the capacitive coupling windows 30-1 and 30-2 is realized as a triplate line.

In this embodiment, the triplate line R3 is used.
And a BRF that utilizes the 1/2 wavelength resonance of R4. That is, the input / output conductor layer 3 is connected through the coupling windows 30-1 and 30-2.
Since 2-1 and 32-2 are coupled to the inner conductor layers 28-1 and 28-2, 1/2 wavelength resonance of the triplate lines R3 and R4 in the signal input from one input / output conductor layer A signal having a frequency in the vicinity of the frequency is coupled to the coupling window 30.
-1 and 30-2 through inner conductor layers 28-1 and 28-2
To the other input / output conductor layer.
At other frequencies, the signal input from one input / output conductor layer is supplied to the other input / output conductor layer via the strip conductor layer 48. Therefore, in this embodiment, a ½ wavelength coupled resonance type BRF is obtained.

This embodiment and the above-mentioned conventional technique are common in that a plurality of inner conductor layers are formed on the same plane and that a dielectric substrate for input / output conductor layers is used. On the other hand,
In this embodiment, since the input / output conductor layers are short-circuited by the strip conductor layer 48, it is not necessary to secure a space between the input / output conductor layers, and the inter-resonator coupling conductor layer and this conductor are formed. Since no dielectric substrate is required, the number of dielectric substrates is smaller than before and a thin and compact microwave filter can be obtained. Furthermore, the dielectric substrate 22-1
And 22-2 to the dielectric substrates 22-3 and 22-
By making the dielectric constant lower than 4, the input / output conductor layers 32-1 and 32-2 can be formed without causing a problem in characteristics.
In addition, the width of the strip conductor layer 48 can be widened to facilitate the formation of the conductor pattern. That is, since the characteristic impedance of the external circuit (for example, 50Ω) is generally higher than the impedance of the resonator, the input / output circuit 40 has a higher dielectric constant than the dielectric substrates 22-3 and 22-4 on the resonator side.
By lowering the dielectric constants of the dielectric substrates 22-1 and 22-2 on the side, the widths of the input / output conductor layers 32-1 and 32-2 and the strip conductor layer 48 can be increased, and the formation thereof can be facilitated. Instead of the strip conductor layer 48, a planar inductor or capacitor may be used.

Fifteenth Embodiment. FIG. 23 shows a first example of the present invention.
The structure of the microwave filter which concerns on 5th Embodiment is shown. In this embodiment, the input / output circuit 40 is realized as a microstrip line instead of a triplate line, and the dielectric chips 50-1 and 50 are provided on the bent portion of the crank so as to face the coupling windows 30-1 and 30-2. -2 is different from the fourteenth embodiment. In addition,
Since the input / output circuit 40 is a microstrip line, the suffix of the reference numeral of each member in the drawing is 1 in contrast to that in FIG.
Deviated. According to this embodiment, the same effect as that of the fourteenth embodiment can be obtained, and the coupling by the coupling windows 30-1 and 30-2 can be adjusted by the dielectric chips 50-1 and 50-2. The effect is that it can be adjusted easily and easily. When using an inductive coupling window, a magnetic chip may be used instead of the dielectric chip. The dielectric constants of the dielectric chips 50-1 and 50-2 are preferably higher than that of the dielectric substrate 22-1. If the dielectric substrate provided with the recess for accommodating the dielectric chip or the magnetic chip is arranged so as to cover the input / output circuit 40, it can be transformed into a triplate line.

(5) Embodiment group having lumped element for forming attenuation pole between input / output / inner conductor layer Sixteenth embodiment. FIG. 24 shows the structure of the microwave filter according to the sixteenth embodiment of the present invention. In this embodiment,
The outer conductor layer 24 covers almost the entire lower surface of the dielectric substrate 22.
, The inner conductor layer 28, the ground portion 36, the input / output conductor layers 32-1 and 32-2, the strip conductor layer 48, and the inductive element 54 on the upper surface, and the side short-circuit conductor layer 26 on the end surface on the back side in the drawing. , Respectively. The inner conductor layer 28 and the outer conductor layer 24 together with the dielectric substrate 22 form a microstrip line R7. Further, one end of the inner conductor layer 28 is short-circuited to the outer conductor layer 24 via the ground portion 36 and the side surface short-circuit conductor layer 26, and the other end is connected to the strip conductor layer 48 via the inductive element 54. The strip conductor layer 48 is short-circuited to the input / output conductor layers 32-1 and 32-2 and constitutes a part of the external line.

The microstrip line R7 composed of the inner conductor layer 28 and the outer conductor layer 24 has a certain frequency f ₁
At 1/4 wavelength. At this frequency,
From the strip conductor layer 48 to the quarter-wave dielectric resonator R7
The impedance when viewed from the side becomes infinite, so that a signal input from one of the input / output conductor layers passes to the other input / output conductor layer. In addition, a certain frequency f ₂ (f ₁ ≠ f ₂ , | f ₁ −f ₂ | <ε, which is slightly higher than the resonance frequency f ₁ ,
At (ε: predetermined minute value), since the resonator R7 is equivalent to the electrostatic capacitance, it resonates in series with the inductive element 54, and the signal is supplied to the resonator R7 side. Thus, in the present embodiment,
A BRF in which the frequency f ₁ is included in the pass band and the frequency f ₂ is included in the stop band is realized. Further, since the difference between the resonance frequency f ₁ of the resonator R7 and the resonance frequency f ₂ of the resonator including the inductive element 54 is very small, the zero point, that is, the passband with a small attenuation determined by the resonator R7, can be obtained. An attenuation pole determined by the second resonator can be arranged in the vicinity (see FIG. 25).

Further, in this embodiment, the second resonator is constructed by using the resonator R7 as one of its constituent elements in addition to the resonator R7. Furthermore, the second
The resonator is a lumped constant circuit. Therefore, as compared with the prior art, the number of inner conductor layers is small, and the inner conductor layer, the input / output conductor layer, and the input / output inner conductor interlayer coupling element can be formed on the same plane, so that the number of dielectric layers is small. Further, since the input / output conductor layers 32-1 and 32-2 are directly connected by the strip conductor layer 48, it is not necessary to secure the space as in the prior art. In addition, it can be realized only by selectively making the surface of the dielectric substrate 22 a conductor, and substrate lamination is not necessary. As a result, the microwave filter is smaller in size, thinner than the above-described respective embodiments, and less expensive than conventional ones. The inductive element 54 may be realized by a chip inductor or the like.
Also, the present embodiment can be applied to 1/2 wavelength resonance instead of 1/4 wavelength resonance.

In designing the microwave filter according to this embodiment, first, the resonance frequencies f ₁ and f ₂ are determined according to the frequencies of the zero point and the pole to be realized. Next, the resonance frequencies f ₁ and f ₂ are

Substituting into the right side of Z ₁ <π ² L (f ₂ −f ₁ ), the characteristic impedance Z _{1 of the} line relating to the resonator R 7 and the inductance L of the inductive element 54 are determined so that this equation holds. To do. Furthermore, as the required specification values of the thus obtained resonant frequency f ₁ and the characteristic impedance Z ₁ is realized, the dielectric constant of the dielectric substrate 22,
The thickness, the width of the inner conductor layer 28, the line length D, etc. are determined. In the case of this embodiment, the line length D is slightly longer than 1/4 of the wavelength at the frequency f ₂ .

This equation is an equation obtained as follows. First, the phase conditions at the resonance frequencies f ₁ and f ₂ are expressed by the following equations, respectively.

[6] _{β 1 D = π / 2 ...} 1/4 -wavelength resonant _{j2πf 2 L + jZ 1 tan (} β 2 D) = 0 ... given by the series resonance. In the above equation, β ₁ and β ₂ are resonance frequencies f ₁
And the wave numbers at f ₂ . If Δf = f ₂ −f ₁ is sufficiently small, it can be considered that Δβ = β ₂ −β ₁ is sufficiently small. Therefore, approximately,

## EQU7 ## tan (β ₂ D) = − 1 / Δβ D holds. By substituting this into the above formula for series resonance and transforming it,

## EQU8 ## Z ₁ = π ² f ₂ L is obtained. Usually, in terms of the area of the dielectric substrate 22 and the formation of a conductor pattern on the dielectric substrate 22, the inductance L has a feasible upper limit. Therefore, this equation is an inequality

## EQU9 ## Convert to Z ₁ <π ² f ₂ L. As long as this constraint expression is satisfied, the inductance L and the characteristic impedance Z ₁ can be realized.

Seventeenth Embodiment. FIG. 26 shows the first aspect of the present invention.
The structure of the microwave filter which concerns on 7th Embodiment is shown. In the present embodiment, a chip-shaped capacitive element 33 is used instead of the inductive element 54 in the 16th embodiment.
Series resonance of the resonator R7 and the capacitor element 33, as shown in FIG. 27, resulting resonator R7 follow lower than the resonance frequency f ₁ of the resonator R7 is at a frequency f ₁ that exhibits inducible. In the case of this embodiment, the equation showing the phase condition is

## EQU10 ## β ₁ D = π / 2 ... 1/4 wavelength resonance 1 / j2πf ₂ C + jZ ₁ tan (β ₂ D) = 0 ... Series resonance. If this equation is transformed by the same method as in the sixteenth embodiment, the following inequality becomes

## EQU11 ## Z ₁ > (f ₁ −f ₂ ) / 4Cf ₂ ² is obtained. As long as this constraint expression is satisfied, the capacitance C and the characteristic impedance Z ₁ can be realized. The capacitive element 33 may be replaced with an element having a conductor pattern.

(6) Embodiment Group Having Loss Reduction Grooves 18th Embodiment. FIG. 28 shows the structure of the microwave filter according to the eighteenth embodiment of the present invention. The structure of this figure is applied to the embodiment having the triplate line structure among the above-described embodiments, particularly to the outer conductor layer. Reference numerals 24-1 and 24-2 are used for the outer conductor layer, and reference numeral 28 is used for the inner conductor layer.
, Respectively, but this is for the purpose of explanation and does not limit the scope of application. This embodiment is characterized in that the grooves 58-1 and 58-2 are provided in the outer conductor layers 24-1 and 24-2 at positions facing the inner conductor layer 28, particularly the central portion in the width direction thereof. Is. As shown in FIGS. 29 and 30, compared with the current distribution when the grooves 58-1 and 58-2 are not provided (FIG. 29), when the grooves 58-1 and 58-2 are provided as in the present embodiment. The current distribution (see FIG. 30) tends to concentrate more in the vicinity of the central portion of the inner conductor layer 28 in the width direction. This is because the grooves 58-1 and 58-2 are provided in a portion facing the widthwise central portion of the inner conductor layer 28, so that the inner conductor layer / outer conductor layer interval in this portion is reduced, and therefore the strength of the electromagnetic field is reduced. Is higher.

Since the current is concentrated near the central portion in this manner, the line loss can be reduced in this embodiment. Here, when the electric resistivity of the inner conductor layer 28 is represented by ρ and the current distribution function in the width direction is represented by I (x), the position x in the width direction is represented.
The line loss per unit width at is ρI (x) ² and therefore the line loss is expressed by the sum in the width direction ∫ρI (x) ² dx. This value becomes the minimum when dI (x) / dx = 0. Therefore, the line loss can be reduced by increasing the degree of concentration of the current in the vicinity of the central portion in the width direction to bring the current distribution closer to a flat distribution as in the present embodiment. Furthermore, the groove 5
By designing the size and shape of 8-1 and 58-2, the degree of loss reduction and the degree of processing can be balanced. Further, even if the impedance changes due to the arrangement of the grooves 58-1 and 58-2, such a change is small and can be ignored, and in some cases, can be included in the line design.

19th to 21st Embodiments. 31 to 33
The structures of microwave filters according to the nineteenth to twenty-first embodiments of the present invention are shown in FIGS. In FIG. 31, the groove 58-1
2 and 58-2 are provided. Thus, the number of grooves can be changed according to the dielectric constant of the dielectric substrate 22 and the like. Further, in FIG. 32, the grooves 58-1 and 58 are provided.
-2 is a semi-circular cross section, which can reduce the wear of the processing tool for forming the groove, for example, the wear of the blade. Further, as shown in FIG. 33, the groove 58 may be provided in the outer conductor layer 56 of the microstrip line instead of the triplate line.
In addition, in these embodiments, the same effect as that of the eighteenth embodiment can be obtained.

(7) Addendum Regarding the possibility of combining the technical means adopted in each of the embodiments, detailed description is omitted for simplification of the description. It should be understood that combinations are possible and that the effects associated with the combinations occur. In addition, in this application, terms such as “design” and “setting” are used for matters related to the structure and characteristics of the resonator and the filter, but this means that the matters are so-called design matters and do not contribute to the inventive step. It is not meant to be mentioned. That is, it should be noted that in the technical field relating to a transmission circuit such as a resonator and a filter, these terms are commonly used in the sense of "determining the essential constituent elements" of the transmission circuit.

Although a flat microwave filter using a dielectric substrate as the dielectric layer is taken as an example, the present invention is strictly a flat plate filter such as one using a dielectric film as the dielectric layer. It can also be realized as a microwave filter that cannot be said (but can be said to be approximately flat).
Further, although the triplate line or the microstrip line is taken as an example, the present invention can be realized as a microwave filter having a flat plate-shaped or film-shaped transmission line structure of another form unless otherwise specified. Further, unless specifically stated otherwise, the embodiments relating to the triplate line may be modified into microstrip lines, and the embodiments relating to the microstrip line may be modified into triplate lines. Also, the first resonator in the single filter is a microstrip line,
A line structure can be mixed, such as a triplate line for the second resonator. Further, unless otherwise specified, the half-wave resonator can be changed to the quarter-wave resonator or vice versa. In addition, although BPF and BRF are taken as an example, filters having other characteristics can be constructed according to the present invention.

Further, the structure in which various conductor layers are adhered and formed on the surface of the dielectric substrate has been exemplified, but various conventionally known methods can be used as the method of adhesion and formation. Further, of the conductor layers, the conductor layers to be sandwiched between the two dielectric substrates at the time of stacking may be adhered and formed on either of the two dielectric substrates. Further, if structurally and characteristically possible, the conductor layer may be realized by a conductor foil, a surface-conducting plastic film or the like. Although the side short-circuit conductor layer is shown as a means for short-circuiting the outer conductor layers, it may be replaced with a through hole that short-circuits the outer conductor layers. In such a configuration, the same effect as when the end face short-circuit conductor is used can be obtained without the end face conductor forming process. This through hole may short-circuit all the outer conductor layers, but for example, a through hole that short-circuits the first outer conductor layer and the second outer conductor layer,
The through hole that short-circuits the second outer conductor layer and the third outer conductor layer may be another through hole. The number of through holes is appropriately determined. In addition, the material of the dielectric and conductor,
Although the frequency band to be used, the specific values of the thickness / dimensions of each layer, etc. have not been mentioned, those skilled in the art can easily implement the present invention based on the description of the present application. In addition, since the coupling window may be a preferably flat means for partially and selectively disturbing the electromagnetic shield by the outer conductor layer, the coupling window is not limited to the small hole formed in the outer conductor layer.

[0097]

As described above, according to the first aspect of the present invention, at least two inner conductor layers formed on different surfaces and at least two inner conductor layers that are combined with any of the inner conductor layers are formed. Of the input / output conductor layer, and an outer conductor layer that electromagnetically shields the inner conductor layer and the input / output conductor layer via a dielectric layer, and further disturbs the electromagnetic shield to electromagnetically couple the inner conductor layer. Since the coupling window to be formed is formed in the outer conductor layer, it is not necessary to separately provide a conductor layer for joining the inner conductor layers. As a result, the structure is simpler and the processing is simpler than in the past, so that the cost of the microwave filter can be reduced. Furthermore,
Since the inner conductor layers form a layer in the thickness direction of the transmission line structure with the outer conductor layer and the dielectric layer separated from each other, even if the number of the inner conductor layers is increased, the thickness is increased but the area is not so large. At the same time, it is possible to electromagnetically shield the input and output conductor layers by the outer conductor layer, and prevent the direct electromagnetic coupling,
As a result, it is possible to prevent the dimensional enlargement that has occurred due to the securing of the space between the input and output conductor layers. As a result, a compact and low-priced microwave filter can be realized as compared with the conventional one. Further, since the position of the input / output conductor layer can be set without considering the positional relationship with other input / output conductor layers, a microwave filter having a higher degree of freedom in design than the conventional one can be obtained.

According to the second structure of the present invention, since the input / output conductor layer and the inner conductor layer are capacitively coupled between the end faces in the first structure, the input / output conductor layer and the inner conductor layer are the same. Since it can be arranged on a plane, a smaller, thinner, and cheaper microwave filter can be realized.

According to the third structure of the present invention, the first
In the above configuration, the inner conductor layer and the input / output conductor layer are arranged so that the input / output conductor layers are close to each other. By a simple method, a microwave filter with high integration capable of performing input and output from the same end face can be inexpensively and easily realized, and the external circuit near the input / output conductor layer can be miniaturized. In addition to these configurations, the same effect as the first configuration can be obtained.

According to the fourth structure of the present invention, in designing the microwave filter according to the first structure, the relative positional relationship between the inner conductor layer and the coupling window is set to one of the input and output conductor layers. Since multiple electromagnetic field propagation paths from one to another coexist at the resonance frequency or in the vicinity thereof, it is possible to design or set the characteristics of the microwave filter in more detail, and design freedom. The degree improves. In particular, according to the fifth configuration of the present invention, in the fourth configuration, the electrical length between the electromagnetic field propagation paths is set so that the phase difference of π or −π [rad] is exhibited at the required attenuation pole frequency. Since the difference is set, a pole or a zero point is provided at the frequency where a pole or zero point is not normally provided without the addition of an external circuit, for example, in the pass band or in the vicinity of the pass band, is provided. It will be possible. Therefore, the arrangement of poles or zeros can be freely set according to the positions of the inner conductor layer and the coupling window,
The degree of freedom in designing the characteristics of the microwave filter is improved, and the size and cost can be reduced by eliminating the external circuit.

According to the sixth structure of the present invention, in designing the microwave filter according to the first structure, the coupling window is made so that the change of the coupling strength with respect to the change of the frequency or the wavelength becomes relatively steep. Since the position is set, a microwave filter having a good cutoff characteristic can be obtained by a simple method.

According to the seventh configuration of the present invention, in designing the microwave filter according to the first configuration, the inner conductor layers are provided with the first coupling window on one side and the second coupling window on the other side. Since the positions, shapes and dimensions of the first and second coupling windows are set so as to be electromagnetically coupled through the microwave filter, the microwave filter can be designed by a simple method of designing the coupling between inner conductor layers by designing the coupling window arrangement and the like. Can be set or determined, which can contribute to improvement in design freedom.

In particular, according to the eighth aspect of the present invention, the seventh
In the above configuration, the electromagnetic coupling by the first coupling window and the second coupling
So that the electromagnetic coupling by the coupling window of
Since the positions, shapes, and dimensions of the first and second coupling windows are set, stronger coupling can be realized as compared with the case where the inner conductor layers are coupled with a single coupling window, and the strength thereof can be increased by the first and second coupling windows.
It can be set by the position, shape and size of the coupling window.

According to the ninth aspect of the present invention, the seventh
In the above configuration, a first coupling window interlinking with the lines of electric force and a second coupling window interlocking with the lines of magnetic force are arranged, respectively, and the capacitive coupling and the second coupling by the first coupling window are arranged. Since the positions, shapes and dimensions of the first and second coupling windows are set so that a pole or a zero point is generated at a desired frequency by combination with the inductive coupling by the window, the characteristics of the microwave filter, particularly the pole, are determined by a simple method. Alternatively, the arrangement of zero points can be arbitrarily set.

According to the tenth structure of the present invention, the input and output conductors are formed on the same plane as the inner conductor layer by capacitively or inductively coupling the end faces of the inner conductor layer formed on the same plane. Since the layer is capacitively coupled or inductively coupled with the inner conductor layer, it is not necessary to provide a dielectric layer for disposing the inter-resonator coupling element, and a compact, simple and low-cost structure can be obtained. Further, the conductor layer coupling and the adjustment of the coupling strength can be realized by a simple means of changing the shape of the inner conductor layer.

According to the eleventh structure of the present invention, in the tenth structure, the input / output coupling element is provided so that capacitive coupling or inductive coupling occurs between the input / output conductor layers.
In addition to the effect similar to that of the configuration described above, the degree of freedom in designing the filter characteristics is improved, and since the input / output conductor layer and the inner conductor layer can be formed on the same plane, a dielectric material dedicated to the input / output conductor layer is provided. A compact, simple, low-cost structure is obtained without the need for layers.

According to the twelfth configuration of the present invention, in the eleventh configuration, one of the input / output conductor layer coupling mode and the coupling mode between the inner conductor layer and the input / output conductor layer is capacitive,
Since the other is made inductive, the same effect as the eleventh configuration can be obtained, and further, between the plural electromagnetic wave propagation paths provided in the eleventh configuration, at the required frequency, A difference in electrical length that causes a phase difference of π or −π [rad] can be imparted. As a result, poles (and zero points) can be provided without an external circuit, and the arrangement can be easily determined by designing each coupling element and input / output conductor layer.

According to the thirteenth structure of the present invention, the mode of the transmission line relating to the coupling between the end faces of the inner conductor layers is changed along the extension direction thereof. As a result of eliminating the elements and having a simple structure, a microwave filter that is easy to design and inexpensive can be obtained. Furthermore, by designing the mode change pattern of the transmission line related to the coupling between the end faces, it becomes possible to design the poles and zeros at arbitrary frequencies, and for example, a microwave filter having a characteristic that exhibits a large amount of attenuation in the vicinity of the passband It will be feasible.

Further, according to the fourteenth structure of the present invention, in the thirteenth structure, the space between the inner conductor layers is partially narrowed near the open end side, so that the inner conductor at that portion is narrowed. The mode of coupling between the end faces of the layers can be a mode in which the electric field is dominant. According to the fifteenth configuration of the present invention, in the fourteenth configuration, since the bridge conductor layer is provided at a site separated from the site related to the constriction, the inner conductor layers are connected to each other. The mode of the coupling between the end faces of the inner conductor layers can be made the mode in which the magnetic field is dominant. With any of these configurations, the mode of the transmission line related to the coupling between the end faces can be changed along the extension direction thereof, and this can be realized by a planar circuit. As a result, a smaller and thinner microwave filter can be obtained. In addition, when both the inner conductor layer spacing narrowing and the bridge conductor layer are adopted,
It is possible to realize an advantage that the degree of freedom in design, that is, the degree of freedom of design, which is not obtained by the configuration in which any one of them is adopted, that is, the pole and the zero point can be set independently of each other.

According to the sixteenth structure of the present invention, at least two inner conductor layers are formed on the same plane, and at least two input / output conductor layers and a space between them are formed on a plane different from these. Since the input / output coupling element is provided, and further the coupling window is provided in the input / output conductor layer and the outer conductor layer between the input / output coupling element and the inner conductor layer, the coupling window and the input / output are provided between the inner conductor layers. A coupled resonant filter coupled via an inter-coupling element is obtained. Since this input / output conductor layer and the input / output coupling element are arranged on the same plane, this filter is small, thin, and low in cost. Furthermore, since the input / output conductor layers are positively connected to each other and used for connecting the inner conductor layers, it is not necessary to secure a space between the input / output conductor layers, and the size and cost are further reduced. In addition, all of these can be realized by the technique of selectively forming the conductor on the surface of the substrate and the technique of lamination, and can be easily implemented, so that the cost is further reduced.

According to the seventeenth structure of the present invention, in the sixteenth structure, the dielectric constant of the dielectric layer used in the first transmission line composed of the inner conductor layer and the outer conductor layer is In comparison, the dielectric constant of the dielectric layer used in the second transmission line, which is composed of the input / output conductor layer, the input / output coupling element, and the outer conductor layer, is set to be low. The width of the conductor can be widened and the forming process can be facilitated. Furthermore, since the impedance design of the first and second transmission lines can be made independent, it becomes easy to design the second transmission line according to the impedance of the external circuit, for example.

According to the eighteenth structure of the present invention, in the fifteenth structure, since the chip-shaped dielectric or magnetic material is interposed in the input / output conductor layer / inner conductor interlayer coupling, the chip-shaped dielectric or magnetic material It becomes possible to adjust the strength of the bond and the like by adjusting the arrangement of the magnetic body.

According to the nineteenth structure of the present invention, the input / output conductor layer and the input / output inner conductor interlayer coupling element are formed on the same plane as the inner conductor layer, and the inner conductor layer is formed by the input / output inner conductor layer coupling element. Since the input and output conductor layers are inductively or capacitively coupled, only two planes in total, that is, the plane forming the inner conductor layer and the plane forming the outer conductor layer, in other words, the front and back of one dielectric layer Utilizing this, the first and second resonators can be realized. Furthermore, since the first resonator operates as a capacitance or an inductance at frequencies other than its resonance frequency f ₁ , the second resonator is connected to the first resonator (or its equivalent capacitance or inductance) in the input / output. It can be realized by a conductor interlayer coupling element (or an inductive coupling or an inductance or a capacitance related to the inductive coupling). Thus, the first
Since the resonator of is used as a part of the second resonator, the number of inner conductor layers is small, and the inner conductor layer, the input / output conductor layer, and the input / output inner conductor interlayer coupling element can be formed on the same plane. Since the number of dielectric layers is small and the input / output conductor layers may be directly connected by conductors, it is not necessary to secure the space between them, and further, it can be realized only by selectively making the substrate surface a conductor, and substrate lamination is not necessary. Therefore, it becomes a microwave filter that is smaller, thinner, and less expensive. In addition, since the difference between the resonance frequencies f ₁ and f ₂ of the _first and _second resonators is very small, according to this configuration, the second resonator is provided near the zero point or pole determined by the first resonator. It is possible to arrange a pole or a zero point determined by the resonator.

According to the twentieth and twenty-first configurations of the present invention, in designing the microwave filter according to the nineteenth configuration, first, the first configuration in the nineteenth configuration is performed according to the frequency of the pole or zero to be realized. And the resonance frequencies f ₁ and f ₂ of the second resonator are determined, and then the characteristic impedance Z ₁ of the first resonator and the inductance L or capacitance C of the input / output inner conductor interlayer coupling element are set as predetermined constraints. Since the formula is determined so as to satisfy the equation, it is possible to realize the liberation of the arrangement of the poles and zeros, the reduction of the attenuation in the pass band, and the steepness of the out-of-band attenuation by a relatively simple means.

According to the twenty-second structure of the present invention, in the first, tenth, thirteenth, sixteenth or nineteenth structure, a part of the portion of the outer conductor layer facing the inner conductor layer is Since the gap with the conductor layer is made narrower than the other parts, current concentration in the part of the inner conductor layer facing the recess of the outer conductor layer, and by this flattening of the width direction current distribution,
The loss in the line can be reduced.

[Brief description of drawings]

FIG. 1 is an exploded perspective view showing a configuration of a microwave filter according to a first embodiment of the present invention.

FIG. 2 is an exploded perspective view showing a configuration of a microwave filter according to a second embodiment of the present invention.

FIG. 3 is an exploded perspective view showing a configuration of a microwave filter according to a third embodiment of the present invention.

FIG. 4 is an exploded perspective view showing a configuration of a microwave filter according to a fourth embodiment of the present invention.

FIG. 5 is an exploded perspective view showing a configuration of a microwave filter according to a fifth embodiment of the present invention.

FIG. 6 is an exploded perspective view showing a configuration of a microwave filter according to a sixth embodiment of the present invention.

FIG. 7 is an exploded perspective view showing a configuration of a microwave filter according to a seventh embodiment of the present invention.

FIG. 8 is an exploded perspective view showing a configuration of a microwave filter according to an eighth embodiment of the present invention.

FIG. 9 is an exploded perspective view showing a configuration of a microwave filter according to a ninth embodiment of the present invention.

FIG. 10 is an exploded perspective view showing a configuration of a microwave filter according to a tenth embodiment of the present invention.

FIG. 11 is an exploded perspective view showing a structure of a microwave filter according to an eleventh embodiment of the present invention.

FIG. 12 is an exploded perspective view showing a structure of a microwave filter according to a first reference example.

FIG. 13 is an exploded perspective view showing a structure of a microwave filter according to a second reference example.

FIG. 14 is an inner conductor layer shape layout diagram showing an equivalent circuit configuration of the microwave filter according to the first reference example.

FIG. 15 is an inner conductor layer shape layout diagram showing an equivalent circuit configuration of a microwave filter according to a second reference example.

FIG. 16 is an inner conductor layer shape layout diagram showing an equivalent circuit configuration of a microwave filter according to an eleventh embodiment of the present invention.

FIG. 17 is a line relation diagram showing an equivalent circuit configuration of the microwave filter according to the first reference example.

FIG. 18 is a line relation diagram showing an equivalent circuit configuration of a microwave filter according to a second reference example.

FIG. 19 is a line relation diagram showing an equivalent circuit configuration of a microwave filter according to an eleventh embodiment of the present invention.

FIG. 20 is an exploded perspective view showing the structure of the microwave filter according to the twelfth embodiment of the present invention.

FIG. 21 is an exploded perspective view showing the structure of the microwave filter according to the thirteenth embodiment of the present invention.

FIG. 22 is an exploded perspective view showing the configuration of the microwave filter according to the fourteenth embodiment of the present invention.

FIG. 23 is an exploded perspective view showing the configuration of the microwave filter according to the fifteenth embodiment of the present invention.

FIG. 24 is an exploded perspective view showing the configuration of the microwave filter according to the sixteenth embodiment of the present invention.

FIG. 25 is a characteristic diagram showing the amount of attenuation near the stop band in this embodiment.

FIG. 26 is an exploded perspective view showing the configuration of the microwave filter according to the seventeenth embodiment of the present invention.

FIG. 27 is a characteristic diagram showing the amount of attenuation near the stop band in this embodiment.

FIG. 28 is an exploded perspective view showing the configuration of the microwave filter according to the eighteenth embodiment of the present invention.

FIG. 29 is a current distribution diagram in a cross section of a transmission line when there is no groove.

FIG. 30 is a current distribution diagram in a transmission line cross section when there is a groove.

FIG. 31 is an exploded perspective view showing the configuration of the microwave filter according to the nineteenth embodiment of the present invention.

FIG. 32 is an exploded perspective view showing the configuration of the microwave filter according to the twentieth embodiment of the present invention.

FIG. 33 is an exploded perspective view showing the configuration of the microwave filter according to the twenty-first embodiment of the present invention.

FIG. 34 is an exploded perspective view showing a configuration of a microwave filter according to a conventional technique.

[Explanation of symbols]

R3 to R6 1/2 wavelength dielectric resonator, R7 to R10
1/4 wavelength dielectric resonator, R11, R12 Transmission line related to bridge-shaped conductor, 22, 22-1 to 22-8 Dielectric substrate, 24-1 to 24-5 Outer conductor layer, 26, 26-
1-26-8 Side short-circuit conductor layer, 28-1 to 28-4, 2
8-1a, 28-1b, 28-2a, 28-2b Inner conductor layer, 30, 30-1 to 30-6 Inter-resonator coupling window, 3
2-1 and 32-2 input / output conductor layers, 33, 33-1 and 3
3-2, 46 capacitive element, 36, 36-1, 36-2
Grounding part, 38 Window connecting part, 39a, 39b Gap, 40
Input / output circuit, 42, 54 inductive element, 44 bridge conductor layer, 48 strip conductor layer, 50-1, 50-2
Dielectric chip, 58, 58-1, 58-2 groove.

Claims (22)

[Claims] 1. At least two inner conductor layers formed on different surfaces, at least two input / output conductor layers coupled to any one of the inner conductor layers, and between the inner conductor layers and the input / output conductor layers. An outer conductor layer for electromagnetic shielding has a flat plate-shaped or film-like transmission line structure laminated via a dielectric layer, and a coupling window for electromagnetically coupling the inner conductor layers by disturbing the electromagnetic shielding, A coupled resonance type microwave filter formed on the outer conductor layer. 2. The microwave filter according to claim 1, wherein the input / output conductor layer and the inner conductor layer to be coupled by the input / output conductor layer are coupled via end-to-end capacitive coupling. 3. The micro conductor according to claim 1, wherein the inner conductor layer and the input / output conductor layer are arranged such that the input / output conductor layers are arranged close to each other with the outer conductor layer interposed therebetween. Wave filter. 4. In designing the microwave filter according to claim 1, the relative positional relationship between the inner conductor layer and the coupling window is changed from one of the input / output conductor layers to another. A design method characterized in that a plurality of electromagnetic field propagation paths are set to coexist at or near the resonance frequency. 5. At the required attenuation pole frequency, π or −π.
The design method according to claim 4, wherein the difference in electrical length between the plurality of electromagnetic field propagation paths is set so as to exhibit a phase difference of [rad]. 6. In designing the microwave filter according to claim 1, the position of the coupling window is set so that the coupling strength by the coupling window changes relatively sharply with respect to a change in frequency or wavelength. A design method characterized by setting. 7. The microwave filter according to claim 1, wherein the inner conductor layers are electromagnetically coupled to each other through a first coupling window on the one hand and a second coupling window on the other hand. So that the positions, shapes and dimensions of the first and second coupling windows, which are the coupling windows, are set. 8. The positions, shapes and dimensions of the first and second coupling windows are set so that the electromagnetic coupling by the first coupling window and the electromagnetic coupling by the second coupling window emphasize each other. The design method according to claim 7, wherein the design method is set. 9. The first coupling window is provided at a position where it intersects with a line of electric force from one of the inner conductor layers of the at least two dielectric resonators to a line of the magnetic force line near the inner conductor layer. The second coupling windows are arranged at intersecting positions, and a combination of the capacitive coupling by the first coupling window and the inductive coupling by the second coupling window causes a pole or a zero point at a required frequency. 8. The designing method according to claim 7, wherein the positions, shapes and dimensions of the first and second coupling windows are set. 10. At least two inner conductor layers which are formed on the same plane and whose end faces are capacitively coupled or inductively coupled to each other, and which are formed on the same plane as the inner conductor layer and are capacitively coupled or inductively coupled to either of them. At least two input / output conductor layers and an outer conductor layer formed on a plane different from that of the inner conductor layer are laminated through a dielectric layer to form a flat plate-shaped or film-shaped transmission line structure. Coupled resonance type microwave filter. 11. The microwave filter according to claim 10, wherein an input / output coupling element for capacitively or inductively coupling the input / output conductor layers is formed on the same plane as the input / output conductor layers. 12. The input / output conductor layers are coupled in any one mode of capacitive coupling and inductive coupling, and the input / output conductor layers and the corresponding inner conductor layers are coupled in the other mode. The microwave filter according to claim 11, wherein the microwave filter is provided. 13. At least two said inner conductor layers formed on the same plane so that inter-end face side coupling is generated between the inner conductor layers, and one of them formed on the same plane as said inner conductor layer and a capacitance. At least two input / output conductor layers that are coupled or inductively coupled to each other and an outer conductor layer formed on a plane different from the inner conductor layer are laminated through a dielectric layer to form a flat plate or film transmission line structure. And, in order to cause signal propagation from one inner conductor layer related to the end-face side coupling to the other inner conductor layer, the mode of the end-face side coupling is changed along the extension direction thereof. Coupled resonance type microwave filter. 14. The inner conductor layer of the above-mentioned inner conductor layer is arranged such that an edge distance between the inner conductor layer and an adjacent inner conductor layer is relatively narrow at a portion close to the open end and relatively wide at a portion separated from the open end. 14. The microwave filter according to claim 13, wherein the conductor width is set. 15. The microwave filter according to claim 13 or 14, wherein the inner conductor layers are connected to each other by a bridge conductor layer at a portion separated from the open end. 16. At least two formed on the same plane
Number of inner conductor layers, at least two input / output conductor layers formed on a plane different from the inner conductor layers, and an input / output coupling element formed on the same plane as the input / output conductor layers and coupling between them And a flat plate shape in which an outer conductor layer formed on a plane different from that of the inner conductor layer and electromagnetically shielding between the input / output conductor layer and the input / output coupling element and the inner conductor layer is laminated via a dielectric layer. Alternatively, a coupling window having a film-like transmission line structure and electromagnetically coupling the input / output conductor layer and the inner conductor layer by disturbing the electromagnetic shield is formed in the outer conductor layer. A coupled resonance type microwave filter characterized by the following. 17. In designing the microwave filter according to claim 16, comparing with a dielectric constant of a dielectric layer used in the first transmission line composed of the inner conductor layer and the outer conductor layer, A design method characterized by lowering the dielectric constant of a dielectric layer used in the second transmission line composed of the input / output conductor layer, the input / output coupling element and the outer conductor layer. 18. A chip-shaped dielectric or magnetic material is arranged in the vicinity of the coupling window and the input / output conductor layer so as to intervene in electromagnetic coupling between the input / output conductor layer and the inner conductor layer. The microwave filter according to claim 16, wherein: 19. An inner conductor layer, an input / output conductor layer formed on the same plane as the inner conductor layer, and an inductive coupling or a capacitance formed on the same plane as the inner conductor layer and the input / output conductor layer. An input / output inner conductor inter-layer coupling element to be coupled, and an outer conductor layer formed on a plane different from the inner conductor layer are laminated via a dielectric layer to form a flat plate or film transmission line structure. In the line structure, a _first resonator composed of the inner conductor layer and the outer conductor layer and resonating at a frequency f ₁ and a frequency f ₂ (composed of the first resonator and the input / output inner conductor interlayer coupling element) However, f ₁ ≠ f ₂ , | f ₁ −f ₂
And a second resonator that resonates with | <ε, ε: a predetermined minute value). 20. In designing the microwave filter according to claim 19, first, the resonance frequency f _{1 of} the first and second resonators is determined according to the frequency of the pole or zero to be realized.
And f ₂ are determined, and the inductance L related to the inductive coupling due to the input / output inner conductor interlayer coupling element and the characteristic impedance Z ₁ of the first resonator are expressed by the following equation: Z ₁ <π ² L A design method characterized by defining | f ₁ −f ₂ |. 21. In designing the microwave filter according to claim 19, first, the resonance frequency f _{1 of} the first and second resonators is determined according to the frequency of the pole or zero to be realized.
And f ₂ are determined, and then the capacitance C related to capacitive coupling by the input / output inner conductor interlayer coupling element and the characteristic impedance Z ₁ of the first resonator are expressed by the following equation: Z ₁ > | A design method characterized in that it is defined so as to satisfy f ₁ −f ₂ | / 4Cf ₂ ² . 22. A part of a portion of the outer conductor layer facing the inner conductor layer is recessed so that a space between the outer conductor layer and the inner conductor layer is narrower than other portions. 1
The microwave filter according to 3, 16 or 19.