JPH09191700A - Control method of induction motor - Google Patents

Abstract

(57) Abstract: To reduce total loss generated in an induction motor from a low speed operation area to a high speed operation area of the induction motor. SOLUTION: A secondary current component calculated using the detected value of the primary current of the induction motor 2, an exciting current component calculated using either the command value of the exciting current or the detected value of the primary current, and an output. Based on the frequency command value, the induction motor 2
Controls the exciting current component that creates the interlinkage magnetic flux. [Effect] From the low speed operation area to the high speed operation area of the induction motor 2, the total loss generated in the induction motor 2 can be reduced and the power supply capacity can be reduced, so that energy consumption can be saved.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a speed control method for an induction motor, and more particularly to a speed control method for an induction motor suitable for reducing various losses generated in the induction motor.

[0002]

2. Description of the Related Art As a conventional technique relating to a control device for an induction motor, a technique described in Japanese Patent Laid-Open No. 61-189193 is known. This control device according to the prior art is provided with a means for controlling the exciting current in the drive control device of the induction motor so that the primary current has a minimum value with respect to the required torque of the induction motor.

[0003]

In the above prior art, since the induction motor is controlled so that the primary current becomes the minimum value, the primary resistance loss can be minimized.
There was a problem that the total loss including secondary resistance loss, hysteresis loss, eddy current loss, etc. could not always be minimized. For this reason, the above-mentioned conventional technique has a problem that it is necessary to supply electric power corresponding to these losses from the drive control device, and the capacity of the control device must be increased. On the induction motor side, there is a possibility of overheating of the rotor.For example, when the induction motor is rotated at high speed, hysteresis loss and eddy current loss increase, and in some cases, the induction motor overheats. It may cause damage.

An object of the present invention is to provide a method of controlling an induction motor capable of reducing the total loss generated in the induction motor from the low speed operation area to the high speed operation area of the induction motor.

[0005]

Means for Solving the Problems A feature of the present invention that achieves the above object is to provide a method for controlling an induction motor in which the induction motor is controlled by a variable frequency voltage source, using the detected value of the primary current of the induction motor to calculate. Based on the secondary current component, the excitation current command value and one of the detection values of the primary current and the excitation current component calculated using one of the detection values of the primary current, and the output frequency command value, the excitation that creates the interlinkage magnetic flux in the induction motor. It is to control the current component.

[0006]

BEST MODE FOR CARRYING OUT THE INVENTION

(Embodiment 1) An induction motor control device according to an embodiment of the present invention will be described below.

First, the induction motor applied to the present embodiment is such that the total loss generated in the induction motor is always the minimum value with respect to the torque required from the load. The method of setting the current will be described.

In general, the induction motor can be represented by an equivalent circuit in which the exciting current I _1d and the torque current I _1q are orthogonal to each other in a vector manner as shown in FIG. 3, and in order to simplify the description. It is assumed that the induction motor has no magnetic saturation, and the magnetic flux Φ and the exciting current I _1d are proportional. In FIG. 2 showing the equivalent circuit, L _σ is a leakage inductance, M ′ is an exciting inductance, r ₁ is a primary resistance, and r is
₂ 'secondary resistance, s is slip.

In the equivalent circuit shown in FIG. 2, the exciting current I _1d , the primary current I ₁ , the primary converted torque current I _1q , and the torque T ₁ have the following relationships expressed by the following equations 1 to 4. To do.

[0010]

[ _Equation 1] I _1dr ² + I _1qr ² = I _1r ² ( _Equation 1)

[0011]

[ _Equation 2] I _1d ² + I _1q ² = I ₁ ² ( _Equation 2)

[0012]

[Equation 3] φ∝I _1q (Equation 3)

[0013]

(Equation 4)

However, in the above equation, the subscript r means the value at the time of rated operation, I _1r is the primary rated current, I _1qr is the secondary rated torque current converted to the primary side, T _r is the rated torque, and I _1dr is It is the rated excitation current.

The phase angle θ _r between the torque current I _1qr and the primary rated current I _1r when the induction motor is operating at the rated torque can be obtained from the following _equation 5.

[0016]

(Equation 5)

Further, the exciting current I in an arbitrary operating state
The ratio of _1d to the rated exciting current I _1dr is set as m as shown in the equation (6).

[0018]

## _EQU6 ## m = I _1d / I _1dr ( _Equation 6) From _Equation 2, _Equation 4, _Equation 5, _Equation 6 to I _1q , I _1qr , I _1d ,
By erasing I _1dr and calculating an equation that holds for the primary current I ₁ , the following equation 7 can be obtained.

[0019]

(Equation 7)

Now, the total loss generated in the induction motor is taken as the evaluation function.

First, the primary resistance loss L _r1 as the resistance loss generated in the winding resistance of the induction motor can be obtained by the following equation using the above-mentioned equation 7.

[0022]

(Equation 8)

The secondary resistance loss L _r2 generated in the rotor copper bar of the induction motor can be obtained by the following equation using the above equations 4 and 5.

[0024]

[Equation 9]

Further, the hysteresis loss L _h generated in the stator core of the induction motor can be obtained by the following equation using the above equations 5 and 6.

[0026]

[Formula 10] L _h = K _h ′ · ω ₁ · B _m ² = K _h · ω ₁ · m ² · I _1r ² · sin ² θ _r ( _Equation 10) In this Equation 10, B _m is the magnetic flux density in the air gap of the induction motor, and K _h ′ and K _h are the stator core. It is a proportional coefficient determined by weight, material, structure, etc.

As can be understood from the equation 10, the hysteresis loss L _h is the square of the exciting current, that is, 2 of m.
It is proportional to the power and is proportional to the output frequency of the drive control device.

Further, an eddy current value I _e is generated in the stator iron core, and this eddy current loss L _e is the same as in the equation (10).
It can be obtained by the following equation using the above equations 5 and 6.

[0029]

[Equation 11] L _e = K _e ′ · ω ₁ ² · B _m ² = K _e · ω ₁ ² · m ² · I _{1 r} ² · sin ² θ _r ( _Equation 11) In this Equation 11, K _e ′ and K _e are the weight and material of the stator core, as well as the hysteresis loss. It is a proportional coefficient determined by the structure.

The above-mentioned four types of losses are the main losses that can be controlled among the total losses of the induction motor, and if the exciting current is controlled so that the sum of these losses is minimized, the induction motor is ideally controlled. You can do it. This total loss L _T is
Equations 8 to 11 can be collectively expressed as follows.

[0031]

(Equation 12)

In order to find the condition that the total loss L _T becomes the minimum, the formula 12 is transformed as follows.

[0033]

(Equation 13)

In this equation 13, the total loss L _T has a minimum value when the following equation 14 is established:
Assuming that the value of m at that time is m _o , when m _o satisfies the value of Equation 14, the total loss L _T generated in the induction motor is the minimum value for the torque T ₁ required by the induction motor. Become.

[0035]

[Equation 14]

Then, the minimum value L _To of the total loss can be obtained by the following equation.

[0037]

(Equation 15)

The present invention controls the exciting current of the induction motor on the basis of the above-described mathematical formulas, which enables the induction motor to be operated at the highest efficiency.

The control device for the induction motor of this embodiment will be described in detail with reference to FIG. In FIG. 1, 2 is an induction motor, 93 is a rotation speed detector, 302 is a voltage type power converter, 304 is a coefficient unit, 305a-c are primary current detectors, 3
Reference numeral 06 is a current detector, 307 is a slip angular frequency command calculator, 308 is an exciting current command value calculator, 309 is an adder,
310 is a first-order delay device, 311 is an integrator, 312 is a function generator, 313 is a 2-phase to 3-phase converter, 314, 316, 31.
Reference numeral 7 is a comparator, 315 is a speed controller, and 318 and 319 are arithmetic units.

The exciting current command value calculator 308, which is a characteristic part of the present embodiment, takes in the inverter output angular frequency ω ₁ * and the torque current value I _q corresponding to the load torque, and uses these data to determine the optimum value. Determine the value of the exciting current.

In FIG. 1, the induction motor 2 is controlled by the output variable frequency power of the voltage type power converter 302. The rotation speed detector (PG) 93 detects the rotation speed of the induction motor 2, and its output signal is transmitted to the coefficient multiplier 304 that multiplies the rotor angular frequency by P, which is the number of pole pairs. In the windings of the U-phase, V-phase, and W-phase of the induction motor 2, the primary current that detects the primary current of each phase and outputs the current feedback signals I _u , I _v , and I _w corresponding to the primary current. Current detector 305a, 3
05b and 305c are provided. The output signals of the primary current detectors 305a, 305b, 305c are given to the current detector 306. The current detector 306 rotates the output signal of the primary current detector at the synchronous angular frequency of the induction motor 2 (the angular frequency of the secondary interlinkage magnetic flux vector) ω ₁ *,
It is converted into two axis components of a rotating coordinate system orthogonal to each other, that is, an exciting current component I _d and a torque current component I _q . The torque current component I _q is given to the slip angular frequency calculator 307, the exciting current command value calculator 308 and the like. The slip angular frequency calculator 307 calculates the slip angular frequency command value Pω _s * using the exciting current command value I _d * and the torque current component I _q output from the exciting current command value calculator 308.
Then, the slip angular frequency command value Pω _s * is sent to the adder 309, where it is added to the output Pω _r of the coefficient multiplier 304 described above. The result of this addition is
It is output as the induction motor synchronous angular frequency ω ₁ *.

The exciting current command value calculator 308 includes an absolute value signal conversion circuit 308a, a command value calculation circuit 308b, a maximum value signal selection circuit 308c, and a maximum value holding register 30.
It is composed of four circuits 8d. The torque current component I _q that is the output signal of the current detector 306 is
This signal is input to the absolute value signal conversion circuit 308a in the exciting current command value calculator 308, and the absolute value signal conversion circuit 308a is input.
Is the absolute value of the torque current component I _q that is always a positive signal |
I _q | is output to the command value calculation circuit 308b. Based on the above-mentioned principle, the command value calculation circuit 308b calculates the excitation current calculation value I _do so that the total loss generated in the induction motor is minimized.

[0043]

(Equation 16)

It is obtained by calculating At that time, the command value calculation circuit 308b takes in the angular frequency ω ₁ * output from the adder 309 through the first-order delay device 310 and uses it for calculation of the excitation current calculation value I _do .

Here, the role of the first-order delay device 310 will be briefly described. If the first-order delay device 310 is not provided, the following phenomenon may occur. That is, when the angular frequency ω ₁ * suddenly fluctuates for some reason and becomes large, the value of the exciting current calculation value I _do determined by the above-mentioned equation 16 becomes small, and this exciting current calculation value I _do Since the slip angular frequency Pω _s * calculated on the basis of becomes larger, as a result, the phenomenon that positive control is performed in which the angular frequency ω ₁ * becomes even larger occurs. Therefore, in some cases, the angular frequency ω ₁ * will diverge. The first-order delay device 310 effectively acts to suppress such a sudden change in the angular frequency ω ₁ *.

The exciting current calculation value I _do obtained by the command value calculation circuit 308b as described above is sent to the maximum value signal selection circuit 308c. Maximum value signal selection circuit 308
c compares the value of the input signal I _dr from the input signal I _do and the maximum value holding register 308d from command value calculating circuit 308b, the value of I _do when I _do is less than I _dr, I _do
There the value of I _dr when larger I _dr, exciting current command value I
_{It is} output as _d * to the slip angular frequency calculator 307 and the like.

The integrator 311 integrates the above-mentioned angular frequency ω ₁ *, outputs the phase angle command θ * of the secondary interlinkage magnetic flux vector, and sends the output signal to the function generator 312. The function generator 312 generates a sine wave signal sin θ * and a cosine wave signal cos θ * corresponding to the phase angle command θ *, and sends these signals to the current detector 306 and the 2-phase / 3-phase converter 313.

The rotor angular frequency ω _r detected by the rotation speed detector 93 is also sent to the comparator 314, and the comparator 3
14 is compared with the rotor angular frequency command value ω _r *. The speed controller 315 amplifies the deviation of the rotor angular frequency from the comparator 314, and the torque current command value I _q *.
Is calculated and output.

The comparator 317 compares the command value I _q * of the torque current with the torque current component I _q which is the output of the current detector 306 and outputs the deviation. Also the comparator
Reference numeral 316 denotes an exciting current component I which is the output of the current detector 306.
_d is compared with the exciting current command value I _d * which is the output of the exciting current command value calculator 308, and the deviation is output.

The computing units 318 and 319 are the comparator 3 respectively.
16 and 317, the excitation current deviation (I _d *
-I _d ) and the torque current deviation (I _q * -I _q ) are amplified so that the exciting current component I _d and the torque current component I _{q of} the induction motor 1 are always the predetermined exciting current command value I _d * and the torque. The excitation voltage axis voltage component V _d * of the primary voltage command and the torque current axis voltage component V _q * of the primary voltage command are controlled so as to be equal to the current command value I _q *.

The 2-phase to 3-phase converter 313 is the same as the arithmetic unit 3 described above.
The voltage component V _{d of the} excitation current axis output from 18, 319
* And the voltage component V _q * of the torque current axis are converted into three-phase voltage command values V _u *, V _v *, V _w * and given to the voltage type power converter 302.

In the induction motor drive control circuit, the torque current is used instead of the torque in the calculation formula used in the command value calculation circuit 308b for determining the excitation current command value. When the exciting current is changed as in the embodiment of the present invention, the load torque required by the induction motor and the torque current are not necessarily in a proportional relationship, and are calculated by the arithmetic expression shown in FIG. The exciting current is not always the optimum value. However, for example, when the current value of the induction motor is small and the output torque is small, the rotation speed of the induction motor decreases and the angular frequency ω _r decreases, so the angular frequency ω ₁ * also decreases. When this angular frequency ω ₁ * decreases, the current value increases, and the torque current I _q also rapidly increases to a value that can produce a torque corresponding to the load torque. Therefore, even if the exciting current is calculated using the torque current I _q instead of the torque, it is practically acceptable. As described above, according to the present embodiment, with respect to the required torque of the induction motor, the primary resistance loss, the secondary resistance loss, and the hysteresis loss generated in the induction motor over the entire range from the low frequency region to the high frequency region. Since the total loss including the eddy current loss can be minimized, the power supply capacity can be reduced, the initial cost of the equipment can be reduced, and the energy consumption can be saved.
Further, since the loss inside the induction motor is small, the heat generated inside the induction motor is small and the temperature rise is suppressed, so that the life of the induction motor can be extended and the reliability can be improved. Play.

Japanese Unexamined Patent Publication No. 60-219983 discloses a magnetizing method based on a secondary current component calculated from a detected primary current value of an induction motor and a magnetizing current component so as to minimize the copper loss of the induction motor. Although it is described that the current component is calculated,
No consideration is given to reducing iron loss. On the other hand, the present embodiment can reduce iron loss such as hysteresis loss as described above.

(Embodiment 2) A control apparatus for an induction motor according to another embodiment of the present invention will be described below.

Generally, as a method of operating the primary voltage by voltage control to control the vector of the induction motor, a command value of the primary voltage having a relationship such that the magnetic flux does not vary even if the primary current varies is referred to as a speed command signal. There is one that calculates based on the detected value of the torque current component and the electric constant of the induction motor and controls the output voltage according to the calculation.

In this control method, current (torque)
In the steady state where does not change abruptly, the rotation speed is stably controlled according to the command value, so there is no problem. But,
During the transition, since the magnetic flux changes due to the influence of the inductance of the induction motor and the primary current, the torque generated by the induction motor may change and the speed control may become unstable.

As a countermeasure against this, a method of detecting the primary current of the induction motor and controlling the voltage so that the exciting current component matches the exciting current command value can be considered. However, when such a feedback loop is provided, a current regulator that does not cause a loop gain setting or an offset is required, and further, a constant is required to be set, which causes a problem that the control configuration and the control design become complicated.

In the present embodiment, a method for suppressing the magnetic flux fluctuations at the time of transition without using the above-mentioned current regulator and controlling the induction motor with high accuracy and high response will be described.

The feature of this embodiment is that the amount of change in the current supplied to the induction motor is detected and the phase of the output voltage of the frequency converter is corrected according to this amount of change.

Here, the power supply current is a current proportional to the primary current or the torque component current, and the amount of change may be obtained from either a command value or an actually measured value.

The principle of this embodiment will be described below with reference to the drawings.

It has been found that the fluctuation of the magnetic flux is proportional to the variation of the q-axis component and the torque current component. The amount of change in the torque current component is proportional to the amount of change in the primary current if the exciting current component is constant.

Therefore, if the output voltage phase of the frequency converter is corrected according to the amount of change in the electric current supplied to the motor such as the torque current component or the primary current, the generation of the magnetic flux of the q-axis component can be suppressed. Therefore, even when the current (torque) changes abruptly, the magnetic flux fluctuation can be suppressed, and the magnetic flux can be held as the command value, so that the speed control with high accuracy and high response can be realized.

FIG. 5 shows a vector diagram of voltage, current and magnetic flux based on the equivalent circuit of the induction motor shown in FIG. FIG.
In, the dq axes are orthogonal coordinates rotating at the synchronous angular frequency ω ₁ . Here, the voltage vector V ₁ is the induced electromotive force E ₁ ′.
And induction motor internal impedance drop (r ₁ I ₁ + ω ₁ · L
It is given by the sum of _σ I ₁ ) and has an internal phase difference angle δ between V ₁ and E ₁ ′ according to the internal impedance drop. Here, when the direction of the vector of E ₁ ′ is aligned with the q-axis, the magnitude V _1a of the primary voltage V ₁ and the internal phase difference angle δ become the components I _1d and I _1q of the induction motor primary current I ₁ and the motor constant. Based on the equations (17) and (18).

[0065]

[Equation 17]

[0066]

V _1a = (E ₁ ′ + ω ₁ · L _σ · I _1d + r ₁ · I _1q ) cosδ − (r ₁ · I _1d −ω ₁ · L _σ · I _1q ) sin δ (Equation 18) where If V _1a and δ are controlled based on the actual values of E ₁ ′, I _1d and I _1q and the motor constant, then E ₁ ′ can be controlled to be constant regardless of the current, that is, the magnetic flux φ _2d can be controlled to be constant.
However, at the time of a transient change in the current (torque), the leakage inductance L _σ causes a delay in the primary current. Therefore, the actual coordinate axis m-t axis is different from the d-q axis coordinate used for control. It deviates by an angle corresponding to the leakage magnetic flux (q-axis magnetic flux Δφ _2q ). Therefore, it is necessary to detect the deviation angle Δθ by the method described below and correct the phase of the voltage with the Δθ.

The axis deviation angle Δθ due to Δφ _2q can be detected as follows. Now, a state equation in which the current and magnetic flux of the induction motor are used as variables is given by the following equation.

[0068]

[Equation 19]

Here, P: differential operator, r ₂ ′: primary conversion secondary resistance, M: mutual inductance, L ₂ : secondary inductance, T ₂ : secondary time constant (L ₂ / r ₂ ), ω _{r is the} rotor angular frequency, ω _{s is the} slip angular frequency, and the variables (vector values) I ₁ , φ ₂ , and V ₁ are expressed by the orthogonal dq axis coordinate system as follows.

[0070]

(Equation 20)

Assuming that the d- and q-axis components V _1d and V _1q of the induction motor primary voltage are controlled by the inverter in proportion to the command value, V _1d and V _1q are

[0072]

(Equation 21)

Therefore, substituting the equation (21) into the equation (19) and assuming that the magnetic flux φ _2d is controlled to be constant, the equation of state regarding the q-axis component is as follows.

[0074]

(Equation 22)

From this, if Pφ _{2q is represented} by I _1q ,

[0076]

(Equation 23)

_Therefore , φ _2q fluctuates by the amount corresponding to the leakage inductance drop of I _1q . Then, the variation amount of φ _2q Δφ _2q
Is expressed by the deviation angle Δθ of the coordinate axes on the dq axis differential marker,

[0078]

(Equation 24)

Is obtained. Here, since Δφ _2q << φ _2d ,

[0080]

(Equation 25)

It follows that Δθ can be detected from the amount of change in ΔI _1q .

Next, a control device for an induction motor, which is an embodiment of the present invention based on the above principle, will be described in detail with reference to FIG. The current detector 4 detects the q-axis component (torque current component) I _1q of the primary current of the induction motor 2 with reference to the phase reference signal θ * obtained by integrating the primary angular frequency command ω ₁ * by the integrator 3. Then, the slip computing unit 50 calculates the slip angular frequency ω _s based on the I _1q , and the ω _s and the speed command ω
_The primary angular frequency ω ₁ * is controlled by addition with _r *. On the other hand, the voltage command calculator 6 has a d-axis component (excitation current) command value I _1d * for generating magnetic flux and the torque current detection value I.
By inputting _1q and ω ₁ *, the magnitude V _1a * of the voltage vector and the internal phase difference angle δ * (the phase difference between the primary voltage and the induced electromotive force) are calculated based on the above-described _Expressions 17 and 18. Here, the phase of the voltage is calculated by inputting the detected torque current value I _1q * into the differentiator 67, and calculating the axis deviation angle Δθ, the phase reference signal θ *, and the internal phase difference angle δ *. Calculated by addition. The obtained voltage phase signal (θ * + δ * −Δθ) and the magnitude V _1a * of the voltage are converted by the coordinate converter 7 into three-phase voltage command values V _u * to V _w *, and the voltage command value causes an inverter. Control 1

By the above control, even when the magnetic flux fluctuation φ _2q occurs when the current (torque) changes suddenly and the actual coordinate value (m-t axis) deviates from the d-q axis, the output value Δθ of the differential calculator 67 Accordingly, the voltage phase is corrected and controlled so that the d-q axes always coincide with the m-t axes.

The control characteristics of this embodiment will be described with reference to FIG. This figure shows the control characteristics during a transition when a step change is applied to the speed command ω _r * of the inverter using the 2.2 KW induction motor as the test machine. FIG. 7A shows the conventional characteristic, and FIG. 8B shows the characteristic of this embodiment.

First, in FIG. 6A, the actual rotation speed ω _r does not follow the change of the speed command ω _r *, but pulsates. This is due to the synergistic action of the pulsation of the voltage command V ₁ *, the phase angle δ * and the currents I _1d and I _1q , and the magnetic fluxes φ _2d and φ _2q , as can be seen from the figure. Eventually, divergence is reached and control becomes difficult.

On the other hand, in the present embodiment, as shown in FIG. 6 (b), a _slight φ _2q is generated during the transition, but the voltage phase is corrected according to the change amount of this φ _2q. Therefore, the magnetic flux φ _2d is controlled to be substantially constant. At this time, as is clear from the waveforms of the currents I _1d , I _1q and the actual speed ω _r , it can be seen that the control is performed stably.

Therefore, according to this embodiment, a high-speed control response to any disturbance can be obtained, and the magnetic flux is always kept constant, so that the torque and speed can be controlled with high accuracy.

(Embodiment 3) An induction motor controller according to another embodiment of the present invention will be described with reference to FIG. In this embodiment, the same parts as those in FIG. 4 are designated by the same reference numerals and the description thereof will be omitted. In the embodiment of FIG. 4, since the present invention is applied in a configuration that does not use the speed detector, the detected current value is used as the input of the differentiator 67 that prevents the magnetic flux fluctuation.
On the other hand, in this embodiment, the rotation speed ω _r is detected by the speed detector 93 attached to the induction motor 2, and the speed controller (ASR) 4 is detected according to the deviation between the rotation speed ω _r and the command value ω _r *.
Based on the torque current command value I _1q * output from 0, the differentiator 67 calculates the axis deviation angle Δθ and corrects the voltage phase.

According to this embodiment, the magnetic flux can be controlled to be constant by correcting the voltage phase, and the rotation speed can be stably controlled by the speed detector and the speed adjuster.

According to the embodiments shown in FIGS. 4 and 7, the fluctuation of the magnetic flux due to the transient current (torque) change can be suppressed, so that the magnetic flux can be held substantially constant, and the speed with high accuracy and high response can be obtained. Control can be realized.

[0091]

According to the present invention, the total loss generated in the induction motor can be reduced and the power supply capacity can be reduced from the low speed operation area to the high speed operation area of the induction motor, thus saving energy consumption. .

[Brief description of the drawings]

FIG. 1 is a configuration diagram of a control device for an induction motor according to a preferred embodiment of the present invention.

FIG. 2 is a diagram showing an equivalent circuit of an induction motor.

3 is a vector diagram of current and voltage in the equivalent circuit of FIG.

FIG. 4 is a configuration diagram of a control device for an induction motor that is another embodiment of the present invention.

5 is a vector diagram of current and voltage in the equivalent circuit of FIG.

FIG. 6 is a diagram showing control characteristics of the present embodiment.

FIG. 7 is a configuration diagram of a control device for an induction motor that is another embodiment of the present invention.

[Explanation of symbols]

2 ... Induction motor, 302 ... Voltage type power converter, 306 ...
Current detector, 308 ... Exciting current command value calculator.

 ─────────────────────────────────────────────────── ─── Continuation of the front page (72) Inventor Takayuki Matsui 4026 Kuji Town, Hitachi City, Ibaraki Prefecture Hitate Manufacturing Co., Ltd.Hitachi Laboratory Ltd. In Hitachi Research Laboratory (72) Inventor Hiroshi Fujii 7-1-1 Higashi Narashino, Narashino, Chiba Prefecture Hitachi Narashino Factory

Claims (1)

[Claims] 1. A method for controlling an induction motor in which an induction motor is controlled by a variable frequency voltage source, wherein a secondary current component calculated by using a detected value of a primary current of the induction motor, a command value of an exciting current, and the primary current. A method for controlling an induction motor, characterized by controlling an excitation current component that creates an interlinkage magnetic flux in the induction motor, based on an excitation current component calculated using one of the detected current values and an output frequency command value. .