JPH08297162A - Bi-static radar equipment - Google Patents

Abstract

PURPOSE: To provide a bi-static radar equipment capable of measuring distance with the accuracy higher than that specified by the sampling frequency. CONSTITUTION: A signal converting means 25 in a receiving device 20 receives the reflected wave 2 transmitted from a transmitting device 10 and reflected on a target 5 and the direct wave from the transmitting device 10 respectively, receives the A/D-converted signals y(nT) and x(nT), and generates the cross correlation function signal Pxy (f, τ) and auto-correlation function signal Pxx (f, τ) having two variables of frequency and time. A delay time estimating means 26 obtains the amplitude modulation fundamental frequency of the reflected wave caused by the fact that the radar cross sectional area of the target 5 is periodically changed based on Pxy (f, τ), discrete Fourier transform on the time variable is applied to Pxy (f, τ) and Pxx (f, τ) for the amplitude modulation fundamental frequency respectively, and the delay time of the reflected wave from the direct wave is estimated based on the phase grade of each signal ratio after transformation.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an apparatus for measuring a target in a bistatic radar.

[0002]

2. Description of the Related Art FIG. 10 is an overall configuration diagram of a conventional bistatic radar device. In the bistatic radar device in FIG. 10, the distance to the target is measured by using the measurement value of the propagation time difference between the target reflected wave 2 of the radio wave 1a transmitted from the transmitter 10 and the direct wave 1b for synchronization. You can The target reflected wave 2 is the antenna 21, the receiver 2
3a and the target reflection signal y via the A / D converter 24a
It is output as (nT). On the other hand, the direct wave 1b passes through the antenna 22, the receiver 23b, and the A / D converter 24b,
It is output as the synchronization signal x (nT). Where n is A
The sampling number of the digital signal in the / D converters 24a and 24b, and T represents the sampling interval. The cross-correlation function calculation means 201 calculates the target reflection signal y (nT).
And the synchronization signal x (nT) are input to calculate the cross-correlation function thereof, and the delay time estimating means 202 estimates the delay time δ from the output of the cross-correlation function calculating means 201. in this way,
The propagation time difference between the target reflected wave 2 and the direct wave 1b is measured by measuring the delay time δ of the target reflected signal y (nT) with respect to the synchronization signal x (nT).

To obtain the delay time δ, RE Boucher an
d JCHassab: ”Analysis of discrete implementation
of generalized cross correlator ”, IEEE Transacti
onson Acoustics, Speech, and Signal Processing, vo
As shown in l.ASSP-29, No.3, pp.609-611 (1981), the cross-correlation function of the target reflection signal y (nT) and the synchronization signal x (nT) is calculated, and One method is to find the time lag that gives the maximum absolute value.

[0004]

The cross-correlation function in a digital signal processing system is a sample value, and its sampling interval is equal to that of a digital signal. If the sampling frequency fs of the A / D converters 24a and 24b cannot be increased due to various circumstances, the sampling interval of the cross-correlation function becomes large. Therefore, the measurement unit (= sampling interval) of the delay time difference between the target reflected signal y (nT) and the synchronization signal x (nT) becomes coarse, and as a result, the accuracy of the propagation path length measurement of the target reflected wave 2 becomes poor.
For example, if fs = 30 [kHz], the ranging accuracy is 1
It becomes 0 [km], which is low in accuracy for a radar that detects a target several tens of kilometers away.

The bistatic radar device of the present invention has been made to solve these problems, and the propagation path length of the target reflected wave 2 is accurately defined, that is, the sampling frequency of the A / D converter. The purpose is to measure the distance with higher accuracy.

[0006]

A bistatic radar device according to the present invention receives a reflected wave from a target and
Receiving means for converting a direct wave from the transmitting device into a second digital baseband signal, and first and second digital baseband signals Signal, frequency −
Signal converting means for converting into a cross-correlation function signal having two variables of time and using the second digital baseband signal to convert into an auto-correlation function signal having two variables of frequency-time, and a cross-correlation function signal. The amplitude modulation fundamental frequency of the reflected wave caused by the periodic change of the target radar reflection cross-section is obtained based on, and the cross-correlation function signal for this amplitude modulation fundamental frequency is subjected to the discrete Fourier transform with respect to the time variable. The first discrete Fourier transform signal is generated by performing the discrete Fourier transform of the autocorrelation function signal for the amplitude modulation fundamental frequency on the time variable by generating the first discrete Fourier transform signal. A delay time estimating means for estimating the delay time of the reflected wave with respect to the direct wave from the phase gradient of the ratio of the discrete Fourier transform signal of 2 Those were example.

The bistatic radar device according to the present invention receives the reflected wave from the target and converts it into the first digital baseband signal, and the synchronizing signal transmitted by wire from the transmitting device. Second receiving means for receiving and converting to a second digital baseband signal;
And a second digital baseband signal are used to convert into a cross-correlation function signal having two variables of frequency-time, and an autocorrelation having two variables of frequency-time is used by using the second digital baseband signal. A signal converting means for converting into a function signal and an amplitude-modulation fundamental frequency of a reflected wave resulting from the periodical change of the target radar reflection cross-section based on the cross-correlation function signal, and the cross-correlation with respect to this amplitude-modulation fundamental frequency. A first discrete Fourier transform signal is generated by performing a discrete Fourier transform of the function signal on the time variable, and a second discrete Fourier transform of the autocorrelation function signal on the amplitude modulation fundamental frequency is performed on the time variable. A signal is generated and the reflected wave is calculated from the phase gradient of the ratio of the first discrete Fourier transform signal and the second discrete Fourier transform signal. It is obtained by a delay time estimating means for estimating a delay time for the period signal.

In the bistatic radar device according to the present invention, the signal converting means has a memory for storing the second digital baseband signal and a second digital baseband signal stored in the first digital baseband signal or the memory. A switch for selecting the signal, a complex conjugating means for outputting a complex conjugate of the first digital baseband signal selected by the switch or the second digital baseband signal stored in the memory, and the second digital baseband A plurality of delaying means for delaying the signal by a predetermined time; a plurality of multiplying means for multiplying the output signals of the plurality of delaying means and the complex conjugate; and a plurality of Fourier transforms for Fourier transforming the output signals of the plurality of multiplying means, respectively. Means for using the first and second digital baseband signals for frequency-time 2 When converting to a cross-correlation function signal having a number, the switch selects the first digital baseband signal and uses the second digital baseband signal to convert to an autocorrelation function signal having two frequency-time variables. If so, the switch is configured to select the second digital baseband signal stored in memory.

In the bistatic radar device according to the present invention, the signal converting means has a memory for storing the second digital baseband signal and the second digital baseband signal stored in the first digital baseband signal or the memory. A first switch for selecting a signal; a complex conjugate means for outputting a complex conjugate of the first digital baseband signal selected by the first switch or the second digital baseband signal stored in the memory; 1 out of the output signals of the plurality of delay means for delaying the two digital baseband signals by a predetermined time, the plurality of multiplication means for multiplying the output signals of the plurality of delay means and the complex conjugate, and the output signals of the plurality of multiplication means A second switch for selecting two output signals and a Fourier transform for the output signal selected by the second switch. D) converting the first and second digital baseband signals into a cross-correlation function signal having two variables of frequency-time, the first switch includes the first digital baseband signal. And the second switch is sequentially switched, the Fourier transforming means Fourier transforms each output signal of the plurality of multiplying means, and the second digital baseband signal is used to generate a self variable having two variables of frequency-time. When converting to a correlation function signal, the first switch selects the second digital baseband signal stored in the memory, the second switch is sequentially switched, and the Fourier transform means outputs each output signal of the plurality of multiplying means. Is configured to be Fourier transformed.

In the bistatic radar device according to the present invention, the delay time estimating means returns the absolute value of the cross-correlation function signal with respect to the axis having a frequency of 0, adds the absolute value, and obtains the amplitude modulation fundamental frequency from the addition result. is there.

[0011]

In the bistatic radar device according to the present invention, the first receiving means receives the reflected wave from the target and
Digital baseband signal, the second receiving means receives the direct wave from the transmitting device and converts it into a second digital baseband signal, and the signal converting means converts the first and second digital baseband signals. The signal is used to convert to a cross-correlation function signal having two variables of frequency-time, and
Using the second digital baseband signal, the frequency −
The signal is converted into an autocorrelation function signal having two variables of time, and the delay time estimation means determines the amplitude modulation fundamental frequency of the reflected wave resulting from the periodic change of the target radar reflection cross section based on the cross-correlation function signal. A first discrete Fourier transform signal is generated by performing a discrete Fourier transform on the cross-correlation function signal with respect to the amplitude modulation fundamental frequency with respect to the time variable, and the autocorrelation function signal with respect to the amplitude modulation fundamental frequency is subjected to the discrete Fourier transform with respect to the time variable. To generate a second discrete Fourier transform signal, and delay the direct wave of the reflected wave from the phase gradient of the ratio of the first discrete Fourier transform signal and the second discrete Fourier transform signal regardless of the sampling frequency. Estimate time.

In the bistatic radar device according to the present invention, the first receiving means receives the reflected wave from the target and converts it into a first digital baseband signal, and the second receiving means synchronizes with the transmitting device. The signal is received and converted into a second digital baseband signal, and the signal converting means includes a first digital baseband signal.
And a second digital baseband signal are used to convert into a cross-correlation function signal having two variables of frequency-time, and an autocorrelation having two variables of frequency-time is used by using the second digital baseband signal. The signal is converted into a function signal, and the delay time estimation means obtains the amplitude modulation fundamental frequency of the reflected wave caused by the periodic change of the target radar reflection cross-section based on the cross-correlation function signal, and A first discrete Fourier transform signal is generated by performing a discrete Fourier transform of the cross-correlation function signal with respect to a time variable, and a second discrete Fourier transform of the autocorrelation function signal with respect to the amplitude modulation fundamental frequency is performed with respect to the time variable. A Fourier transform signal is generated, and from the phase gradient of the ratio of the first discrete Fourier transform signal and the second discrete Fourier transform signal, Regardless pulling frequency to estimate the delay time for the reflected wave of the synchronizing signal.

In the bistatic radar device according to the present invention, in the signal conversion means, the memory stores the second digital baseband signal, and the switch stores the second digital baseband signal or the second digital baseband signal stored in the memory. The baseband signal is selected, the complex conjugating means outputs the first digital baseband signal selected by the switch or the complex conjugate of the second digital baseband signal stored in the memory, and the plurality of delaying means output the second conjugate signal. The digital baseband signal of is delayed by a predetermined time, the multiplying means multiplies the output signals of the plurality of delaying means and the complex conjugate, and the plurality of Fourier transforming means Fourier-transforms the output signals of the plurality of multiplying means, respectively. Cross-correlation function signal having two variables of frequency-time using the first and second digital baseband signals When converting, the switch selects the first digital baseband signal, and when using the second digital baseband signal, the switch stores it in the memory when converting it to an autocorrelation function signal having two variables of frequency-time. The selected second digital baseband signal.

In the bistatic radar device according to the present invention, in the signal conversion means, the memory stores the second digital baseband signal, and the first switch stores the first digital baseband signal or the memory. Selecting two digital baseband signals, the complex conjugating means outputting a complex conjugate of the first digital baseband signal selected by the first switch or the second digital baseband signal stored in the memory, Delaying means delays the second digital baseband signal by a predetermined time, a plurality of multiplying means multiplies the output signals of the plurality of delaying means by the complex conjugate, and the second switch outputs the plurality of multiplying means. One output signal is selected from the signals, and the Fourier transform means Fourier transforms the output signal selected by the second switch. When converting the first and second digital baseband signals into a cross-correlation function signal having two variables of frequency-time, the first switch selects the first digital baseband signal and the second switch. , The Fourier transform means Fourier transforms the respective output signals of the plurality of multiplying means, and the second digital baseband signal is used to transform the signal into an autocorrelation function signal having two variables of frequency-time. , The first switch selects the second digital baseband signal stored in the memory, and sequentially switches the second switch, and the Fourier transform means Fourier transforms the output signals of the plurality of multiplying means.

In the bistatic radar device according to the present invention, the delay time estimating means returns the absolute value of the cross-correlation function signal with respect to the axis having a frequency of 0 and adds it, and obtains the amplitude modulation fundamental frequency from the addition result.

[0016]

EXAMPLES Example 1. Embodiments of the present invention will be described below with reference to the drawings. FIG. 1 is an overall configuration diagram of the bistatic radar device of the first embodiment. In FIG. 1, reference numeral 10 denotes a transmitter, which is a modulation signal generator 11 that generates a modulation signal, a transmitter 12 that frequency-converts a modulation signal generated by the modulation signal generator 11 into a carrier frequency, and amplifies the transmission signal.
It is composed of a transmitting antenna 13 which radiates an output signal of 2 into the air.

Reference numeral 5 is a target, and the radar reflection cross-sectional area is assumed to change periodically. For example, when a wing-shaped object such as a propeller rotates, the radar reflection cross-sectional area changes periodically. 1
Reference symbol a is a transmission wave traveling from the transmission device 10 to the target 5, and reference symbol 2 is a reflection wave of the transmission wave 1a reflected by the target 5 and traveling toward the reception device 20. 1b directly from the transmitter 10 to the receiver 20
The transmission wave 1b is referred to as a direct wave here.

The receiving device 20 is installed at a predetermined distance from the transmitting device 10. 21 is a reflected wave 2 from the target 5
Is a main receiving antenna for receiving the
It is an auxiliary receiving antenna for receiving the transmission wave 1b transmitted from the device to the receiving device. This auxiliary receiving antenna 22 is oriented so that the antenna gain in the direction of the transmitting device 10 becomes high in order to receive the transmission wave 1b from the transmitting device 10. Reference numerals 23a and 23b denote receivers for amplifying a signal received by the main receiving antenna 21 or the auxiliary receiving antenna 22, and converting the signal to an intermediate frequency or performing phase detection. Reference numerals 24a and 24b denote receivers 23a and 23b. An A / D converter for converting an output signal into a digital signal. Below, the sampling interval at the time of this A / D conversion is set to T, and the reciprocal sampling frequency is set to fs. The digital baseband signals which are the output signals of the A / D converters 24a and 24b are respectively converted into y (nT) and x.
(NT), and x (nT) is used as a synchronization signal in the subsequent processing.

Reference numeral 25 is a signal converting means for converting a target reflected wave reception signal (hereinafter simply referred to as a target reflected signal) y (nT) and a synchronizing signal x (nT) into a signal having two variables of frequency-time delay. To do. Reference numeral 26 is a delay time estimating means, which is the output signals P _xy (f, τ) and P _xx (f, of the signal converting means 25.
τ), the propagation path length of the reflected wave 2 that is transmitted as the transmission wave 1a from the transmission device 10 and is reflected by the target 5 and received, and the direct wave 1b that directly propagates from the transmission device 10 toward the reception device 20. Y (n) due to the difference from the propagation path length
Estimate the delay time difference between T) and x (nT). Here, f is the amplitude modulation frequency, and τ is the number of delay samples given to the synchronizing signal x (nT) in the signal converting means 25.

In the above-described embodiment, the A / D conversion is performed after the phase detection, but the A / D conversion of the intermediate frequency signal and the conversion to the digital intermediate frequency signal may be performed after the phase detection by the digital signal processing. good.

FIG. 2 is an example of an internal configuration diagram of the signal converting means 25 in FIG. In FIG. 2, 31 is a memory for temporarily storing the synchronization signal x (nT), and 32 is a memory.
Is a first switch for switching the synchronization signal x (nT) and the target reflection signal y (nT), and 33 is the target reflection signal y
(NT) or a complex conjugate device that outputs a complex conjugate of the synchronization signal x (nT), 34 is a delay device for one sampling interval, 3
Reference numeral 5 is a multiplier, and 36 is a discrete Fourier transformer (DFT). If the number of samples of the signal to be subjected to the discrete Fourier transform is a power of 2, for example, a fast Fourier transformer may be used as the discrete Fourier transformer 36.

Further, in FIG. 3, (a) is a transmission wave 1a,
1b shows the signal waveform, and (b) shows the signal waveform of the reflected wave 2. Further, FIG. 4 shows the output signal P _xy of the signal conversion means 25.
An example of the absolute value of (f, τ) is shown, and FIG. 5 is a flow chart showing the operation of the delay time estimating means 26.

Next, the operation of the first embodiment will be described with reference to FIGS. In FIG. 1, a transmitter 1
0 transmits a modulated signal as shown in FIG. 3A to the target 5 as transmission waves 1a and 1b. Receiver 20
Receives the reflected wave 2 from the target 5 at the main receiving antenna 21 and directly transmits the transmitted wave 1b at the auxiliary receiving antenna 22.
To receive. The receivers 23a and 23b amplify the signal received by the main receiving antenna 21 or the auxiliary receiving antenna 22, and sequentially perform conversion to an intermediate frequency signal and phase detection. A /
The D converters 24a and 24b convert the output signals of the receivers 23a and 23b into digital baseband signals, respectively, and target reflection signals y (nT) and synchronization signals x, respectively.
(NT) is output to the signal conversion means 25.

FIG. 3 shows an example of a signal waveform transmitted from the transmitter 10 and a target reflection signal from the target 5. The transmission signal waveform of (a) is a frequency-modulated signal with a constant envelope, and thus some kind of modulation must be performed in order to measure the distance. The target reflection signal waveform of (b) is amplitude-modulated by the target radar reflection cross-section changing periodically.

The signal converting means 25 shown in FIG. 2 is substantially the same as the signal converting means including a demodulation circuit for a received signal and a Doppler filter bank in a radar which uses a code-modulated continuous wave or pulse signal as a transmitted wave. But memory 3
The difference is that 1 and the first switch 32 are present. When the first switch 32 is set to J1, the synchronization signal x (nT) has the delay device 34 and the multiplier 3 by the number of samples required for processing by the discrete Fourier transform device 36.
5, and is temporarily stored in the memory 31. Target reflection signal y input from the first switch 32
(NT) is complex-conjugated by the complex conjugator 33, and the multiplier 3
In step 5, the signal is multiplied by the synchronizing signal transferred to the delay device 34 and the multiplier 35, input to the discrete Fourier converter 36, and P
_xy (f, τ) is obtained.

When the processing of the discrete Fourier transform 36 is completed, the first switch 32 is switched to J2, and the synchronization signal x (nT) stored in the memory 31 is transferred to the delay device 34 and the multiplier 35. At the same time, it is transferred to the complex conjugator 33 and complex-conjugated. The complex-conjugated signal is multiplied by the synchronizing signal transferred to the delay unit 34 and the multiplier 35 in the multiplier 35, input to the discrete Fourier transform unit 36, and P _xx
(F, τ) is obtained. When the delay time estimation means 26, which is the post-processing of the signal conversion means 25, obtains the estimated value of the delay time, the processing of one cycle of the present device is completed. Next, when the estimated value of the delay time is obtained, the contents of the memory 31 are erased, the switch 32 is switched to J1 again, and the above processing is repeated.

As described above, the memory 31 and the switch 32 in the signal converting means 25 are composed of the discrete Fourier converter 3
It operates in synchronization with the processing timing of 6 and the overall processing timing. The output signal of the signal converting means 25 is
It becomes a two-dimensional signal having two variables of frequency-time delay similar to the case of the pulse Doppler radar, and the synchronizing signal x (nT) is used as a reference signal for conversion.

First, as shown in FIG. 2, the first switch 3
2 is connected to the input terminal side J1 of the target reflection signal y (nT). Focusing on the amplitude modulation due to the periodic change of the radar reflection cross section of the target reflection signal, the received signal y
If there is clutter or direct wave leakage at (nT),
It is possible to distinguish these from the target reflection signal.

The target reflection signal y (nT) input via the first switch 32 is complex-conjugated by the complex conjugator 33, and delayed by the delay device 34 with the synchronizing signal x (nT) and the multiplier 35. Since it is multiplied by, the input signal q _m (nT) of the discrete Fourier transformer 36 can be expressed as in equation (1). q _m (nT) = x ((n−m) T) y ^* (nT) (1) where m = 0, 1, ..., L−1, and L is the target reflection signal y.
It is an integer determined by the maximum detection distance of (nT). This subscript m represents the amount of delay by the delay unit 34 in units of sampling intervals. At this time, the output signal P _xy (kΔf, mT) of the discrete Fourier transformer 36 becomes as shown in Expression (2).

[0030]

[Equation 1]

FIG. 4 shows the function P _xy (kΔf, m) of the equation (2).
It is an example of the absolute value of T). In FIG. 4, P _xy (kΔf, m
The portion of N / 2 <k ≦ N−1 in T) is moved to f <0 and drawn. It also assumes that the target has not moved. 91 is a direct wave 1b leaking into the main receiving antenna 21.
Since the amplitude modulation due to the periodic change of the radar reflection cross-section is not caused in the peak due to the component of, the peak occurs at f = τ = 0. Reference numeral 92 is a peak due to clutter, which is also not subjected to amplitude modulation due to the periodic change of the radar reflection cross-section, so on f = 0, τ is at a position corresponding to the path length difference between the clutter and the direct wave 1b. Peaks occur.

Reference numeral 93 is a peak due to the target reflection signal which is amplitude-modulated due to the periodic change of the target radar reflection cross-section, and with respect to f, the amplitude modulation basic frequency f = ± f _AM ,
Regarding τ, a peak occurs at a position corresponding to the path length difference between the target reflected wave 2 and the direct wave 1b. Further, a peak may occur due to the harmonic component of the amplitude modulation. By the Fourier transform of equation (2), the amplitude modulation component of the target reflection signal y (nT) caused by the periodic change of the target radar reflection cross section,
Clutter components that have not undergone such amplitude modulation can be separated on the frequency axis. Where not all k = 0,
It is not necessary to calculate P _xy (kΔf, mT) for 1, ..., N−1, and only the necessary frequency range may be calculated.

Since the target reflection signal and the unnecessary signal are separated by utilizing the amplitude modulation component as described above, it is desirable that the transmission signal has a constant envelope.

The calculation of equation (2) is equivalent to obtaining the radar ambiguity function. In the radar ambiguity function, a peak occurs at the Doppler frequency corresponding to the target velocity and the delay time corresponding to the distance, and the absolute value of the radar ambiguity function takes the maximum value at that point. The difference is that such a peak of P _xy (kΔf, mT) | is discarded and the propagation path length of the target reflected wave 2 is obtained from the peak on f = ± f _AM .

Next, the delay time estimating means 26 outputs the output signals P _xy (kΔf, mT) and P _xx (kΔ) of the signal converting means 25.
f, mT), a method of estimating the propagation time difference resulting from the propagation path length difference between the target reflected wave 2 and the direct wave 1b with accuracy not limited to the sampling interval T will be described.

Generally, two signals a (t) and b (t) are considered and b (t) = a (t-δ). Here, the delay time δ is a time lag that gives the maximum absolute value of the cross-correlation function of a (t) and b (t). And this is obtained in the frequency domain as shown below. Further, the power spectrum S _aa (f) of a (t) and the mutual spectrum S _ab (f) of a (t) and b (t) have the relationship of the following expression (3).

[0037]

[Equation 2]

Here, Φ _aa (τ) is the autocorrelation function of a (t), and Φ _ab (τ) is the cross-correlation function of a (t) and b (t). The following expression (4) is obtained from the expression (3).

[0039]

(Equation 3)

As shown in the equation (4), the delay time δ is calculated from the phase gradient of the ratio of the cross spectrum to the power spectrum.
Is required.

A similar relation is that P _xy (kΔf, m in equation (2)
T) and P _xx (f, τ) defined by the equation (5) are also established. This will be shown below, but here, for convenience, the discrete variables kΔf and mT are continuous variables f and τ, respectively.
Notated with.

[0042]

[Equation 4]

For simplicity, y (t) = x (t-τ
₀ ). At this time, the equation (2) becomes like the equation (6).

[0044]

(Equation 5)

As shown in the equation (6), that is, P _xy (f,
τ) is equal to P _xx (f, τ) shifted by τ ₀ with respect to τ and multiplied by a complex number. Therefore,
When f = f _AM and both sides of Expression (6) are Fourier-transformed with respect to τ, a relational expression of Expression (7) similar to Expression (3) is obtained. That is, the delay time τ is calculated from the phase gradient of the Fourier transform ratio of τ between P _xy (f _AM , τ) and P _xx (f _AM , τ).
₀ is required. Note that Fτ [·] in the equation (7) represents a Fourier transform regarding τ.

[0046]

(Equation 6)

FIG. 5 is a flow chart showing the operation of the delay time estimating means 26 specifically showing the above operation for a discrete time signal. The operation of FIG. 5 will be described below.
First, in step 101, the amplitude modulation fundamental frequency f _AM of the target reflection signal due to the periodical change of the radar reflection cross section of the target 5 is obtained. Next, in step 102, P _xy (f _AM , mT) is subjected to DFT or FFT with respect to the delay variable m in order to obtain the one corresponding to F τ [P _xy (f _AM , τ)] in the equation (7), The result is DFT [P _xy
(F _AM , mT)].

Then, in step 103, equation (7)
The value corresponding to F τ [P _xy (f _AM , τ)] is calculated, but first, the first switch 32 is switched to J2.
Then, in step 104, the signal value x (nT) stored in the memory 31 is read out, and in step 105, P _xx
Calculate (f _AM , mT). This is
The output signal y (nT) of the filter 34 is converted into the synchronization signal x (n
T) and perform the calculation in the same manner as in equation (2). However, in the equation (2), k = 0, 1, ...
The calculation of _xx (f _AM , mT) may be performed only for a specific k satisfying f _AM = kΔf.

After obtaining P _xx (f _AM , mT), in step 106, P _xx (f _AM , mT) is DF with respect to the delay variable m.
T or FFT is performed and the result is DFT [P _xx (f
_AM , mT)].

Then, in step 107, equation (4)
And the following equation (8) corresponding to equation (7),
[P _xy (f _AM , mT)] / DFT [P _xx (f _AM , mT)
The estimated delay value τ ₀ is obtained from the phase gradient of T)].

[0051]

(Equation 7)

This delay estimation method uses the A / D converter 24a,
There is an advantage that the sampling interval T of 24b does not affect the distance measurement accuracy of the propagation path length measurement. That is, the delay estimation value τ ₀ is not limited to an integral multiple of the sampling interval T. Therefore, the delay time can be estimated with accuracy independent of the sampling interval, and accurate propagation path length measurement can be performed.

From the estimated value of the delay time, the difference in propagation path length between the target reflected wave 2 and the direct wave 1b can be known. Further, the position of the target can be known by measuring the direction (azimuth and elevation) of the target reflected wave 2.

As described above, the delay time due to the difference in propagation path length between the target reflected wave 2 and the direct wave 1b can be accurately measured in a time unit shorter than the receiving system sampling interval.

Example 2. FIG. 6 shows the internal configuration of the signal converting means 25 in the second embodiment of the bistatic radar device of the present invention. In FIG. 6, 37 is a second switch,
38 is a switch control means for controlling the switching timing of the second switch 37.

In the signal converting means 25 of FIG. 2 in the first embodiment, since the discrete Fourier transformer is connected to the output of each of the L multipliers 35, L discrete Fourier transformers are required. . In this embodiment, by switching the second switch 37, only one discrete Fourier transformer needs to be used.

First, in the second switch 37, the terminals T1 and T0 are connected. Then, P _xy (kΔf, 0) is calculated by the discrete Fourier transformer 36 from q ₀ (nT) by the equation (1). P _xy (k
After the calculation of Δf, 0) is completed, the second switch 37 is switched to connect the terminal T2 and the terminal T0, and then P _xy (kΔf, T) is transformed from q ₁ (nT) by the equation (1) by the discrete Fourier transform. Calculate with the instrument 36. Similarly, the calculation of P _xy (kΔf, (L-1) T) from q _L-1 (nT) is repeated until it is completed. After the calculation of P _xy (kΔf, mT) is completed, it is passed to the delay time estimation means 26. The same applies to the calculation of P _xx (kΔf, mT). As described above, since only one discrete Fourier transformer 36 is used in this embodiment, there is an advantage that the circuit scale becomes small. The other effects are similar to those of the first embodiment.

Example 3. In the first and second embodiments, the output signal P _xy (f of the signal converting means 25 is estimated in the delay time estimation with respect to the synchronization signal x (nT) of the target reflected signal due to the difference in the propagation path length of the target reflected wave 2 and the direct wave 1b. , Τ) f
The area <0 was not used. Therefore, this embodiment 3
In the delay time estimation at P _xy (f, τ)
The area of f <0 is also positively used. Less than,
The method will be described.

In step 101 of FIG. 5, the fundamental frequency f _AM of the amplitude modulation of the target reflection signal due to the periodic change of the target radar reflection cross-section is searched, but before step 101, it is shown in FIG. The operation of step 131 is performed. FIG. 8 schematically shows this operation. The region of absolute value f <0 of the output signal P _xy (kΔf, mT) of the signal conversion means 25 is returned to the axis of f = 0 and added to the region of f> 0. The addition result is | P _xy '(kΔf, m
T) |, then | P _xy '(kΔf, mT) | = | P _xy (kΔf, mT) | + | P _xy (−kΔf, mT) | (9). Then, in step 101 of FIG. 5, | P
Instead of _xy (kΔf, mT) |, | P _xy 'in equation (9)
(KΔf, mT) | is used.

When the target is not moving, the peak of | P _xy (f, τ) | due to the amplitude modulation component of the target reflection signal due to the periodic change of the radar reflection cross section of the target is as shown in FIG. Appear symmetrically with respect to f = 0, but generally with respect to side lobes, P _xy (f, τ) ≠ P _xy (−f,
τ), the signal-to-noise power ratio can be increased. Therefore, more accurate f _AM estimation is possible.

Example 4. FIG. 9 is an overall configuration diagram showing a third embodiment of the present invention, in which a synchronization signal is transmitted from the transmission device 10 to the reception device 20 by wire in order to synchronize the transmission device 10 and the reception device 20. . In FIG. 9, 1
2b is a transmitter for transmitting the synchronization signal from the transmission device 10 to the reception device 20 by wire, 3 is a transmission line for transmitting the synchronization signal, and 23b is a synchronization signal sent from the transmission device 10 by wire. Is a receiver for receiving. Since the synchronization signal is transmitted by wire, unlike the case of FIG.
Has no antenna to receive direct waves.

The operation of FIG. 9 is the same as the first embodiment except that the synchronizing signal is sent from the transmitter 10 to the receiver 20 by wire.
Of course, the second embodiment and the third embodiment can be applied to the fourth embodiment. Since the synchronization signal is not transmitted wirelessly in the fourth embodiment, the influence of the direct wave leaking into the main receiving antenna 21 is smaller than that in the first embodiment.

[0063]

As described above, according to the present invention, when the target radar reflection cross-section periodically changes, the fact that the target reflection signal undergoes amplitude modulation is used to change the unnecessary signal component to the target reflection signal. By separating, the arrival time difference between the target reflection signal and the synchronization signal can be estimated at a time interval shorter than the sampling interval of the A / D converter. Therefore, it is possible to obtain the bistatic radar device capable of accurately measuring the path length of the target reflected wave from the arrival time difference between the target reflected signal and the synchronization signal. Especially, the present invention is particularly effective when the sampling frequency is low.

Further, in the present invention, since the direct wave from the transmitting device to the receiving device is transmitted by wire, it is not necessary to provide the receiving device with an antenna dedicated to the direct wave, and the influence of the direct wave on the reflected wave is reduced. It is possible to accurately measure the path length of the target reflected wave.

Further, according to the present invention, since only one discrete Fourier transformer is used in the signal converting means, there is an effect that the circuit scale becomes small.

Further, in the present invention, since the absolute value of the output signal of the signal converting means is folded and added with respect to the frequency 0 in the delay time estimating means, the signal-to-noise power ratio can be increased and a more accurate target can be obtained. The fundamental frequency of the amplitude modulation of the reflected signal can be extracted.

[Brief description of drawings]

FIG. 1 is an overall configuration diagram showing a bistatic radar device according to a first embodiment of the present invention.

FIG. 2 is an internal configuration diagram of a signal conversion unit according to the first embodiment of the present invention.

FIG. 3 is an example of transmission signal and target reflection signal waveforms.

FIG. 4 is a signal converting means output signal P in the present invention.
It is an example of the absolute value of _xy (kΔf, mT).

FIG. 5 is a flowchart showing the operation of the delay time estimating means in the first embodiment of the present invention.

FIG. 6 is an internal configuration diagram of a signal converting means according to a second embodiment of the present invention.

FIG. 7 is a flow chart showing the operation of means for folding back and adding the absolute value of the output signal of the signal converting means with respect to frequency 0 in the third embodiment of the present invention.

FIG. 8 is a diagram schematically showing an operation of means for folding back and adding an absolute value of a predetermined one of the output signals of the signal converting means according to the third embodiment of the present invention with respect to frequency 0;

FIG. 9 is an overall configuration diagram showing a bistatic radar device according to a fourth embodiment of the present invention.

FIG. 10 is an overall configuration diagram showing a conventional bistatic radar device.

[Explanation of symbols]

 1a, 1b Transmitted wave 2 Target reflected wave 3 Transmission line 5 Target 10 Transmitter 13 Transmitter antenna 20 Receiver 21 Main receiver antenna 22 Auxiliary receiver antenna 24a, 24b A / D converter 32 First switch 33 Complex conjugate device 34 Delay 35 35 Multiplier 36 Discrete Fourier Transform 37 Second switch

Claims (5)

[Claims] 1. A transmission device that radiates a transmission wave to a target, and a reception device that is installed at a predetermined distance from the transmission device and that receives a direct wave that directly arrives from the transmission device and a reflected wave from the target. In the bistatic radar device, the receiving device includes a first receiving unit that receives the reflected wave and converts it into a first digital baseband signal, and a second digital baseband signal that receives the direct wave. Second receiving means for converting into a signal, and using the first and second digital baseband signals, converting into a cross-correlation function signal having two variables of frequency-time, and the second digital baseband. Signal conversion means for converting the signal into an autocorrelation function signal having two variables of frequency-time, and the target radar reflection cross section based on the cross-correlation function signal. A first discrete Fourier transform signal is obtained by obtaining an amplitude modulation fundamental frequency of the reflected wave caused by a temporal change and performing a discrete Fourier transform on the cross-correlation function signal with respect to the amplitude modulation fundamental frequency with respect to a time variable. A second discrete Fourier transform signal is generated by performing a discrete Fourier transform of the autocorrelation function signal for the amplitude modulation fundamental frequency with respect to a time variable, and the first discrete Fourier transform signal and the second discrete Fourier transform signal are generated. And a delay time estimating means for estimating a delay time of the reflected wave with respect to the direct wave from a phase gradient of a ratio of a Fourier transform signal. 2. A transmission device that radiates a transmission wave to a target, a synchronization signal that is installed at a predetermined distance from the transmission device, and is transmitted from the transmission device by wire, and a reflection wave from the target. In the bistatic radar device, the receiving device includes: a first receiving unit that receives the reflected wave and converts the reflected wave into a first digital baseband signal; and a second receiving unit that receives the synchronization signal. Second receiving means for converting into a digital baseband signal, and using the first and second digital baseband signals, converting into a cross-correlation function signal having two variables of frequency-time, and the above-mentioned second A signal converting means for converting an autocorrelation function signal having two variables of frequency-time using a digital baseband signal, and the target radar radar based on the cross-correlation function signal. A first discrete Fourier transform is performed by obtaining an amplitude modulation fundamental frequency of the reflected wave caused by the periodical change of the cross-sectional area and performing a discrete Fourier transform on the cross-correlation function signal with respect to the amplitude modulation fundamental frequency with respect to a time variable. A second discrete Fourier transform signal is generated by generating a transform signal and performing a discrete Fourier transform on the autocorrelation function signal with respect to the amplitude modulation fundamental frequency with respect to a time variable, to generate the second discrete Fourier transform signal and the first discrete Fourier transform signal. And a delay time estimating means for estimating the delay time of the reflected wave with respect to the synchronizing signal from the phase gradient of the ratio of the discrete Fourier transform signal of 2. 3. The signal converting means selects a memory for storing the second digital baseband signal, and the first digital baseband signal or the second digital baseband signal stored in the memory. Switch, a complex conjugating means for outputting a complex conjugate of the first digital baseband signal selected by the switch or the second digital baseband signal stored in the memory, and the second digital baseband. A plurality of delaying means for delaying the signal by a predetermined time; a plurality of multiplying means for multiplying the output signals of the plurality of delaying means and the complex conjugate; and a plurality of Fourier transforming the output signals of the plurality of multiplying means, respectively. Fourier transform means of
When the digital baseband signal of is converted into the above cross-correlation function signal having two variables of frequency-time,
When the switch selects the first digital baseband signal and converts the second digital baseband signal into an autocorrelation function signal having two variables of frequency-time, the switch stores in the memory. 3. The bistatic radar device according to claim 1 or 2, wherein the stored second digital baseband signal is selected. 4. The signal converting means selects a memory for storing the second digital baseband signal, and the first digital baseband signal or the second digital baseband signal stored in the memory. A first switch and a second digital baseband signal selected by the first switch or a second stored in the memory
A complex conjugate means for outputting a complex conjugate of the digital baseband signal, a plurality of delay means for delaying the second digital baseband signal by a predetermined time, an output signal of the plurality of delay means and the complex conjugate. A plurality of multiplying means for multiplying by, a second switch for selecting one output signal from the output signals of the plurality of multiplying means, and a Fourier transform for Fourier transforming the output signal selected by the second switch. And a first means and a second means.
When the digital baseband signal of is converted into the above cross-correlation function signal having two variables of frequency-time,
The first switch selects the first digital baseband signal, switches the second switch in sequence, and the Fourier transforming means Fourier transforms the output signals of the plurality of multiplying means to obtain the second transforming signal. When the digital baseband signal is used for conversion into the autocorrelation function signal having two frequency-time variables, the first switch selects the second digital baseband signal stored in the memory and 3. The bistatic radar device according to claim 1 or 2, wherein the second switch is sequentially switched, and the Fourier transforming means Fourier transforms the output signals of the plurality of multiplying means. 5. The delay time estimating means returns the absolute value of the cross-correlation function signal with respect to an axis having a frequency of 0, adds the absolute values, and obtains the amplitude modulation fundamental frequency from the addition result. The bistatic radar device according to any one of claims 1 to 4.