JPH06343021A - Filter circuit - Google Patents

Abstract

(57) [Abstract] [Purpose] A first characteristic control means capable of switching a filter characteristic such as a cutoff frequency and a second characteristic capable of compensating for a characteristic variation at a switching destination. And a filter circuit including a control unit. [Configuration] When the switch 6 (first characteristic control means) is switched, the operating differential amplifier is switched, so that the mutual conductance g _m is switched and the cutoff frequency of the filter is switched. Further, by varying the variable voltage source 2 (second characteristic control means), the mutual conductance g _m changes, so that characteristic variation can be compensated.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an improvement of a filter circuit, and more particularly to an improvement of a filter circuit suitable for signal processing for image quality correction in a television receiver or the like. Is.

[0002]

2. Description of the Related Art A technique for constructing a filter into an IC is described in, for example, Japanese Patent Publication No. 1-54884. This is a filter configured by a combination of a variable gain amplifier and a capacitor, but the cutoff frequency as a filter characteristic can be changed by controlling the gain of the variable gain amplifier. When the filter is configured as an IC, since the resistance value and the capacitance value inside the IC have manufacturing variations, the cutoff frequency, which is the filter characteristic, also varies. The gain of the variable gain amplifier is controlled to compensate for variations.

FIG. 10 is a circuit diagram showing an example of such a conventional filter circuit. A signal from the input signal source INPUT is output from the output terminal OUT. C is a capacitor that constitutes a filter. By adjusting the variable voltage source VV, the gain of the variable gain amplifier forming the filter can be controlled to compensate the variation in the cutoff frequency due to the manufacturing variation in the resistance value and the capacitance value inside the IC configuring the filter. It is like this.

[0004]

In the filter according to the above-mentioned prior art, for frequency switching for switching the cutoff frequency from a certain frequency to another frequency, for fine adjustment for compensating the variation in the cutoff frequency, Since the same gain control means is also used in the above, if the gain control means is configured so that the cutoff frequency can be arbitrarily switched, in such a gain control means, variation compensation of the cutoff frequency can be compensated. If the gain control means is configured so that it is not possible and the variation of the cutoff frequency can be compensated, such a gain control means cannot change the cutoff frequency arbitrarily.

By the way, it is possible to switch a filter characteristic such as a cutoff frequency characteristic from a certain characteristic to another characteristic, and when there is a variation in the characteristic after switching, this characteristic variation can also be compensated. Filters may be required. For example, in a television receiver which receives both a high-definition broadcast television signal and a standard television signal converted into double scan lines, since these two signals have different original signal bands, image quality correction (for example, contour When you try to apply
It is necessary to switch the center frequency for image quality correction according to which television signal is received.

When the image quality correction circuit is composed of an IC, a phase shift circuit, which is one of the filters, is often used as a substitute for the required delay line. Since the frequency is determined by the delay time of the phase shift circuit, this delay time (that is, the filter characteristic) must be switched. Further, if there is characteristic variation at the switching destination, it is required to compensate for it.

It is an object of the present invention to provide a filter circuit which can be required under the circumstances as described above and which can switch the filter characteristic and can compensate for the characteristic variation at the switching destination. To do.

[0008]

In order to achieve the above object, in the present invention, in a filter circuit constituted by a combination of a variable conductance amplifier as a voltage controlled current source and a capacitor,

As the differential amplifier constituting the variable conductance amplifier, at least two first and second differential amplifiers having a common load are provided and depending on which one is selected to operate. First characteristic control means for switching the mutual conductance of the variable conductance amplifier to switch filter characteristics such as a cutoff frequency;

Of the two differential amplifiers, the output current of the differential amplifier on the selected and operating side is variably controlled, so that the mutual conductance of the variable conductance amplifier is variably controlled to cut off. Second characteristic control means for varying filter characteristics such as frequency.

Further, in a filter circuit composed of a combination of a variable conductance amplifier as a voltage controlled current source and a capacitor, at least two capacitors, a first capacitor and a second capacitor, are provided as the capacitors, and First characteristic control means for switching filter characteristics such as a cutoff frequency depending on which is selected to function;

A second characteristic in which the mutual conductance of the variable conductance amplifier is variably controlled by variably controlling the output current of the differential amplifier constituting the variable conductance amplifier, and the filter characteristic such as the cutoff frequency is varied. And a control means.

[0013]

It is possible to switch the filter characteristic by using the first characteristic control means and compensate for the variation of the filter characteristic at the switching destination by using the second characteristic control means. In this way, it is possible to arbitrarily switch the filter characteristics such as the cutoff frequency characteristics, and it is possible to realize a filter in which characteristic variations are compensated for at each switching destination.

[0014]

Embodiments of the present invention will be described below with reference to the drawings.
explain. FIG. 1 is a circuit diagram showing an embodiment of the present invention.

In the figure, 1 is an independent voltage source for supplying a power supply voltage to the circuit, 2 is a variable independent voltage source, and a differential pair composed of transistors 9 and 10 and transistor 1 are provided.
It is a variable independent voltage source that inputs to the differential pair composed of 5 and 16, and can change the voltage in both positive and negative directions. Reference numeral 3 is an independent voltage source for biasing the differential pair.

An input signal voltage source 4 is input to the bases of the transistors 17 and 21.
The emitter of the transistor 17 is connected to the emitter of the transistor 18 via the resistor 25, the emitter of the transistor 17 is connected to the collector of the current source transistor 19, and the emitter of the transistor 18 is connected to the collector of the current source transistor 20. And constitutes the first differential amplifier.

The emitter of the transistor 21 is connected to the emitter of the transistor 22 via the resistor 26, the emitter of the transistor 21 is connected to the collector of the current source transistor 23, and the emitter of the transistor 22 is connected to the current source transistor 24. It is connected to the collector and constitutes a second differential amplifier.

The loads of the first differential amplifier and the second differential amplifier are common, that is, the transistor 1
The collectors of 7 and 21 are common, and the collectors of transistors 18 and 22 are common. Transistor 1
The collectors of 7 and 21 are connected to the independent voltage source 1, and the collectors of the transistors 18 and 22 are connected to the emitters of the transistors 15 and 16 that form a differential pair.

The collector of the transistor 15 is connected to the independent voltage source 1, and the collector of the transistor 16 is PN.
The output current i _{o of} the first differential amplifier or the second differential amplifier is output from the contact point of the collector of the P transistor 14 and the collector of the PNP transistor 14 and the collector of the transistor 16.

This output current i _o is determined by the load capacitor 2
7 and let the capacitance value of the capacitor 27 be C, the output voltage v _o = i _o / sC (where s is a Laplace operator, and if the frequency response is stationary, replace s → jω. Is supplied to the base of the transistor 28, and the diodes 29, 30, 31 for level shifting are provided.
Transistor 1 forming a first differential amplifier via
Negative feedback is input to the base of 8 and the base of the transistor 22 which constitutes the second differential amplifier.

32 is a transistor 28 and a diode 2
A bias current source for driving 9, 30, 31. The current I ₁ flowing from the current source 5 is supplied to the collector of the transistor 7 or 8 via the switch 6.
The transistor 7 has a collector and a base connected to each other, and this contact is connected to the base of the transistor 11 and the bases of the current source transistors 19 and 20 of the first differential amplifier to form a current mirror circuit. is doing.

Similarly, the transistor 8 is also connected to the collector and the base, and its contact is connected to the base of the transistor 12 and the bases of the current source transistors 23 and 24 of the second differential amplifier to make a current. Configure a mirror circuit. The collectors of the transistors 11 and 12 are connected to the emitters of the transistors 9 and 10 that form a differential pair.
The collector of is connected to the independent voltage source 1
The collector of 0 is connected to the collector of the PNP transistor 13.

The transistor 13 has its collector and base connected to each other, and its contact is connected to the base of the PNP transistor 14, and the transistors 13 and 14 form a current mirror circuit. The bases of the differential pair transistors 9 and 10 are the same as the differential pair transistor 1.
The bases of the transistors 9 and 15 are connected to one terminal of the variable voltage source 2, and the bases of the transistors 10 and 16 are connected to the other side of the variable independent voltage source 2. Connected to the terminal. Reference numeral 33 is a terminal for inputting a switching signal for controlling the switch 6.

Next, the operation of this embodiment will be described. The first characteristic control means used for switching the filter characteristic is
It is first clarified that the second characteristic control means, which is a switching means of the switch 6 and is used for compensating for variations in filter characteristics, is a variable control in the variable independent voltage source 2.

In FIG. 1, for simplification of description, the value of the resistor 25 is R ₁ and the value of the resistor 26 is R ₂ , and these are the emitter resistances r _e = kT / qI _{E of the} transistors.
And the value of the variable voltage source 2 is 0 V (where k is the Boltzmann constant, T is the absolute temperature, q is the elementary charge, and _IE is the emitter current of the transistor). .

When the switch 6 is tilted to the left, the current source 5
As a result, the current I ₁ flows through the transistor 7, and the current I ₁ also flows through the collectors of the transistors 11, 19 and 20 which form a current mirror with the current I ₁ . On the other hand, since no current flows through the transistor 8, no current also flows through the transistors 12, 23 and 24 which form a current mirror with the transistor 8. Therefore, the first differential amplifier including the transistors 17 and 18 operates, but the second differential amplifier including the transistors 21 and 22 does not operate.

At this time, the alternating current i _o1 output from the collector of the transistor 18 is i _o1 = (v _i −v _o ) / R
_{Is 1} . This i _o1 is due to the transistors 15 and 16.
Since the variable voltage source 2 is 0V, the base voltages of the transistors 15 and 16 are equal, the shunt ratio is 1: 1, and the AC currents flowing in the transistors 15 and 16 are i _o1 / It is 2.

Therefore, the collector of the transistor 16 and P
AC current i output from the contact of the NP transistor 14
_{Since the} AC current supplied from the collector of the transistor 14 is zero, the AC current output from the collector of the transistor 16 is all i _o = i _o1 / 2 = (v
_i −v _o ) / 2R ₁ . Since the input voltage of this differential amplifier is (v _i −v _o ), this differential amplifier has a current i _o = (v _i −v _o ) / with respect to the input (v _i −v _o ).
It is a voltage-controlled current source that outputs 2R ₁ , and the mutual conductance g _{m1 at} this time is ½R ₁ .

When the switch 6 is tilted to the right, the current I ₁ flows from the current source 5 to the transistor 8, and the transistors 12, 23, 24 forming a current mirror with the current I _1.
Also flows. On the other hand, since no current flows through the transistor 7, the transistor 1 that forms a current mirror with this does not flow.
No current flows through 1, 19, 20 and the first differential amplifier does not operate. Therefore, according to the same idea as when the switch 6 is tilted to the left, the output current i _o is i _o = (v _i −v
_o ) / 2R ₂ and the mutual conductance g _{m2 at} this time is 1 / 2R ₂ .

Also, the mutual conductance g _{m1 of} this circuit
Alternatively, g _m2 can be changed by changing the value of the variable voltage source 2 in both positive and negative directions centering on 0V. When the variable voltage source 2 is sufficiently low in the negative direction, the transistor 16 is cut off and no current flows. Therefore, at this time, the mutual conductance g _m is zero.

On the contrary, when the variable voltage source 2 is sufficiently high in the positive direction, i _o1 or i _o2 all flow through the transistor 16,
i _o = i _o1 = (v _i −v _o ) / R ₁ or i _o = i _o2
= Become _{(v i -v o) / R} 2, the value of g _m in this case is 1 / R ₁ or 1 / R _2. Therefore, g _m1 is 1/2
In the range of 0 to 1 / R ₁ centered on R _1, g _m2 is 1 / 2R ₂
It is possible to change it in the range of 0 to 1 / R ₂ centering on.

[0032] The present embodiment as described above, that the switching signal inputted from the terminal 33, it is possible to switch the value of the transconductance g _m, changing the value of g _m by changing the variable voltage source 2 You can

It should be noted, by changing the variable voltage source 2, focusing on direct current even I _1/2 sucked from the collector of the transistor 16, the DC varies to 0 to I _1, which is sucked from the collector of the transistor 10 The current also changes, and thus the direct current flowing through the PNP transistor 13 also changes. Since the PNP transistors 13 and 14 form a current mirror, the direct current flowing from the collector of the transistor 14 also changes.

[0034] That is, until 0 to I ₁ around the I _1/2, exactly changed in the same way as direct current drawn from the collector of the transistor 16. Therefore, the current output from the contact between the collector of the transistor 14 and the collector of the transistor 16 is only the alternating current i _o , and no direct current is output. Therefore, the base DC voltage of the transistor 17 or 21 and the base DC voltage of the transistor 18 or 22 are always equal, and even if the variable voltage source 2 is changed, the output voltage does not change with respect to the input voltage.

FIG. 2 is a circuit diagram showing an equivalent circuit of the embodiment shown in FIG. In FIG. 2, a voltage controlled current source (variable conductance amplifier) 101 is the current source 5, switch 6, transistors 7 to 24, resistor 25,
26 is represented collectively as an equivalent circuit. For simplicity, the emitter follower and the level shift circuit including the transistor 28 are omitted.

When the transfer function is calculated by the equivalent circuit of FIG. 2, H (s) = (g _m1 / C) / (s + g _m1 / C) or H (s) = (g _m2 / C) / (s + g _m2 / C). That is, this is a first-order LPF (low-pass filter), and the transfer function can be selected by the switching signal input from the terminal 33. Therefore, the cutoff frequency can be switched to either f _c = gm ₁ / 2πC or f _c = gm ₂ / 2πC. Moreover, since the mutual conductance gm ₁ or gm ₂ can be arbitrarily changed by the variable voltage source 2, variations in the resistance value and the capacitance value can be compensated by controlling the variable power source 2.

In this embodiment, the positive side input terminal of the voltage controlled current source (variable conductance amplifier) 101 is grounded, and the input signal source 4 is connected to the output terminal of the voltage controlled current source 101 via the capacitor 27. By HPF
(High-pass filter), and the cutoff frequency can be switched like the LPF.

Next, FIG. 3 is a circuit diagram in which the present invention is applied to a primary phase shift circuit. That is, in the embodiment shown in FIG. 2, the phase inversion circuit 102 is provided, one end of the capacitor 27 is the output end of the phase inversion circuit 102, and the other end is the output of the voltage controlled current source 101 as in the case of FIG. The transfer function is H (s) = (-s + gm
_{1 / C) / (s +} gm 2 / C) or, H (s) = (-s
+ Gm ₂ / C) / (s + gm ₂ / C), which works as a phase shift circuit.

[0039] In this case, for sufficiently low frequencies than the cutoff frequency can be used as a delay circuit, when the delay time tau _d, tau _d is τ _d = 2 (C / g
m ₁ ) or τ _d = 2 (C / gm ₂ ), and terminal 3
Either delay time can be selected by the switching signal input from the 3rd step.

FIG. 4 is a circuit diagram in which the present invention is applied to a secondary phase shift circuit. That is, the secondary phase shift circuit shown in FIG.
It can be used as a delay circuit up to a frequency much higher than that of the primary phase shift circuit shown in FIG. 3, and its delay time is similar to that of the primary phase shift circuit shown in FIG. 3, and the delay time can be switched. .

In the circuit shown in FIG. 4, a BPF (bandpass filter) and a trap circuit can be realized by changing the connection position of the input signal 4 and the connection position of the capacitor 27 (as a matter of course, a secondary circuit). LPF and HPF are also feasible). It is also possible to construct a third-order or higher-order filter.

FIG. 5 is a circuit diagram showing another embodiment of the present invention. 5, parts common to those in FIG. 1 are designated by the same symbols as those in FIG. Reference numeral 1 is an independent voltage source that supplies a power supply voltage, 2 is a variable independent voltage source that is input to the bases of the transistors 9, 10 and 15, 16 that form a differential pair, and 3 is an independent voltage source that biases the transistors. Reference numeral 4 denotes an independent voltage source that generates an input signal.

The input signal source 4 is the base of the transistor 17.
The transistor 17 is connected to the resistor 25 and the collector of the current source transistor 19. Further, the transistor 17 is connected to the emitter of the transistor 18 and the collector of the current source transistor 20 via the resistor 25 to form a differential amplifier, the collector of the transistor 17 is connected to the independent voltage source 1, and the collector of the transistor 18 is It is connected to the emitters of transistors 15 and 16 forming a differential pair.

The base of the transistor 15 is the variable voltage source 2
, The base of the transistor 16 is connected to the other end of the variable voltage source 2, the collector of the transistor 15 is connected to the independent voltage source 1, and the collector of the transistor 16 is connected to the collector of the PNP transistor 14. The output current i _{o of the} differential amplifier is output from the connection point between the collector of the transistor 16 and the collector of the transistor 14.

The connection point between the collector of the transistor 16 and the collector of the transistor 14 is the capacitors 40 and 41.
Is connected to the base of the transistor 28, the collector of the transistor 28 is connected to the independent voltage source 1, and the emitter is connected in series to three diodes 29 and 3.
Via 0 and 31, it is connected to the base of the transistor 18 forming the differential amplifier and the current source 32.

The current source 32 is a bias current source for driving the transistor 28 and the diodes 29, 30, 31. The current source 5 is connected to the collector of the transistor 7, the collector and base of the transistor 7 are connected, and this connection point is connected to the bases of the transistors 11, 19 and 20, which are current mirrors. -Constitutes a circuit. The collector of the transistor 11 is connected to the emitters of the transistors 9 and 10 forming a differential pair, the collector of the transistor 9 is connected to the independent voltage source 1, and the collector of the transistor 10 is connected to the collector of the PNP transistor 13.

The bases of the transistors 9 and 10 are the same as the bases of the transistors 15 and 16 which also constitute the differential amplifier, that is, the base of the transistor 9 is one end of the variable voltage source 2. In addition, the base of the transistor 10 is connected to the other end of the variable voltage source 2. The collector of the PNP transistor 13 is also connected to the base of the transistor 13, and this connection point is connected to the base of the PNP transistor 14, so that the transistors 13 and 14 form a current mirror circuit.

The current source 43 is connected to the collector of the transistor 38 or 39 through the switch 44, the collectors of the transistors 38 and 39 are also connected to their respective bases, and the base of the transistor 38 is the transistor 36.
, And the base of the transistor 39 is connected to the base of the transistor 37, respectively, and these respectively form a current mirror circuit.

The DC voltage source 42 is connected to the bases of the transistors 34 and 35, and their collectors are connected to the independent voltage source 1. The emitter of the transistor 34 is connected to the collector of the current source transistor 36 and the transistor 3
The emitter of 5 is the collector of the current source transistor 37,
Connected to each other, these are emitter followers as seen from the voltage source 42.

Further, the emitter of the transistor 34 is connected to the collector of the transistor 16 and P via the capacitor 40.
Connected to the connection point of the collector of the NP transistor 14,
The emitter of the transistor 35 is connected to the connection point of the collector of the transistor 16 and the collector of the PNP transistor 14 via the capacitor 41. Reference numeral 33 is an input terminal of a signal for controlling the switch 44.

The emitters of the transistors forming the current mirror are all grounded in the case of the NPN transistor and all connected to the voltage source 1 in the case of the PNP transistor.

Next, the operation of the present embodiment will be described. The first characteristic control means used for switching the filter characteristic is as follows.
It is first described that the second characteristic control means, which is the switching means of the switch 44 and is used for compensating the variation in the filter characteristics, is the variable control in the variable independent voltage source 2.

Referring to FIG. 5 to 25 constitute a voltage controlled current source, and the point different from the voltage controlled current source in the embodiment of FIG. 1 is that there is a switch 6 and a second differential amplifier composed of 21 to 24 and 26 in FIG. Do not do it. Therefore, the transconductance g _m of this voltage controlled current source shown in FIG. 5 is g _m = 1 when the value of the resistor 25 is R.
By changing the variable voltage source 2 centering on / 2R, it is possible to change in the range of 0 to 1 / R, but it is impossible to switch g _m as in the embodiment of FIG. Is.

FIG. 6 is a circuit diagram showing an equivalent circuit of the embodiment shown in FIG. The voltage-controlled current source (variable conductance amplifier) 103 in FIG. 6 represents the parts 5 to 25 in FIG. 5 collectively as an equivalent circuit. This will be described below with reference to FIG. When the switch 44 is tilted to the left, the current I ₂ flows from the current source 43 to the transistor 38, and the same current also flows through the transistor 36 forming a current mirror with the current I ₂ .

On the other hand, since no current flows in the transistor 39, the transistor 37 which forms a current mirror with this does not flow.
However, no current flows. Therefore, the transistor 34 is turned on, but the transistor 35 is not turned on and is in a cutoff state. Therefore, the output current i _o = g _{m of the} voltage controlled current source 103
All of (v _i −v _o ) flows to the capacitor 40. The transistor 34 functions as an emitter follower, and its base voltage is the DC voltage from the DC voltage source 42.
The emitter voltage also becomes direct current, and is therefore equivalent to ground in terms of alternating current.

When the capacitance value of the capacitor 40 is C ₁ ,
The output voltage at this time is v ₀ = i _o / sC ₁ , and the transfer function is H (s) = (g _m / C ₁ ) / (s + g _m /
C ₁ ). On the other hand, when the switch 44 is tilted to the right, the operation is the reverse of the above, and a current flows through the transistor 39, which forms a current mirror with the transistor 37.
The same current flows through.

On the other hand, since no current flows through the transistor 38, no current flows through the transistor 36 which forms a current mirror with the transistor 38. Therefore, the transistor 35 is turned on, but the transistor 34 is in the cutoff state. Therefore, at this time, the output current i _o of the voltage controlled current source 103 is entirely
And the output voltage is v _o = i _o / sC ₂ where C ₂ is the capacitance value of the capacitor 41. The transfer function in this case is H (s) = (g _m / C ₂ ) / (s + g _m / C ₂ ).
Becomes

As described above, the present embodiment shown in FIG. 6 is a first-order LPF, and its cutoff frequency is f _c = g _m / 2πC ₁ by controlling the switch 44, or
It is possible to switch to f _c = g _m / 2πC ₂ . or,
Variations in resistance value and capacitance value can be compensated by controlling the variable voltage source 2.

FIG. 7 is a circuit diagram in which this embodiment is applied to an HPF. In this case, the transfer function is H (s) = s / (s
+ G _m / C ₁ ) or H (s) = s / (s + g _m / C
₂ ), the cutoff frequency can be switched as in the example of FIG.

FIG. 8 is a circuit diagram in which this embodiment is applied to a phase shift circuit, and 102 in FIG. 8 is the same as the phase inversion circuit in FIG. The circuit shown in FIG. 8 can be used as a delay circuit for frequencies sufficiently lower than the cutoff frequency. The transfer function in this case is H (s) =
_{(-S + g m / C 1} ) / (s + g m / C 1) or, H (
s) = (− s + g _m / C ₂ ) / (s + g _m / C ₂ ), and at a frequency sufficiently lower than the cutoff frequency, the delay time τ _d is τ _d = 2 (C ₁ / g _m ) Or τ _d = 2 (C ₂ / g _m ), and switching can be performed by controlling the switch 44.

FIG. 9 is a circuit diagram in which this embodiment is applied to a secondary phase shift circuit, and it is possible to use it as a delay circuit up to a frequency much higher than that in the case of FIG. The delay time in this case is also the same as in the case of FIG. 8 and can be switched.

Further, the embodiment shown in FIG. 6 can also realize a BPF and a trap circuit by using a second-order filter (as a matter of course, a second-order LPF and HPF can also be realized). It is also possible to construct a filter of higher order than the third order.

[0063]

According to the present invention, the filter characteristics such as the cut-off frequency can be switched, and after the switching, there is a variation in the resistance value and the capacitance value in the IC that constitutes the filter circuit. If there are characteristic variations, there is an advantage that a filter circuit that can compensate for these variations can be provided.

In particular, in the case of a television receiver capable of receiving both a high-definition broadcast television signal and a standard television signal which has been double-scanned, which television signal is to be received? Also,
It is useful to use the filter circuit according to the present invention when it is desired to properly perform image quality correction (for example, contour correction).

[Brief description of drawings]

FIG. 1 is a circuit diagram showing an embodiment (primary LPF) of the present invention.

FIG. 2 is a circuit diagram showing an equivalent circuit of the embodiment shown in FIG.

FIG. 3 is a circuit diagram showing a primary phase shift circuit configured as an application example of the embodiment shown in FIG. 1 by using the equivalent circuit of FIG.

4 is a circuit diagram showing a secondary phase shift circuit configured as an application example of the embodiment shown in FIG. 1 by using the equivalent circuit of FIG.

FIG. 5 is a circuit diagram showing another embodiment (primary LPF) of the present invention.

FIG. 6 is a circuit diagram showing an equivalent circuit of the embodiment shown in FIG.

7 is a circuit diagram showing a primary HPF configured as an application example of the embodiment shown in FIG. 5 by using the equivalent circuit of FIG.

8 is a circuit diagram showing a primary phase shift circuit configured as an application example of the embodiment shown in FIG. 5 by using the equivalent circuit of FIG.

9 is a circuit diagram showing a secondary phase shift circuit configured as an application example of the embodiment shown in FIG. 5 by using the equivalent circuit of FIG.

FIG. 10 is a circuit diagram showing an example of a conventional filter circuit.

[Explanation of symbols]

1-3, 42 ... Independent voltage source, 4 ... Input signal source, 5, 3
2, 43 ... Independent current source, 6, 44 ... Switch, 7-1
2, 15 to 24 ... NPN transistors forming a voltage controlled current source, 13, 14 ... PN forming a voltage controlled current source
P-transistors, 25, 26 ... Resistors for determining g _m of the voltage-controlled current source, 27, 27 ', 40, 41, 4
0 ', 41' ... Capacitor, 28 ... NPN transistor for impedance conversion, 29-31 ... Diode for level shift, 33 ... Terminal for inputting signal for controlling switch, 34-39, 34 '~ 37' ... NPN transistor forming a circuit for switching capacitors,
101, 101 '... voltage controlled current source can be switched g _m, 102 ... phase inverter, 103, 103'
… Voltage controlled current source.

 ─────────────────────────────────────────────────── ─── Continuation of the front page (72) Inventor Chiharu Ishino 292 Yoshida-cho, Totsuka-ku, Yokohama-shi, Kanagawa Prefecture Hitachi Ltd. AV equipment division

Claims (2)

[Claims] 1. A filter circuit comprising a combination of a variable conductance amplifier as a voltage controlled current source and a capacitor, wherein at least a first load and a second load are used as a differential amplifier constituting the variable conductance amplifier. 2 in common
A first characteristic control means for switching the mutual conductance of the variable conductance amplifier to switch filter characteristics such as a cutoff frequency according to which one of the differential amplifiers is selected and operated. By variably controlling the output current of the differential amplifier on the selected and operating side of the two differential amplifiers,
A second characteristic control unit that variably controls the mutual conductance of the variable conductance amplifier to vary a filter characteristic such as a cut-off frequency. 2. A filter circuit comprising a combination of a variable conductance amplifier as a voltage controlled current source and a capacitor, wherein at least a first and a second capacitor are provided as the capacitor.
The first characteristic control means for switching the filter characteristics such as the cutoff frequency and the output current of the differential amplifier constituting the variable conductance amplifier are provided depending on which one of the capacitors is provided and which is selected to function. A second characteristic control unit that variably controls the mutual conductance of the variable conductance amplifier to vary a filter characteristic such as a cut-off frequency.