JPH06292361A - Resonance-type dc-dc converter - Google Patents

Abstract

PURPOSE:To prevent the breakdown of a switching element due to the recovery current of a diode by connecting an inductance between the connection point of the first and second switching elements of an inverter and/or the connection point of third and fourth switching elements and the neutral point of a DC power supply. CONSTITUTION:An inductance is connected, as an auxiliary circuit, either or both between the connection point of first and second switching elements 20a and 20b of an inverter 4 and the neutral point of a DC power supply 1 and between the connection point of third and fourth switching elements 20c and 20d and the neutral point of the DC power supply, 1 thus constantly maintaining the operation mode of phase difference so tat the current flowing to arms 10a-10d of the inverter 4 is negative on turn-on and positive on turn-off and hence preventing the breakdown of the switching elements 20a-20d due to the recovery current of diodes 3a-3d.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a phase difference control type resonance type D which sends an AC voltage from a DC power source through an inverter to a transformer, rectifies its output and supplies the DC voltage to a load.
In particular, the present invention relates to a C-DC converter, which reduces loss in the switching element to achieve high efficiency, reduces the rate of change of voltage applied to each switching element of the inverter, and reduces noise. The present invention relates to a resonance type DC-DC converter designed to prevent switching elements from being damaged due to a short circuit of a lossless snubber circuit at the start of operation.

[0002]

2. Description of the Related Art As one of DC power supply devices used for X-ray equipment, a DC-type power supply device has been used for downsizing and weight saving and high efficiency.
A switching type power supply device called a DC converter is widely used. However, since the switching type power supply device that is widely used at present is basically PWM-controlled, the voltage and current waveforms to be handled are square waves, and there are the following problems.

(1) Switching noise is likely to occur.

(2) The loss in the switching element during switching is large.

(3) The upper limit of the operating frequency of the converter largely depends on the switching time of the switching element.

As a solution to these problems, in recent years, a DC-DC converter which reduces the load on the switching element at the time of switching by inserting a resonant element into a part of the converter to make the voltage waveform or the current waveform sinusoidal. Development is in progress. There is a phase difference control method as a method for controlling the output voltage of such a resonance type DC-DC converter, and as a resonance type DC-DC converter using this type of conventional control method, Japanese Patent Laid-Open No. 63-190556. Some are described in the official gazette.

This resonance type DC-DC converter is shown in FIG.
As shown in, the DC power supply 1 and the positive electrode of the DC power supply 1 +
Has a first series connection body composed of a first switching element 2a connected to the negative pole and a second switching element 2b connected to the negative pole −, and a third series connected to the positive pole +
And a second series connection body parallelly connected to the first series connection body, the second series connection body being composed of the switching element 2c and the fourth switching element 2d connected to the negative electrode.
An inverter 4 having first to fourth diodes 3a to 3d respectively connected in antiparallel to the fourth switching elements 2a to 2d and converting the direct current from the direct current power supply 1 into an alternating current.
, An inductance 5 and a capacitance 6 connected in series on the output side of the inverter 4, a transformer 7 connected in series with the inductance 5 and the capacitance 6 and insulated from the output, and the output of the transformer 7 is converted into a direct current. A rectifier 8 for supplying a direct current output to the load, and a voltage applied to the load 9 connected to the output side of the rectifier 8 and a setting signal for the current flowing through the load 9 to set the first to fourth switching elements 2a. .About.2d on / off timing control means (not shown), and supplies a DC output to the load at a desired voltage and current.

In FIG. 6, the first to fourth switching elements 2a to 2d and the diodes 3a to 3d respectively include a first arm 10a and a second arm 10 respectively.
b, the third arm 10c, and the fourth arm 10d are configured. Further, the rectifier 8 is configured so that the four diodes 11a, 11b, 11c, 11d perform full-wave rectification of the input voltage. Further, reference numeral 12 denotes the capacitance of the high voltage cable for applying the output voltage of the rectifier 8 to the load 9, and functions as a capacitance that smoothes the output voltage from the rectifier 8.

Next, the operation of the conventional resonance type DC-DC converter configured as described above will be briefly described with reference to FIG. In FIG. 7, (a), (b),
(C) and (d) are the ON and OFF periods of the first switching element 2a, the fourth switching element 2d, the second switching element 2b, and the third switching element 2c of the inverter 4 shown in FIG. 6, respectively. Is shown. And
As is apparent from FIG. 7, the first switching element 2a and the fourth switching element 2d are turned on with a phase difference α, and the second switching element 2b and the third switching element 2d are turned on.
The switching element 2c is also turned on with a phase difference α. Furthermore, the first switching element 2a
The second switching element 2b and the third switching element 2c and the fourth switching element 2d are alternately turned on with a phase difference of 180 °.

In the above operation, the periods (Tb3 to Tb4) shown in (a) and (b) in which the first switching element 2a and the fourth switching element 2d are simultaneously turned on, and (c) and (d). ) 2nd switching element 2b
Since electric power is supplied from the DC power supply 1 shown in FIG. 6 only during a period (Tb6 to Tb7) in which the third switching element 2c and the third switching element 2c are simultaneously turned on, the output power waveform Vt of the inverter 4 is shown in FIG. 7 (j). Thus, the voltage is a square wave having positive and negative peak values only during the above period.

Therefore, the first switching element 2a
Difference α between the second switching element 2b and the fourth switching element 2d or the second switching element 2b and the third switching element 2d
By changing the phase difference α with respect to c, it is possible to change the period in which the respective switching elements 2a to 2d are simultaneously turned on, and it is possible to control the power supplied to the load 9 shown in FIG.

The relationship between the output voltage Vt and the phase difference α between the corresponding switching elements in this case is shown in FIG. In FIG. 8, the horizontal axis represents the phase difference α and the vertical axis represents the output voltage Vt to the load 9, and the phase difference α and the output voltage Vt.
The relation with t is the resistance value R1, R2, R3 (R1>
6 is a graph showing a predetermined curve with R2> R3) as a parameter.

In the X-ray high voltage device using the X-ray tube as the load 9, the voltage applied to the X-ray tube (tube voltage) is required to rise quickly. For this reason, a method of starting X-ray irradiation in a state where the voltage of the DC power supply 1 (normally obtained by rectifying a commercial power supply) in FIG. 3 is set to a certain constant value is adopted.

[0014]

The problems in the conventional resonance type DC-DC converter having the above-mentioned structure will be described below. First, a first arm 10a composed of a first switching element 2a whose turn-on is not delayed and a first diode 3a antiparallel-connected thereto, and a second switching element 2b and a first switching element 2b antiparallel-connected thereto. Consider the operation of the second arm 10b consisting of two diodes 3b.

As shown in FIG. 7E, the first arm 1
The current I1 flowing through 0a is the first switching element 2a.
It is negative at the time point Tb1 when the ON signal is input to. Therefore, at this time, the first switching element 2a
The voltage applied to is only the ON voltage of the first diode 3a and is almost zero. The loss of the switching element 2a when the current changes from negative to positive and the current starts to flow through the first switching element 2a is zero because it is the product of the voltage and the current at that time. However, the first
At the time Tb4 when the switching element 2a of FIG. 7 is turned off, the current I1 flowing through the first arm 10a is
It is positive as shown in (e).

The operation until the first switching element 2a starts to turn off and the current becomes zero is shown in FIG. 9. As shown in this figure, the voltage of the switching element 2a before the current becomes zero. As a result, the current and voltage cause a loss in the first switching element 2a corresponding to a shaded area. Such an operation is also the same for the second arm 10b.

In order to reduce the loss as described above, for example, as shown in FIG. 10A, a capacitance 14 and a resistor 15 are connected in series, or as shown in FIG. 10B, the capacitance 14 and the resistor 15 are connected in series. A circuit called a lossless snubber circuit having a configuration in which 15 and a diode 16 are combined is used by being connected in parallel to the switching element 13 such as a transistor.

When such a lossless snubber circuit is provided in parallel with the first and second switching elements 2a and 2b, the rise of the voltage when each switching element 2a and 2b is turned off is suppressed, and at the time of turn off. It was possible to reduce the switching loss.

However, in the lossless snubber circuit as described above, when the switching element 13 shown in FIG. 10 is off, the charge accumulated in the capacitance 14 is turned on when the switching element 13 is turned on. And is discharged through the resistor 15,
The resistor 15 causes a loss. And this resistance 15
Is for controlling the maximum value of the current at this time, an excessive current flows without the resistor 15 and the switching element 1
3 will be destroyed.

Since the loss due to the resistor 15 occurs each time the switching element 13 is repeatedly turned on and off, in the inverter 4 shown in FIG. 6, the loss of each switching element 2a, 2b becomes equal to the operating frequency of the inverter 4. Increase in proportion. In particular, in such a resonance type DC-DC converter, it is common to increase the operating frequency in order to reduce the size and weight of the device, resulting in an extremely large switching loss.

Next, in the configuration and operation shown in FIGS. 6 and 7, the third switching element 2 whose turn-on is delayed is delayed.
c and a third arm 10c composed of a third diode 3c connected in anti-parallel thereto, and a fourth switching element 2d and a fourth diode 3d connected in anti-parallel thereto.
Consider the operation of the fourth arm 10d consisting of Figure 7
In the example shown in FIG. 7, the current I4 of the fourth arm 10d becomes zero at time Tb5 within the period Tb3 to Tb6 during which the ON signal of the fourth switching element 2d shown in FIG. f)), a negative current flows after that time Tb5. That is, a current flows through the fourth diode 3d that is connected in anti-parallel in the fourth arm 10d. After that, at time Tb6, the ON signal to the fourth switching element 2d disappears as shown in FIG. 7B, and the third switching element 2c starts to turn on as shown in FIG. 7D. As a result, the current that has been flowing through the fourth diode 3d until then is commutated to the third switching element 2c, and the fourth diode 3d is reverse biased and turned off.

However, at this time, the fourth diode 3
d cannot be instantly turned off, and a current called a recovery current flows through the diode 3d until the junction capacitance of the PN junction is charged. Therefore, while the recovery current is flowing, the third arm 10 shown in FIG.
c and the fourth arm 10d are equivalent to a short-circuited state, and an excessive current flows to increase the switching loss, as well as the third and fourth switching elements 2c, 2
Sometimes d was destroyed. Such an operation is the same when the third switching element 2c is turned off.

As described above, in the phase difference control in the conventional resonance type DC-DC converter, the operation of the first and second arms 10a and 10b of the inverter 4 shown in FIG. 6 and the operation of the third and fourth arms are performed. This is different from the operation of the arms 10c and 10d. In the first and second arms 10a and 10b, the switching loss due to the lossless snubber circuit (see FIG. 10) increases or the third and fourth arms 10c and 1d.
At 0d, there has been a problem that the switching elements 2c, 2d are destroyed by arm short circuit due to the recovery current of the diodes 3c, 3d connected in antiparallel to the switching elements 2c, 2d.

Besides, each of the switching elements 2a-2
Since the voltage applied to d is large, each switching element 2a-2
Since the time change rate of the current flowing through d is large, the electromagnetic interference noise generated becomes large, which may adversely affect the control system. In particular, such a resonance type DC-DC
When the converter is used as a power supply device such as an X-ray CT device (X-ray computed tomography device) as an X-ray device, a diagnostic image is formed in the immediate vicinity of the inverter 4 that switches several hundred volts and several hundred amperes. Therefore, it is necessary to detect a very small signal (several microvolts), and the influence of the electromagnetic interference noise becomes very large, and there has been a strong demand for improvement.

The present invention addresses such a problem, the change rate of the voltage applied to each switching element of the inverter is small, the electromagnetic interference noise can be reduced, and the loss in the switching element is reduced to improve the efficiency. Resonant DC-D that can be achieved and can prevent the breakdown of the switching element due to the recovery current of the diode and the short circuit of the lossless snubber circuit at the start of the inverter operation.
An object is to provide a C converter.

[0026]

The above problems are as follows. (1) The current flowing through each switching element of the inverter becomes negative at the time of turn-on (the current is flowing through the diode connected in antiparallel to the switching element), and , Maintaining an operation mode of phase difference and frequency that flows in the positive direction at turn-on, (2) in parallel with each switching element in order to suppress a sharp rise in the voltage applied to the switching element when the switching element is off. Connect a lossless snubber circuit with only capacitance. (3) Set the phase difference to 180 ° from the time when the inverter input voltage is zero, and use the first switching element and the third switching element during the first half of the inverter cycle. Of the second switching element and the fourth switching element during the latter half cycle at the same time.
By turning on the switching elements at the same time and repeating this operation until the DC output supply to the load is started, the voltage charged in the lossless snubber capacitance at the start of inverter operation becomes almost zero and the overcurrent flowing in the switching element It is solved by suppressing.

The object of the present invention is to provide the above (1), (2),
This is achieved by performing switching (called soft switching) control that satisfies the condition (3). That is,
A direct current power supply having a neutral point, a first series connection body composed of a first switching element connected to the positive electrode of the direct current power supply and a second switching element connected to the negative electrode, and connected to the positive electrode. A third switching element and a fourth switching element connected to the negative electrode, and a second series connection body connected in parallel to the first series connection body, and the first to fourth switching elements. An inverter for converting direct current from the direct current power source into alternating current, which has first to fourth diodes respectively connected in anti-parallel to
It has an inverter output circuit including at least a transformer connected to the output side of this inverter, and a rectifier connected to the transformer and converting its output to direct current, and connected to the output side of the rectifier. In a resonance type DC-DC converter that supplies a DC output to a load at a desired voltage and current, first to fourth lossless snubber capacitances respectively connected in parallel to the first to fourth switching elements, and the first to fourth lossless snubber capacitances. Between the connection point of the first and second switching elements and the neutral point of the DC power supply, and the third and fourth
An auxiliary circuit having an inductance connected to either or both of the connection point of the switching element and the neutral point of the DC power source, and the first and second switching elements are at a 180 ° position. The switching elements are alternately turned on with a phase difference, the third and fourth switching elements are also turned on with a phase difference of 180 °, and the ON time of the switching element is slightly less than 1/2 of the operating cycle of the inverter. A small dead time is provided between the ON times of the first and second switching elements and between the ON times of the third and fourth switching elements, and the first switching element is turned on. Phase difference at which the fourth switching element is turned on and the third switching element is turned on after the second switching element is turned on The electric power supplied to the load is controlled by changing the phase difference of each of the switching elements, and at the time when the ON signal is input to the first to fourth switching elements, the first to fourth diodes are energized. The first to fourth switching elements are driven with a frequency and a phase difference that are equal to the above, and the above-mentioned level is maintained in the operation start period from the rise of the voltage of the DC power supply to the output DC voltage of the rectifier reaching a predetermined voltage. This is achieved by providing a control means for driving each of the switching elements in a state where the phase difference is maximum.

Further, it is achieved by providing control means for short-circuiting the primary side of the transformer or the output terminals of the inverter during the operation start period.

The value of the capacitance used for the lossless snubber must be set within the range where charging / discharging can be completed during the dead time.

[0030]

According to the above construction, the first and second inverters are provided.
An auxiliary circuit between the connection point of the switching element and the neutral point of the DC power source, and between the connection point of the third and fourth switching elements and the neutral point of the DC power source, or both. As a result, the current flowing in each arm of the inverter becomes negative at the time of turn-on, and the operation mode of the phase difference becomes positive at the time of turn-off, and the operation mode of the phase difference can always be maintained, and the destruction of the switching element due to the recovery current of the diode is prevented. .

Further, since capacitances are connected in parallel as lossless snubber circuits to the first to fourth switching elements of the inverter, the switching elements are charged and discharged by charging and discharging the lossless snubber capacitance during the dead time period. It is possible to realize a soft switching element having a small voltage change rate with time. Therefore, the change rate of the voltage applied to the switching element is small, the noise is reduced, and the loss in the switching element is reduced to improve the efficiency.

Further, when the switching element is turned on, from the time when the voltage of the DC power supply is raised to when the output DC voltage of the rectifier reaches a predetermined voltage, and when the switching element is turned on, the diode connected in anti-parallel connection is always used. Since a current flows through the lossless snubber capacitance, the lossless snubber capacitance is not unnecessarily charged. Therefore, the lossless snubber capacitance is prevented from being short-circuited, the overcurrent of each switching element of the inverter is eliminated, and the destruction of the switching elements is prevented.

[0033]

Embodiments of the present invention will be described below with reference to the drawings. FIG. 1 is a circuit diagram showing an embodiment of a resonance type DC-DC converter according to the present invention. This resonance type DC-D
The C converter serves as a power converter that sends an AC voltage from a DC power source to a transformer via an inverter, rectifies the output of the AC power source, and supplies the DC voltage to a load. As shown in FIG. , Inverter 4, and inductance 5
And a capacitance 6, a high-voltage transformer 7, a high-voltage rectifier 8, an X-ray tube 17 as a load, a frequency phase control circuit 71 as an inverter control means, and a frequency, phase, etc. in the frequency phase control circuit 71. And a drive circuit 72-74 for driving the inverter 4 by receiving a signal from the frequency / phase control circuit 71. It is called a device.

The DC power source 1 is, for example, a secondary battery, and in FIG. 1, two power sources E / 2 are shown symmetrically for the sake of convenience. The inverter 4 receives DC from the DC power supply 1 and converts it into AC, and is connected to the transistor 20 a as the first switching element connected to the positive electrode + of the DC power supply 1 and the negative electrode − of the DC power supply 1. Transistor 20 as a second switching element
and a transistor 20c serving as a third switching element connected to the positive electrode + and a first series connection body made of b.
And a second series-connected body, which is connected in parallel to the first series-connected body and which is composed of a transistor 20d as a fourth switching element connected to the negative electrode, and antiparallel-connected to the transistors 20a to 20d, respectively. It also includes first to fourth diodes 3a to 3d.

The above transistors 20a to 20d are provided.
Are turned on by passing a base current, respectively. And the first transistor 2
0a and the first diode 3a form the first arm 10a.
To the second transistor 20b and the second diode 3b.
To connect the second arm 10b to the third transistor 20c.
And the third diode 3c form a third arm 10c, and the fourth transistor 20d and the fourth diode 3d form a fourth arm 10d.

An inductance 5 is connected to the output side of the inverter 4, and a capacitance 6 is connected in series to the inductance 5. The inductance 5 and the capacitance 6 form a resonance circuit. A primary winding of a high-voltage transformer 7 is connected in series to the inductance 5 and the capacitance 6, and the high-voltage transformer 7 boosts the output voltage from the inverter 4 and isolates it from the output.

The high voltage rectifier 8 is for full-wave rectifying the output voltage from the high voltage transformer 7 and converting it into a direct current, and is composed of four diodes 11a to 11d as shown in FIG. Further, an X-ray tube 17 is connected to the output side of the high voltage rectifier 8 as a load. Reference numeral 12 is a capacitance that smoothes the output voltage of the high voltage rectifier 8.

Here, in the present invention, the first to fourth transistors 20a to 20d of the inverter 4 are
First to fourth used as lossless snubber circuit
Lossless snubber capacitances 22a to 22d are respectively connected in parallel, and between the connection point of the first and second transistors 20a and 20b and the neutral point (potential E / 2) of the DC power source 1 and the third point. Between the connection point of the fourth transistors 20c and 20d and the neutral point of the DC power supply 1, an inductance 23 is provided as an auxiliary circuit.
a and 23b are connected.

The phase determination circuit 70 is divided into first to first by a signal corresponding to a voltage applied to the X-ray tube 17 (hereinafter referred to as a tube voltage) and a current flowing in the X-ray tube 17 (hereinafter referred to as a tube current).
The operating phase of the fourth transistors 20a to 20d is determined. The frequency determining circuit 76 determines the operating frequencies of the first to fourth transistors 20a to 20d according to the signal corresponding to the tube voltage. Frequency phase control circuit 7
1 outputs a signal corresponding to the phase and frequency at which the first to fourth transistors 20a to 20d operate in accordance with the signals from the phase determining circuit 70 and the frequency determining circuit 76 as an X-ray exposure signal. The drive circuits 72 to 75 are arranged in accordance with the signals from the frequency phase control circuit 71, respectively.
It drives the transistors 20a to 20d.

Next, the resonance type DC-configured as described above.
The operation of the DC converter will be described. First, the main circuit components (DC power supply 1, inverter 4, inductance 5, capacitance 6, high-voltage transformer 7, high-voltage rectifier 8, X-ray tube 17) in the resonance type DC-DC converter shown in FIG. The equivalent circuit shown in FIG. That is, the transistors 20a to 20d of the inverter 4 are
Similar to the case shown in FIG. 6, each of the first switching elements 2
a, the second switching element 2b, the third switching element 2c, and the fourth switching element 2d, and the X-ray tube 17 is represented as a load 9. Therefore, by using the equivalent circuit shown in FIG. 2, the principle of operation of the main circuit constituent part is shown in FIG.
And FIG. 4 will be described.

In the equivalent circuit of FIG. 2, attention is paid to the first arm 10a and the second arm 10b of the inverter 4. Then, the inductance 23a is set to the current Ia flowing to the DC power supply 1 side, and the first and second arms 10 are
The current output from the a and 10b to the high voltage transformer 7 side is I
Let r. In this state, the current I1 flowing through the first switching element 2a when the first switching element 2a is on is expressed by I1 = Ia + Ir (1).

Here, since the first switching element 2a and the second switching element 2b are turned on and off with a duty cycle of about 50%, the current I in the steady state is
The waveform of a becomes a triangular wave as shown in FIG.
When the switching element 2a is turned off (FIG. 3 (a))
The current Ia has the maximum value Ia (max). That is,
The current I1 (0) at turn-off is given by I1 (0) = Ia (max) + Ir (0) (2) from the above equation (1). However, Ir (0) means the current Ir at turn-off.

At this time, the gradient of the current Ia is the value La of the inductance 23a and the potential E / 2 at the neutral point of the DC power supply 1.
The maximum value Ia (max) above is also determined by L
It depends on a and E / 2. Therefore, it is possible to always set the magnitude of the current I1 (0) at turn-off to a certain value or more. That is, Ia (max) can be set so that I1 (0) = Ia (max) + Ir (0)> (constant value) (3).

With this setting, the currents flowing through the arms 10a and 10b at the time of turn-on in the switching elements 2a and 2b are negative (reversely connected in parallel to the switching elements 2a and 2b, respectively) as described below. In addition, a current is flowing through the respective diodes 3a and 3b). At this time, as shown in FIGS. 3A and 3B, between the time when the first switching element 2a is turned off and the time when the second switching element 2b is turned on,
A dead time period Td in which both of the switching elements 2a and 2b are turned off is set.

From the above, the dead time period T
Soft switching can be realized by effectively using the capacitances 22a and 22b as the lossless snubber circuit shown in FIG. About this, FIG.
(D) shows that the switching element 2a is turned off (b) from the state (a) in which the first switching element 2a is on,
It shows a mode until the second switching element 2b is turned on (d) after a predetermined dead time (b) to (c).

First, in FIG. 4A, only the first switching element 2a is turned on, and as shown in FIG. 3E, the current Ia increases almost linearly regardless of the polarity of the current Ir. To do. Further, the inclination depends on the value La of the inductance 23a and the power supply potential E / 2. At this time, the voltage Vc1 of the first capacitance 22a = 0 volt,
The voltage Vc2 of the capacitance 22b is equal to E volt.

Next, in FIG. 4B, both switching elements 2a and 2b are turned off. At this time,
As shown in (e), the current Ia = Ia (max), and the current I1 (0) of the switching element 2a at turn-off is sufficiently large according to the setting of this maximum value Ia (max) and the above equation (3). It can be a positive current. Therefore, due to the resonance phenomenon of the inductance 23a and the other inductance 5 as the auxiliary circuit and the capacitances 22a and 22b as the lossless snubber circuit shown in FIG. 2, the capacitance 22a is charged and the capacitance 22b is discharged.

Next, in FIG. 4C, charging / discharging of the capacitances 22a and 22b is completed, the voltage Vc1 of the first capacitance 22a goes from 0 to E volt, and the voltage Vc2 of the first capacitance 22b becomes E. → Change to 0 volts,
The second diode 3b becomes conductive. At this time, the current Ia flowing through the inductance 23a starts to decrease due to the voltage of -E / 2.

Then, in FIG. 4D, an ON signal is given to the second switching element 2b, and the current (Ia + I)
When the polarity of r) is reversed from positive to negative, the second switching element 2b is replaced with the one shown in FIGS.
Proceed as with. Note that changes in the voltages Vc1 and Vc2 of the first and second capacitances 22a and 22b during the operations of FIGS. 4A to 4D are shown in the graph of FIG. The above-described operation is the same on the side of the third arm 10c and the side of the fourth arm 10d of the inverter 4 shown in FIG.

From the above, the current I1 at turn-off is
The value La of the inductance 23a as the auxiliary circuit is set so that (0) is equal to or more than the value required for charging and discharging the first and second capacitances 22a and 22b, and the value is charged during the dead time shown in FIG. 4 (e). By appropriately selecting the values of the capacitances 22a and 22b that will complete the discharge, FIG.
Soft switching can be realized by effectively using the capacitances 22a to 22d as the lossless snubber circuit for all the switching elements 2a to 2d shown in FIG.

In the X-ray high voltage apparatus, since the rising speed of the tube voltage is required to be high, the output voltage of the rectifier circuit (not shown, which corresponds to the DC power supply 1), that is, the inverter 4 is usually used. The exposure is started by operating the switching elements 2a to 2d of the inverter 4 with the input voltage of 1 set to a certain constant value. However, in such an operating method, when the lossless snubber circuit is used, the capacitance 2 of the capacitance 2 at the time of start-up (when the operation of the inverter 4 starts)
2a to 22d are always short-circuited, and switching elements 2a to 2d
Destruction of d may occur.

Therefore, in the present invention, the operation of the inverter 4 is started in a state where the output voltage of the rectifier circuit (DC power supply 1) is zero to the maximum phase difference, and the output voltage of the rectifier circuit thereafter becomes the set voltage. It continues its action until it is reached.
After that, the phase difference of the inverter 4 is reduced in synchronization with the exposure start signal, power is supplied to the load 9, and X-rays are output.

On the basis of the above-described operation principle, the switching elements 2a to 2d are connected to the switching elements 2a through the inductances 23a and 23b which are auxiliary circuits.
Current flows through the diodes 3a to 3d that are connected in anti-parallel to the switching elements 2a to 2d, so that zero volt turn-on is realized in each of the switching elements 2a to 2d, and the lossless snubber capacitance 2
It is possible to prevent a short circuit between 2a to 22d.

According to this method, since the phase difference is maximum before the start of exposure, no current flows in the load 9 (power is not supplied), and only the inductances 23a and 23b, which are auxiliary circuits, receive current. Flow, switching element 2
It can be operated without risk of destroying a to 2d.

Next, returning to the embodiment shown in FIG. 1, the operation of this embodiment will be described with reference to FIGS. 5, 8, 11 and 12. The switching element 2a shown in FIG.
2d correspond to the transistors 20a to 20d in FIG.

First, as can be seen from the time chart of FIG. 11, when the voltage of the DC power supply (inverter input voltage) is raised by the X-ray exposure preparation signal, the first half of the inverter cycle is the first transistor 20a and the first half cycle. The third transistor 20c is turned on at the same time, and the second transistor 20b and the fourth transistor 20 are turned on at the same time in the latter half cycle, and even if the inverter input voltage reaches a steady state, the above operation is repeated until the X-ray irradiation starts. The first to fourth lossless snubber capacitances 20a to 20d are prevented from being charged with voltage.

When the tube voltage and the tube current applied to the X-ray tube 17 as a load are determined, the tube voltage setting signal G1 corresponding to the tube voltage and the tube current setting signal G2 corresponding to the tube current.
Is input to the phase determination circuit 70 and the frequency determination circuit 76.

The phase determining circuit 70 obtains the load resistance value from the tube voltage setting signal G1 and the tube current setting signal G2,
From the load resistance value and the tube voltage, each transistor 20a of the inverter 4 is used by using the relationship of the graph shown in FIG.
The operating phase difference α of ˜20d is determined. The frequency determining circuit 76 determines the operating frequency of the inverter 4 from the tube voltage setting signal G1 using the relationship shown in the graph of FIG.

A signal corresponding to the above operation phase difference and operation frequency is input to the frequency phase control circuit 71, and the frequency phase control circuit 71 uses the signals to output each of the transistors 20a ...
20d creates a drive signal that turns on and off.

Next, an X-ray exposure signal G3 is sent from the controller (not shown in FIG. 1) to the frequency phase control circuit 71.
11 is input to the drive circuit 72, a signal for turning on the transistor 20a of the inverter 4 is output to the drive circuit 72.
The transistor 20a is turned on as shown in FIG. As a result, the + side (upper side in the figure) DC power supply 1 and inductance 2
An auxiliary current flows through the circuit 3a. A signal for turning on the transistor 20d is output to the drive circuit 75 after the time when the phase difference becomes α, the transistor 20d is turned on as shown in FIG. 11, and the − side (lower side in the figure) DC power source 1
At the same time when an auxiliary current flows through the circuit of the inductance 23b and the resonance circuit, a voltage of the sum of the + side (upper side in the figure) DC power source 1 and the − side (lower side in the figure) DC power source 1 is applied,
Resonance current begins to flow. When a half cycle has elapsed and a signal for turning off the transistor 20a and turning on the transistor 20b is output from the frequency phase control circuit 71, the drive circuit 72 is output.
As a result, the transistor 20a is turned off, the voltage across the transistor 20a is suppressed and increased by the lossless snubber capacitance 22a, and the current in the inductance 23a is diverted to the diode 3b that is connected in antiparallel with the transistor 20b. The reversal of the resonant current that is clamped to zero and flowing in the load 17 causes the transistor 20b to turn on at zero voltage.

The transistor 20d is turned on by 180 °,
When a signal for turning this off is output from the frequency phase control circuit 71, the transistor 20d is turned off by the drive circuit 75, and the voltage across the transistor 20d is the lossless snubber capacitance 2.
The current of the inductance 23b is commutated to the diode 3c connected in anti-parallel with the transistor 20c while being suppressed by 2d and rising, the voltage across the transistor 20c is clamped to almost zero, and the resonance current flowing in the load 17 is reduced. The inversion causes transistor 20c to turn on at zero voltage.

As a result, the transistors 20c and 20
b is turned on, a current flows in the resonance circuit in the opposite direction, and thereafter, the transistors 20a to 20d are sequentially driven according to the signal from the frequency phase control circuit 71. As a result, the resonance current It as shown in FIG. 5 (i) flows through the high voltage transformer 7 shown in FIG. 1, and the set tube voltage is applied to the X-ray tube 17 and the tube current flows. At this time, looking at the currents I1 to I4 flowing through the first arm 10a to the fourth arm 10d, as shown in FIGS. 5 (e) to 5 (h), the operations described in FIGS. According to the principle, it can be seen that all of the respective transistors 20a to 20d have a negative value at turn-on and a positive value at turn-off. Further, it can be seen that the time change rate of the voltage applied to each of the transistors 20a to 20d is small as described with reference to FIG. In the time chart shown in FIG. 5, the dead time between the transistors 20a to 20d is omitted for convenience.

The phase determining circuit 70 and the frequency determining circuit 76 can be easily configured by using a memory, a function generator, an operational amplifier or the like in which the relationships shown in FIG. 8 and FIG. 12 are tabulated, and therefore detailed description will be omitted. To do.

FIG. 13 shows a resonance type DC-DC converter according to a second embodiment of the present invention. In this example, a switch (switching element) is provided in the auxiliary circuit, and the description is omitted because it is the same as FIG. 1 except for this auxiliary circuit.

In the above-described first embodiment, during the operation of the inverter 4, the alternating current always continues to flow in the auxiliary circuit. The values of the inductances 23a and 22b of the auxiliary circuit must be designed assuming the worst case of the phase difference and the load condition, but depending on the phase difference and the type of the X-ray tube 17 (load 9),
An unnecessary current will flow, and the transistors 20a ...
There is a wasteful state in terms of the conduction loss of 20d and the loss of the inductances 23a and 22b. Therefore, in the second embodiment, the switching elements 110 to 113 are provided in the auxiliary circuit.
Is provided, and the circuit configuration is such that the current is supplied only in the switching period of the transistors 20a to 20d in accordance with the current state of the transistors 20a to 20d, as much as necessary for realizing the soft switching. With this configuration, it is possible to operate in a more efficient state than in the first embodiment, regardless of the phase difference and load range. In addition, in FIG.
In, 100 to 103 are diodes.

The effective driving method of this auxiliary circuit is as follows. That is, the current when the transistor 20a is turned off is the lossless snubber capacitance 22a.
Since it should be a value more than necessary for charging and discharging of
The switching element 110 is turned on when the transistor 20a is turned off, and the switching element 111 is turned on when the transistor 20b is turned off.
Turning off the switching element 113, and turning off the transistor 20d turns on the switching element 1
It suffices to make 12 conductive. By such a method, an efficient operation can always be realized even for an X-ray high voltage device having an X-ray tube 17 (load range) of a very wide range.

FIG. 14 is a circuit diagram showing a third embodiment of the resonance type DC-DC converter according to the present invention. Here, FIG.
A switch 200 is added to the same configuration as the above.
In this case, the switch 200 has an operation start period from when the voltage of the DC power supply 1 is raised to when the output DC voltage of the rectifier 8 reaches a predetermined voltage (the output DC voltage of the rectifier 8 before the start of X-ray exposure). During the period from when the voltage reaches a predetermined voltage to zero), the primary side of the transformer 7 is short-circuited (Fig. 1).
5).

According to this, regardless of the operation of the frequency phase control circuit 71, there is no room for unnecessary charge to the lossless snubber capacitance during the above period, and the short circuit of the lossless snubber capacitances 22a to 22d is prevented. There is no overcurrent in each of the switching elements 20a to 20d of the inverter 4, and the switching elements 20a to 20d are removed.
The destruction of d is prevented.

FIG. 16 is a circuit diagram showing a specific configuration example of the switch 200. Here, a thyristor 201, 204 connected in anti-parallel is used as a main switch for short-circuiting the primary side of the transformer 7, and the switch 202, 205 is turned on before the start of X-ray irradiation to turn on the thyristor 201. , 204 to make current flow through them to make the primary side of the high-voltage transformer 7 short-circuited, and turn off the switches 202 and 205 by the X-ray exposure start signal G3 to turn off the thyristors 201 and 204. Conduction, X-ray tube 17
Inverter output to (load) (DC voltage output from rectifier 8)
Is to give. In FIG. 16, 203,
Reference numerals 206 respectively indicate protection resistors.

FIG. 17 is a circuit diagram showing a fourth embodiment of the resonance type DC-DC converter according to the present invention, in which the switch 200 is connected between the output terminals of the inverter 4 in this case. However, the same effect as that of the above-described third embodiment can be obtained. Also in the above-described third and fourth embodiments, the diodes 100 to 1 are provided in the auxiliary circuit as shown in FIG. 13, that is, in the inductances 23a and 23b.
03 and switching elements 110 to 113 can be applied as an auxiliary circuit.

In each of the above-mentioned embodiments, the transistor is used as the switching element used in the inverter 4 and the auxiliary circuit. However, the present invention is not limited to this, and a GTO may be used. IGBT,
A switching element such as an SI transistor or SI thyristor may be used. In addition, the DC power supply 1 of the inverter 4
May be a battery or a rectified commercial power source. Further, the load is not limited to the X-ray tube 17 and may be another general load. Further, the auxiliary circuit may be provided only on one side, for example, only on the inductance 23a side when the load range or the range of the phase difference to be controlled is narrow. Further, the frequency phase control circuit 71 is generally a normal proportional-integral control, but it is also possible to convert it once to a digital value and apply control by software.

[0072]

As described above, according to the present invention, between the connection point of the first and second switching elements of the inverter and the neutral point of the DC power source, and the third and fourth switching elements. By connecting an inductance as an auxiliary circuit to either or both of the connection point and the neutral point of the DC power supply, the current flowing through each arm of the inverter becomes negative at turn-on and becomes positive at turn-off. There is an effect that the operation mode can be always maintained and the breakdown of the switching element due to the recovery current of the diode can be prevented.

By connecting capacitors in parallel to the first to fourth switching elements of the inverter as a lossless snubber circuit, each switching element is charged and discharged by charging and discharging the lossless snubber capacitance during the dead time period. It is possible to realize a soft switching element having a small voltage change rate with time. Therefore, there is an effect that the rate of change of the voltage applied to the switching element is small, noise can be reduced, and loss in the switching element can be reduced to achieve high efficiency.

Further, since the lossless snubber capacitance is not unnecessarily charged at the time of starting the operation of the inverter, it is possible to prevent the lossless snubber capacitance from being short-circuited to prevent the overcurrent of each switching element of the inverter and prevent the destruction of the switching elements. There is also an effect that you can.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing an embodiment of a resonance type DC-DC converter according to the present invention.

FIG. 2 is an equivalent circuit diagram of a main circuit configuration unit in the above resonance type DC-DC converter.

FIG. 3 is a time chart for explaining the operating principle of the main circuit configuration unit of the above.

FIG. 4 is a diagram for explaining operation modes of a first switching element and a second switching element in the same main circuit configuration unit.

5 is a time chart for explaining the operation of the circuit shown in FIG. 1. FIG.

FIG. 6 is a circuit diagram of a conventional resonance type DC-DC converter.

FIG. 7 is a time chart for explaining the operation of the conventional resonance type DC-DC converter.

FIG. 8 is a graph showing the relationship between the phase difference and the output voltage in the conventional resonance type DC-DC converter of the phase difference control method, using the load resistance as a parameter.

FIG. 9 is a diagram showing a turn-off waveform when a lossless snubber circuit is not used.

FIG. 10 is a diagram showing an example of a conventional lossless snubber circuit.

11 is a time chart for explaining drive control of each switching element of the inverter from preparation for X-ray irradiation of the X-ray tube in FIG. 1 to steady operation after X-ray irradiation.

FIG. 12 is a diagram showing a relationship between an inverter frequency and an output voltage with a phase difference α as a parameter.

FIG. 13 is a circuit diagram showing a second embodiment of a resonance type DC-DC converter according to the present invention.

FIG. 14 is a circuit diagram showing a third embodiment of a resonant DC-DC converter according to the present invention.

FIG. 15 is a time chart showing the operation of each part of the X-ray tube in FIG. 14 from preparation for X-ray irradiation to normal operation after X-ray irradiation.

16 is a circuit diagram showing a specific example of switches in FIG.

FIG. 17 is a circuit diagram showing a fourth embodiment of a resonant DC-DC converter according to the present invention.

[Explanation of symbols]

1 ... DC power supply 3a-3d ... 1st-4th diode 4 ... Inverter 5 ... Inductance 6,12 ... Capacitance 7 ... High voltage transformer 8 ... High voltage rectifier 10a-10d ... 1st-4th arm 17 ... X-ray tube (load) 20a-20d ...... 1st-4th transistor (1st-4th)
Fourth switching element) 22a to 22d ... Lossless snubber capacitance 23a, 23b ... Inductance (auxiliary circuit) 70 ... Phase determination circuit 71 ... Frequency phase control circuit 72-75 ... Transistor drive circuit 76 ... Frequency Decision circuit 200 ... Switch

 ─────────────────────────────────────────────────── ─── Continued Front Page (72) Inventor Keinobu Hatakeyama 1-1-14 Kanda, Uchida, Chiyoda-ku, Tokyo Inside Hitachi Medical Co., Ltd.

Claims (3)

[Claims] 1. A direct current power supply having a neutral point, and a first series connection body comprising a first switching element connected to a positive electrode of the direct current power supply and a second switching element connected to a negative electrode of the direct current power supply. It has a second series connection body which is composed of a third switching element connected to the positive electrode and a fourth switching element connected to the negative electrode and which is connected in parallel to the first series connection body. Fourth
1 to 1 which are respectively connected in anti-parallel to the switching elements of
An inverter having a fourth diode for converting DC from the DC power supply into AC, an inverter output circuit including at least a transformer connected to the output side of the inverter, and an output connected to the transformer to A rectifier for converting to a direct current, wherein the load connected to the output side of the rectifier supplies a direct current output with a desired voltage and current, the resonance type DC-DC converter comprising: First to fourth lossless snubber capacitances respectively connected in parallel to the switching elements, between the connection point of the first and second switching elements and the neutral point of the DC power source, and the third and fourth points. An auxiliary circuit having an inductance connected to either or both of a connection point of the switching element and the neutral point of the DC power supply; The second switching element is alternately turned on with a phase difference of 180 °, the third and fourth switching elements are also turned on with a phase difference of 180 °, and the ON time of the switching element is the operation of the inverter. It is set to be slightly smaller than 1/2 of the period, and the time between the on-points of the first and second switching elements and the third
And a dead time is provided between the ON times of the fourth switching element and the phase difference in which the fourth switching element is turned on after the first switching element is turned on and the second switching element is turned on. To control the power supplied to the load by changing the phase difference at which the third switching element turns on, and at the time when the ON signal is input to the first to fourth switching elements, ~ The above-mentioned first to second at the frequency and phase difference during which the fourth diode is energized
The fourth switching element is driven to drive each of the switching elements in the maximum phase difference during the operation start period from the rise of the voltage of the DC power supply to the output DC voltage of the rectifier reaching a predetermined voltage. A resonance type DC-DC converter comprising: a control unit for driving. 2. A direct current power supply having a neutral point, and a first series connection body composed of a first switching element connected to a positive electrode of the direct current power supply and a second switching element connected to a negative electrode of the direct current power supply. It has a second series connection body which is composed of a third switching element connected to the positive electrode and a fourth switching element connected to the negative electrode and which is connected in parallel to the first series connection body. Fourth
1 to 1 which are respectively connected in anti-parallel to the switching elements of
An inverter having a fourth diode for converting DC from the DC power supply into AC, an inverter output circuit including at least a transformer connected to the output side of the inverter, and an output connected to the transformer to A rectifier for converting to a direct current, wherein the load connected to the output side of the rectifier supplies a direct current output with a desired voltage and current, the resonance type DC-DC converter comprising: First to fourth lossless snubber capacitances respectively connected in parallel to the switching elements, between the connection point of the first and second switching elements and the neutral point of the DC power source, and the third and fourth points. An auxiliary circuit having an inductance connected to either or both of a connection point of the switching element and the neutral point of the DC power supply; The second switching element is alternately turned on with a phase difference of 180 °, the third and fourth switching elements are also turned on with a phase difference of 180 °, and the ON time of the switching element depends on the operation of the inverter. It is set to be slightly smaller than 1/2 of the period, and the time between the on-points of the first and second switching elements and the third
And a dead time is provided between the ON times of the fourth switching element and the phase difference in which the fourth switching element is turned on after the first switching element is turned on and the second switching element is turned on. To control the power supplied to the load by changing the phase difference at which the third switching element is turned on, and at the time when the ON signal is input to the first to fourth switching elements, ~ The above-mentioned first to second at the frequency and phase difference during which the fourth diode is energized.
The fourth switching element is driven, and the primary side of the transformer or the output terminals of the inverter are connected to each other in the operation start period from the rise of the voltage of the DC power supply to the time when the output DC voltage of the rectifier reaches a predetermined voltage. Resonant D comprising: a control means for driving each of the switching elements with the phase difference being maximized while being short-circuited.
C-DC converter. 3. The resonance type DC-DC converter according to claim 1, wherein the load is an X-ray tube.