JPH06225538A - Inverter device - Google Patents

Abstract

PURPOSE:To provide an inverter device in which the rush current at power on does not flow, the input power factor is high, and a high frequency current can be suppressed, and the number of switching elements is small, and which is small-light, low-priced, little in occurrence of electric noise, and high in circuit efficiency. CONSTITUTION:The output of a full-wave rectifying circuit DB is intermitted with a switching means S1, and the energy accumulated in an inductor L1 is discharged to a capacitor C1 through a switching means S2. The current flowing from the power source into the capacitor C1 is intermitted with a switching means S3. A load current is made a resonant current by means of an inductor L2 and a capacitor CO. Hereby, the input current distortion is reduced, and the input power factor can be raised. The rush current at power on does not flow. High frequency noise is reduced, and switching loss is reduced, and circuit efficiency becomes high, and a heat sink becomes needless, whereby it can constituted in small size at low cost.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching type inverter device using an AC power source as an input,
More specifically, it relates to prevention of inrush current at power-on, improvement of input power factor, and measures against harmonic distortion of input current.

[0002]

2. Description of the Related Art FIG. 19 is a circuit diagram of a power supply device proposed as Japanese Patent Application No. 4-38212. The circuit configuration will be described below. AC power supply Vs is power switch SW
Is connected to the AC input terminal of the full-wave rectifier circuit DB. A small capacity capacitor Cf for filtering is connected in parallel to the AC input terminal of the full-wave rectifier circuit DB. The inductor L1 is connected to the DC output terminal of the full-wave rectifier circuit DB via the switching element Q1. A smoothing capacitor C1 is connected to both ends of the inductor L1 through a backflow blocking diode D2. The primary winding N1 of the high frequency transformer T is connected to both ends of a capacitor C1 via a diode D3 and a switching element Q1. To the secondary winding N2 of the high frequency transformer T, a parallel circuit of a capacitor C5 and a load Z is connected via a series circuit of a diode D0 and an inductor L0. The diode D1 is connected to the series circuit of the inductor L0 and the capacitor C5 with the polarity shown. Although not particularly shown, snubber circuits are connected to both ends of the switching element Q1.

FIG. 20 shows operation waveforms of the circuit of FIG. In the figure, I1 is a current waveform flowing in the switching element Q1, and I2 is a current waveform flowing in the diode D2. When the switching element Q1 is turned on, the rectified output voltage of the full-wave rectifier circuit DB is applied to the inductor L1. Let the inductance value of inductor L1 be L, and full-wave rectifier circuit DB
Of the rectified output voltage of Vd and the ON voltage of the switching element Q1 are Vs, the current I1 flowing through the inductor L1 is
Increases linearly with a slope of di / dt = (Vd−Vs) / L.

Next, when the switching element Q1 is turned off, the electromagnetic energy accumulated in the inductor L1 is released to the capacitor C1 via the diode D2 and the capacitor C1 is charged. The voltage charged in the capacitor C1 can be increased or decreased by controlling the ON time of the switching element Q1.

Next, when the switching element Q1 is turned on while the capacitor C1 is charged, the primary winding N1 of the high frequency transformer T, the diode D3,
A current flows through the switching element Q1. At this time, the diode D is attached to the secondary winding N2 of the high frequency transformer T.
A voltage with a polarity that forward biases 0 is generated. For this reason,
The diode D0 becomes conductive, and a current flows through a parallel circuit of the capacitor C5 and the load Z via the inductor L0.

Next, when the switching element Q1 is turned off, a counter electromotive force is generated in the primary winding N1 of the high frequency transformer T. At this time, a voltage having a polarity that reversely biases the diode D0 is generated in the secondary winding N2 of the high frequency transformer T. Therefore, the diode D0 becomes non-conductive, and the energy stored in the inductor L0 is transferred to the diode D1.
Is discharged to the parallel circuit of the capacitor C5 and the load Z via. Therefore, the ripple of the voltage across the load Z is reduced. The voltage charged in the capacitor C5 can be increased or decreased by controlling the ON time of the switching element Q1.

In this conventional example, since the chopper circuit is provided, the input power factor is a high power factor, and by adopting a low pass filter such as the capacitor Cf on the input side, the input current harmonics are increased. There are few wave components. Further, since the diode D3 is inserted, there is an advantage that no rush current (inrush current) flows from the commercial AC power supply Vs to the capacitor C1 via the inductor L1 when the power is turned on. However, in this conventional example, a resonance inverter is not used, and the voltage waveform and the current waveform across the switching element are enlarged as shown in FIG. Due to the rising and falling characteristics of the switching element Q1, there is a delay in the waveform change, and when the switching element Q1 is off and on, there is a portion where the voltage waveform and the current waveform overlap, as indicated by the diagonal lines. This overlapped portion is a power loss at the time of switching, the loss is large, the circuit efficiency is deteriorated, the heat generation amount is increased, and a heat radiation measure is required. For this reason, structural ingenuity such as providing a heat radiating plate on the switching element is required, resulting in a large size and a high cost. Further, there is a disadvantage that high-frequency electric noise is generated due to a sharp rise / fall of the voltage generated when the switching element Q1 is turned on / off. In particular, when the discharge lamp load is lit with a rectangular wave, the wiring to the lamp and the lamp itself serve as an antenna and radiate high frequency noise. In order to prevent this, it is conceivable to electromagnetically shield the lamp or the like, but this is not preferable in terms of increased cost and reduced brightness.

Next, according to the inverter device disclosed in Japanese Patent Laid-Open No. 1-160367, an inrush current at the time of turning on the power source is prevented, and a high-input power factor monolithic resonance inverter is obtained. Since the voltage supplied to the load has a ripple component that is synchronized with the input frequency of the AC power supply,
There is a problem that the output voltage does not become a completely flat and stable voltage without ripples and the light output includes flicker when the discharge lamp is a load. In addition, JP-A-4-127875
The inverter device described in the publication is not a resonant inverter.

[0009]

SUMMARY OF THE INVENTION The present invention has been made in view of the above points, and an object of the present invention is to prevent a rush current from flowing at the time of turning on a power source and to provide a high input power factor. By inserting a low-pass filter on the input side, harmonic current can be suppressed, the number of switching elements is small,
An object of the present invention is to provide an inverter device that is small, lightweight, inexpensive, generates little electrical noise, and has high circuit efficiency.

[0010]

In order to solve the above-mentioned problems, in the inverter device of the present invention, as shown in FIG. 1, a full-wave rectification circuit DB for full-wave rectifying the AC power supply Vs.
A first switching means S1 for connecting and disconnecting the DC output of the full-wave rectification circuit DB at a high frequency; and a first inductor connected to the DC output end of the full-wave rectification circuit DB via the first switching means S1. L1, a second switching means S2 which is alternately turned on / off with the first switching means S1, and a first smoothing capacitor connected to the first inductor L1 via the second switching means S2. C1, a second inductor L2 connected to the first capacitor C1 via the first switching means S1 and the third switching means S3, and a second inductor L2 connected to resonate with the second inductor L2. Second capacitor C0 and second
Load circuit Z driven by receiving the resonance voltage of the capacitor C0, and the third switching means S3 is configured to block the current flowing from the full-wave rectifier circuit DB to the smoothing first capacitor C1. It is characterized by that.

[0011]

According to the present invention, the AC voltage of the AC power supply Vs is full-wave rectified by the full-wave rectifier circuit DB, and the output voltage is intermittently applied at high frequency by the switching means S1 and applied to the first inductor L1. Accumulate energy,
Since this electromagnetic energy is released to the first smoothing capacitor C1 via the second switching means S2, the period during which the input current from the AC power supply Vs is flowing can be lengthened and the input current distortion can be increased. And the input power factor can be increased. The path through which the current directly flows from the DC output terminal of the full-wave rectifier circuit DB to the smoothing capacitor C1 is the third switching means S3.
As a result, the inrush current does not flow when the power switch SW is turned on. Furthermore, since the load current becomes a resonance current due to the second inductor L2 and the second capacitor C0, high-frequency noise is reduced, switching loss is reduced, circuit efficiency is increased, and a heat sink or the like is not required. As a result, it is inexpensive and can be made compact.

[0012]

1 is a circuit diagram of a first embodiment of the present invention.
The circuit configuration will be described below. The AC power supply Vs is connected to the AC input terminal of the full-wave rectifier circuit DB via the power switch SW. A small capacity capacitor Cf for filtering is connected in parallel to the AC input terminal of the full-wave rectifier circuit DB. At the DC output terminal of the full-wave rectifier circuit DB,
The inductor L1 is connected via the switching means S1. A smoothing capacitor C1 is connected to both ends of the inductor L1 via a switching means Q2. The inductor L2 is connected to both ends of the capacitor C1 via the switching means S3 and S1. A capacitor C0 is provided in the series circuit of the switching means S3 and S1.
And a load Z parallel circuit is connected. Inductor L2
And capacitor C0 are switching means S1, S2, S3.
Constants are set to resonate at the operating frequency of the load Z
Are connected so that power is supplied from both ends of the capacitor C0. The load Z is set to an impedance value that does not significantly impair the resonance between the inductor L2 and the capacitor C0. In addition, when the free wheel current flows,
Since the electrolytic capacitor C1 has a high internal impedance with respect to high frequencies, a high-frequency bypass capacitor Ch may be connected in parallel at both ends of the smoothing capacitor C1.

The operation waveforms of the circuit of FIG. 1 are shown in FIG. In the figure, (a) shows the waveform of the control signal of the switching means S1,
(B) is the waveform of the control signal of the switching means S2,
(C) and (d) are waveforms of the control signal of the switching means S3. As shown in FIGS. 2A and 2B, the switching means S1 and S2 are alternately driven on / off.
First, when the switching means S1 is on and the switching means S2 is off, the rectified output voltage of the full-wave rectifier circuit DB is applied to the inductor L1, and electromagnetic energy is accumulated in the inductor L1. Next, the switching means S
When 1 is off and the switching means S2 is on, the electromagnetic energy accumulated in the inductor L1 is released to the capacitor C1 via the switching means S2, and the capacitor C1 is charged. The voltage charged in the capacitor C1 can be increased or decreased by controlling the ON time of the switching means S1.

Next, when the switching means S1 is turned on while the capacitor C1 is charged, if the switching means S3 is turned on, the voltage of the capacitor C1 is applied to the inductor L2. When the switching means S1 is turned off, a resonance current flows in the resonance circuit of the inductor L2 and the capacitor C0. In this way, the switching means S
1 is used as a switching means of the one-stone resonant inverter circuit composed of an inductor L2, a capacitor C0 and a load Z, an AC power supply Vs, a full-wave rectification circuit D.
B, an inductor L1, a switching device S2, and a capacitor C1 are commonly used as a switching device of a chopper circuit.

Next, regarding the switching means S3,
As shown in FIG. 2C or 2D, there are two control methods.
There is. As shown in FIG. 2C, the switching means S3
In case of ON / OFF operation in synchronization with the switching means S1
In case of power on, inrush to capacitor C1 at power-on
The current can be prevented. That is, switching means
When S3 is turned on, switching means S1 is turned off.
The switching means S3 and the inductor
Inrush current flows to capacitor C1 via L2
There is no such thing. Further, the switching means S1 is turned off.
Sometimes the switching means S3 is also off
Then, through the switching means S3 and the inductor L2,
No inrush current flows through the capacitor C1.
Also, as shown in FIG. 2D, the capacitor C1 is a switch.
Switching means S1, inductor L1, switching means S2
When it takes to be fully charged by the chopper operation of
Interval t ₀ When the switching means S3 is turned on after the passage of
Since the capacitor C1 has already been charged, the AC
Power supply Vs to full-wave rectifier circuit DB, switching means S
3, inductor L2, capacitor C1, inductor L
1, the current flowing in the path of the full-wave rectifier circuit DB is suppressed,
Inrush current can be prevented.

FIG. 3 is a circuit diagram of the second embodiment of the present invention. This embodiment is a concrete embodiment of the embodiment shown in FIG. In the figure, Q1 is a switching element, which is a power bipolar transistor, a power MOSFET, or an IGBT.
And so on. D2 and D3 are diodes, which are used as switching means S2 and S3, respectively. The diode D4 is preferably connected in antiparallel to the series circuit of the diode D3 and the switching element Q1. The diode D4 has a function of preventing application of a reverse bias voltage to the switching element Q1,
It may be inserted to prevent deterioration of the switching element Q1 or to increase the degree of freedom in designing the inverter.

The operation of this circuit will be described below. The input inrush current when the power switch SW is turned on is the full-wave rectification circuit DB, the diode D3, the inductor L.
2, the capacitor C1 and the inductor L1 try to flow, but the diode D3 prevents the current from flowing in the inrush current. Further, since the chopper circuit is configured by the full-wave rectifier circuit DB, the switching element Q1, the inductor L1, the diode D2, and the smoothing capacitor C1, a high input power factor can be ensured, and further, the capacitor Cf on the power input terminal side can be secured. By inserting such a low-pass filter, the input current is smoothed into a sine wave, and the harmonic current is reduced.

Operation waveform diagrams of the circuit of FIG. 3 are shown in FIGS. 4 and 5. In the figure, v ₀ is the voltage across the discharge lamp FL, i ₀ is the current flowing through the capacitor C ₀ , i ₄ is the reverse current of the diode D ₄ , i ₃ is the forward current of the diode D ₃ , and i ₂ is the forward direction of the diode D _2. Current, i ₁ is full-wave rectifier circuit DB
Is the output current of. Further, (a) is a voltage v ₀ across the discharge lamp FL, (b) is a current waveform of each part of the inverter part,
(C) is a current waveform of each part of the chopper part. Load Z
Since the impedance value of is a value that does not impair the conditions of the resonance system, the current flowing through the load Z has substantially the same waveform as the current i ₀ flowing through the capacitor C0. A current having a rest section flows through the load Z, and a high-frequency half-wave current is supplied. FIG. 4 is an operation waveform diagram in the case where there is a diode D4 for flowing a free wheel current. The voltage v ₀ across the series connection circuit of the switching element Q1 and the diode D3
Is due to the resonance of the inductor L2 and the capacitor C0,
It becomes a half-sine wave. The current flowing through the switching element Q1 is the sum of the waveforms of i ₁ and i ₃ , and this current (i ₁ +
The degree of overlap between i ₃ ) and the voltage v ₀ is small, and the loss of the switching Q1 is significantly smaller than that in the case of rectangular wave lighting. Further, since the rise and fall of the voltage v ₀ are not steep, it can be seen that the generation of high frequency noise is suppressed.

FIG. 5 is an operation waveform diagram in the case where there is no diode D4 for flowing a free wheel current. In this case, it can be seen that there is a portion where the voltage v ₀ across the series connection circuit of the switching element Q1 and the diode D3 becomes a negative voltage, that is, there is a period in which a reverse voltage is applied to the switching element Q1. Considering the application of the reverse bias voltage to the switching element Q1, the diode D4 can be omitted, so that the number of parts can be reduced, which is advantageous in terms of cost.

FIG. 6 is a circuit diagram of the third embodiment of the present invention. In this embodiment, in the second embodiment, the capacitor C
The power source side terminals of the filament of the discharge lamp FL are connected to both ends of 0 through a series circuit of a capacitor C3 and an inductor L3. The capacitor C2 is connected to the non-power supply side terminal of the filament of the discharge lamp FL. The capacitor C3 is for cutting the DC component, and the capacitors C0,
The capacitance is larger than that of C2 and does not contribute to resonance. The capacitor C2 is a resonance capacitor, and also has a function of supplying a preheating current to the filament of the discharge lamp FL.
The inductor L3 is for resonance and serves as a current limiting element (ballast) for the lamp current of the discharge lamp FL. The inductor L2 has a larger inductance value than the inductor L3, and exhibits a constant current action in the present embodiment, so that stable power is supplied to the tank circuit configured by the resonance circuit including the inductor L3 and the capacitors C0 and C2. It has the function of supplying.

The operation waveforms of the embodiment of FIG. 6 are shown in FIG.
The load circuit including the discharge lamp FL is designed to handle inductive loads as a whole.
I am configuring. Switching when the discharge lamp FL is preheated
The frequency is set sufficiently higher than the resonance frequency,
Resonance becomes weak. At this time, inductor L2,
Filament of C3, inductor L3, discharge lamp FL
The condenser, the condenser C2, and the filament of the discharge lamp FL.
Preheating current flows in the path. After a certain time, switching element
Reduce the switching frequency of the child Q1 to close to the resonance frequency.
The resonance effect is strengthened by attaching the both ends of the discharge lamp FL.
A high voltage is applied to the lamp to turn on the discharge lamp FL. Release
The inductor L3 acts as a ballast to maintain the lighting of the electric lamp FL.
And turn on stably. Obtained on both ends of the capacitor C0
Voltage v ₀ Is discontinuous, but the coupling capacitor C3
Cuts the DC component by using the inductor L3 for resonance.
By utilizing the resonance action of the capacitor C2, discharge
Electric current (electric power) is continuously supplied to the lamp FL. When lit
Voltage v of the discharge lamp FL at₂ Is the inductor L3
Resonance of the capacitor C3 and inductor L3
Connection voltage v₃ Compared to, although it contains some distortion
It is close to the sine wave voltage. For discharge lamp FL
The flowing current i is also a sine wave. This allows
Radiation noise from the wiring to the electric lamp FL and from the discharge lamp FL itself
Can be reduced compared to when square wave is lit and high frequency pulse generation
It can be seen that the components can be reduced. Charge of capacitor C1
Pressure is pure DC with no ripple component due to chopper action
Power source of the discharge lamp FL
Since it has a flat, stable light can be emitted.

FIG. 8 is a circuit diagram of the fourth embodiment of the present invention, and FIG. 9 is a circuit diagram of the fifth embodiment. In the embodiment of FIG. 8, the load circuit connected in parallel to both ends of the capacitor C0 in the embodiment of FIG. 6 is connected in parallel to both ends of the inductor L2. In the embodiment of FIG. 9, the capacitor C0 is also connected in parallel to both ends of the inductor L2. In the latter embodiment, a low-pass filter composed of a capacitor Cf and an inductor Lf is inserted on the AC input side of the full-wave rectification circuit DB, and the input current is has a smooth waveform with little distortion, and harmonic components are reduced. It is something.

FIG. 10 is a circuit diagram of a sixth embodiment of the present invention. In this embodiment, the inductor L4 is connected in parallel with the capacitor C2 connected in parallel to the non-power supply side terminal of the filament of the discharge lamp FL. The inductor L4 is also used as a bypass inductor to prevent a direct current component from flowing through the discharge lamp FL. The capacitors C0 and C2 and the inductor L3 form a tank circuit and act as a resonance circuit of the inverter. The advantage of this embodiment is that the large-capacity capacitor C3 for cutting direct current can be omitted. Naturally, the inductor L3 also acts as a ballast. Further, by inserting the inductor L3 in the illustrated portion, there is an effect of preventing high frequency pulse noise during the switching of the switching element Q1 and the diode D3 from returning to the discharge lamp FL.

FIG. 11 is a circuit diagram of the seventh embodiment of the present invention. In this embodiment, the switching element Q1 is commonly used as the switching means for the chopper circuit and the inverter circuit when the load is light such as during preheating. On the other hand, under heavy load such as lighting, the switching element Q1 is overloaded, and practically few power transistors can be adopted.
Or, since there may be almost no case, as a solution, another switching element Q2 is newly added to the diode D.
3 and the switching element Q1. When the diode D4 is connected in anti-parallel to the switching element Q2, a power MOSFET is used as the switching element Q2, so that the switching element Q2 can be apparently composed of one element. This is because the reverse diode that is parasitic between the drain and the source of the power MOSFET can be used as the diode D4.

The operation waveforms of the circuit of FIG. 11 are shown in FIG. Since the load is light during the preheating period, switching element Q1
Is used as switching means for the chopper circuit and the inverter circuit, and the switching element Q2 is turned off.
Since the load is heavy during the lighting period of the discharge lamp, the switching element Q2 is operated exclusively for the inverter and the switching element Q1 is operated exclusively for the chopper. The switch S4 is a switch for controlling the preheating current,
It is turned on during the preheating period and turned off during the lighting period.

The advantages of this embodiment are that the circuit efficiency is high and the degree of freedom in design is improved. When the switching element Q1 is shared as the switching means of the chopper circuit and the inverter circuit, there was a power loss due to the forward voltage drop of the diode D3. The power loss at D3 is reduced.
Further, since the current capacity of the switching element Q1 may be small, the selection range thereof is widened, and even if the two switching elements Q1 and Q2 are combined, it is cheaper than the configuration in which the switching element Q1 is shared. May be.

FIG. 13 is a circuit diagram of the eighth embodiment of the present invention. In this embodiment, the secondary winding of the inductor L1 is used as the preheating power source for the filament of the discharge lamp FL, and the preheating current is supplied via the impedance element Z0.

14 to 16 are circuit diagrams of ninth to eleventh embodiments of the present invention. In these embodiments, the insulating transformer T is used, and the connection position of the resonance capacitor C0 is different. In the embodiment of FIG. 14, a resonance capacitor C0 is connected in parallel with the primary winding N1 of the transformer T.
In the embodiment of FIG. 15, the resonance capacitor C0 is connected in series with the primary winding N1 of the transformer T. In the embodiment shown in FIG. 16, the resonance capacitor C0 is connected in parallel with the load Z. In these embodiments, the size is increased by using the insulating transformer T, but it is safe because an electric shock accident under the load Z can be prevented. In particular, when a discharge lamp such as a fluorescent lamp is used as the load Z, an electric shock accident when replacing the lamp can be prevented.

FIG. 17 is a circuit diagram of the twelfth embodiment of the present invention. In this embodiment, a large capacity capacitor C4 is connected in parallel to the diode D3, and a small capacity capacitor C0 is connected in parallel to the switching element Q1 so that the capacitor C4 has the turn-off function of the diode D3. This eliminates the recovery loss due to the reverse recovery current of the diode D3, and the diode D3 can be an inexpensive one for low frequencies.

FIG. 18 is a circuit diagram of the thirteenth embodiment of the present invention. In this embodiment, the resonance capacitor C0 is connected between the negative side of the full-wave rectifier circuit DB and the anode of the diode D3, and the inductor L3 and L1 and the capacitor C0 form a resonance circuit. in this way,
By also using the inductor L1 for the chopper as the resonance inductance component of the inverter circuit,
The size of the inductor L3 can be reduced.

[0031]

According to the present invention, an AC power supply is rectified by a full-wave rectification circuit, and the rectified output is intermittently cut off at high frequencies by the first switching means to store electromagnetic energy in the first inductor. Since this electromagnetic energy is entirely injected into the smoothing capacitor via the second switching means, the input current waveform can be sinusoidal, and the input power factor can be increased. . Further, since the third switching means for preventing the output current of the full-wave rectifier circuit from directly flowing into the smoothing capacitor is provided, there is an effect that the inrush power does not flow when the power is turned on. Further, since the load current is made to be the resonance current by the second inductor and the second capacitor, noise radiation is reduced, switching loss is also reduced, and the heat radiation structure can be simplified, so that the inverter device is small and lightweight. There is an effect that can be realized.

When a discharge lamp is used as the load, a stable output voltage without ripple of any magnitude can be supplied to the load by controlling the ON time of the first switching means, and the light emission is flat and of good quality. There is an effect that a light output can be obtained.

[Brief description of drawings]

FIG. 1 is a circuit diagram of a first embodiment of the present invention.

FIG. 2 is an operation waveform diagram of the first embodiment of the present invention.

FIG. 3 is a circuit diagram of a second embodiment of the present invention.

FIG. 4 is a first operation waveform diagram of the second embodiment of the present invention.

FIG. 5 is a second operation waveform diagram of the second embodiment of the present invention.

FIG. 6 is a circuit diagram of a third embodiment of the present invention.

FIG. 7 is an operation waveform diagram of the third embodiment of the present invention.

FIG. 8 is a circuit diagram of a fourth embodiment of the present invention.

FIG. 9 is a circuit diagram of a fifth embodiment of the present invention.

FIG. 10 is a circuit diagram of a sixth embodiment of the present invention.

FIG. 11 is a circuit diagram of a seventh embodiment of the present invention.

FIG. 12 is an operation waveform diagram of the seventh embodiment of the present invention.

FIG. 13 is a circuit diagram of an eighth embodiment of the present invention.

FIG. 14 is a circuit diagram of a ninth embodiment of the present invention.

FIG. 15 is a circuit diagram of a tenth embodiment of the present invention.

FIG. 16 is a circuit diagram of an eleventh embodiment of the present invention.

FIG. 17 is a circuit diagram of a twelfth embodiment of the present invention.

FIG. 18 is a circuit diagram of a thirteenth embodiment of the present invention.

FIG. 19 is a circuit diagram of a conventional example.

FIG. 20 is an operation waveform diagram of a conventional example.

FIG. 21 is an operation waveform diagram at the time of switching in the conventional example.

[Explanation of symbols]

 S1 1st switching means S2 2nd switching means S3 3rd switching means DB Full wave rectification circuit Vs AC power supply L1 1st inductor L2 2nd inductor C1 1st capacitor C0 2nd capacitor Z load

Claims (11)

[Claims] 1. A full-wave rectifier circuit for full-wave rectifying an AC power supply, a first switching means for intermittently connecting and disconnecting a DC output of the full-wave rectifier circuit at high frequencies, and a full-wave rectifier circuit via the first switching means. A first inductor connected to the DC output terminal of the second switching device, a second switching device that is alternately turned on and off with the first switching device, and a second inductor connected to the first inductor via the second switching device. A first smoothing capacitor, a second inductor connected to the first capacitor via the first switching means and the third switching means, and a second inductor connected to resonate with the second inductor. The second capacitor and a load circuit driven by receiving the resonance voltage of the second capacitor.
Inverter device, wherein the switching means is configured to block a current flowing from the full-wave rectifier circuit to the smoothing first capacitor. 2. A full-wave rectifier circuit for full-wave rectifying an AC power supply, a switching element for intermittently connecting and disconnecting the DC output of the full-wave rectifier circuit at a high frequency, and a DC output terminal of the full-wave rectifier circuit via the switching element. A connected first inductor and a smoothing first capacitor connected to the first inductor via a backflow blocking first diode;
A second inductor connected to the first capacitor via the switching element and a second diode for blocking backflow; a second capacitor connected to resonate with the second inductor; And a load circuit driven by receiving the resonance voltage of the capacitor, and the second diode is connected in a direction to block a current flowing from the full-wave rectifier circuit to the smoothing capacitor. . 3. A full-wave rectifier circuit for full-wave rectifying an AC power source, a switching element for intermittently switching the DC output of the full-wave rectifier circuit at high frequencies, and a DC output terminal of the full-wave rectifier circuit via the switching element. A connected first inductor and a smoothing first capacitor connected to the first inductor via a backflow blocking first diode;
A high frequency transformer having a primary winding connected to the first capacitor via the switching element and a second diode for blocking reverse current, a load circuit connected to a secondary winding of the high frequency transformer, and a second It is composed of an inductor and a second capacitor connected so as to resonate with the second inductor, and the second diode is connected in a direction of blocking a current flowing from the full-wave rectifier circuit to the smoothing capacitor. An inverter device characterized in that 4. The inverter device according to claim 2, wherein a third diode is connected in antiparallel to both ends of the series connection of the second diode and the switching element. 5. The load circuit includes a series circuit of a DC cutting capacitor, a ballast inductor and a discharge lamp, and a preheating current conducting capacitor is connected in parallel to a non-power supply side of a filament of the discharge lamp, The inverter device according to claim 2, 3 or 4, wherein the ballast inductor and the preheating current carrying capacitor constitute a resonance circuit. 6. The load circuit includes a series circuit of a ballast inductor and a discharge lamp, and a preheating current conducting capacitor and a DC bypass inductor are connected in parallel to the non-power supply side of the filament of the discharge lamp. The inverter device according to claim 2, wherein the ballast inductor and the preheating current carrying capacitor form a resonance circuit. 7. The load circuit includes a series circuit of a third inductor and a discharge lamp, and the third inductor is interposed between the second diode and the discharge lamp. Inverter device described. 8. A third diode is connected in anti-parallel to the switching element, and a reverse recovery capacitor is connected in parallel to the second diode in any one of claims 2 to 7. Inverter device according to. 9. A second switching element that operates in synchronization with the first switching element when a load current is large is connected in parallel across the series circuit of the second diode and the switching element. Item 9. The inverter device according to any one of items 2 to 8. 10. The inverter device according to claim 2, wherein a preheating winding for supplying a preheating current to the filament of the discharge lamp is provided in the first inductor. 11. A second capacitor for resonance is connected in parallel with a series circuit of a second diode, a switching element and a first inductor.
0. The inverter device according to any one of 0.