JPH01264009A - Active filter automatic adjustment circuit - Google Patents

Abstract

PURPOSE:To attain automatic adjustment of a cut-off frequency characteristic of an active filter without increasing the circuit scale by supplying an output signal of a low pass filter as a control voltage controlling the cut-off frequency to the active filter. CONSTITUTION:An output signal of a low pass filter 14 is fed to an active filter 12 as a control voltage to control the cut-off frequency of the active filter 12. That is, the cut-off frequency of a phase shifter 12 is designed in advance so that the phase shift of the phase shifter 12 is 90 deg. at the FM carrier frequency to absorb the dispersion automatically even if the cut-off frequency is fluctuated due to the dispersion of components at manufacture of an IC. Thus, an audio FM detection circuit 8 is provided with a function to apply FM demodulation and to adjust automatically the dispersion of the cut-off frequency of the phase shifter 12, then it is not required to add an automatic adjustment circuit provided with a pseudo active filter or the like.

Description

[Detailed Description of the Invention] [Field of Industrial Application] The present invention automatically adjusts deviations in frequency characteristics in active filters used as phase shifters, traps, equalizers, delay circuits, etc. The present invention relates to an automatic adjustment circuit, and particularly to an automatic adjustment circuit for an active filter suitable for automatically adjusting variations in cutoff frequency that tend to occur when an active filter is integrated into an IC.

[Conventional technology]

In recent years, as IC processing technology has improved, the integration density of ICs has tended to increase, and more functions have tended to be incorporated into one IC.

Along with this, many filters have been incorporated into ICs in the form of active filters.

When incorporating an active filter into an IC, variations occur in the resistance value and capacitance within the IC during the IC manufacturing process, and variations also occur in the cutoff frequency of the active filter, which is determined directly by the resistance value and capacitance. Therefore, when implementing an active filter into an IC, it is necessary to adjust the variation in cutoff frequency. Various circuits have been devised to automatically adjust this variation. Related devices of this type include, for example, JP-A-751-174810 and JP-A-62-11.
Examples include circuits and devices described in Publication No. 7407.

[Problem that the invention seeks to solve]

Generally, within an IC, the absolute values of each element vary, but it is known that the rate of variation of each element is approximately equal. In other words, the amounts of deviation due to variations in the cutoff frequencies of several active filters with substantially equal characteristics existing in the same IC are all approximately equal. From this, the same -IC
When adjusting the cutoff frequency of several active filters that have substantially the same characteristics, the cutoff frequency of one of the active filters is adjusted, and the control signal used after that adjustment is applied to the other active filters. When feeding active filters, the cutoff frequencies of these active filters are adjusted in the same way.

When performing automatic adjustment of the active filter in the conventional technology, a pseudo active filter having substantially the same characteristics as the active filter existing in the IC is provided, the cutoff frequency of the pseudo active filter is adjusted, and at this time, All active filters are adjusted by the control signals used.

When adjusting the pseudo active filter, first a reference signal is input to the pseudo active filter, and its output signal and the reference signal are input to a level comparator. The level comparator compares the output signal level of the pseudo active filter and the reference signal level, and outputs a signal according to the level difference. The amount of attenuation of the reference signal by the pseudo-active filter changes depending on the variation in the characteristics of the pseudo-active filter, so by comparing the output level of the pseudo-active filter and the reference signal level, the output of the υ level comparator is determined by the pseudo-active filter. The amount of deviation is detected. Then, when the output signal of the level comparator is supplied to the pseudo active filter as a cutoff frequency control signal of the pseudo active filter, the characteristics of the pseudo active filter are moved in a direction in which deviations are eliminated, and the characteristics of the pseudo active filter are adjusted.

Then, by supplying the output signal of the level comparator used for adjusting the pseudo active filter to several active filters having substantially the same characteristics as the pseudo active filter, the cutoff frequencies of these filters are also adjusted.

However, in order to automatically adjust the active filter as described above, circuits such as a pseudo active filter, an oscillator that generates a reference signal, and a level comparator are required, which inconveniently increases the overall scale of the IC circuit. occurs.

An object of the present invention is to solve the above problems and provide a circuit that can automatically adjust the cutoff frequency characteristics of an active filter without increasing the circuit scale of an IC.

[Means for solving problems]

The above purpose is to take an audio FM signal as input, and to
A phase shifter whose phase shift amount changes from a certain reference frequency according to the frequency of the signal, and a signal voltage that is inputted with the output signal of the phase shifter and the audio FM signal and that corresponds to the phase shift difference between both signals. an audio FM detection circuit that extracts the output of the phase shift comparator as an FM demodulated audio signal (this is called a quadreature audio detection circuit).
The phase shifter is configured with an active filter of a type whose cutoff frequency is controlled by an applied voltage, and the output signal of the phase shift comparator is used as an input signal, and the cutoff frequency is controlled by an applied voltage. A low-pass filter is provided for extracting the signal of the active filter, and the output signal of the low-pass filter is supplied to the active filter as a control voltage for controlling the cut-off frequency of the active filter, and the cut-off frequency of the other active filter is This is achieved by also using it as a control voltage.

. [Operation] The audio FM detection circuit described above feeds back the output of the phase comparator as a control signal that controls the cutoff frequency of the phase shifter (active filter), thereby enabling PLL detection.
(phase-locked loop) circuit.

However, since the audio frequency band corresponding to the FM modulated signal is removed from the control 114+ signal fed back to the phase shifter by the low-pass filter, the circuit follows the input as a PLL circuit. This is performed only on the audio FM signal with the FM modulated signal component removed, that is, on the FM carrier frequency, and as a result, the phase shifter is always controlled so that the amount of phase shift is 90 degrees at the FM carrier frequency. Ru.

Therefore, if the cutoff frequency of the phase shifter is designed in advance so that the phase shift amount of the phase shifter is 90 degrees at the FM carrier frequency, the cutoff frequency will not fluctuate due to element variations during IC manufacturing. It also automatically operates to absorb this variation. That is, the audio FM
The detection circuit performs FM demodulation and at the same time has the function of automatically adjusting the dispersion of the cutoff frequency of the phase shifter. If utilized, there is no need to add an automatic adjustment circuit equipped with a pseudo active filter or the like as in the conventional case.

〔Example〕

An embodiment of the present invention will be explained below with reference to FIG.

FIG. 1 is a block diagram showing one embodiment of the present invention. That is, since a color television receiver employs an audio FM detection circuit (quadreature audio detection circuit), this embodiment utilizes this to implement the present invention.

In the same figure, 1 is an antenna, 2 is an antenna that selects any station among the broadcast signals received by the antenna 1, and a video intermediate F17.
A tuner circuit for controlling the J-wave signal Kanayama, and 3 an F filter that limits the video intermediate frequency signal to a predetermined band characteristic. 5
is an amplifier circuit that amplifies the output signal from the IF filter, 6
is an amplifier circuit 5) An audio detection circuit that reproduces an audio intermediate frequency signal from the output video intermediate frequency signal, 7 is an amplifier that amplifies the output signal from the audio detection circuit 6, and 8 is an audio signal that is reproduced from the audio intermediate frequency signal. The audio FM detection circuit according to the present invention (
(automatic adjustment circuit for active filter).

In the audio FM detection circuit (also automatic adjustment circuit for the active filter (12)) 8, 9 is an input terminal into which an audio intermediate frequency signal is input, 1o is an output terminal that outputs the reproduced audio signal, and 11 is an active filter C. $ phase vessel 12. Trap circuit 17. Equalizer circuit 18. 12 is a phase shifter, 13 is a phase comparator, 14 is a low-pass filter (hereinafter referred to as LPF),
15 is a band pass filter l (abbreviated as lower BPF).

16 is the video intermediate frequency signal υ output from the amplifier circuit 5.
17 is a trap circuit that attenuates the audio intermediate frequency signal included in the video signal output from the detection circuit 16; 18 is a band characteristic of the amplifier 5; Detection circuit 1 by
6, a frequency characteristic correction circuit (hereinafter abbreviated as an equalizer circuit) for correcting the sloped characteristic near the color subcarrier frequency of the output video signal; 19, the equalizer circuit 13 corrects the characteristic that the color subcarrier frequency of the video signal has a flat characteristic; The trap circuit 17, the equalizer circuit, and the delay circuit 19 have the same cutoff frequency controlled by the control voltage output from the output terminal 11 of the audio FM detection circuit 8. As already mentioned, it is configured as an active filter within the chip.

20 and 21 are the audio FMm wave circuit 8 and the delay circuit 1
Output terminals 22 and 23 are input terminals for inputting video and audio signals from outside the color television receiver; 22 and 23 are input terminals for inputting video and audio signals from outside the color television receiver; 25 is a BPF that extracts only the color signal from the video signal; 26 is a switch that selects three color difference signals (
RY), (G-Y) and CB-Y); 27 is a trap circuit that attenuates the color signal in the video signal and extracts the luminance signal; 28 is a luminance signal and color signal processing circuit; A delay circuit corrects a delay time difference caused by different paths, 29 a luminance signal processing circuit that adjusts the brightness and contrast of the luminance signal, 60 five color difference signals (R-, -Y), (G-Y), (B-Y) and a matrix circuit that outputs the luminance signal Y and the three primary color signals R, G, and B; 32 is an amplifier circuit that drives a color cathode ray tube;

33 is a switch for switching the audio signal supplied from the BPF 15 and the input terminal 25; 35 is a speaker; 54
3 is an amplifier circuit for driving the speaker 55, and 36 is an input terminal for inputting a control signal for controlling the switches 24 and 33.

In FIG. 1, the important circuit parts related to the present invention are 8.
2, the audio FM detection circuit (and active filter automatic adjustment circuit) 8 will be explained in detail below.

In the audio FM detection circuit 8, multiple audio intermediate frequency signals input to the input terminal 9 are passed through a phase shifter (active filter) 1.
2 and is also supplied as one input of the phase comparator 15.

The phase shifter 12 changes the phase shift 1 according to the frequency of the input audio intermediate frequency signal (for example, if the audio intermediate frequency signal is 4.
If the frequency is 5 MHz, it will be a certain standard phase shift amount, but if the same wave number is higher than that, the phase will be delayed, if it is a lower frequency, it will be advanced), and the output is passed to the phase comparator 15.
supply as the other input. The phase ratio [513] outputs a voltage according to the phase difference between the input audio intermediate frequency signal and the output of the phase shifter 12, and outputs this output as L P F 14. and supplied to BPF15. The BPF 15 extracts only the audio signal from the output of the phase comparator 15 and supplies it to the output terminal 10, and the LPF 4 extracts the signal below the audio signal frequency band from the output of the phase comparator 15 and supplies it to the phase shifter 12 and the output terminal. Supply to 11.

In the audio FM detection circuit 8, the phase shifter 12 is constituted by an active filter, as already mentioned many times, and its cutoff frequency varies depending on the voltage.

Diagram M2 is a characteristic diagram showing the phase characteristics of the phase shifter 12 when the control voltage for controlling the cutoff 1ili1 wave number is va. The phase characteristic of the phase shifter 12 is designed as shown in characteristic a in FIG. 2, and the amount of phase shift is 90 degrees (reference value) for an input signal of 4.5 MHz. Characteristics A and C in FIG. 2 are characteristic examples when the cutoff frequency of the phase shifter 12 deviates due to element variations.

Further, as for the phase shift W12, if the control voltage is increased, the cut-off frequency is decreased, and if the side leg voltage t is decreased, the cut-off frequency is increased. As the phase difference between the two input signals increases, the output voltage of the phase comparator 15 also increases, and when the phase difference between the input signals becomes 90 degrees, the phase comparator 15 outputs a voltage of va.

When the phase shifter 12 and the phase comparator 13 are designed as described above, the phase shifter 122, the phase comparator 13, and the LPF
The circuit constituted by l4 receives 4,5 Ml1z
operates as a PLL circuit that follows the audio intermediate frequency signal carrier wave, and the input signal phase difference of the phase comparator 13 is 4.5M1.
Since the cutoff frequency of the phase shifter 12 is controlled to be expressed at 90 degrees with respect to the carrier wave of lz, even if the cutoff frequency of the phase shifter 12 deviates from the design value, it is adjusted to the design value. This operation will be explained in detail below.

Now, when a single frequency signal of 4.5M& is input to the audio FM detection circuit 8, the operation is as follows.

First, if the phase characteristic of the phase shifter 120 is as designed and the control voltage is Va, the characteristic color shown in FIG. 2 is obtained.
When a 5MHz signal is input to the audio FM detection circuit 8,
The phase shifter 12 shifts the input signal by 90 degrees, and the phase comparator 1
Since the phase difference between the two signals input to 5 is 90 degrees, the voltage of va is LPFl4 at the output of the phase comparator 13.
When the signal is fed back to the phase shifter 12 via the phase shifter 12, the phase characteristic of the phase shifter 120 is maintained as the characteristic &f) shown in FIG.

Next, if the phase characteristics of the phase shifter 120 are shifted in the direction of increasing the cut-off station wave number due to element variations, for example, when the control voltage is va, the characteristics shown in FIG. 2 are obtained. 4.
When a 5MHz signal is input to the audio FM detection circuit 8,
In the phase shifter 12, the input signal is smaller than 90 degrees [(90-Δ
φ) degree, and therefore the output of the phase comparator 13 has v
A lower voltage than a is output. Since the output voltage of the phase comparator 13 is lower than Va, this signal is input to I and PF14.
When the signal is fed back as a control signal for the cutoff frequency of the phase shifter 12, the cutoff frequency of the phase shifter 12 is lowered to have the characteristic shown in characteristic 1 in FIG.

In addition, if the phase characteristic of the phase shifter 12 deviates in the direction of lowering the cutoff frequency (due to element variations, etc.), for example, when the control voltage is VaO, and the characteristic becomes the characteristic shown in characteristic C in FIG. The output of the device 13 is a voltage higher than va. When this voltage is fed back to the phase shifter 12, the cut-off frequency of the phase shifter 12 is increased, resulting in the characteristic shown in FIG.

When a 4.5 MHz signal is input to the audio FM detection circuit 8 in this way, even if the phase characteristics of the phase shifter 120 deviate due to element variations during manufacturing, the characteristics are adjusted by the control signal and become the designed characteristics. .

Next, the signal input to the audio FM detection circuit 8 is 4.5M.
In the case of the l1z signal, when an audio intermediate frequency signal which is an actual input signal is input, the following operation is performed.

In the audio FM detection circuit 8, the phase shifter 12 and the phase comparator 13 constitute the quadrature chair detection circuit as described above.The audio signal is FM demodulated, and the audio intermediate station wave signal with a carrier frequency of 4.5 Mflz is output. When input, an audio signal is reproduced at the output of the phase comparator 15.

In addition, the control signal fed back to the phase shifter 12 is the phase comparator NJ13.
This is a signal from which the audio frequency band has been removed by LPF I 4 from the output signal of . Since the audio signal removed (by LPF14) corresponds to the FM modulated wave of the input audio intermediate frequency signal, the control signal fed back to the phase shifter 12 removes the modulated signal component from the input audio intermediate frequency signal. Round things, Tsuma? The signal is the same as when only the 4.5 Mfiz carrier wave is input.

Therefore, the same control signal as when a 45 MHz single frequency signal is input to the audio FM detection circuit 8 is sent to the phase shifter 1.
2, the phase characteristic of the phase shifter 12 is expanded and adjusted even when an audio intermediate frequency signal is input.

As mentioned above, the audio FM detection circuit 8 performs FM demodulation and also adjusts the cutoff frequency of the phase shifter (active filter) 12. Therefore, there is no need to provide a new circuit for adjusting the cutoff frequency.

In addition, the control signal used to control the cutoff frequency of the phase shifter 12 can be applied to another active filter (another active filter manufactured using the same chip) having similar characteristics to the phase shifter 12, such as the trap shown in FIG. By supplying the signal as a signal for controlling the cutoff frequencies of the circuit 17, the equalizer circuit 18, the delay circuit 19, etc., the cutoff frequencies of these active filters are also adjusted.

FIG. 6 shows an audio F-wave circuit 8 as another embodiment of the present invention.
Items that are the same as or have the same function are given the same code.
．． , a detailed explanation thereof will be omitted.

In FIG. 3, reference numeral 57 denotes an amplifier to which the output signal of the LPF 14 is input, amplifies this input signal, and supplies the amplified signal to the phase shifter 12 and the output terminal 11.

In the embodiment of FIG. 6, amplifier 57 amplifies the output signal from LPF 14 which is fed back to phase shifter 12. This increases the loop gain of the PLL circuit that controls the cutoff frequency of the phase shifter 12. By increasing the loop gain, the error between the cutoff frequency and the design value when the phase shifter 12 is adjusted can be reduced.

Also, in the embodiment shown in FIG. 6, the basic operation when performing FM demodulation and simultaneously adjusting the cutoff frequency of the phase shifter 12 is the same as in the embodiment shown in FIG. 1, so the effect of the present invention is obvious. is.

FIG. 4 is a block diagram showing still another embodiment of the present invention. In FIG. 4, components that are the same as or have the same functions as the audio FM detection circuit 8 of FIG. 1 are denoted by the same reference numerals, and detailed explanation thereof will be omitted.

In FIG. 4, the output of the phase comparator 13 is directly connected to the phase shifter 1.
2 and simultaneously supplied to LPF14 and LPF1
The output of 4 is supplied to output terminal 11.

In the embodiment shown in FIG. 4, since the output signal of the phase comparator 13 is directly fed back as a signal for controlling the cutoff frequency of the phase shifter 12, the audio intermediate frequency signal input from the input terminal 9 On the other hand, the phase difference between the two signals input to the phase comparator '15 is always controlled to be 90 degrees. Therefore, the circuit constituted by the phase shifter 12 and the phase comparator 15 operates as a PLL circuit that follows the input audio intermediate frequency signal. Therefore, when an audio intermediate frequency signal is input to the audio FM detection circuit 8, an FM demodulated audio signal is reproduced at the output of the phase comparator 16.

Further, the signal supplied from the output signal of the phase comparator 15 to the output terminal 11 via the LPF 14 is a signal corresponding to the input carrier wave, as in the case of the audio FM detection circuit 8 of FIG.

On the other hand, since the phase difference between the two signals input to the phase comparator 13 is always controlled to be 90 degrees, the phase difference between the two signals input to the phase comparator 13 is also 90 degrees with respect to the input carrier wave. becomes. Therefore, the signal supplied to the multi-output terminal 11 from the LPF 14 is a signal that shifts the phase of the phase shifter 12 by 90 degrees, and adjusts the cutoff frequency of the active filter connected to the tb output terminal 11. I'll stop by the signal.

From this, the cutoff frequency of the active filter connected to the output terminal 11 can be adjusted.

Then, when the frequency of the input audio intermediate station wave signal becomes equal to the frequency of its carrier wave, the phase shifter 12 cuts off Ii! Even if the i1 wave number varies, the amount of phase shift is controlled to be 90 degrees, so an effect equivalent to adjusting the cutoff frequency can be obtained.

From the above, the effects of the present invention can also be obtained in the embodiment shown in FIG.

Next, in order to give an example of the phase shifter 12 used in the embodiments of FIGS. 1, 5, and 4, FIG. 5 shows an integrator 59 whose integral gain can be changed depending on the voltage. FIGS. 6 and 7 show an example of a phase shifter configured with the integrator shown in FIG. 5, in which the number of cutoff stations i changes by changing the integral gain of the integrator.

FIG. 5 shows an example of a table integrator whose integral gain can be changed by voltage. In FIG. 5, a resistor R1 is connected to the emitter of a transistor Q1, a resistor R2 is connected to an emitter of a transistor Q2, the other ends of the resistors R1 and R2 are respectively connected to a constant current source A1, and the transistors Q1 and Q2 are connected to a differential They constitute a pair.

Further, the base of the transistor Q1 is connected to the input terminal T1 and is applied with the input signal voltage Via, and the base of the transistor Q2 is applied with the bias voltage VB1. Furthermore, the collector current of transistor Q2 is the differential pair transistors Qs and Q4 whose emitters are connected to the collector of Q2.
It is divided into two parts.

A bias voltage VB2 is applied to the base of the transistor Q5, and a control voltage V is connected to the base of the transistor Q4 to a control terminal T2 to adjust the degree of the shunting. is applied to the control terminal T2.

The collector of the transistor Q4 is connected to the collector of the transistor Q5 and also to the load capacitor C whose other end is grounded. Further, the collector is also connected to an output terminal T3, and has an output voltage V. produce ut.

The base of the transistor Q5 is connected to the base and collector of the transistor Q6 to perform a current mirror operation, and a current substantially equal to the collector current of the transistor Q6 flows as a substantially constant current to the collector of the transistor Q5.

On the other hand, transistor Q7. The respective emitters of Qa are connected together and connected to a constant current source A2. Constant current [The current value of A2 is 1 of the current Io of the constant current source A1]
The value of I o /2 is set as /2.

The bases of the master transistors Qy + Qa are connected to the transistors Q5. It is connected to the Q40 base and has a similar current shunting effect. Therefore transistor Q
The collector current of Q4, the collector current of Qa, and the collector current of Q6 are approximately equal to each other.

In the integrator 39 configured as follows, the input voltage at the input terminal T1 is Vi. , the output voltage of output terminal T3 is Vout
, ) transistor Q2 by transistor Qs, Q4
RFX
and transistor Q1. If it is sufficiently larger than the emitter resistance of Q2 and the capacitance of the capacitor is C, then
The transfer function Ha(S) of this circuit is Ha(S)=□ ・・・・・・(1)SC
・2RE S nira plus operator, indicating that this circuit is an integrator.

The integral gain T of this integrator is expressed by the above equation (1). ζ Since the shunt ratio changes depending on the control voltage VC applied to the control terminal T2, the integral gain, which is proportional to the magnitude of the shunt ratio, can be changed.

The detailed operation of this integrator can be found in Japanese Patent Application No. 1983-
It is described in the integral circuit of the 2016 specification.

Figure 6 shows a phase shifter 1 used in Figures 1, 3, and 4.
As an example of No. 2, an example of a circuit constructed using the integrator 69 shown in FIG. 5 will be given.

In FIG. 6, 41 is an input terminal, 42 and 44 are subtracters that output the difference voltage between two input signals, and 43 is an integrator whose integral gain changes depending on the control voltage as shown in the example of FIG. The terminals (39) and 45 are output terminals.

The signal input through the input terminal 41 is sent to the subtracters 42 and 4.
4, the output of the subtracter 42 is supplied to the integrator 43, the output of the integrator 43 is supplied to the other input terminal of the subtracter 44, and the output of the subtracter 44 is The signal is supplied to the output terminal 45 and one input terminal of the subtracter, thus configuring a phase shifter.

Next, the operation of this phase shifter will be explained. The voltage of the input signal input through the input terminal 41 is Vln1, and the angular frequency is ω.
The integral gain of the integrator 43 is To, and the voltage of the output signal taken out from the output terminal 45 is V. Let it be ut.

When a signal of the input voltage Vin (from the input terminal 41) is input, the subtracter 42 outputs the input voltage Vin and the output voltage V. The integrator 43 outputs a voltage obtained by integrating the output of the subtracter 42, and the subtracter 44 outputs a voltage difference between the output of the integrator 43 and the input voltage Vin.

From the above, input voltage Vin and output voltage V. The relationship between ut is expressed by the following equation (3).

O Vout--(Vin-VOui)-Mln・(3)j
ω Then, the transfer function H1(S) of the phase shifter is obtained from equation (3) and is expressed by equation (4) below.

In addition, the cutoff frequency ω0 is determined from the above equation (4) by the lower equation p (
5).

ω. -T.・・・・・・(
5) It is shown that the circuit shown in Fig. 6 is a phase shifter in addition to the above (4-type). Also, in the above equation (5), since the integral gain To of the integrator 43 can be controlled by the voltage, The cutoff frequency, which is equal to the integral gain, is also controlled by the voltage.

From the above, the circuit shown in FIG. 6 is a phase shifter whose cutoff frequency changes depending on the voltage.

Next, in FIG. 7, another example of a phase shifter constructed using the integrator shown in FIG. 5 will be described. And the example shown in Figure 7 is 2
This is an example in which the following LPF is used as a phase shifter.

In FIG. 7, 51L input terminal, 52 and 54 are subtracters, 53 and 55 are integrators (59) whose integral gain changes (depending on the control voltage as shown in the example of FIG. 5), 56 is an attenuator, and 57 is the output terminal.

The signal input to the input terminal 51 is supplied to one input terminal of a subtracter 52, the output of the subtractor 52 is supplied to an integrator 53, and the output of the integrator 53 is supplied to one input terminal of a subtracter 54. the output of the subtractor 54 is supplied to an integrator 55, the output of the integrator 55 is supplied to an output terminal 57, an attenuator 56 and the other input terminal of the subtractor 52, and the output of the attenuator 56 is It is supplied to the other input terminal of the subtracter 54.

Next, the operation of this phase shifter will be explained.

The voltage of the input signal input to the input terminal 51 is η. , the angular frequency is ω, the integral gains of the integrators 53 and 55 are respectively T 1 e T 2 , and the attenuation amount of the attenuator 56 is A (the input signal is multiplied by 1/A and output from the input/output terminal 57). Let the voltage of the output signal taken out be vOut.

When a signal with an input voltage Min smaller than the input terminal 51 is input, the subtracter 52 outputs the difference voltage between the input voltage Min and the output voltage Vout, and the integrator 55 outputs a voltage obtained by integrating the output of the subtracter 52, Also, the attenuator 56 outputs a voltage ■. Attenuate ut and V. The subtracter 54 outputs the voltage difference between the output of the integrator 55 and the output of the attenuator 56, and the integrator 55 outputs the voltage obtained by integrating the output of the subtracter 54.

From the above, input voltage vin and output voltage V. The relationship between ut is expressed by the following equation (6).

......(6) And according to the above equation (6), the transfer function H2(S) of this phase shifter is
is expressed by the following formula (7 n).

In addition, the cutoff frequency ω0 is determined by the above equation (7) and the lower equation υ (
8).

・. −Vπ core ・・・・・・(8) Above (
8) At second, the integral gain of integrators 53 and 55 is 1
Since T2 and T2 can be controlled by voltage, the cutoff frequency of the circuit shown in FIG. 6 can also be controlled by voltage.

Furthermore, in this phase shifter, Q, which represents the slope of the phase characteristic near the cutoff frequency, is determined using the above equation (7) and is expressed by the following equation (9).

According to the above equation (9), Q changes depending on the attenuation amount A of the attenuator 56. Therefore, the phase shifter of FIG. 6 is replaced by the phase shifter of FIG.

When used in the phase shifter shown in FIG. 3 or FIG. 4, increasing the attenuation type increases the loop gain of the circuit operating as a PLL, and the cutoff frequency and design when the phase shifter 12 is adjusted. The effect is that the error between the values is reduced.

〔Effect of the invention〕

According to the present invention, the active filter automatic adjustment circuit can be used also as the audio FM detection circuit that reproduces the audio signal from the audio FM signal, so the active filter can be automatically adjusted without providing a new circuit. .

[Brief explanation of the drawing]

FIG. 1 is a block diagram showing one embodiment of the present invention, FIG. 2 is a characteristic diagram showing the phase characteristics of the phaser 120 in FIG. 1, and FIGS. 3 and 4 are other embodiments of the present invention. A block diagram showing an example, FIG. 5 is a specific circuit diagram of an integrator used in configuring the phase shifter 12 in FIG. 1, and FIG. 6 is an example of the configuration of the phase shifter 12 in FIG. 1. FIG. 7 is a block diagram showing another example of the configuration of the phase shifter 12 in FIG. 1. Explanation of symbols 8...Audio FM detection circuit (also active filter automatic adjustment circuit), 9...Input terminal, 10...
...output terminal, 11...output terminal, 12.
... Phase shifter, 13 ... Phase comparator, 14
...LPF, 15...BF

Claims (1)

[Claims] 1. A phase shifter that receives an audio FM signal as input and whose phase shift amount changes from a certain reference value according to the frequency of the audio FM signal; and a phase shift comparator that receives an audio FM signal and outputs a signal voltage according to the phase shift difference between both signals, and extracts the output of the phase shift comparator as an FM demodulated audio signal. In this method, the phase shifter is constituted by an active filter of a type whose cutoff frequency is controlled by an applied voltage, and the output signal of the phase shift comparator is used as an input signal, and a signal below the audio frequency band is used as an input signal. An automatic adjustment circuit for an active filter, comprising: a low-pass filter for extracting a low-pass filter; and an output signal of the low-pass filter is supplied to the active filter as a control voltage for controlling a cut-off frequency of the active filter. 2. A phase shifter that inputs an audio FM signal and whose phase shift amount changes from a certain reference value according to the frequency of the audio FM signal, and inputs the output signal of the phase shifter and the audio FM signal. and a phase shift comparator that outputs a signal voltage according to a phase shift difference between both signals, and an audio FM detection circuit that extracts the output of the phase shift comparator as an FM demodulated audio signal, the phase shifter is constituted by an active filter whose cutoff frequency is controlled by an applied voltage, and the output of the phase shift comparator is applied to the active filter as a control voltage for controlling the cutoff frequency. Using the output signal of the phase comparator as an input signal, a low-pass filter is provided for extracting signals below the audio frequency band from the output signal, and the output signal of the low-pass filter is directed to another active filter as its cutoff frequency control voltage. An automatic adjustment circuit for an active filter.