JPH01194883A - Controlling method for induction motor - Google Patents

Abstract

PURPOSE:To enable slip frequency to be controlled to always come to a proper value even if a primary resistance is fluctuated, by taking a signal in relation to a secondary resistance fluctuation of a motor from a voltage fluctuation quantity, and by controlling converter output frequency according to said signal. CONSTITUTION:The output of current detectors 3U-3W is fed to a coordinate converter 5 via a three-phase/two-phase converter 4. The outputs of the coordinate converter 5 and current regulators 15, 16 is fed to an arithmetic circuit 18. Output voltage fluctuation component due to the change of a secondary resistance of a motor 2 from the arithmetic unit 18 is applied to an adder 23, and the frequency of the motor 2 from an adder 26 is sent to a coordinate reference generator 13 and a voltage command arithmetic circuit 14. From the voltage command arithmetic circuit 14, the voltage component command of a revolving-field coordinate system is applied to adders 21, 22 and the output is sent to a coordinate converter 17. By the voltage command of a stator coordinate system from the coordinate converter 17, via a PWM controlling circuit 19, a power converter 1 is controlled.

Description

DETAILED DESCRIPTION OF THE INVENTION [Field of Industrial Application] The present invention relates to a control method for an induction motor, and more particularly to a control method for controlling the voltage and torque of a motor with high precision.

[Conventional technology]

So-called vector control is known in which the current of an induction motor is divided into an excitation component and a torque component and each is controlled independently to perform speed control with high speed response and high accuracy. In this system, the motor current is controlled based on each current component command, but whether or not each component is controlled within the motor according to the command value depends on the control accuracy of the slip frequency. In other words, the motor current supplied from the frequency converter is a vector composition of the excitation component and the torque generation component, but this is considered to be decomposed into the excitation component and the torque generation component each within the motor using the slip frequency as a medium. Therefore, unless the slip frequency is appropriate, each component cannot be controlled according to the command value.

The cause of the unsuitable frequency is that the secondary resistance of the motor fluctuates depending on the temperature. This effect causes a torque response delay and pulsation when the load changes, and also causes problems such as the inability to control the motor voltage according to the command value.

Therefore, conventionally, as described in JP-A-59-156184, there has been a method of detecting the blade voltage of the inverter and correcting the secondary resistance setting value based on this voltage detection signal to compensate for the slip frequency. Proposed.

[Problem to be solved by the invention]

However, the above-mentioned conventional technology requires a detector to detect the output voltage of the inverter, and has a problem in that the hardware configuration is complicated. In addition, in order to estimate the secondary resistance change of the motor from the cutting voltage, it is necessary to subtract the voltage drop due to the primary resistance included in the output voltage and find the voltage component due to the secondary resistance change. Since it changes with temperature, it is difficult to obtain it correctly, and there is a problem in that it is not possible to compensate for the slip frequency properly.

SUMMARY OF THE INVENTION An object of the present invention is to provide a control method that eliminates the need for an output voltage detector and can always control the frequency to an appropriate value even if the primary resistance varies.

[Means to solve the problem]

The above purpose is to calculate the amount of voltage fluctuation from the reference value based on the deviation between the excitation current command of the motor and the detected value of the excitation current in the rotating magnetic field coordinate system, and the deviation between the torque current command and the detected value of the torque current. This is achieved by extracting a signal related to the secondary resistance fluctuation of the motor from this voltage fluctuation amount and controlling the converter output frequency according to this signal.

[Effect]

In a device that controls the motor voltage so that the deviation between the excitation current command of the motor and the detected value of the excitation current in the rotating magnetic field coordinate system and the deviation between the torque current command and the detected value of the torque current are O, the constant of the motor is The amount of voltage fluctuation due to the change includes a voltage drop component due to a change in the primary resistance and a voltage fluctuation component due to a change in the secondary resistance. Therefore, in the present invention, based on the amount of variation in the voltage command of the output voltage, the variation component of the motor voltage is decomposed into a component that is in phase with the motor current vector consisting of the excitation current and the torque current and a component that is orthogonal to the motor current vector (voltage drop due to resistance). However, the signal related to the secondary resistance fluctuation is extracted from this orthogonal output voltage component, and the reference value of the secondary resistance is corrected so that it becomes O to compensate for the slip frequency, so the primary resistance unaffected by fluctuations in

〔Example〕

An embodiment of the present invention will be described below with reference to FIG. The induction motor 2 is supplied with power from a power conversion circuit 1.

The three-phase alternating current flowing through the motor 2 is detected by a current detector 3tJ.

Detected by 3V, 3W. This three-phase alternating current is converted into a two-phase alternating current by a three-phase to two-phase converter 4, and its output is input to a coordinate converter 5, where it is vector decomposed into excitation current and torque current components in a rotating magnetic field coordinate system. The detected values are matched with respective command signals by adders 6 and 7. The rotational speed of the electric motor 2 is detected by a speed detector 8, and its output is compared with the command signal in an adder 9 and inputted to an adder 10. The output of the adder 9 is input to the speed controller 11, and the torque command Iq# of the electric motor 2 is inputted to the speed controller 11.
is output, and its output is input to the slip calculator 12 and the adder 7. The output Ws” of the slip computing unit 12 is output from the adder 10.
is added to the rotational speed W, and the output is added to the adder 2
3 is input. The output of the adder 6 is the deviation between the excitation current command Id rate and the detected value I6 of the coordinate converter 5, and the output is input to the current regulator 15, and the rotating magnetic field coordinate is adjusted so that the deviation becomes 0. The system voltage component command ΔV- is added to the adder 21
Output to. The output of the adder 7 outputs the deviation between the torque current command knee and the detected value Iq of the coordinate converter 5, and the output is input to the current regulator 16, and the rotating magnetic field coordinate system is adjusted so that the deviation becomes 0. The voltage component command Δvq* is outputted to the adder 22. The outputs from the current regulators 15 and 16 and the output from the coordinate converter 5 are input to the arithmetic circuit 18, and the output voltage fluctuation component ΔV due to the change in the secondary resistance of the motor 2 is output to the adder 23. This ΔV is added to the output wit of the adder 10 in the adder 23 to output the frequency W 1'' of the motor 2, and the output is input to the coordinate reference generator 13 and the voltage command calculation circuit 14. The command calculation circuit 14 outputs voltage component commands vd and vq* of the rotating magnetic field coordinate system based on the excitation current command I, the torque current command IqI, and the frequency w1*, and the output is sent to the adder 21.2.
2 is input. The outputs of these adders 21 and 22 are input to the coordinate converter 17, and based on the frequency w1** of the electric motor 2, voltage commands v1, 4, . Vv4y
Outputs V-.

These voltage commands V-, Vv*p V- are input to a pulse width modulation (PWM) control circuit 19, and output V u HV v is subjected to PWM control based on the carrier wave frequency.

vw is output to the power converter 1. In addition, the coordinate converter 5
, 17 are provided by a coordinate reference generator 13.

First, the operation of the above configuration will be briefly explained.

In this embodiment, the voltage commands y44 and vq are determined based on the excitation current command Id and the torque current command Iq in the rotating magnetic field coordinate system.
* is calculated, and a feedback control system is configured for the excitation current and torque current, and the voltage command v-2vq is adjusted by the output of each current regulator so that the deviation between the command value and the actual value becomes 0. * has been corrected, resulting in 71 voltage commands.

In the coordinate converter 17, yq** is converted to 3 in the stator coordinate system.
It is converted into a phase AC voltage command, and the voltage supplied to the electric motor 2 is controlled by the power converter 1 based on the converted three-phase AC voltage command.

Next, the details and operation of the arithmetic circuit 18 related to the present invention will be explained in the second section.
This will be explained using FIG. The electric motor 2 is normally operated with a voltage determined by the output V-, yq* of the voltage command calculation circuit 14. However, as the temperature of the motor changes with operation, the primary resistance and secondary resistance change, and currents equal to the excitation current command I- and the torque current command Iq no longer flow through the motor. As a result, the output of the current control system is Δ
V, *.

Δvq* occurs. This voltage command ΔV-2Δvq vehicle includes an output voltage fluctuation component due to a primary resistance change and an output voltage fluctuation component due to a secondary resistance change.

Therefore, in the present invention, these voltage commands Δy, *.

The variation of the output voltage based on Δvq is expressed as the excitation current ■,
The slip frequency change due to the secondary resistance change is detected by vector decomposition into a resistance voltage drop component in phase with the current vector consisting of the current vector and the torque current (9), and an output voltage component ΔV orthogonal to this current vector.

This orthogonal output voltage component can be determined from the vector diagram of current and voltage in the rotating magnetic field coordinate system shown in FIG. The output voltage component ΔV perpendicular to the current vector can be expressed by the following equation.

Δv, 1=Δv sin′cosO=−Δv dh 1sin O+Δvq*9cosO=
(-Δva*-Iq+Δv q$+ I a)/I
i-(+) where: If: Ila + Iq,
cos=Ia/It.

sjnθ=Iq/rt.

Further, ΔV 4 "p ΔVql includes an output voltage component due to a variation in the primary resistance and an output voltage component due to a variation in the secondary resistance. That is, ΔVdl and Δvq* can be expressed by the following equation.

Here, Δv4. Δ■9 is an output voltage component due to fluctuations in the secondary resistance.

Therefore, from equations (1) and (2), Δv1 can be rearranged as shown in the following equation.

Δv up=(-(ArtIa+8vd)I+(Δrl'
9+Δvq)・IJ/If=(−ΔVa”Iq+Δvq・I,+)/r
t·(3) From equation (3), there is no effect on ΔV due to the variation in the primary resistance, and only the output voltage variation component due to the variation in the secondary resistance is determined. This equation (3) is the calculation content of the calculation circuit 18. Also, between ΔV and 2 water resistance r2,
There is a relationship shown in FIGS. 3(a) and 3(b). Third
In Figure (a), when the motor 2 is in an electric state (Iq > 0),
FIG. 3(b) shows the case in the regenerative state (I q < O). In FIG. 3(a), when the water resistance r2 of the motor 2 increases by the reference value r2, the polarity above ΔV becomes negative, and conversely, when it decreases from rz, the polarity above ΔV becomes positive and the polarity is reversed.

On the other hand, in the regenerative state shown in FIG. 3(b), the polarity on V with respect to the change in the water resistance r2 is opposite to that in the electric state. Therefore, since the slip frequency is proportional to the magnitude of r2, in the motorized state, the primary frequency of the motor 2 is increased when the output ΔV of the arithmetic circuit 18 is of negative polarity, and when the output ΔV is of positive polarity, the primary frequency of the motor 2 is increased. If the primary angular frequency w1 is corrected and controlled according to Δv1 so that the primary frequency is decreased, the slip of the electric motor 2 can be controlled to an appropriate value. Further, in the regenerative state, the polarity on ΔV can be reversed to make a similar correction.

FIG. 4 shows another embodiment of the invention. 1 shown in Figure 1.
Components that are the same as those in the embodiment are given the same numbers, and therefore their description will be omitted. The difference from the first embodiment is that the current regulator is 1.5.
．． Based on the voltage commands ΔV- and Δvq* from 16, the excitation current command Ia'' and the torque current command Iq*, the variation of the motor voltage is converted into a component in phase with the motor current vector and a component orthogonal to the motor current vector, as in the previous embodiment. The point is that the frequency change due to the secondary resistance change is detected by vector decomposition.

This orthogonal output voltage component Δv is expressed by the following equation.

Δv1 = (-ΔVa"Iq increase+ΔvqII-L"
) / I t”・(4) Here, ■1ψ=JT'7 finished ma animals.

This equation (4) is the calculation content of the calculation circuit 18 of this embodiment.

This embodiment also provides the same effects as the first embodiment.

FIG. 5 shows another embodiment of the invention. Components that are the same as those in the first embodiment are given the same numbers, and therefore their description will be omitted. 1st
The difference from the embodiment is that the variation of the output voltage is changed based on the voltage command Δv-4ΔVqI from the current regulators 1.5 and 1.6 into an output that is in phase with the current vector consisting of the exciting current ■ and the torque current regulator 1. The vector is decomposed into an output voltage component and an output voltage component Δvl orthogonal to this current vector, and the change Δr2 in the secondary resistance of the motor 2 is calculated from this Δ■ to correct the coefficient ks of the sublime calculation circuit 12. This is what I decided to do. The change amount Δr2 in the secondary resistance is expressed by the function F of the following equation.

Here, φa""M'Id", r2" is the reference value of the secondary resistance, W is the primary angular frequency, M is the excitation inductance of the motor 2, and k is the proportional gain.

Further, the coefficient 8 of the slip calculation circuit 12 is expressed by the following equation.

φ-L.

Here, T-r = M + Q2, Q2 is 2
This is the leakage inductance on the next side. (1), (5)
, (6) is the calculation content of the calculation circuit 18 of this embodiment.

FIG. 6 shows another embodiment of the invention. First, the same numbers as in the embodiment are given to the same parts, so the explanation will be omitted. The difference from the first embodiment is that, based on the voltage commands Δv4* and Δ■q from the current regulators 15 and 16, the fluctuation of the output voltage is in phase with the current vector consisting of the excitation current ■ and the torque current ■. vector decomposition into an output voltage component and an output voltage component ΔV orthogonal to this current vector, calculate the change Δrz in the secondary resistance of the motor 2 from this ΔV, and calculate the secondary resistance used in the slip calculation circuit 12. The point is that the size has been corrected. In this embodiment, the magnitude of the secondary resistance l”
2$6 is calculated by the adder 23 as the sum of the preset secondary resistance reference value rz and Δr2 calculated from the above ΔV and L, and this calculated r2** is the secondary resistance of the motor 2. In order to match, in the electric state (1, > O) Δ
When V and L have negative polarity, rz is increased to increase the slip angular frequency, and when ΔV is positive, r20 is decreased to decrease the slip angular frequency.
” is corrected, and as a result, ΔV becomes O.Integrator 24
stores the result of identifying Δr2 as described above, and operates to correct Δr2 when ΔV↓ is no longer O.

According to this embodiment, since the fluctuation amount ΔrZ of the secondary resistance is automatically calculated through correction of the slip angular frequency Wsl,
This has the effect that the secondary resistance can always be identified with high accuracy.

According to this embodiment, since the magnitude of the secondary resistance of the motor 2 can be identified, the rotor temperature of the motor can be estimated from this magnitude, and when the temperature exceeds a predetermined value, the current limit value of the inverter main circuit can be lowered. , or by shutting off the power, it is effective to protect against overheating.

FIG. 7 shows another embodiment of the invention. Components that are the same as those in the first embodiment are given the same numbers, and therefore their description will be omitted. 1st
The difference from the embodiment is that the output voltage is adjusted based on the voltage commands v-* and Vq** from the adders 21 and 22, and the excitation current I
an output voltage component that is in phase with the current vector consisting of a and the torque current generator 9, and an output voltage component V that is orthogonal to this current vector.
The point is that the frequency change due to the secondary resistance change of the electric motor 2 is detected from this V-factor by vector decomposition. This orthogonal output voltage component V is expressed by the following equation.

V engineering==(va**・Iq+Vq**' Id)/I
t - (7) Here, 11 = 6 - r 璽士ding.

On the other hand, in a normal state, the electric motor 2 is operated by the voltage command calculation circuit 1.
It is operated by the voltage determined by the outputs v, *, vq of 4. The calculation content of the voltage command calculation circuit 14 is expressed by the following equation.

(8) Here, Lsσ is the sum of the leakage inductance on the primary side and the secondary side of the motor 2, Lr=M+Qz, Qz is the leakage inductance on the secondary side, and rl is the primary resistance.

Therefore, under normal conditions, the motor voltage component vlo orthogonal to the current vector of the motor 2 is expressed by the following equation.

v on o = (Vdl'Iq+vq" ・ x, t
) / I ILr 11 (9) Therefore, the amount of variation ΔV in V due to the temperature change of the electric motor 2 is expressed by the following equation from equations (7) and (9).

ΔVl = VI VLO =(-(Δri・I+i+Δvll)・engine.

+ (ΔrtIq+Δvq) I a) / I
t= (-ΔV,・Iq+Δvq・Id)/It・
(10) Equations (7), (9), and (10) are the calculation contents of the calculation circuit 18 of this embodiment. From this, the Δ■ process is not affected by fluctuations in the primary resistance, and by correcting the primary angular frequency according to ΔV, the same effect as in the first embodiment can be obtained.

Although each embodiment has been described above using analog circuit blocks to make the explanation easier to understand, it goes without saying that the embodiments can also be implemented by digital control using a microprocessor.

〔Effect of the invention〕

According to the present invention, even if the primary resistance and the resistance due to the on-delay of the PWM inverter change, the motor is not affected by the setting error of the reference value of the secondary resistance from the actual value and the fluctuation of the secondary resistance. This has the effect of preventing fluctuations in torque and voltage.

According to the present invention, in order to prevent a DC short circuit caused by simultaneous conduction of switching elements connected in series in an inverter, a period in which both elements are non-conductive is provided at the time of switching between conduction and non-conduction. The internal voltage drop of the inverter is related to the polarity of the output current and is the same in phase relationship as the primary resistance drop of the motor.

Therefore, according to the present invention, not only the effect of the primary resistance drop but also the effect of the above-mentioned inverter internal voltage drop can be compensated at the same time.

[Brief explanation of the drawing]

FIG. 1 is a control configuration diagram showing an embodiment of the present invention, FIG. 2 is a vector diagram for explaining the present invention in detail, FIG. 3 is a characteristic diagram for explaining the present invention in detail, and FIG. Figure, Figure 5,
FIGS. 6 and 7 are control configuration diagrams showing other embodiments of the present invention.

Claims (1)

[Scope of Claims] 1. In a device that controls an induction motor using a frequency conversion device, the difference between the excitation current command and the excitation current detection signal of the induction motor and the difference between the torque current command and the torque current detection signal in the rotating magnetic field coordinate system. Obtain a voltage command in the rotating magnetic field coordinate system according to the difference, convert that voltage command to an AC voltage command in the stator coordinate system, and control the voltage supplied to the induction motor based on this converted AC voltage command. A method for controlling an induction motor, comprising means for modifying the output frequency of the frequency conversion device based on the voltage command of the rotating magnetic field coordinate system. 2. In a device that controls an induction motor using a frequency conversion device, the rotating magnetic field is adjusted according to the difference between the excitation current command and the excitation current detection signal and the difference between the torque current command and the torque current detection signal of the induction motor in the rotating magnetic field coordinate system. means for obtaining a voltage command in a coordinate system, converting it into an alternating current voltage command in a stator coordinate system, and controlling the voltage supplied to the induction motor based on the voltage command in the rotating magnetic field coordinate system; Based on the excitation current and torque current, calculate a component of a motor voltage vector orthogonal to the motor current vector,
A method for controlling an induction motor, characterized in that the output frequency of the frequency converter is corrected based on the frequency converter. 3. In a device that controls an induction motor using a frequency conversion device, the rotating magnetic field is adjusted according to the difference between the excitation current command and the excitation current detection signal and the difference between the torque current command and the torque current detection signal of the induction motor in the rotating magnetic field coordinate system. means for obtaining a voltage command in a coordinate system, converting it into an alternating current voltage command in a stator coordinate system, and controlling the voltage supplied to the induction motor based on this, the q-axis voltage in the rotating magnetic field coordinate system; The command is multiplied by the excitation current (d-axis current), and d of the same coordinate system is
A method for controlling an induction motor, characterized in that the shaft voltage command and the torque current (q-axis current) are multiplied, and the output frequency of the frequency converter is corrected based on the difference between them. 4. In a device that controls an induction motor using a frequency conversion device, the rotating magnetic field is adjusted according to the difference between the excitation current command and the excitation current detection signal and the difference between the torque current command and the torque current detection signal of the induction motor in the rotating magnetic field coordinate system. means for obtaining a voltage command in a coordinate system, converting it into an alternating current voltage command in a stator coordinate system, and controlling the voltage supplied to the induction motor based on the voltage command in the rotating magnetic field coordinate system; Based on the excitation current and torque current, calculate a component of a motor voltage vector orthogonal to the motor current vector,
A method for controlling an induction motor, characterized in that a secondary resistance or a secondary time constant of the induction motor is calculated based on the calculated value, and a slip frequency of the induction motor is controlled based on the calculated value. 5. In a device that controls an induction motor using a frequency conversion device, the rotating magnetic field is adjusted according to the difference between the exciting current command and the exciting current detection signal of the induction motor and the difference between the torque current command and the torque current detection signal in the rotating magnetic field coordinate system. means for obtaining a voltage command in a coordinate system, converting it into an alternating current voltage command in a stator coordinate system, and controlling the voltage supplied to the induction motor based on this, the q-axis voltage in the rotating magnetic field coordinate system; The command is multiplied by the excitation current (d-axis current), and d of the same coordinate system is
The shaft voltage command is multiplied by the torque current (q-axis current), the secondary resistance or secondary time constant of the induction motor is calculated based on the difference between them, and the entire induction motor is calculated based on this calculated value. A method for controlling an induction motor, characterized in that the frequency is controlled by 6. The temperature of the secondary winding is estimated based on the change in the secondary resistance or the secondary time constant calculated in claim 4 or 5, and the temperature of the induction winding is estimated based on this temperature estimate. A method for protecting an induction motor, characterized in that the motor is prevented from overheating.