JP7603906B1 - AC/DC converters, rotating machine drives, and refrigeration cycle application equipment - Google Patents

Abstract

The AC-DC converter (2) includes a rectifier circuit (20) having a switching element (215) that rectifies the power supply voltage applied from the AC power supply (1), a capacitor (216) that is connected to the DC bus (9a, 9b) and smoothes the output voltage of the rectifier circuit (20), a reactor (212) that is disposed closer to the AC power supply (1) than the capacitor (216), and a control unit (6) that generates a switching signal for controlling the switching element (215). The control unit (6) generates a switching signal so that the harmonic components contained in the power supply current flowing between the AC power supply (1) and the rectifier circuit (20) comply with the harmonic standard value of the power supply current. The switching element (215) performs switching operation at least once per half cycle of the power supply voltage.

Description

The present disclosure relates to an AC/DC conversion device that converts AC power into desired DC power, as well as a rotating machine drive device and a refrigeration cycle application device that are equipped with the AC/DC conversion device.

When obtaining DC voltage from an AC power source, it is common to use a power factor correction circuit. A power factor correction circuit has the functions of controlling the bus voltage at a constant level and controlling the power supply current so as to comply with harmonic standards. A power factor correction circuit and one of its control methods, the "simple switching method" (also called the "partial switching method"), is a method in which switching is performed at least once per half cycle of the power supply voltage, which is the voltage of the AC power supply, and has the characteristic of being able to control the bus voltage to be lower than the peak value of the power supply voltage. However, when the bus voltage is set lower than the peak value of the AC power supply with the simple switching method, the operating circuit switches from a boost chopper to a capacitor-input type diode rectifier, which creates the problem of distorting the power supply current.

To address this issue, the conventional technology shown in Patent Document 1 below determines whether the combination of reactor capacity and switching timing complies with harmonic standards by repeating the design for each load power.

JP 2000-125545 A

However, the conventional technology described in Patent Document 1 is a method for checking whether or not compliance with harmonic standards can be achieved through repeated trials, which has the problem that the number of trials increases exponentially as the number of pulses increases. In addition, there is no clear guideline for quantitatively and uniquely designing control gains, which has the problem that it takes a long time to complete the design.

The present disclosure has been made in consideration of the above, and aims to reduce the time required for design by obtaining an AC-DC conversion device that can comply with harmonic standards without relying on trial-and-error adjustments.

In order to solve the above-mentioned problems and achieve the object, the AC-DC conversion device according to the present disclosure includes a rectifier circuit, a capacitor, a reactor, and a control unit. The rectifier circuit has at least one switching element and rectifies the power supply voltage applied from the AC power supply. The capacitor is connected to the DC bus and smoothes the output voltage of the rectifier circuit. The reactor is arranged on the AC power supply side of the capacitor. When generating a switching signal for controlling the switching element arranged on the AC power supply side of the capacitor, the control unit generates the switching signal so that the harmonic components contained in the power supply current flowing between the AC power supply and the rectifier circuit comply with the harmonic standard value of the power supply current. The switching element performs switching operation at least once per half cycle of the power supply voltage.

The AC-DC conversion device disclosed herein has the advantage of being able to comply with harmonic standards without relying on trial-and-error adjustments, thereby reducing the time required for design.

FIG. 1 is a block diagram showing a configuration example of a rotary machine driving device according to a first embodiment; A circuit diagram showing a configuration example of an AC-DC converter according to a first embodiment. FIG. 1 is a block diagram showing a configuration example of a control unit according to a first embodiment; FIG. 1 is a block diagram showing a configuration example of a voltage control unit according to a first embodiment; FIG. 1 is a block diagram showing a configuration example of a current control unit according to a first embodiment; FIG. 6 is a block diagram showing a configuration example in which a target value filter is introduced into the current control unit shown in FIG. FIG. 7 is a block diagram for explaining a transfer function of a current control system in an AC-DC converter including the current control unit shown in FIG. FIG. 1 is a block diagram showing a configuration example of a switching signal generating unit according to a first embodiment; FIG. 7 is a diagram showing an example of the operation waveforms of a power supply voltage, a bus voltage, and a power supply current when PI control is applied to the current controller shown in FIG. FIG. 7 is a diagram showing an example of the operation waveforms of the power supply voltage, the bus voltage, and the power supply current when the PS control is applied to the current controller shown in FIG. FIG. 7 is a diagram showing an example of current harmonic characteristics when PS control is applied to the current controller shown in FIG. FIG. 7 is a diagram showing an example of the relationship between the ratio of the sum of harmonic components and the operating power when PS control is applied to the current controller shown in FIG. FIG. 1 is a diagram showing a configuration example of an AC-DC converter according to a second embodiment; FIG. 13 is a diagram showing a configuration example of a refrigeration cycle application device according to a third embodiment.

The following provides a detailed description of the AC/DC conversion device, rotating machine drive device, and refrigeration cycle application equipment relating to embodiments of the present disclosure, with reference to the attached drawings.

Embodiment 1.
1 is a block diagram showing a configuration example of a rotating machine driving device 8 according to embodiment 1. The rotating machine driving device 8 is connected to an AC power source 1 and a load 4 including a motor 41. The rotating machine driving device 8 includes an AC/DC converter 2 and a DC/AC converter 3. When the rotating machine driving device 8 is used in an air conditioner, the load 4 is a compressor or a fan, and the motor 41 is a compressor motor or a fan motor.

Figure 2 is a circuit diagram showing an example configuration of the AC-DC converter 2 according to embodiment 1. The AC-DC converter 2 according to embodiment 1 includes, as its main components, a control unit 6, a rectifier circuit 20, a reactor 212, and a capacitor 216. The AC-DC converter 2 also includes, as voltage or current detection means, a current detection unit 211 and voltage detection units 217a and 217b. In this document, when the voltage detection units 217a and 217b are to be distinguished from each other without reference numbers, the voltage detection unit 217b is referred to as the "first voltage detection unit" and the voltage detection unit 217a is referred to as the "second voltage detection unit".

The rectifier circuit 20 includes single-phase diode bridge cells 213a and 213b in which four diodes are bridge-connected, and a switching element 215 connected in parallel to both ends of the single-phase diode bridge cell 213b. The single-phase diode bridge cells 213a and 213b are connected in parallel to each other with the AC power source 1. The rectifier circuit 20 as shown in FIG. 2 is called a "simple switching circuit." The single-phase diode bridge cell 213b and the switching element 215 constitute a switching cell 225. The switching element 215 performs a switching operation at least once per half cycle of the power supply voltage.

The capacitor 216 is connected between the DC bus 9a and the DC bus 9b. The reactor 212 is disposed closer to the AC power source than the capacitor 216. The rectifier circuit 20 receives the power source voltage applied from the AC power source 1 via the reactor 212 and rectifies the received power source voltage. The capacitor 216 smoothes the output voltage of the rectifier circuit 20.

The voltage detection unit 217b detects the bus voltage, which is the voltage of the DC buses 9a and 9b to which the capacitor 216 is connected. The voltage detection unit 217a detects the power supply voltage. The current detection unit 211 detects the power supply current flowing between the AC power supply 1 and the rectifier circuit 20.

The control unit 6 receives the detection values of the voltage detection units 217a, 217b and the current detection unit 211. The control unit 6 generates a switching signal for controlling the on and off of the switching element 215 based on each detection value.

An example of the switching element 215 is an insulated gate bipolar transistor (IGBT) as shown in the figure, but is not limited to an IGBT. Any element capable of switching operation may be used as the switching element 215. Another example of the switching element 215 is a metal-oxide-semiconductor field-effect transistor (MOSFET).

2 is configured as a closed loop using the detection values of the voltage detection units 217a, 217b and the current detection unit 211, but may be configured as an open loop using target values, estimated values, etc. When the AC-DC conversion device 2 is configured as an open loop, it is also possible to control the switching element 215 without using the detection values of the voltage detection units 217a, 217b and the current detection unit 211.

Figure 3 is a block diagram showing an example configuration of the control unit 6 according to the first embodiment. The control unit 6 includes a voltage control unit 61, a current control unit 62, and a switching signal generation unit 63. The voltage control unit 61 generates a first current command value using a first voltage command value. The current control unit 62 generates a second voltage command value using the first current command value. The switching signal generation unit 63 generates a switching signal using the second voltage command value.

FIG. 4 is a block diagram showing an example of the configuration of the voltage control unit 61 according to the first embodiment. The voltage control unit 61 includes a voltage controller 611 and a subtractor 612. The voltage control unit 61 generates a first current command value using a first voltage command value that is a command value of the bus voltage. In more detail, the subtractor 612 generates a voltage deviation that is the difference between the first voltage command value and the detected voltage detected by the voltage detection unit 217b. The voltage controller 611 generates the first current command value using the voltage deviation output from the subtractor 612. The voltage controller 611 can be configured, for example, as a PI (Proportional Integral) controller.

When the voltage controller 611 is configured as a PI controller, the transfer function G _AVR(s) can be expressed by the following equation (1).

Here, "AVR" in the transfer function G _AVR(s) is an abbreviation for "Automatic Voltage Regulator." In addition, in the above formula (1), K _pAVR is a proportional gain, K _iAVR is an integral gain, and s is a Laplace operator. In the PI controller, the proportional gain K _pAVR and the integral gain K _iAVR can be determined arbitrarily. Note that the proportional gain K _pAVR may be set to zero to configure an I controller, or the integral gain K _iAVR may be set to zero to configure a P controller.

FIG. 5 is a block diagram showing an example of the configuration of the current control unit 62 according to the first embodiment. The current control unit 62 includes a current controller 621, a subtractor 622, and a multiplier 623. The current control unit 62 generates a second voltage command value using a second current command value that causes the first current command value to follow a sine wave. In more detail, the multiplier 623 multiplies the first current command value by an excitation signal. The excitation signal is a sine wave synchronized with the phase of the power supply voltage. The sine wave is generated based on the voltage phase, which is the phase of the detected voltage detected by the voltage detection unit 217a. The output of the multiplier 623 is input to the subtractor 622 as a second current command value. The subtractor 622 generates a current deviation, which is the difference between the second current command value and the detected current detected by the current detection unit 211. The current controller 621 generates a second voltage command value using the current deviation output from the subtractor 622. The current controller 621 can be configured, for example, by a PS (Proportional Sinusoidal) controller.

When the current controller 621 is configured as a PS controller, the transfer function G _ACR(s) can be expressed by the following equation (2).

Here, "ACR" in the transfer function G _ACR(s) is an abbreviation for "Automatic Current Regulator." In addition, in the above formula (2), K _pACR is a proportional gain, K _sACR is an S control gain, _ωn is an angular frequency, and s is a Laplace operator. In the PS controller, the proportional gain K _pACR , the S control gain K _{sACR ,} and the angular frequency _ωn can be arbitrarily determined.

The PS controller is a controller in which an S controller, which is a Laplace transform expression of a sine wave function or a cosine wave function, is inserted in parallel to a P controller. The S controller is a controller with improved tracking performance for a sine wave input with an angular frequency _ωn . The reason why the S controller has improved tracking performance for an input pulsating with an angular frequency _ωn can be explained by the internal model principle. The internal model principle is that if the denominator of the controller has the same factor as the denominator polynomial of the command value expressed by the Laplace transform, the command value can be tracked without deviation. The current controller 621 may be configured as a PIS controller by inserting an I controller in parallel to the PS controller.

Figure 6 is a block diagram showing a configuration example in which a target value filter 624 is introduced into the current control unit 62 shown in Figure 5. Also, Figure 7 is a block diagram used to explain the transfer function of the current control system 7 in the AC-DC converter 2 including the current control unit 62 shown in Figure 6.

In Fig. 6, a target value filter 624 is inserted in front of the current control unit 62 shown in Fig. 5. The target value filter 624 is inserted to adjust the response of the transfer function of the current control unit 62 shown in Fig. 5. Specifically, the response of the transfer function of the current control unit 62 is adjusted by making the zero points of the transfer function of the current control unit 62 be cancelled by the poles of the target value filter 624.

Here, the closed loop transfer function G _close(s) in the current control system 7 in FIG.

In the above formula (3), _Gc is the transfer function of an arbitrary controller 72, and _Gp is the transfer function of an arbitrary controlled plant 73. When the above formula (3) has a zero point, the response of the closed loop cannot be determined by the poles alone. Therefore, a target value filter 74 having the same pole as the zero point of the closed loop transfer function Gclose _(s) is inserted in front of the closed loop. Then, a pole-zero cancellation is performed between the transfer function _GF(s) of the target value filter 74 and the closed loop transfer function Gclose( _s) , and the desired response is realized by canceling the influence of the zero point. The transfer function G'close _(s) after the insertion of the target value filter 74 can be expressed by the following formula (4).

In the above formula (4), G _X is an arbitrary transfer function. The transfer function G _X may be the value of the zeroth power term of the Laplace operator s in the denominator polynomial of the transfer function G' _close(s) , or may have an arbitrary zero point. By applying the target value filter 74 having such a transfer function G _F(s) , the controller response can be adjusted. Note that a PI controller may be used as a controller that obtains the same effect as the target value filter 74.

Figure 8 is a block diagram showing an example configuration of the switching signal generating unit 63 according to the first embodiment. The switching signal generating unit 63 includes a scaler 630, a carrier signal generator 631, and a comparator 632. The scaler 630 scales the second voltage command value input to the switching signal generating unit 63. If the control gain of the current controller 621 that outputs the second voltage command value is designed taking scaler into consideration, the scaler 630 is not necessary.

The carrier signal generator 631 generates a carrier signal used to generate a switching signal and outputs it to the comparator 632. The carrier signal is a bipolar or unipolar signal having a triangular waveform that operates at an arbitrary frequency. The comparator 632 generates a switching signal based on the output of the standardizer 630 and the output of the carrier signal generator 631. In more detail, the comparator 632 compares the output value of the standardizer 630 with the output value of the carrier signal generator 631, and if the output value of the standardizer 630 is greater than the output value of the carrier signal generator 631, it outputs a signal that turns on the switching element 215. Also, if the output value of the standardizer 630 is smaller than the output value of the carrier signal generator 631, the comparator 632 outputs a signal that turns off the switching element 215. In addition, the configuration may be reversed to this operation, in which an ON operation signal is generated when the output value of the scaler 630 is smaller than the output value of the carrier signal generator 631, and an OFF operation signal is generated when the output value of the scaler 630 is greater than the output value of the carrier signal generator 631.

In addition, the carrier signal generated by the carrier signal generator 631 does not have to be triangular wave-shaped and can have any configuration as long as it can generate a switching signal from the second voltage command value.

Next, the points to consider when configuring the switching signal generating unit 63 will be explained. To suppress the switching loss of the switching element 215 in the switching cell 225, it is desirable to have a small number of switching operations. To reduce the number of switching operations, the switching frequency can be lowered. On the other hand, if the switching frequency is lowered too much, the frequency components of the carrier signal will be included in the components up to the 40th order of the fundamental frequency defined by the current harmonics standard.

Here, the fundamental frequency is the frequency of the power supply voltage, which in Japan is 50 Hz or 60 Hz. The 40th order of 50 Hz is 2 kHz, and the 40th order of 60 Hz is 2.4 kHz. Therefore, when the frequency of the power supply voltage is 50 Hz, the carrier frequency, which is the frequency of the carrier signal, must be 2 kHz or higher. Also, when the frequency of the power supply voltage is 60 Hz, the carrier frequency must be 2.4 kHz or higher. This can be expressed mathematically as in the following equation (5).

In the above formula (5), _fsw is the carrier frequency, _fsi is the frequency of a sideband wave relative to the carrier frequency _fsw , _fs is the frequency of the power supply voltage, and n is an integer of 1 or more.

In the first embodiment, the carrier frequency _fsw , which is the switching frequency when the switching element 215 performs a switching operation, may be determined so that the component of the sideband frequency _fsi satisfies the harmonic standard. Specifically, the carrier frequency fsw is determined so as to satisfy the following formula (6).

In the above formula (6), _Nsw is the number of switching times in one cycle of the power supply voltage. Also, ( _fsw / _fs ) shown on the right side is the ratio of the carrier frequency _fsw to the frequency _fs of the power supply voltage, and is the frequency obtained by dividing the carrier frequency _fsw by the frequency _fs of the power supply voltage, i.e., normalized by the frequency _fs of the power supply voltage. That is, in the first embodiment, the number of switching times _Nsw in one cycle of the power supply voltage is determined to be 1 or more and equal to or less than the value ( _fsw / _fs ) obtained by dividing the carrier frequency _fsw by the frequency _fs of the power supply voltage.

In the first embodiment, the above formula (6) is shown as a constraint on the number of switching times _Nsw in one cycle of the power supply voltage, but it is not necessary to satisfy this in all operation regions of the load 4. For example, when the load 4 operates in a light load region, the switching operation of the switching element 215 may be stopped, so the above formula (6) does not have to be satisfied.

Next, the operating characteristics when PS control is applied to the current controller 621 are clarified by comparing them with the conditions when a PI controller is applied to the current controller 621.

Figure 9 is a diagram showing example operating waveforms of the power supply voltage, bus voltage, and power supply current when PI control is applied to the current controller 621 shown in Figure 6. The upper part of Figure 9 shows the waveforms of the absolute values of the bus voltage and power supply voltage. The lower part of Figure 9 shows the waveforms of the detected power supply current, the fundamental wave component of the detected power supply current, and the power supply current command value. The detected power supply current is the detected waveform of the power supply current detected by the current detection unit 211.

The operating conditions in Figure 9 are those in which the bus voltage is below the peak absolute value of the power supply voltage, as shown in the upper part of Figure 9. Under these conditions, the rectifier circuit 20 operates as a capacitor-input type diode rectifier circuit rather than as a boost circuit, making it impossible to control the bus voltage. At this time, excessive values accumulate in the PI control integrator, causing a wind-up phenomenon and preventing the detected power supply current from following the power supply current command value.

In contrast, Fig. 10 is a diagram showing example operating waveforms of the power supply voltage, bus voltage, and power supply current when PS control is applied to the current controller 621 shown in Fig. 6. As in Fig. 9, the upper part of Fig. 10 shows the waveforms of the absolute values of the bus voltage and power supply voltage, and the middle part of Fig. 10 shows the waveforms of the detected power supply current, the fundamental wave component of the detected power supply current, and the power supply current command value. In addition, the lower part of Fig. 10 shows a switching signal for controlling the switching element 215.

As shown in the middle part of Figure 10, the fundamental wave component of the detected power supply current is almost equal to the power supply current command value. This shows that if PS control with high tracking performance for sine wave input is applied to the current controller 621, the fundamental wave of the detected power supply current can track the power supply current command value even under conditions where the bus voltage is equal to or less than the peak value of the absolute value of the power supply voltage. Note that Figure 10 shows the results for operating conditions where the bus voltage is equal to or less than the peak value of the absolute value of the power supply voltage, but it goes without saying that the fundamental wave of the detected power supply current will track the power supply current command value even under operating conditions where the bus voltage exceeds the peak value of the absolute value of the power supply voltage.

Figure 11 is a diagram showing an example of current harmonic characteristics when PS control is applied to the current controller 621 shown in Figure 6. The current harmonic standard used in Figure 11 is IEC 61000-3-2 Class A. Note that IEC 61000-3-2 Class A is an example of a current harmonic standard, and is not limited to this standard.

In Figure 11, the solid lines show the standard values for current harmonics from the 2nd to the 40th orders as specified in IEC 61000-3-2 Class A. Also in Figure 11, the dashed lines show the effective values of the 2nd to the 40th harmonic components during rated operation. The harmonic components during rated operation represent the components remaining after subtracting the fundamental component of the power supply current from the power supply current that flows when the AC-DC converter 2 is operated at rated power. In this paper, the 2nd to the 40th harmonics are defined as "low-order harmonics".

According to Fig. 11, the waveforms of the dashed lines are lower than those of the solid lines from the 2nd to the 40th orders. This shows that if PS control, which has high tracking performance for sine wave input, is applied to the current controller 621, the low-order harmonics contained in the power supply current will comply with the harmonic standard value of the power supply current. Note that Fig. 11 shows the harmonic components during rated operation as an example, but it goes without saying that the effect of suppressing harmonic components by PS control can be obtained even when not operating at rated speed.

Figure 12 is a diagram showing an example of the relationship between the ratio of the sum of harmonic components and the operating power when PS control is applied to the current controller 621 shown in Figure 6. The ratio of the sum of harmonic components here means the ratio of the sum of second and higher harmonic components to the fundamental wave of the power supply current.

It can be seen from Figure 12 that the proportion of the sum of the harmonic components is 10% or more in all operating regions shown in Figure 12. The fact that the proportion of the sum of the harmonic components is 10% or more is unique to the method of embodiment 1 in which PS control is applied to the current controller 621 to control the switching element 215. When the operating power is small, the proportion of the sum of the harmonic components increases, but the harmonic standard value when the operating power is small also increases, and as shown in Figure 11, the effective value of the current harmonics does not exceed the harmonic standard value of the power supply current.

The control method of the first embodiment is characterized in that a current controller 621 is applied to the control unit 6, and then PS control, which has high tracking performance for a sine wave input, is applied to the current controller 621. It is considered that the application of PS control to improve tracking performance for a sine wave input is relatively common, but the application of PS control to comply with harmonic standards is considered to be a novel method that has not been used before.

As described above, the AC-DC converter according to the first embodiment includes a rectifier circuit that rectifies the power supply voltage applied from the AC power supply, a capacitor that smoothes the output voltage of the rectifier circuit, and a reactor that is arranged on the AC power supply side of the capacitor. The rectifier circuit has at least one switching element that is arranged on the AC power supply side of the capacitor. When generating a switching signal for controlling the switching element, the control unit generates a switching signal so that the harmonic components contained in the power supply current flowing between the AC power supply and the rectifier circuit comply with the harmonic standard value of the power supply current. The switching element performs switching operation at least once per half cycle of the power supply voltage. According to the AC-DC converter according to the first embodiment, it is possible to comply with the harmonic standard without relying on a trial-and-error adjustment method that confirms whether or not compliance with the harmonic standard can be achieved by repeated trials. Furthermore, according to the AC-DC converter according to the first embodiment, even under operating conditions in which the bus voltage exceeds the peak value of the absolute value of the power supply voltage, it is possible to comply with the harmonic standard while improving the input power factor.

In order to realize the above function, the control unit according to the first embodiment is configured to include a voltage control unit that generates a first current command value using a first voltage command value, which is a command value for the bus voltage, a current control unit that generates a second voltage command value using a second current command value that makes the first current command value follow a sine wave, and a switching signal generation unit that generates a switching signal using the second voltage command value. The current control unit has a current controller, and PS control is applied to the current controller. In addition, the input stage of the current control unit is configured to include a filter that operates the zero point of a feedback loop that feeds back the second voltage command value output by the current control unit to the input stage of the second current command value. By using a control unit configured in this way, it is no longer necessary to repeat the design for each load power to determine whether the combination of reactor capacity and switching timing can comply with the harmonic standard, so that trial and error adjustment work is eliminated, and it is possible to shorten the time required for design work and the time until design is completed.

It is desirable for the switching signal generating unit to generate a switching signal so that the number of switching operations in one power supply voltage cycle when the switching element performs switching operation is at least once and the carrier frequency, which is the frequency of the carrier signal, is equal to or less than the value normalized by the frequency of the power supply voltage. By generating such a switching signal, it is possible to suppress switching losses in the switching cell.

When the switching element is controlled using the control unit configured as described above, low-order harmonics are superimposed on the power supply current flowing between the AC power supply and the AC-DC converter. The sum of the harmonic components of the power supply current is 10% or more of the fundamental wave of the power supply current.

In addition, when the control system of the AC-DC converter is configured as a closed loop, a current detection unit that detects the power supply current flowing between the AC power supply and the rectifier circuit, a first voltage detection unit that detects the bus voltage, and a second voltage detection unit that detects the power supply voltage are provided. When the control system of the AC-DC converter is configured as a closed loop, the first current command value is generated using a voltage deviation that is the difference between the first voltage command value and the detected voltage detected by the first voltage detection unit, and the sine wave that excites the first current command value is generated based on the voltage phase detected by the second voltage detection unit. In addition, the second voltage command value is generated using a current deviation that is the difference between the second current command value and the detected current detected by the current detection unit.

Embodiment 2.
In the first embodiment, a configuration has been described in which PS control is applied to the current controller 621 of the current control unit 62 for the purpose of improving the tracking performance for a sine wave command. In the second embodiment, a configuration will be described in which PIR (Proportional Integral Resonant) control is applied to the current controller 621 of the current control unit 62 for the same purpose.

Figure 13 is a diagram showing an example configuration of an AC-DC conversion device 2 according to embodiment 2. Components having the same or equivalent functions as those in Figure 2 are indicated with the same reference numerals. Note that descriptions of contents that overlap with embodiment 1 will be omitted as appropriate.

The rectifier circuit 20 shown in Fig. 13 has a configuration called a full PAM (Pulse Amplitude Modulation) circuit. The rectifier circuit 20 includes a single-phase diode bridge cell 213a, a switching element 215, and a diode 218. In the single-phase diode bridge cell 213a, the switching element 215 performs a switching operation at least once per half cycle of the power supply voltage.

2 is disposed between the single-phase diode bridge cell 213a and the diode 218. The switching element 215 is connected in parallel to the single-phase diode bridge cell 213a and the capacitor 216 between the single-phase diode bridge cell 213a and the capacitor 216. The current detection unit 211 described in the configuration of FIG. 2 is disposed between the single-phase diode bridge cell 213a and the diode 218.

In the configurations of Figures 2 and 13, the arrangement positions and connection forms of the switching element 215 and the reactor 212 are different, but in both configurations, the switching element 215 and the reactor 212 are arranged on the AC power source 1 side relative to the capacitor 216.

In FIG. 13, the switching element 215 is shown as an IGBT, but any element capable of switching operation may be used. In addition, although the AC-DC converter 2 shown in FIG. 13 is configured as a closed loop, it may also be configured as an open loop. When the AC-DC converter 2 is configured as an open loop, it is not necessary to use the detection values of the voltage detection units 217a, 217b and the current detection unit 211.

In addition, the full PAM circuit of Fig. 13 may be used in the first embodiment, and the simple switching circuit of Fig. 2 may be used in the second embodiment. That is, when the AC-DC converter 2 has a step-up function or a step-down function, the rectifier circuit 20 has at least one switching element, but the control method described in this paper is applicable even if the control method for controlling the switching element is different.

Next, the transfer function of the PIR controller will be described. First, the transfer function G _PIR(s) of the PIR controller can be expressed by the following equation (7).

In the above formula (7), _Kp is a proportional gain, _Ki is an integral gain, _Kr is a resonance control gain, _ω1 is the angular frequency of the current control response, and _ω2 is the angular frequency of the sine wave command to be followed. A PIR controller having such a transfer function G _PIR(s) may be applied to the current controller 621 in Fig. 5 or 6. The same effect as that of PS control can be obtained by using a PIR controller.

As described above, in the control unit according to the second embodiment, PIR control is applied to the current controller provided in the current control unit. Even when PIR control is applied instead of PS control, the same effect as in the first embodiment can be obtained.

Embodiment 3
14 is a diagram showing a configuration example of a refrigeration cycle-applied device 900 according to embodiment 3. The refrigeration cycle-applied device 900 according to embodiment 3 includes the rotating machine drive device 8 described in embodiment 1. The refrigeration cycle-applied device 900 according to embodiment 3 can be applied to products including a refrigeration cycle, such as air conditioners, refrigerators, freezers, and heat pump water heaters.

The refrigeration cycle application device 900 includes a compressor 42 incorporating the motor 41 in the first embodiment, a four-way valve 902, an indoor heat exchanger 906, an expansion valve 908, and an outdoor heat exchanger 910 attached via a refrigerant piping 912. Inside the compressor 42, a compression mechanism 904 that compresses the refrigerant and a motor 41 that operates the compression mechanism 904 are provided. The refrigeration cycle application device 900 can perform heating or cooling operation by switching the four-way valve 902.

The compression mechanism 904 is driven by a motor 41 that is variable speed controlled. During heating operation, as shown by the solid arrow, the refrigerant is pressurized by the compression mechanism 904 and sent out, and returns to the compression mechanism 904 through the four-way valve 902, the indoor heat exchanger 906, the expansion valve 908, the outdoor heat exchanger 910, and the four-way valve 902. During cooling operation, as shown by the dashed arrow, the refrigerant is pressurized by the compression mechanism 904 and sent out, and returns to the compression mechanism 904 through the four-way valve 902, the outdoor heat exchanger 910, the expansion valve 908, the indoor heat exchanger 906, and the four-way valve 902. During heating operation, the indoor heat exchanger 906 acts as a condenser to release heat, and the outdoor heat exchanger 910 acts as an evaporator to absorb heat. During cooling operation, the outdoor heat exchanger 910 acts as a condenser to release heat, and the indoor heat exchanger 906 acts as an evaporator to absorb heat. The expansion valve 908 reduces the pressure of the refrigerant to expand it.

Although the refrigeration cycle application equipment 900 according to embodiment 3 has been described as being equipped with the rotating machine drive device 8 described in embodiment 1, the present invention is not limited to this. It may also be equipped with a rotating machine drive device 8 equipped with the rectifier circuit 20 described in embodiment 2. Furthermore, the rotating machine drive device 8 may be equipped with a rectifier circuit other than the rectifier circuit 20 described in embodiments 1 and 2, as long as the control method of embodiment 1 or embodiment 2 can be applied.

The configurations shown in the above embodiments are merely examples, and may be combined with other known technologies, or the embodiments may be combined with each other. In addition, parts of the configurations may be omitted or modified without departing from the spirit of the invention. For example, the above-mentioned control method may also be applied to a DC-AC converter.

1 AC power source, 2 AC/DC converter, 3 DC/AC converter, 4 Load, 6 Control unit, 7 Current control system, 8 Rotating machine drive unit, 9a, 9b DC bus, 20 Rectifier circuit, 41 Motor, 42 Compressor, 61 Voltage control unit, 62 Current control unit, 63 Switching signal generation unit, 72 Controller, 73 Control target plant, 74, 624 Target value filter, 211 Current detection unit, 212 Reactor, 213a, 213b Single-phase diode bridge cell, 215 Switching element, 216 Capacitor, 217a, 217b Voltage detection unit, 218 Diode, 225 Switching cell, 611 Voltage controller, 612, 622 Subtractor, 621 Current controller, 623 Multiplier, 630 Standardizer, 631 Carrier signal generator, 632 Comparator, 900 refrigeration cycle application device, 902 four-way valve, 904 compression mechanism, 906 indoor heat exchanger, 908 expansion valve, 910 outdoor heat exchanger, 912 refrigerant piping.

Claims (8)

a rectifier circuit having at least one switching element and rectifying a power supply voltage applied from an AC power supply;
a capacitor connected to a DC bus for smoothing an output voltage of the rectifier circuit;
a reactor that is disposed closer to the AC power source than the capacitor;
a control unit that generates a switching signal for controlling the switching element;
Equipped with
the switching element is disposed closer to the AC power supply than the capacitor;
The control unit generates the switching signal so that harmonic components included in a power supply current flowing between the AC power supply and the rectifier circuit comply with a harmonic standard value of the power supply current, and the number of switching operations in one period of a power supply voltage when the switching element performs a switching operation is one or more times and is equal to or less than a value obtained by dividing a carrier frequency, which is a frequency of a carrier signal, by a frequency of the power supply voltage.
1. An AC/DC converter comprising: The control unit is
a voltage control unit that generates a first current command value using a first voltage command value that is a command value of a bus voltage that is a voltage of the DC bus;
a current control unit that generates a second voltage command value using a second current command value obtained by making the first current command value follow a sine wave;
a switching signal generating unit that generates the switching signal by using the second voltage command value;
2. The AC/DC converter according to claim 1, further comprising: 3. The AC-DC converter according to claim 2, wherein the current control unit includes a current controller configured as a proportional sine wave controller, a proportional integral sine wave controller, or a proportional integral resonant controller . a current detection unit that detects the power supply current;
A first voltage detection unit that detects the bus voltage;
a second voltage detection unit that detects the power supply voltage;
Equipped with
the first current command value is generated using a voltage deviation that is a difference between the first voltage command value and a detection voltage detected by the first voltage detection unit,
the sine wave is generated based on a voltage phase detected by the second voltage detection unit,
3. The AC-DC converter according to claim 2 , wherein the second voltage command value is generated using a current deviation that is a difference between the second current command value and a current detected by the current detection unit. 3. The AC-DC converter according to claim 2, wherein low-order harmonics are superimposed on the power supply current. 3. The AC-DC converter according to claim 2, wherein the current control unit includes a filter that operates a zero point of a feedback loop that feeds back the second voltage command value output by the current control unit to an input stage of the second current command value. A rotating machine drive device equipped with an AC/DC converter according to any one of claims 1 to 6. A refrigeration cycle application device equipped with an AC-DC converter according to any one of claims 1 to 6.

Images