JP7585563B2 - Current and Leakage Sensors - Google Patents

Description

The present invention relates to a magnetic balance type current sensor that detects the current to be measured under conditions where the magnetic flux density of the magnetic core is close to zero, and a leakage current sensor that uses the current sensor.

Magnetic balance type current sensors have been known for some time (see, for example, Patent Documents 1 to 6).

Of these patent documents 1 to 6, patent document 1 discloses a current sensor in which two coils, a circulating coil and a detection coil, are wound around a magnetic core.

Patent Document 2 also discloses a current sensor in which a gap is provided in a part of a closed magnetic circuit, a ferromagnetic core is inserted into the gap, a feedback coil is wound around the closed magnetic circuit, and an excitation coil and a receiving coil are wound around the ferromagnetic core.

Patent document 3 also discloses a current sensor in which a feedback coil is wound around a magnetic core and a Hall element is placed in a notch in the magnetic core.

Furthermore, Patent Document 4 discloses a current sensor that, like Patent Document 3, has a coil wound around a magnetic core and a Hall element disposed in a portion of the magnetic core.

Furthermore, Patent Document 5 discloses a configuration in which two coils, an excitation winding and a detection winding, are wound around a magnetic core, and a configuration in which an excitation winding is wound around a magnetic core and a magnetic sensor such as a Hall element is disposed.

Patent document 6 also shows a configuration in which only one excitation coil is wound around the magnetic circuit in the drawings. However, because the measurable current range is severely limited when only an excitation coil is wound around the magnetic circuit, it is described in text that the circuit is provided with a configuration for canceling excitation. To achieve this, it is necessary to add a coil for excitation cancellation, which ultimately requires two coils.

As described above, these conventional magnetic balance type current sensors have a magnetic core through which an electric wire carrying the current to be detected is inserted, and a configuration in which multiple coils are wound around the magnetic core, or a configuration in which coils are wound around the magnetic core and a magnetic sensor such as a Hall element is disposed.

Japanese Unexamined Patent Publication No. 2-291973 JP 11-281678 A JP 2005-55300 A JP 2015-34758 A Patent No. 5449222 Patent No. 5606521

Conventional magnetic balance type current sensors require multiple elements, such as a magnetic core, multiple coils such as an excitation coil and a feedback coil, or an excitation coil and a magnetic sensor. This has drawbacks such as a complex circuit configuration and a large current sensor.

The present invention aims to provide a magnetic balance type current sensor that has a simplified configuration and is highly sensitive, and a leakage current sensor that uses the current sensor.

The current sensor of the present invention comprises:
a current transformer having a magnetic core forming a closed magnetic circuit and an excitation winding wound around the magnetic core, the closed magnetic circuit being passed through by an electric wire through which a current to be measured flows;
a drive circuit that applies an excitation voltage to a first end of the excitation winding;
a current-voltage conversion circuit connected to a second end of the excitation winding for converting a current flowing through the excitation winding into a voltage;
a first waveform conversion circuit that converts an output of the current/voltage conversion circuit into a first rectangular wave;
a second waveform conversion circuit that converts the output of the current-to-voltage conversion circuit into a second rectangular wave with hysteresis;
a reference value generating circuit that generates a reference value to be referred to as both the threshold value of the first waveform transformation circuit and the central threshold value of the second waveform transformation circuit based on the first rectangular wave;
The drive circuit generates an excitation voltage derived from a second rectangular wave.

The current sensor of the present invention achieves self-oscillation through the above-mentioned configuration. This realizes a highly sensitive magnetic balance type current sensor with a simplified configuration.

In the current sensor of the present invention,
a frequency measurement circuit for measuring a repetition frequency of the first rectangular wave;
It is preferable to further include a threshold generating circuit that generates an upper threshold value and a lower threshold value to be referenced by the second waveform transforming circuit, the width between which is increased when the repetition frequency is high, based on the repetition frequency measured by the frequency measuring circuit.

In this way, by widening the interval between the upper and lower threshold values when the repetition frequency is high, the time until accurate current measurement is possible is shortened when the current being measured is relatively large.

In the current sensor of the present invention,
a duty ratio measuring circuit for measuring a duty ratio of the first rectangular wave;
It is preferable that the reference value generating circuit generates a reference value according to the duty ratio measured by the duty ratio measuring circuit.

With this duty ratio measurement circuit, the reference value can be adjusted so that the second square wave is obtained even when the current being measured is so large that the second square wave described above is not generated.

Furthermore, in the current sensor of the present invention, it is preferable to have a first output circuit that outputs a current value of the current to be measured according to the reference value generated by the reference value generation circuit.

In the case of the current sensor of the present invention, the reference value when the magnetic equilibrium state is converged to corresponds to the current value of the current to be measured.

Furthermore, it is preferable that the current sensor of the present invention is provided with an inflection point measurement circuit that measures the voltage at the inflection point of the output waveform of the current-voltage conversion circuit.

The voltage at the inflection point of the output waveform of the current-voltage conversion circuit above represents the current value of the current being measured. Therefore, by providing this inflection point measurement circuit, the voltage at the inflection point of the output waveform above can be measured before the magnetic equilibrium state is reached, and at a point approaching that state, enabling faster measurements.

Here, it is preferable that the inflection point measurement circuit measures both the first voltage at the inflection point when the voltage rises and the second voltage at the inflection point when the voltage falls in the output waveform of the current-voltage conversion circuit, and determines the average voltage of the first voltage and the second voltage as the voltage at the inflection point of the output waveform.

The first voltage and the second voltage do not match due to the magnetization of the magnetic core. Therefore, by setting the above average voltage as the voltage at the inflection point of the output waveform, it becomes possible to measure the current with even higher accuracy.

Here, when an inflection point measurement circuit is provided, it is preferable to provide a second output circuit that outputs the current value of the current to be measured according to the inflection point voltage obtained by the inflection point measurement circuit.

When an inflection point measurement circuit is provided, the voltage of the inflection point of the output waveform obtained by the current-voltage conversion circuit corresponds to the current value of the current being measured. Therefore, when the second output circuit is provided, the current value of the current being measured can be output at an earlier timing than when the above-mentioned first output circuit is provided.

In addition, the leakage current sensor of the present invention that achieves the above object comprises:
a current transformer having a magnetic core forming a closed magnetic circuit and an excitation winding wound around the magnetic core, with a current passing through each of the wires and the total current value of the wires representing the leakage current;
a drive circuit that applies an excitation voltage to a first end of the excitation winding;
a current-voltage conversion circuit connected to a second end of the excitation winding for converting a current flowing through the excitation winding into a voltage;
a first waveform conversion circuit that converts an output of the current/voltage conversion circuit into a first rectangular wave;
a second waveform conversion circuit that converts the output of the current-to-voltage conversion circuit into a second rectangular wave with hysteresis;
a reference value generating circuit that generates a reference value to be referred to as both the threshold value of the first waveform transformation circuit and the central threshold value of the second waveform transformation circuit based on the first rectangular wave;
The drive circuit generates the excitation voltage based on a second rectangular wave, and further
a frequency measurement circuit for measuring a repetition frequency of the first rectangular wave;
and a determination circuit for determining whether or not the repetition frequency measured by the frequency measurement circuit exceeds a predetermined trip frequency.

When detection of the current to be detected begins, the higher the current value of the leakage current, the higher the repetition frequency observed. Therefore, by determining that the repetition frequency exceeds a predetermined trip frequency, it is possible to quickly determine that a leakage current has occurred.

The present invention provides a magnetic balance type current sensor that has a simplified configuration and is highly sensitive, and a leakage current sensor that uses the current sensor.

1 is a circuit block diagram of a current sensor according to a first embodiment of the present invention; 13 is a diagram showing waveforms at points A to E when the differential current ΔI is zero. 3 is a diagram showing waveforms at points A to E immediately after the differential current ΔI shown in FIG. 2 changes from a state in which it is zero to ΔI=−1 A (ampere). This is a diagram showing the waveforms at points A to E after ΔI has changed to −1 A (ampere) and sufficient time has passed. This is a diagram showing the waveforms at points A to E immediately after the differential current ΔI shown in FIG. 2 changes from zero to ΔI=−2 A (amperes), which is a larger change than in FIG. This is a diagram showing the waveforms at points A to E after ΔI has changed to −2A (amperes) and sufficient time has passed. FIG. 5 is a circuit block diagram of a current sensor according to a second embodiment of the present invention. 8 is a diagram showing waveforms at each point when the threshold width is fixed to a narrow threshold width ΔTh1 in the current sensor 1B of the second embodiment shown in FIG. 7. 8 is a diagram showing waveforms at each point when the threshold width is fixed to a wide threshold width ΔTh2 in the current sensor 1B of the second embodiment shown in FIG. 7. FIG. 8 is a flowchart showing an operation sequence of the current sensor 1B according to the second embodiment shown in FIG. 7 . FIG. 11 is a circuit block diagram of a current sensor according to a third embodiment of the present invention. FIG. 4 is a diagram showing a list of thresholds stored in a threshold generating circuit. FIG. 13 is a flowchart showing an operation sequence of the current sensor 1C according to the third embodiment shown in FIG. 12. FIG. 13 is a diagram illustrating a method for adjusting a threshold central value. FIG. 12 is a diagram showing waveforms at points A to F shown in FIG. 11 at the start of measurement. 13 is a diagram showing waveforms at points A to F after the threshold width is expanded to 100 V. FIG. 13 is a diagram showing waveforms at points A to F when the threshold central value is made to coincide with the voltage of the inflection point. The threshold width is narrowed to 10 V, and the waveform in the final state after the inflection point voltage is repeated several times is shown. FIG. 13 is a diagram showing a method for detecting an inflection point. FIG. 11 is a circuit block diagram of a current sensor according to a fourth embodiment of the present invention.

The following describes an embodiment of the present invention.

FIG. 1 is a circuit block diagram of a current sensor according to a first embodiment of the present invention.
The current sensor 1A includes a current transformer 10A. The current transformer 10A has a magnetic core M that forms a closed magnetic circuit and an excitation winding L1 wound around the magnetic core M. In this embodiment, two electric wires L2 and L3 pass through the magnetic core M. A current I1 flows through one of the two electric wires L2 and L3, the electric wire L2. A current I2 flows through the other electric wire L3. The directions of the currents I1 and I2 are opposite to each other. The current sensor 1A detects the total current value flowing through the two electric wires L2 and L3. However, the directions of the currents I1 and I2 flowing through the two electric wires L2 and L3 are opposite to each other. Therefore, the currents I1 and I2 include + and - signs according to the current direction, and here, the difference current ΔI = I1 + I2 = |I1| - |I2|
is measured.

Here, this first embodiment does not include an element for detecting electric leakage. However, as described below, the current sensor of the present invention can be used as an electric leakage sensor, or as a current sensor that also functions as an electric leakage sensor. Therefore, here, the current transformer 10A of the same configuration is configured to measure the differential current ΔI so that it can be used as is in the explanation of an electric leakage sensor.

The two electric wires L2 and L3 each have a forward current I1 and a return current I2 flowing through them. Therefore, when there is no leakage current, |I1| = |I2|, and the differential current ΔI = zero. When a leakage current occurs, |I1| ≠ |I2|, and the differential current ΔI is no longer zero. This current sensor 1A is a sensor that detects this differential current ΔI.

The current sensor 1A also includes a drive circuit 20A, an I/V conversion circuit 30A, a comparator 40A, a hysteresis comparator 50A, a low-pass filter 60A, and an AD conversion circuit 70A.

The drive circuit 20A applies an excitation voltage to the first end of the excitation winding L1. Here, this excitation voltage is a square wave, and will be referred to as an excitation square wave hereinafter. This drive circuit 20A corresponds to an example of the drive circuit of the present invention.

The I/V conversion circuit 30A is connected to the second end of the excitation winding L1 and converts the current flowing through the excitation winding L1 into a voltage. This I/V conversion circuit 30A corresponds to an example of the current-voltage conversion circuit of the present invention.

The comparator 40A converts the output of the I/V conversion circuit 30A into a square wave. This comparator 40A corresponds to an example of the first waveform conversion circuit according to the present invention. The square wave output from the comparator 40A corresponds to an example of the first square wave according to the present invention.

Furthermore, the hysteresis comparator 50A converts the output of the I/V conversion circuit 30A into a square wave with hysteresis. This hysteresis comparator 40A corresponds to an example of the second waveform conversion circuit according to the present invention. Furthermore, the square wave output from this hysteresis comparator 50A corresponds to an example of the second square wave according to the present invention.

The low-pass filter 60A smoothes the first rectangular wave output from the comparator 40A to such an extent that the repetition frequency component of the first rectangular wave is almost eliminated. However, when the duty ratio of the first rectangular wave changes, the output of the low-pass filter 60A changes over a time that is sufficiently longer than the repetition period of the rectangular wave. The output of the low-pass filter 60A is input to the comparator 40A and the hysteresis comparator 50A. In the comparator 40A, the output of the low-pass filter 60A becomes the threshold value when converting the output of the I/V conversion circuit 30A into a rectangular wave. In the hysteresis comparator 50A, the output of the low-pass filter 60A becomes the threshold center value when converting the output of the I/V conversion circuit 30A into a rectangular wave.

Here, the width of the hysteresis in the hysteresis comparator 50A, i.e., the threshold width, is ΔTh. Also, the output of the low-pass filter 60A is Th. At this time, when the output of the I/V conversion circuit 30A is rising, the hysteresis comparator 50A changes its output to H level at the timing when the output reaches the upper threshold limit (Th+ΔTh/2). Also, when the output of the I/V conversion circuit 30A is falling, the hysteresis comparator 50A changes its output to L level at the timing when the output reaches the lower threshold limit (Th-ΔTh/2). In this way, the hysteresis comparator 50A generates the second rectangular wave.

The second rectangular wave, which is the output of the hysteresis comparator 50A, is input to the drive circuit 20A. The drive circuit 20A generates an excitation rectangular wave by lowering the impedance of the input second rectangular wave. This excitation rectangular wave is applied to the first end of the excitation winding L1.

Here, the low-pass filter 60A corresponds to an example of the reference value generating circuit according to the present invention. Also, the output Th of the low-pass filter 60A corresponds to an example of the reference value according to the present invention.

The current sensor of the present invention achieves self-oscillation through the above configuration. This allows for a magnetically balanced current sensor with a simple configuration.

Before explaining the operation of the current sensor 1A in FIG. 1, the current measurement principle of the present invention will be explained below.

When the current flowing through the magnetic core M is I and the total magnetic flux flowing within the magnetic core M is Φ, the total magnetic flux Φ and the current I are proportional to the inductance L as follows:
Φ=L・I...(1)
It can be expressed as follows: Φ: total magnetic flux, I: current, and L: inductance.

In the case of the current transformer 10A shown in FIG. 1, when the current flowing through the excitation winding L is IL1 and the number of turns of the excitation winding L1 is N, the current I is expressed as follows:
I = IL1 · N
Here, it is assumed that the currents I2=I3=0.

Here, when the magnetic flux Φ changes, an electromotive force V is generated. That is,
V=-dΦ/dt...(2)
holds true.

Here, when equation (1) is differentiated with respect to time,
dΦ/dt=L・dI/dt...(3)
From equations (2) and (3),
dI/dt=(1/L)・(dΦ/dt)
=-(1/L)・V...(4)
If we integrate equation (4) with respect to time to find the current I, we get
I=-(1/L)・∫Vdt...(5)
If the voltage V is constant,
I=-(V/L)・t...(6)
It becomes.

Here, it is driven by a square wave voltage V. In that case, if we assume that the inductance L is constant, then from equation (6), the waveform of the current I will be a triangular wave.

On the other hand, when the magnetic flux density of the magnetic core M is B, the magnetic flux is H, and the magnetic permeability is μ,
B=μ・H...(7)
Therefore,
μ=B/H...(8)
From this formula (8), it can be seen that the magnetic permeability μ is the gradient of the BH curve.

Moreover, the inductance L is
L=μ・(N・N・S/d)...(9)
Here, N, S, and d are the number of turns, the cross-sectional area, and the length, respectively, of the excitation winding L1. In other words, N, S, and d are constants determined by the shape of the excitation winding L1.

Therefore, from equation (9), we can see that inductance L and magnetic permeability μ are proportional.

Here, as can be seen from equation (8), the magnetic permeability μ is the slope of the BH curve. As will be described later, this BH curve changes greatly depending on the currents I1 and I2 flowing through the electric wires L2 and L3, and the magnetic permeability μ also changes greatly accordingly. Therefore, as can be seen from equation (9), the inductance L also changes greatly. Furthermore, referring to equation (6), when the inductance L changes greatly, the current I changes greatly. This change in the current I appears as a change in the repetition frequency in the second rectangular wave that is the output of the hysteresis comparator 50A of the current sensor 1A shown in FIG. 1. This is because when the current I is large, it passes between the lower threshold value and the upper threshold value of the hysteresis comparator 50A at high speed. In the hysteresis comparator 50A, the H level and the L level are inverted each time the lower threshold value or the upper threshold value is reached. Therefore, when the current I is large, a second rectangular wave with a high repetition frequency is generated. In this way, when the current I is large, a second rectangular wave and an excitation rectangular wave with a high repetition frequency are obtained, the repetition frequency of the current IL1 flowing through the excitation winding L1 increases, and the frequency of the first rectangular wave, which is the output of the comparator 40A, also increases.

The present invention applies this principle to create a current sensor and a leakage current sensor.

The symbols A to E in Figures 2 to 6 described below represent the voltage waveforms or current waveforms at the points in Figure 1 marked with the corresponding symbols A to E.

Figure 2 shows the waveforms at points A to E when the differential current ΔI is zero.

Here, FIG. 2(A) shows the waveform of the current IL1 flowing through the excitation winding L1 wound around the magnetic core M.

Figure 2(B) also shows the excitation square wave voltage (A) output from the drive circuit 20A and applied to the first end of the excitation winding L1. The waveform of this voltage (A) coincides with the waveform of the second square wave output from the hysteresis comparator 50A. Figure 2(B) also shows the virtual ground voltage (D) of the negative input terminal of the I/V conversion circuit 30A.

Figure 2(C) also shows the output voltage (B) of the I/V conversion circuit 30A, the output voltage (E) of the comparator 40A, and the output voltage (C) of the low-pass filter 60A.

Furthermore, Figure 2 (D) shows the B-H curve of the magnetic core M.

Note that while Figure 2 is used for the explanation here, the method of assigning the symbols A to E is the same for Figures 2 to 6.

Figure 2 shows the waveforms at points A to E when the differential current ΔI is zero, which is the initial state. The B-H curve shown in Figure 2 (D) is a curve that is rotationally symmetric about the magnetic field H = 0.00 [A/m].

Figure 3 shows the waveforms at points A to E immediately after the differential current ΔI shown in Figure 2 changes from zero to ΔI = -1 A (amperes).

Looking at the B-H curve in Figure 3 (D), we can see that compared to the B-H curve in Figure 2 (D), the magnetic field H has shifted significantly to the negative side, and the slope has become much smaller. The smaller slope means that the magnetic permeability μ has changed to a smaller value, according to equation (8). This means that the inductance L has changed to a smaller value, according to equation (9), and furthermore, according to equation (6), the current I has increased. As mentioned above, this increase in current I causes the repetition frequency to become higher. Compared to Figure 2, Figure 3 has a higher frequency.

Figure 3 also shows the waveforms at points A to E immediately after the change to ΔI = -1A (amperes). As a result, the change is not yet reflected in the output of low-pass filter 60A, and voltage (C) remains at 0V, just as in Figure 2.

Meanwhile, in response to the change in the BH curve, the current flowing through the excitation winding L1 changes, and as shown in the output voltage (B) of the I/V conversion circuit 30A, the average value deviates from zero V and the waveform has a DC component. For this reason, the output rectangular wave of the comparator 40A no longer has a duty ratio of 50%, as shown by the voltage (E). If a sufficient amount of time passes in this state, the fact that the duty ratio is no longer 50% is reflected in the output of the low-pass filter 60A.

Figure 4 shows the waveforms at points A to E after ΔI has changed to -1A (amperes) and sufficient time has passed.

Here, the above-mentioned deviation of the duty ratio from 50% is reflected in the output of low-pass filter 60A, and voltage (C) changes to -2V. And, as a result of the change in voltage (C), voltage (E) returns to a square wave with a duty ratio of 50%. In addition, the repetition frequency of both voltage (A) and voltage (E) drops to the same frequency as in FIG. 2. Furthermore, the B-H curve also returns to the same shape as the B-H curve in FIG. 1, with the magnetic field H = 0.00 at its center. When this state is reached, the output voltage (C) of low-pass filter 60A passes through AD conversion circuit 70A and is output as a current measurement value of differential current ΔI = -1A (amperes).

Figure 5 shows the waveforms at points A to E immediately after the differential current ΔI shown in Figure 2 changes from zero to ΔI = -2A (amperes), which is a larger change than in Figure 3.

Comparing Figure 5 with Figure 3, the B-H curve has shifted even more significantly to the negative side, and its slope has become even smaller. Also, the repetition frequency has become even higher.

Figure 6 shows the waveforms at points A to E after ΔI has changed to -2A (amperes) and sufficient time has passed.

Here too, the repetition frequency has dropped to the same frequency as in Figure 2. Furthermore, the B-H curve has returned to the same shape as the B-H curve in Figure 1, with the magnetic field H = 0.00 at its center. However, here, the output voltage (C) of the low-pass filter 60A is -4V. This voltage (C) is passed through the AD conversion circuit 70A and output as a current measurement value of differential current ΔI = -2A (amperes).

Here, we have described the case of differential current ΔI of -1A (amperes) and -2A (amperes), which is within the range that can be measured by current sensor 1A shown in Figure 1. If the differential current ΔI becomes even larger, for example, ΔI = -4A (amperes), the slope of the B-H curve becomes even smaller, and it becomes impossible to return from the state shown in Figure 3 to the state shown in Figure 4, or from the state shown in Figure 5 to the state shown in Figure 6. In other words, in the case of current sensor 1A in Figure 1, this level of differential current ΔI is the measurement limit. Here, it is possible to adjust the measurement limit by changing the circuit constants that are not explicitly shown in Figure 1. However, in that case, it will cause a decrease in measurement accuracy.

Note that, here, we have illustrated the cases of a negative differential current ΔI, where ΔI = -1A (amperes) and ΔI = -2A (amperes), but the same is true for a positive differential current ΔI. In the case of a positive differential current ΔI, the voltage (C) after sufficient time has passed shows a positive value, indicating that the differential current ΔI flows on the positive side.

Figure 7 is a circuit block diagram of a current sensor according to a second embodiment of the present invention.

This current sensor 1B is equipped with a current transformer 10B. This current transformer 10B has the same configuration as the current transformer 10A used in the current sensor 1A of the first embodiment shown in FIG. 1.

The current sensor 1B also includes a drive circuit 20B, an I/V conversion circuit 30B, a comparator 40B, a hysteresis comparator 50B, a low-pass filter 60B, and an AD conversion circuit 70B.

Of these, the drive circuit 20B, the I/V conversion circuit 30B, and the comparator 40B correspond to the drive circuit 20A, the I/V conversion circuit 30A, and the comparator 40A, respectively, of the current sensor 1A of the first embodiment shown in FIG. 1. A duplicated explanation of these will be omitted.

The hysteresis comparator 50B corresponds to the hysteresis comparator 50A of the current sensor 1A of the first embodiment. However, the configuration is different. This difference will be described later.

Furthermore, the low-pass filter 60B and the AD conversion circuit 70B correspond to the low-pass filter 60A and the AD conversion circuit 70A, respectively, of the current sensor 1A of the first embodiment shown in FIG. 1. Duplicate explanations of these will also be omitted.

The current sensor 1B of the second embodiment shown in FIG. 7 further includes a frequency measurement circuit 80B and a threshold generation circuit 90B.

The frequency measurement circuit 80B measures the repetition frequency F of the first square wave output from the comparator 40B.

The threshold generating circuit 90B generates an upper threshold value and a lower threshold value that are referenced by the hysteresis comparator 50B. The threshold generating circuit 90B receives the threshold center value, which is the output of the low-pass filter 60B, and the repetition frequency F measured by the frequency measuring circuit 90B.

This threshold generation circuit 90B is preset with information on two threshold widths ΔTh1 and ΔTh2 (where ΔTh1<ΔTh2). In this threshold generation circuit 90B, when the repetition frequency F measured by the frequency measurement circuit 90B is lower than the predetermined frequency threshold Fth1, the upper threshold value and the lower threshold value are generated based on the narrower threshold width ΔTh1. On the other hand, when the frequency F is higher than the frequency threshold Fth1, the upper threshold value and the lower threshold value are generated based on the wider threshold width ΔTh2. Specifically, if the threshold center value, which is the output of the low-pass filter 60B, is Th, when the frequency F<Fth1, the upper threshold value (Th+ΔTh1/2) and the lower threshold value (Th-ΔTh1/2) are generated. On the other hand, when the frequency F>Fth1, the upper threshold value (Th+ΔTh2/2) and the lower threshold value (Th-ΔTh2/2) are generated. The reason for changing the upper and lower threshold limits in two stages will be explained later.

The upper and lower threshold values generated in this manner are set in the hysteresis comparator 50B.

The hysteresis comparator 50B of the current sensor 1B of this second embodiment includes a threshold determination circuit 51 and a voltage inversion circuit 52.

The threshold determination circuit 51 receives the upper and lower threshold values generated by the threshold generation circuit 90B, and also receives the output voltage (B) of the I/V conversion circuit 30B, which represents the waveform of the current flowing through the excitation winding L1. The threshold inversion circuit 51 compares the output voltage (B) with the upper threshold value when the waveform V(B) rises, and notifies the voltage inversion circuit 52 that the output voltage (B) has reached the upper threshold value at the timing when the output voltage (B) has risen and reached the upper threshold value. The threshold inversion circuit 51 also compares the output voltage (B) with the lower threshold value when the output voltage (B) falls, and notifies the voltage inversion circuit 52 that the output voltage (B) has reached the lower threshold value at the timing when the output voltage (B) has fallen and reached the lower threshold value.

The voltage inversion circuit 52 generates a second rectangular wave by inverting the voltage at the timing when it receives a notification from the threshold inversion circuit 51. This second rectangular wave is input to the drive circuit 20B and applied to the excitation winding L1 as an excitation rectangular wave represented by a voltage (A).

The current sensor 1B of the second embodiment shown in FIG. 7 also has a function of detecting leakage current. When a large leakage current occurs, it is more important to notify the occurrence of the leakage current as soon as possible than to measure the exact current value of the leakage current.

The current sensor 1B of the second embodiment is provided with a frequency determination circuit 100B as an element for quickly detecting leakage current. As described above, the larger the differential current ΔI, the higher the repetition frequency observed. The differential current ΔI is, in other words, the leakage current. Information on the repetition frequency F measured by the frequency measurement circuit 80B is also input to the frequency determination circuit 100B. This frequency determination circuit 100B determines whether the repetition frequency F measured by the frequency measurement circuit 80B exceeds the trip frequency Ftht, which is the criterion for leakage current. If the repetition frequency F exceeds the trip frequency Ftht (F>Ftht), a trip output indicating the occurrence of leakage current is immediately generated.

The exact current value of the leakage current, i.e., the differential current ΔI, is then measured once the operation of the current sensor 1B has converged.

Here, we explain why the threshold width is widened when the repetition frequency F is high.

The current sensor 1A of the first embodiment shown in FIG. 1 does not have a configuration for changing the threshold width. That is, the hysteresis comparator 50A of the current sensor 1A of the first embodiment is set with a single fixed threshold width ΔTh. The threshold center value Th is the voltage (C) of the output of the low-pass filter 60A, and as explained above, it changes to 0V in FIG. 2, -2V in FIG. 4, and -4V in FIG. 6. That is, the current sensor 1A of the first embodiment operates with a threshold upper limit value (Th+ΔTh/2) and a threshold lower limit value (Th-ΔTh/2) centered around the changing threshold center value Th. The threshold width ΔTh is always constant.

Figure 8 shows the waveforms at each point in the current sensor 1B of the second embodiment shown in Figure 7 when the threshold width is fixed to a narrow threshold width ΔTh1.

The symbols A to F in this Figure 8 and the following Figure 9 represent the voltage or current waveforms at the points in Figure 7 marked with the corresponding symbols A to F.

Figure 8 shows the subsequent changes in the waveforms at each point when the differential current ΔI is changed in steps from ΔI = 0 A to ΔI = 8 A. The threshold width ΔTh1 is the width between the upper threshold voltage (C) and the lower threshold voltage (D), and is fixed at ΔTh1 = 10 V.

Figure 8 (D) shows the B-H curve with the inside hatched. This means that the B-H curve changes over time, and the changed B-H curve is shown overlaid.

Figure 9 shows the waveforms at each point when the threshold width ΔTh2 is fixed to a wide range in the current sensor 1B of the second embodiment shown in Figure 7.

As with FIG. 8, FIG. 9 shows the subsequent changes in the waveforms at each point when the differential current ΔI is changed in steps from ΔI = 0 A to ΔI = 8 A. The threshold width ΔTh2 is the width between voltages V(C) and V(D) and is fixed at ΔTh2 = 20 V.

As can be seen by comparing Fig. 9 with Fig. 8, in relation to the rate of change of the current I in the above-mentioned equation (6), a lower repetition frequency is observed in the initial stage in Fig. 9. In addition, the time until the waveform converges to a stable state where it does not change any more is shortened.

The reason for widening the threshold width when the repetition frequency F is high will be further explained with reference to the waveform of the current sensor 1A of the first embodiment. Specifically, the B-H curves in FIG. 2, which shows the initial state of the differential current ΔI=0A, FIG. 3, which shows the state immediately after changing to ΔI=-1A, and FIG. 5, which shows the state immediately after changing to ΔI=-2A, are compared. As can be seen from comparing these B-H curves, the larger the differential current ΔI, the more the center value of the B-H curve deviates from the magnetic field H=0.00 [A/m], and the smaller its slope, i.e., the smaller the magnetic permeability μ. The larger this magnetic permeability μ, the smoother the transition from the state in FIG. 3 to the convergence state in FIG. 4, or from the state in FIG. 5 to the convergence state in FIG. 6, is. As mentioned above, only the waveforms when changing to ΔI=-1A and when changing to ΔI=-2A are shown here. As mentioned above, when ΔI is changed to -4A, the slope of the BH curve, i.e., the magnetic permeability μ, becomes too small, and it is no longer possible to transition to the convergence state shown in Figure 4 or Figure 6.

Here, widening the threshold width means using a wide range of the BH curve, that is, the range approaching the magnetic field H = 0.00 or the range including the magnetic field H = 0.00. In other words, widening the threshold width means that it is possible to use the steeper part of the BH curve, that is, the larger magnetic permeability μ, compared to when the threshold width is not widened. In other words, widening the threshold width makes it possible to transition from the state of FIG. 3 or FIG. 5 to the convergence state of FIG. 4 or FIG. 6, even with a large differential current ΔI, compared to when the threshold width is not widened. And the output voltage of the low-pass filter 60A when transitioning to the convergence state, that is, voltage (C) = -2V in FIG. 4 or voltage (C) = -4V in FIG. 6, corresponds to the differential current ΔI.

However, if the threshold width is widened, the voltage (C) will be measured from within that wide threshold width, which may result in a decrease in measurement accuracy. In other words, for highly accurate current measurement, a narrow threshold width is preferable.

Therefore, the current sensor 1B of the second embodiment shown in FIG. 7 employs a sequence in which, even if the threshold width is widened to ΔTh2, when the convergence state is approached, the threshold width is narrowed again to ΔTh1 and the current measurement is performed.

Figure 10 is a flowchart showing the operation sequence of the current sensor 1B of the second embodiment shown in Figure 7.

First, an initial value of the threshold central value is set in the hysteresis comparator 50B (step SB1). This threshold central value is the output of the low-pass filter 60B, and after operation starts, the threshold central value changes according to changes in the output of the low-pass filter 60B.

Next, an initial value of the threshold width is set in the hysteresis comparator 50B (step SB2). Here, the threshold width is configured to be changed in two stages, ΔTh1 and ΔTh2 (where ΔTh1<ΔTh2), and the narrowest threshold width ΔTh1 is used as the initial value of this threshold width. In the case of a configuration in which the threshold width is changed in three or more stages, the smallest threshold width is used as the initial value of the threshold width.

In this way, once the initial value of the threshold center value (step SB1) and the initial value of the threshold width (step SB2) are set, the repetition frequency F is measured in the frequency measurement circuit 80B (step SB3). Then, if the measured repetition frequency F is a frequency higher than the frequency threshold Fth1, the threshold width is expanded to ΔTh2 (step SB4), and the repetition frequency F is measured again.

In addition, if the measured repetition frequency F is higher than the trip frequency Ftht, which is the leakage current criterion, a trip signal is immediately output.

Here, the output voltage, i.e., the output of low-pass filter 60B, is measured (step SB5), the measured output voltage is converted to a current value (step SB6), and the current value is output.

Once the output voltage measurement is complete (step SB5), the process returns to step SB2, where the initial threshold width, i.e., the narrow threshold width ΔTh1, is set again. If the repetition frequency F is still equal to or greater than the threshold Fth1 (step SB3), the threshold width is changed again to the wide threshold width ΔTh2 (step SB4). This process is repeated until the repetition frequency F falls below the threshold Fth1 (step SB3), at which point the threshold width is no longer changed to the wide threshold width ΔTh2, and the output voltage is measured at the narrow threshold width ΔTh1 (step SB5). In this way, the current value output converges from a value measured with low accuracy to a value measured with high accuracy.

Figure 11 is a circuit block diagram of a current sensor according to a third embodiment of the present invention.

Here, the current sensor 1B of the second embodiment shown in FIG. 7 will be described as a comparison example.

This current sensor 1C is equipped with a current transformer 10C. This current transformer 10C has the same configuration as the current transformer 10B used in the current sensor 1B of the second embodiment shown in FIG. 7.

The current sensor 1C also includes a drive circuit 20C, an I/V conversion circuit 30C, a comparator 40C, and a hysteresis comparator 50C.

The drive circuit 20C, I/V conversion circuit 30C, comparator 40C, and hysteresis comparator 50C correspond to the corresponding elements of the current sensor 1B of the second embodiment shown in FIG. 7. A duplicated explanation of these elements will be omitted.

The current sensor 1C of the third embodiment shown in FIG. 11 further includes a frequency measurement circuit 80C, a duty ratio measurement circuit 110C, a threshold generation circuit 120C, a DA conversion circuit 130C, an AD conversion circuit 140C, an inflection point measurement circuit 150C, and an output interface 160C.

The frequency measurement circuit 80C, like the frequency measurement circuit 80B of the current sensor 1B of the second embodiment shown in FIG. 7, measures the repetition frequency F of the first rectangular wave output from the comparator 40C.

In addition, the duty ratio measurement circuit 110C measures the duty ratio of the first rectangular wave output from the comparator 40C.

This duty ratio will be explained with reference to the waveform diagram in the first embodiment.

Figure 3 (C) shows the voltage (E), which is the square wave output from comparator 40A. This voltage (E) is a square wave with a duty ratio exceeding 50%. Figure 3 shows the waveform when the differential current ΔI is negative (here, ΔI = -1A). In this way, when the differential current ΔI is negative, a square wave with a duty ratio exceeding 50% is observed. On the other hand, although not shown, when the differential current ΔI is positive, a square wave with a duty ratio of less than 50% is observed. In this way, by measuring this duty ratio, the direction of flow of the differential current ΔI can be determined. Here, the direction of flow of the differential current ΔI is detected using the duty ratio measurement circuit 110C.

The repetition frequency F and duty ratio measured by the frequency measurement circuit 80C and the duty ratio measurement circuit 110C are input to the threshold generation circuit 120C. The repetition frequency F measured by the frequency measurement circuit 80C is also input to the output interface 160C.

The output of the I/V conversion circuit 30C is input to the comparator 40C and also to the AD conversion circuit 140C where it is converted to a digital signal. The inflection point measurement circuit 150C detects the inflection point and calculates the voltage of the inflection point by calculation using the digital signal. This calculated inflection point voltage corresponds to the current value of the differential current ΔI. The calculation in the inflection point measurement circuit 150C will be explained later. The inflection point voltage measured by the inflection point measurement circuit 160C is input to the threshold generation circuit 120C and the output interface 160C.

Figure 12 shows a list of thresholds stored in the threshold generation circuit.

This list includes the initial threshold center value Th1, the initial threshold width ΔTh1, and the threshold widths ΔTh2, ΔTh3, ΔTh4, ... for each frequency range F1-F2, F2-F3, F3-F4, ....

The frequency ranges in this list are ranges divided by the repetition frequency F measured by the frequency measurement circuit 80C. F1, F2, F3, F4, etc. represent increasing repetition frequencies F.

The threshold widths ΔTh2, ΔTh3, ΔTh4, ... for each frequency range F1-F2, F2-F3, F3-F4, ... are all wider than the initial threshold width ΔTh1. The threshold widths ΔTh2, ΔTh3, ΔTh4, ... for each frequency range F1-F2, F2-F3, F3-F4, ... are wider the higher the repetition frequency F. Here, F1<F2<F3<F4< ..., and therefore ΔTh1<ΔTh2<ΔTh3<ΔTh4< ....

Let's return to Figure 11 to continue the explanation.

The threshold generation circuit 120C generates a central threshold value, an upper threshold value, and a lower threshold value. The algorithm for generating these central threshold value, upper threshold value, and lower threshold value will be explained later.

The central threshold value, upper threshold value, and lower threshold value generated by the threshold generation circuit 120C are input to the DA conversion circuit 130C and converted to analog signals. The upper threshold value and lower threshold value of the central threshold value, upper threshold value, and lower threshold value as analog signals are input to the hysteresis comparator 50C. These upper threshold value and lower threshold value are scaled to fit the AD conversion circuit 140C and input to the AD conversion circuit 140C as the H-side reference voltage and L-side reference voltage of the AD conversion circuit 140C. As a result, the AD conversion circuit 140C obtains a digital signal whose full scale is between the upper threshold value and the lower threshold value.

The threshold center value, which is one of the threshold center value, upper threshold value, and lower threshold value converted to analog signals by the DA conversion circuit 130C, is input to the comparator 40C. The comparator 40C binarizes the output of the I/V conversion circuit 30C using the threshold center value as a threshold value, thereby generating a first rectangular wave.

In the case of the current sensor 1C of the third embodiment shown in FIG. 11, the threshold generation circuit 120C corresponds to an example of the reference value generation circuit of the present invention.

As described above, the output interface 160C receives a signal representing the inflection point voltage obtained by the inflection point measurement circuit 150C and a signal representing the repetition frequency F obtained by the frequency measurement circuit 150C.

The voltage at the inflection point corresponds to the current value of the differential current ΔI, and in the output interface 160C, the voltage at the inflection point is converted to the current value of the differential current ΔI, resulting in a current value output.

In addition, the output interface 160C determines whether the repetition frequency F exceeds the trip frequency Ftht, which is the criterion for a ground fault. If the repetition frequency F exceeds the trip frequency Ftht (F>Ftht), a trip output is immediately generated to indicate that a ground fault has occurred.

Figure 13 is a flowchart showing the operation sequence of the current sensor 1C of the third embodiment shown in Figure 12.

First, an initial value Th1 (see FIG. 12) of the threshold center value is set (step SC1). An initial value ΔTh1 of the threshold width is set (step SC2). The initial value ΔTh1 of the threshold width is a threshold width narrower than any of the threshold widths ΔTh2, ΔTh3, ΔTh4, ... for each frequency domain. Specifically, in the threshold generation circuit 120C (FIG. 12), the upper threshold value (Th1+ΔTh1/2) and the lower threshold value (Th1-ΔTh1/2) are generated based on these initial values Th1 and ΔTh1.

These upper threshold value (Th1+ΔTh1/2) and lower threshold value (Th1-ΔTh1/2) are set in the hysteresis comparator 50C via the DA conversion circuit 130C. In addition, these upper threshold value (Th1+ΔTh1/2) and lower threshold value (Th1-ΔTh1/2) are scaled and set in the AD conversion circuit 140C.

In addition, the initial value Th1 of the threshold center value is set in the comparator 40C via the DA conversion circuit 130C.

Next, the repetition frequency F is measured in the frequency measurement circuit 80C (step SC3). If the measured repetition frequency F is higher than the trip frequency Ftht, which is the reference for an earth leakage current, a trip signal is immediately output.

When the measured repetition frequency F is higher than the predetermined frequency threshold F1, the threshold width is expanded to a threshold width corresponding to the frequency range including the measured repetition frequency F according to the list shown in FIG. 12 (step SC4). When the measured repetition frequency F is, for example, a frequency included in the frequency range of F1 to F2, the threshold width is expanded to ΔTh2. Specifically, the threshold generation circuit 120C generates a threshold upper limit value (Th1+ΔTh2/2) and a threshold lower limit value (Th1-ΔTh2/2) using the expanded threshold width ΔTh2. The generated threshold upper limit value (Th1+ΔTh2/2) and threshold lower limit value (Th1-ΔTh2/2) are then set in the hysteresis comparator 50C.

Next, the inflection points are detected (step SC5).

The inflection point measurement circuit 150C receives the output of the I/V conversion circuit 30C via the AD conversion circuit 140C, and detects an inflection point based on the input output of the I/V conversion circuit 30C, and measures the voltage of the detected inflection point. The output of the I/V conversion circuit 30C has a waveform shown in FIG. 16(C), FIG. 17(C), or FIG. 18(C) as voltage (B). The inflection points are the points of this waveform shown by circle R1 where the waveform transitions from an upward convex to a downward convex, and the points of this waveform shown by circle R2 where the waveform transitions from a downward convex to an upward convex. The voltages of these inflection points correspond to the current value of the differential current ΔI. A specific method of detecting the inflection points will be described later.

Here, we assume that the inflection point is undetectable.

The output of the I/V conversion circuit 30C is input to the AD conversion circuit 140C. As described above, the AD conversion circuit 140C performs AD conversion on signals within a voltage region between an upper threshold value and a lower threshold value. However, if the current value of the differential current ΔI is too large, the output waveform of the I/V conversion circuit 30C may deviate from between the upper threshold value and the lower threshold value even if the threshold width is widened. In this case, the output waveform of the I/V conversion circuit 30C is not reflected in the digital signal, making it impossible to detect the inflection point.

When it is not possible to detect an inflection point, the duty ratio measured by the duty ratio measurement circuit 110C is referenced (step SC6). As described above, when this duty ratio exceeds 50%, it indicates that a negative differential current ΔI is flowing. Also, when this duty ratio is less than 50%, it indicates that a positive differential current ΔI is flowing.

Then, the threshold central value is adjusted according to the duty ratio, i.e., according to the direction of the differential current ΔI (step SC7).

Figure 14 shows the method for adjusting the threshold center value.

Figure 14 (A) shows the adjustment method when the duty ratio exceeds 50%, i.e., when a negative differential current ΔI flows.

Suppose the threshold center value before adjustment was at the voltage level of Figure 14 (A) (1). The threshold center value is in the center, and the upper threshold value and lower threshold value are at equal intervals from the threshold center value. In step SC7 of Figure 13, the threshold center value at the level of (A) (1) is changed to the same voltage level as the threshold lower limit value before adjustment, as shown in (A) (2). Accordingly, the threshold upper limit value and threshold lower limit value are also adjusted by the same voltage width.

After adjusting the threshold center value in this way in step SC7 of FIG. 13, an attempt is made to measure the voltage at the inflection point again. If the adjustment is still insufficient, another similar adjustment is made, as shown in FIG. 14 (A) (2) to (A) (3). Although not shown, if the adjustment is still insufficient, the same adjustment is repeated.

On the other hand, Figure 14 (B) shows the adjustment method when the duty ratio is 50% short, i.e., when a positive differential current ΔI flows.

Let us assume that the threshold center value before adjustment was at the voltage level in Figure 14(B)(1). The threshold center value is in the center, and the upper and lower threshold values are at equal intervals from the threshold center value. This Figure 14(B)(1) is shown separately from Figure 14(A)(1), but shows the same voltage levels as Figure 14(A)(1).

In step SC7 of FIG. 13, the central threshold value at the level of (B)(1) is changed to the same voltage level as the upper threshold value before adjustment, as shown in (B)(2). Accordingly, the upper threshold value and the lower threshold value are also adjusted by the same voltage range.

After adjusting the threshold center value in this way in step SC7 of FIG. 13, an attempt is made to measure the voltage at the inflection point again. If the adjustment is still insufficient, another similar adjustment is made, as shown in FIG. 14(B)(2) to (B)(3). Although not shown in the figure, if the adjustment is still insufficient, the same adjustment is repeated.

By performing such an adjustment in step SC7 of FIG. 13, a digital signal reflecting the output of the I/V conversion circuit 30C can be obtained, making it possible to detect the inflection point.

When an inflection point can be detected (step SC5), the voltage of the inflection point is measured (step SC8), the measured voltage of the inflection point is converted to a current value in the output interface 60C of FIG. 13 (step SC9), and the current value is output.

However, the current value output at this stage is a current value measured with the threshold width greatly widened. A wide threshold width means that the AD conversion circuit 140C converts an analog signal within that wide threshold width into a digital signal with a certain number of bits, resulting in a digital signal with low resolution. As a result, the voltage value of the inflection point measured based on this low-resolution digital signal and the current value converted from this are low-precision values. Therefore, here, the following sequence is further continued to improve the measurement accuracy.

That is, in step SC10, the threshold center value is adjusted to a voltage that matches the voltage of the inflection point measured in step SC8. Then, returning to step SC5, the threshold width is changed to the initial value ΔTh1, which is the narrowest threshold width. Then, in the AD conversion circuit 140C, the analog signal within that narrow threshold width is converted into a digital signal with a predetermined number of bits. That is, a high-resolution digital signal is obtained. Then, the voltage of the inflection point is measured based on this high-resolution digital signal (step SC8), converted into a current value (step SC9), and the current value is output.

However, at this point, the threshold central value is still adjusted to match the low-precision inflection point voltage measured when the threshold width was wide, so there is a risk that it will not match the exact voltage of the inflection point. Therefore, the sequence continues, the threshold central value is further adjusted so that it matches the inflection point voltage measured by narrowing the threshold width (step SC10), and the inflection point voltage is measured. By repeating this process, the threshold central value matches the exact voltage of the inflection point, and an accurate current value is output.

Next, examples of waveforms at each point of the current sensor 1C of the third embodiment shown in Figure 1 are shown.

Figure 15 shows the waveforms at points A to F in Figure 11 at the start of measurement.

Figure 15 shows the waveform immediately after the differential current ΔI = -30 A is applied. The initial value Th1 of the threshold center value is Th1 = 0 V, and the initial value ΔTh1 of the threshold width is the initial value ΔTh1 = 10 V. Therefore, the upper threshold limit is +5 V, and the lower threshold limit is -5 V.

Here, FIG. 15(A) shows the waveform of the current IL1 flowing through the excitation winding L1 wound around the magnetic core M.

Figure 15(B) shows the voltage (A) of the excitation square wave output from the drive circuit 20C and applied to the first end of the excitation winding L1. Figure 15(B) also shows the voltage (D) which is the output square wave of the comparator 40C.

In addition, FIG. 15(C) shows the output voltage (B) of the I/V conversion circuit 30C, as well as voltages (F), (C), and (D) as the central threshold value, upper threshold value, and lower threshold value, respectively.

Furthermore, Figure 15 (D) shows the BH curve of the magnetic core M.

Note that while Figure 15 is used for the explanation here, the way in which the symbols A to F are assigned is the same for Figures 15 to 18.

The I/V conversion circuit 30C has a different current-to-voltage conversion constant from the I/V conversion circuit 30A in the first embodiment shown in FIG. 1. That is, in the first embodiment, when the differential current ΔI is ΔI=-1A, the voltage (C) representing the threshold center value is -2V. In contrast, in the third embodiment of FIG. 11, when the differential current ΔI is ΔI=-1A, the voltage (F) representing the threshold center value is -1V. FIGS. 15 to 18 illustrate a differential current ΔI=-30A. Therefore, here, as shown in FIG. 18, the voltage (F) representing the threshold center value is obtained as voltage (F)=-30V.

In the state shown in Figure 15, the differential current ΔI = 30 A cannot be handled, and it is impossible to detect the inflection point.

Figure 16 shows the waveforms at points A to F after the threshold width is expanded to 100 V.

In Figure 16, the differential current ΔI = -30 A, and the central threshold value remains at the initial value Th1 = 0 V. Therefore, the upper threshold value is +50 V, and the lower threshold value is -50 V.

When the threshold width is widened in this way, the repetition frequency decreases, and as shown in Figure 16 (C), the output waveform (voltage (B)) of the I/V conversion circuit 30C appears. Here, because the output waveform (voltage (B)) of the I/V conversion circuit 30C appears, it is possible to detect the inflection point and measure the voltage at the inflection point even at this stage. However, at this stage, the threshold width is wide at 100V, and therefore the measurement is of low accuracy. The threshold center value (voltage (F)) is outside the inflection point (the points of circles R1 and R2).

Here, the output waveform (voltage (B)) of the I/V conversion circuit 30C is captured with the threshold center value remaining at 0V, i.e., without changing the threshold center value. However, if the differential current ΔI is even larger, for example ΔI = -60A, the output waveform (voltage (B)) of the I/V conversion circuit 30C cannot be captured if the threshold center value remains at 0V even if the threshold width is widened to 100V. In that case, the threshold center value is adjusted as described with reference to step SC7 in FIG. 13 and FIG. 14.

Figure 17 shows the waveforms at points A to F when the threshold center value is made to match the voltage at the inflection point. However, in this case, the threshold width is still 100 V.

In the case of Figure 16, the threshold center value (voltage (F)) remains at 0V and is off the inflection point (points on circles R1 and R2). In contrast, in Figure 17, the threshold center value is adjusted to a voltage that matches the voltage of the inflection point.

Figure 18 shows the final waveform after narrowing the threshold width to 10 V and repeating the voltage measurement of the inflection point several times.

The center threshold value is -30V.

Figure 19 shows how to detect an inflection point.

Here, Figure 19 (A) shows the waveform of one cycle of the current IL1 flowing through the excitation winding L1. Figure 19 also shows the BH curve.

The B-H curve, as shown in Figure 19 (B), is a curve with hysteresis. Therefore, there are two points on the B-H curve where the magnetic flux density B = 0 [T]: the point indicated by circle C1 and the point indicated by circle C2. The inflection point of the current IL1, indicated by circle R1, where the current transitions from an upward convex to a downward convex, corresponds to the point indicated by circle C1 on the B-H curve. Also, the inflection point of the current IL1, indicated by circle R2, where the current transitions from a downward convex to an upward convex, corresponds to the point indicated by circle C2 on the B-H curve. The current values at these two inflection points are slightly different from each other. Therefore, here, each of these two inflection points is detected and the average value is calculated.

Figure 19(C) shows one cycle of the waveform of current IL1, divided into a monotonically decreasing part and a monotonically increasing part. Here, in the case of Figure 19(A), the vertical axis is the current value, and in the case of Figure 19(B), the vertical axis is the voltage value. This means that what is being detected is the inflection point of current IL1, and the actual calculation is based on the voltage waveform after passing through the I/V conversion circuit 30C.

To detect the inflection point, as shown in Figure 19 (C), the waveform of one cycle of current IL1 is divided into a monotonically decreasing part and a monotonically increasing part, and noise removal processing such as smoothing is performed.

Figure 19 (D) shows a waveform in which differentiation has been applied to the monotonically decreasing and monotonically increasing parts shown in Figure 19 (C).

This differentiation process results in a monotonically decreasing portion that is an upward convex curve, and a monotonically increasing portion that is a downward convex curve. For the monotonically decreasing portion, the upward apex P2 of the upward convex curve corresponds to the inflection point, and for the monotonically increasing portion, the downward apex P1 of the downward convex curve corresponds to the inflection point. The voltages at these inflection points are then read. The average of the voltage values at both of the read inflection points is then calculated, and this calculated average is used as the voltage measurement value at the inflection point. This voltage measurement value is then converted to a current value to obtain the current value of the differential current ΔI.

This concludes the explanation of the third embodiment shown in Figure 11.

Figure 20 is a circuit block diagram of a current sensor according to a fourth embodiment of the present invention.

The current sensor 1B of the second embodiment shown in FIG. 7 and the current sensor 1C of the third embodiment shown in FIG. 11 are configured to also function as a leakage current sensor. This is because the current sensor 1B of the second embodiment and the current sensor 1C of the third embodiment are provided with frequency measurement circuits 80B, 80C as elements of the current sensor. In contrast, the current sensor 1A of the first embodiment shown in FIG. 1 is a current sensor 1A that does not require a frequency measurement circuit. For this reason, the current sensor 1A of the first embodiment does not function as a leakage current sensor. The current sensor 1D of the fourth embodiment shown in FIG. 20 is configured to also function as a leakage current sensor by adding a frequency measurement circuit and the like to the current sensor 1A of the first embodiment.

The current sensor 1D of the fourth embodiment includes a current transformer 10D, a drive circuit 20D, an I/V conversion circuit 30D, a comparator 40D, a hysteresis comparator 50D, a low-pass filter 60D, and an AD conversion circuit 70D. These elements correspond to the current transformer 10A, the drive circuit 20A, the I/V conversion circuit 30A, the comparator 40A, the hysteresis comparator 50A, the low-pass filter 60A, and the AD conversion circuit 70A in the first embodiment shown in FIG. 1, respectively. A duplicated description of these elements will be omitted.

The current sensor 1D of this fourth embodiment further includes a frequency measurement circuit 80D and a frequency determination circuit 100D.

The frequency measurement circuit 80D measures the repetition frequency F of the first rectangular wave output from the comparator 40D.

Furthermore, the frequency determination circuit 100D determines whether the repetition frequency F measured by the frequency measurement circuit 80D exceeds the trip frequency Ftht, which is the criterion for a ground fault. If the repetition frequency F exceeds the trip frequency Ftht (F>Ftht), a trip output is immediately generated to indicate that a ground fault has occurred.

The exact current value of the leakage current, i.e., the differential current ΔI, is then measured once the operation of the current sensor 1D has converged.

In this way, by adding an element for detecting electrical leakage to the first embodiment shown in FIG. 1, it is possible to make the current sensor function as an electrical leakage sensor as well.

In each embodiment, a current transformer with two electric wires L2, L3 is used, with the current sensor also being used as a leakage current sensor. However, when used as a current sensor that only detects a current value, only one electric wire may be arranged, and the current flowing through that one electric wire may be detected. Also, when using this current sensor 1 as a leakage current sensor, for example, three electric wires through which three-phase AC flows may be arranged. In this way, there is no limit to the number of electric wires that pass through the magnetic core M.

The waveform diagrams in each of the above-mentioned embodiments are waveform diagrams obtained as a result of simulation. For this reason, voltage values that are not used in normal signal processing circuits, such as a threshold width of 100V, are shown. However, when actually configuring a signal processing circuit, the voltage levels are scaled to be suitable for signal processing. Then, the drive circuits 20A to 20D adjust the voltage level to be applied to the excitation winding L1.

In addition, in each of the above embodiments, each drive circuit 20A to 20D applies the rectangular wave output from each hysteresis comparator 50A to 50D to the excitation winding as a drive rectangular wave without changing the rectangular wave. However, the drive voltage output from each drive circuit 20A to 20D and applied to the excitation winding does not necessarily have to be a rectangular wave, and it may be a drive voltage derived from the rectangular wave output from each hysteresis comparator 50A to 50D. For example, the drive voltage may be a triangular wave or sine wave, etc., obtained by converting the rectangular wave output from each hysteresis comparator 50A to 50D.

1A, 1B, 1C, 1D Current sensor 10A, 10B, 10C, 10D Current transformer 20A, 20B, 20C, 20D Drive circuit 30A, 30B, 30C, 30D I/V conversion circuit 40A, 40B, 40C, 40D Comparator 50A, 50B, 50C, 50D Hysteresis comparator 51 Threshold determination circuit 52 Voltage inversion circuit 60A, 60B, 60D Low pass filter 70A, 70B, 70D AD conversion circuit 80B, 80C, 80D Frequency measurement circuit 90B Threshold generation circuit 100B, 100D Frequency determination circuit 110C Duty ratio measurement circuit 120C Threshold generation circuit 130C DA conversion circuit 140C AD conversion circuit 150C Inflection point measurement circuit 160C Output interface

Claims (6)

a current transformer having a magnetic core forming a closed magnetic circuit and an excitation winding wound around the magnetic core, the closed magnetic circuit being passed through by an electric wire through which a current to be measured flows;
a drive circuit that applies an excitation voltage to a first end of the excitation winding;
a current-voltage conversion circuit connected to a second end of the excitation winding for converting a current flowing through the excitation winding into a voltage;
a first waveform conversion circuit that converts an output of the current/voltage conversion circuit into a first rectangular wave;
a second waveform conversion circuit that converts the output of the current-to-voltage conversion circuit into a second rectangular wave with hysteresis;
a duty ratio measurement circuit that measures a duty ratio of the first rectangular wave based on the first rectangular wave;
a threshold generating circuit that generates a central threshold value, an upper threshold value, and a lower threshold value based on a duty ratio of the first rectangular wave;
a DA conversion circuit that converts the central threshold value, the upper threshold value, and the lower threshold value into analog signals;
an AD conversion circuit that converts an output of the current/voltage conversion circuit into a digital signal;
an inflection point measurement circuit for measuring a voltage at an inflection point of the output waveform of the current-to-voltage conversion circuit converted into the digital signal;
The first waveform transformation circuit includes:
The threshold center value converted into the analog signal is input,
The output of the current/voltage conversion circuit is binarized using the threshold center value as a threshold value, thereby converting it into the first rectangular wave;
The second waveform transformation circuit includes:
The upper threshold value and the lower threshold value converted into the analog signals are input,
converting an output of the current-voltage conversion circuit into the second rectangular wave based on the input upper threshold value and lower threshold value;
The AD conversion circuit includes:
the upper threshold value and the lower threshold value converted into the analog signals scaled so as to be compatible with the AD conversion circuit are input as an H-side reference voltage and an L-side reference voltage of the AD conversion circuit,
converting an output of the current/voltage conversion circuit into the digital signal having a full scale between the upper threshold value and the lower threshold value;
the voltage of the inflection point measured by the inflection point measurement circuit corresponds to the current value of the current to be measured,
The threshold generating circuit includes:
when the inflection point voltage cannot be detected by the inflection point measurement circuit, adjusting the central threshold value, the upper threshold value, and the lower threshold value according to the duty ratio of the first rectangular wave measured by the duty ratio measurement circuit;
the adjustment is performed by changing the threshold center value before the adjustment to a voltage level equal to the threshold upper limit value or the threshold lower limit value before the adjustment that exist at equal intervals with the threshold center value as a center in accordance with a duty ratio of the first rectangular wave, and also adjusting the threshold upper limit value and the threshold lower limit value before the adjustment by the same voltage width;
The inflection point measurement circuit includes:
After the adjustment, the voltage at the inflection point is measured again;
the adjustment by the threshold generation circuit and the measurement of the voltage of the inflection point by the inflection point measurement circuit after the adjustment are repeated until the voltage of the inflection point is detected;
The current sensor, wherein the drive circuit generates the excitation voltage derived from the second rectangular wave.
2. The current sensor according to claim 1, wherein the inflection point measurement circuit measures both a first voltage at an inflection point when the voltage rises and a second voltage at an inflection point when the voltage falls of the output waveform of the current -to-voltage conversion circuit converted into the digital signal, and determines an average voltage of the first voltage and the second voltage as the voltage of the inflection point of the output waveform.
3. The current sensor according to claim 2 , further comprising an output circuit that outputs a current value of the current to be measured according to the voltage of the inflection point obtained by the inflection point measurement circuit.
a current transformer having a magnetic core forming a closed magnetic circuit and an excitation winding wound around the magnetic core, with a current passing through each of the wires and the total current value of the wires representing the leakage current;
a drive circuit that applies an excitation voltage to a first end of the excitation winding;
a current-voltage conversion circuit connected to a second end of the excitation winding for converting a current flowing through the excitation winding into a voltage;
a first waveform conversion circuit that converts an output of the current/voltage conversion circuit into a first rectangular wave;
a second waveform conversion circuit that converts the output of the current-to-voltage conversion circuit into a second rectangular wave with hysteresis;
a duty ratio measurement circuit that measures a duty ratio of the first rectangular wave based on the first rectangular wave;
a threshold generating circuit that generates a central threshold value, an upper threshold value, and a lower threshold value based on a duty ratio of the first rectangular wave;
a DA conversion circuit that converts the central threshold value, the upper threshold value, and the lower threshold value into analog signals;
an AD conversion circuit that converts an output of the current/voltage conversion circuit into a digital signal;
an inflection point measurement circuit for measuring a voltage at an inflection point of the output waveform of the current-to-voltage conversion circuit converted into the digital signal;
The first waveform transformation circuit includes:
The threshold center value converted into the analog signal is input,
The output of the current/voltage conversion circuit is binarized using the threshold center value as a threshold value, thereby converting it into the first rectangular wave;
The second waveform transformation circuit includes:
The upper threshold value and the lower threshold value converted into the analog signals are input,
converting an output of the current-voltage conversion circuit into the second rectangular wave based on the input upper threshold value and lower threshold value;
The AD conversion circuit includes:
the upper threshold value and the lower threshold value converted into the analog signals scaled so as to be compatible with the AD conversion circuit are input as an H-side reference voltage and an L-side reference voltage of the AD conversion circuit,
converting an output of the current/voltage conversion circuit into the digital signal having a full scale between the upper threshold value and the lower threshold value;
the voltage of the inflection point measured by the inflection point measurement circuit corresponds to a current value of the leakage current,
The threshold generating circuit includes:
when the inflection point voltage cannot be detected by the inflection point measurement circuit, adjusting the central threshold value, the upper threshold value, and the lower threshold value according to the duty ratio of the first rectangular wave measured by the duty ratio measurement circuit;
the adjustment is performed by changing the threshold center value before the adjustment to a voltage level equal to the threshold upper limit value or the threshold lower limit value before the adjustment that exist at equal intervals with the threshold center value as a center in accordance with a duty ratio of the first rectangular wave, and also adjusting the threshold upper limit value and the threshold lower limit value before the adjustment by the same voltage width;
The inflection point measurement circuit includes:
After the adjustment, the voltage at the inflection point is measured again;
the adjustment by the threshold generation circuit and the measurement of the voltage of the inflection point by the inflection point measurement circuit after the adjustment are repeated until the voltage of the inflection point is detected;
The drive circuit generates the excitation voltage derived from the second rectangular wave, and further
a frequency measurement circuit for measuring a repetition frequency of the first rectangular wave;
and a determination circuit for determining whether or not the repetition frequency measured by the frequency measurement circuit exceeds a predetermined trip frequency.
The leakage sensor according to claim 4, characterized in that the inflection point measurement circuit measures both a first voltage at an inflection point when the voltage rises and a second voltage at an inflection point when the voltage drops in the output waveform of the current -to-voltage conversion circuit converted into the digital signal, and determines the average voltage of the first voltage and the second voltage as the voltage at the inflection point of the output waveform.
6. The leakage current sensor according to claim 5 , further comprising an output circuit that outputs a current value of the leakage current corresponding to the voltage of the inflection point obtained by the inflection point measurement circuit.

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