JP7011806B2 - Dielectric material evaluation device - Google Patents

Description

The present invention relates to a dielectric material evaluation device for measuring the dielectric properties of a dielectric material such as semiconductor ceramics represented by barium titanate.

With the miniaturization and higher performance of mobile communication devices such as smartphones, the need for highly accurate measurement of the physical characteristics of dielectric substrates for microwave circuit configurations, especially the dielectric constants such as the relative permittivity, has rapidly increased. ing.
In Patent Document 1, a ring-shaped resonator is integrally formed on the first and second dielectric substrates, and the resonance frequency and no-load Q value of both are measured to obtain the electromagnetic property values of the derivative substrate. It is stated that the request is made.
In Patent Document 2, a conductor film divided into an inner conductor and an outer conductor by a ring-shaped gap and a derivative thin film laminated on the upper surface and the lower surface of the conductor film are provided on a derivative support substrate, and the inner conductor and the outer conductor are provided. It is described that the measurement of the relative permittivity and the induced tangent is made more accurate by obtaining the resonance mode in which an electric field is generated from one to the other.

Japanese Patent No. 4373902 Japanese Patent No. 4540596

However, in the measuring devices of Patent Documents 1 and 2, it is necessary to prepare a measurement target equipped with a dedicated ring-shaped resonator in advance and search for experimental conditions for generating the ring resonance, which requires time and cost. Not only that, there was a problem that it was not possible to obtain highly accurate measurement results.

なお、Γは反射係数、Ｚ₀は線路の特性インピーダンス（通常５０Ω）、Ｚは誘電体設置部の特性インピーダンスである。 As a solution to this, as shown in FIG. 1, by installing a dielectric material (DUT) as a measurement target on the ring resonator a, the effective relative permittivity is changed and the resonance frequency is changed. A method of seeing is conceivable.
However, when the permittivity of the object to be measured is high, especially when the relative permittivity is 20 or more, the effective relative permittivity in the ring circuit of the ring resonator a changes significantly. According to 1), the influence of reflection in the circuit becomes large, high-precision measurement cannot be performed, and an error of about 50% may occur in the evaluation value of the relative permittivity.

Γ is the reflection coefficient, Z ₀ is the characteristic impedance of the line (usually 50 Ω), and Z is the characteristic impedance of the dielectric installation portion.

Therefore, an object of the present invention is to make it possible to accurately and easily measure the effective relative permittivity and the relative permittivity even for a measurement object having a small size and a high relative permittivity.

In order to solve the above problems, the relative permittivity measuring device of the present invention is a support substrate and at least two conductors formed on the support substrate, which intersect with each other and one of which is a substrate to be measured. At least two conductors, the other of which is isolated from the board to be measured, and the input and output ports provided at both ends of the other conductor to which the probe is connected. It is composed of and.

According to the present invention, even if the object is small and has a high dielectric constant, the relative permittivity is simply arranged on one of the conductors formed on the support substrate so as to be separated from the other conductor. Can be measured and evaluated accurately and easily.

FIG. 1 is a diagram showing a problem in measuring a change in resonance frequency by changing an effective relative permittivity by installing a DUT on a ring resonator a. FIG. 2 is a diagram showing a basic configuration of the first embodiment. FIG. 3 is a diagram showing a route of a high frequency signal from the input side port 3a to the output side port 3b. FIG. 4 is a diagram showing a support structure of the DUT 4 according to the second embodiment. FIG. 5 is a diagram showing advantages when a coplanar line is adopted based on the third embodiment. FIG. 6 is a diagram showing a circuit structure based on the fourth embodiment. FIG. 7 is a diagram showing an outline of an experimental device based on the present invention. FIG. 8 is a diagram showing the experimental results.

Hereinafter, embodiments of the present invention will be described with reference to the drawings.

[Example 1]
The basic configuration of the embodiment will be described with reference to FIG.
On the surface of the support substrate 1, a T-shaped conductor line 2 including a horizontal line portion 2a and a vertical line portion 2b extending in the vertical direction from the central portion thereof is formed. The support substrate 1 is made of, for example, alumina, and the T-shaped line 2 is formed by gold plating or the like.
An input side port 3a and an output side port 3b to which a probe is electrically connected are provided at both ends of the horizontal line portion 2a, and the vertical line portion 2b is provided with a measurement object DUT4 so as not to come into contact with the horizontal line portion 2a. Is placed.
Here, as compared with the state without DUT4 (the atmosphere is a dielectric), the DUT4, which is a dielectric material, is placed on the horizontal line portion 2a of the T-shaped conductor line 2, so that the effective relative permittivity of the circuit is increased. Changes. As a result, the wavelength of the high-frequency signal changes due to the wavelength shortening effect.
That is, as shown in FIG. 3, the high-frequency signal input from the input-side port 3a directly reaches the output-side port 3b through the horizontal line portion 2a, and a part of the DUT4 is mounted, which has a wavelength shortening effect. It is divided into those reaching the output side port 3b via the vertical line portion 2b including the receiving region.
Therefore, by measuring the high frequency output from the output side port 3b and evaluating the amount of change in the resonance frequency depending on the presence or absence of the DUT 4, the effective relative permittivity of the DUT is used using the following equations (2) and (3). Can be calculated.
L = (2n-1) λ / 4 ... (2)
λ = λ ₀ / (ε _eff) ^1/2・ ・ ・ ・ ・ ・ (3)
However, L is the vertical line length, λ is the wavelength after the wavelength is shortened by the dielectric, λ ₀ is the wavelength of the incident signal, and ε _eff is the effective relative permittivity.

Generally, the resonance of the T-type filter circuit composed of the T-shaped conductor 2 occurs at a wavelength (frequency) satisfying the equation (2), but the gap between the T-shaped conductor 2 and the DUT 4 Since there is a disturbance such as a deviation in the arrangement of the DUT 4 in the T-shaped conductor line 2, it may not always be possible to obtain an accurate effective relative permittivity ε _eff by the equation (2).
Therefore, in this embodiment, the evaluation value of the relative permittivity is calculated based on the measurement result of the calibration line and the resonance frequency obtained by using the standard sample having the known relative permittivity, and the result is the known relative permittivity. The evaluation is based on how much the deviation is.

The gap d between the horizontal line portion 2a and the DUT 4 placed on the vertical line portion 2b needs to have a gap of at least about 1 mm in order to prevent the influence of the parasitic impedance due to the interaction.
Further, in this embodiment, the center line of the DUT 4 is aligned with the vertical line portion 2b and placed so that the upper side is parallel to the horizontal line portion 2a, but the DUT 4 covers the vertical line portion 2b. As described above, as long as the gap with the horizontal line portion 2a is equal to or greater than a predetermined value, even if the center line of the DUT 4 is slightly inclined with respect to the vertical line portion 2b, the measurement accuracy is not significantly affected. However, when the correction using the calibration curve is performed as described above, the arrangement of the standard sample and the DUT4 to be measured is always the same for the horizontal line portion 2a and the vertical line portion 2b in order to maintain the measurement accuracy constant. , Need to be kept constant.

[Example 2]
There is a gap (air gap) between the DUT 4 and the T-shaped conductor 2 on the support substrate 1 due to the thickness of the T-shaped conductor 2, the pressing pressure at the time of installation, the surface roughness of both, and the like. It causes measurement error.
Therefore, in this embodiment, as shown in FIG. 4, a jig 6 for supporting the support substrate 1 is provided on the pedestal 5, and through holes 7a and 7b are formed in the support substrate 1. The DUT 4 is placed so as to cover the through holes 7a and 7b, the space formed by the jig 6 is evacuated, and the suction pressure thereof is kept constant so that the DUT 4 is in close contact with the support substrate 1 and air is applied. Measurement error can be minimized by maintaining the gap at a constant value as close to zero as possible.

これに対し、コプレーナ線路を採用すると、図５に示したとおり電界は上部誘電体と基板に均一に分布する。そのため、下記の式（５）に基づき、誤差の原因となるパラメータが少ない、シンプルな演算式により実効比誘電率を高精度に解析することが可能となる。

また、マイクロストリップ線路では線路の厚み分だけ支持基板１とＤＵＴ４の間に隙間ができるため、実施例２において貫通孔を設けて真空引きした場合もよりＤＵＴ４とＴ字型導線路２との間に空隙ができやすくなる。コプレーナ線路ならばより安定的に吸着をさせることができるため、マイクロストリップ線路よりも安定的な空隙を得ることが可能となる。 [Example 3]
In this embodiment, in addition to the second embodiment, by adopting a coplanar line having two slots in the T-shaped conductor line 2, it is possible to analyze the effective relative permittivity with higher accuracy.
That is, as shown in FIG. 5, the effective relative permittivity ε _eff when the microstrip line is adopted for the T-shaped conductor line 2 as in the first and second embodiments is calculated by the following equation (4). Need to calculate. In the equation, ε _r is the relative permittivity of the substrate which is the dielectric material to be measured, h is the thickness of the substrate, and W is the line width of the T-shaped conductor line 2.

On the other hand, when the coplanar line is adopted, the electric field is uniformly distributed on the upper dielectric and the substrate as shown in FIG. Therefore, based on the following equation (5), it is possible to analyze the effective relative permittivity with high accuracy by a simple arithmetic expression with few parameters that cause an error.

Further, in the microstrip line, a gap is created between the support substrate 1 and the DUT 4 by the thickness of the line. Therefore, even when a through hole is provided and a vacuum is drawn in the second embodiment, the space between the DUT 4 and the T-shaped conductor line 2 It becomes easy to create a gap in the railroad track. Since the coplanar line can be adsorbed more stably, it is possible to obtain a more stable void than the microstrip line.

[Example 4]
In this embodiment, as shown in FIG. 6, the coplanar line structure shown in the third embodiment has a circuit structure in which an inductance is connected to the tip of the line 2b. Assume an equivalent circuit in which the capacitance C ₁ of the line and the capacitance C ₂ connected in series to the inductance L and the inductance L are connected in parallel. In this equivalent circuit, two resonances are obtained, and the resonance frequencies are ω _r1 and ω _r2 , respectively. At this time, ω _r 1 does not include C ₁ . The capacitance C ₁ of the line changes depending on the installation position of the DUT 4 installed on the line capacitance C 1. That is, the value of C ₁ changes depending on the line length covered by the DUT 4 in the line 2b. Therefore, by using ω _r1 that does not contain C ₁ , the position dependence of DUT 4 on the support substrate 1 can be reduced.

[Example 5]
In this embodiment, a time domain (TD) analysis is adopted in which the magnitude of the signal (reflection coefficient) that is incident from the input side port 3a and then returns to the same port is analyzed in the time domain. By analyzing the signal amount at each time, it is possible to analyze at which position on the circuit the reflection point is located, and if the signal passing through the circuit directly under DUT4 is small, the shift of the resonance frequency becomes small. In the embodiment, the resonance frequency is corrected by using the magnitude of the signal that has passed through the depth of the T-shaped vertical portion (the signal that has actually passed through the circuit directly under the DUT).

Here, an experiment on a relative permittivity using the present invention and an evaluation result thereof will be described with reference to FIGS. 7 and 8. In this experiment, as shown in FIG. 7, a vector network analyzer E8631A and a 1 mm frequency expansion unit were used. The measurement frequency was 10 MHz to 110 GHz, the IF bandwidth was 100 Hz, and the source power was -17 dBm.
A material having a nominal relative permittivity ε _{r, norm} of 9.8, 38, 110 is installed as DUT4, and among a plurality of resonances obtained by measuring the transmission coefficient, the resonance frequency in the lowest frequency band is the first resonance frequency f. It was set to _{r, 1st, and εr} .
Next, the TD analysis of the reflectance coefficient is performed to obtain the maximum value A _εr obtained in the range of 0.1-0.3 nsec. In addition, fr _{, 1st, 1} and A ₁ when DUT 4 is not installed are also obtained. Since the relative permittivity of air is 1, it corresponds to the evaluation result of the relative permittivity 1.
Then, the resonance frequencies _{fr, 1st, εr, and mod} modified based on the equation (6) are obtained.
_{fr, 1st, εr, mod} = fr _{, 1st, εr} × A _εr / A ₁ (6)
The obtained fr _{, 1st, εr, mod} are plotted against the square root √ε _{r, norm} of the nominal relative permittivity, and a calibration curve (y = Ax + B) is obtained by linear approximation.
Then, the evaluation values ε _{r and eval} of the relative permittivity are obtained by the following equation (7).
ε _{r, eval} = ((f _{r, 1st, εr, mod} −b) / a) ² (7)
When the difference between ε _{r, eval} and ε _{r, norm} was treated as an evaluation error, evaluation was possible within a range of ± 1 even with a DUT having a relative permittivity of 38 as shown in FIG.

As described above, according to the present invention, even a small object to be measured having a high dielectric constant is arranged on one of the conductors formed on the support substrate so as to be isolated from the other conductor. Since the relative permittivity of the substrate can be accurately measured by such a simple operation, it can be expected to be widely adopted in the semiconductor manufacturing process and the like.

1 ... Support board 2 ... T-shaped lead line 2a ... Horizontal line part 2b ... Vertical line part 3a ... Input side port 3b ... Output side port 4 ... DUT
5 ... Pedestal 6 ... Jigs 7a, 7b ... Through holes

Claims (6)

Support board and
Two conductors formed on the support substrate, which intersect with each other, on which a dielectric material to be measured is placed, and on the other, which is isolated from the dielectric material. ,
A dielectric material evaluation device provided at both ends of the other conductor, comprising an input side port and an output side port to which a probe is connected. A calculation means for calculating the evaluation value of the measurement result for the dielectric material to be measured is provided based on the calibration curve obtained by placing a dielectric material having a known relative permittivity on the conductor. The dielectric material evaluation device according to claim 1, wherein the dielectric material is evaluated. The support substrate is supported by a jig arranged on the pedestal, and the dielectric material to be measured is adsorbed through the through hole formed in the support substrate to make the gap with the support substrate uniform. The dielectric material evaluation apparatus according to claim 1 or 2. The dielectric material evaluation device according to any one of claims 1 to 3, wherein the conductor is a coplanar line having two slots. The dielectric material evaluation device according to claim 4, wherein the capacitance C1 of the coplanar line is an equivalent circuit connected in parallel to the inductance L and the capacitance C2 connected in series to the inductance L. It is provided with a correction means for correcting the calculation of calculating the evaluation value of the measurement result for the dielectric material to be measured by analyzing the magnitude of the signal incident from the input side port and returning to the input side port in the time domain. The dielectric material evaluation device according to any one of claims 1 to 5, wherein the dielectric material evaluation device is characterized by the above.

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