JP6676745B2 - Wireless communication system and wireless communication method - Google Patents

Description

  The present invention relates to a wireless communication system and a wireless communication method.

[Preface]
At present, with the explosive spread of smartphones, highly convenient microwave band frequency resources are depleted. As countermeasures, a shift from a third-generation mobile phone to a fourth-generation mobile phone and allocation of a new frequency band are being performed. However, since there are many operators who want to provide services, frequency resources allocated to each operator are limited.

  In the mobile phone service, studies are underway to improve the frequency use efficiency of a multi-antenna system using a plurality of antenna elements. The already widely used wireless standard IEEE (The Institute of Electrical and Electronics Engineers, Inc.) 802.11n employs a multiple input multiple output (MIMO) transmission technique using a plurality of antenna elements for both transmission and reception. To perform spatial multiplexing transmission. Thus, in IEEE 802.11n, the transmission capacity is increased to improve the frequency use efficiency. Note that the term MIMO is generally used on the assumption that both a transmitting station and a receiving station include a plurality of antenna elements. If the receiving side is a single antenna element, the term MISO (Multiple Input Single Output) is used instead of MIMO. However, hereinafter, the term MIMO is used to mean all of them.

  Further, as a recent communication technology, as in OFDM (Orthogonal Frequency Division Multiplexing) modulation method and SC-FDE (Single Carrier Frequency Domain Equalization) method, the signal is divided into a plurality of frequency components (subcarriers) on a frequency axis. A method of performing signal processing is general. In the following, description will be made using the term “subcarrier” on the premise of a general method common to both OFDM and SC-FDE without distinction between OFDM and SC-FDE.

In the MIMO transmission technology, more efficient transmission can be performed by knowing the transmission path information between the transmitting station and the receiving station. As the simplest example, the case of a plurality of antennas on the receiving side has been described. However, when the transmitting side is provided with N antenna elements and the receiving side is provided with only one antenna element, the N antenna elements are used. Directivity control is performed on the transmitting side so that the transmitted signal is in-phase synthesized by the receiving-side antenna element. As a result, the line gain can be increased. Specifically, when the channel information between the j-th antenna element of the transmitting station and the antenna element of the receiving station in the k-th subcarrier is defined as h _j ^(k) , the following equation ( The transmission weight w _j ^(k) of ¹⁾ is calculated, and a product obtained by multiplying the transmission weight w _j ^(k) is transmitted from each antenna element (equal gain combining). Note that the channel information in this specification is strictly speaking of an amplifier, a filter, etc. in an RF (Radio Frequency) circuit of a transmission system and a reception system in consideration of a circuit configuration in actual operation and a procedure of signal processing and control. The information includes rotation of the complex phase and fluctuation information of the amplitude.

A vector (h ₁ ^(k) ,..., H _j ^(k) ,..., H _N ^(k) ) whose components are channel information corresponding to the first antenna element to the N-th antenna element on the transmission side is a channel vector h. ^(K) . Also, vectors (w ₁ ^(k) ,..., W _j ^(k) ,..., W _N ^(k) ) ^T (T) having transmission weights corresponding to the N-th antenna element from the first antenna element on the transmission side as components. Represents transposition.) Is referred to as a transmission weight vector w ^(k) . Strictly speaking, the channel vector in the downlink → h ^(k) (the symbol “→” before “h ^(k) ” is a symbol given above h to represent the vector) is a line Vector, transmit weight vector → w ^(k) should be expressed as a column vector. However, in the following, for simplicity, the symbol “→” is omitted and the row vector and the column vector are described without distinction. In the following description, the notation relating to the received signal Rx, the transmitted signal Tx, and the noise n should also be clearly indicated as a vector by adding “→”. However, since there is no other confusing notation, “→” is used. The description is omitted. The reception signal Rx is given by the following equation (2) with respect to the transmission signal Tx and the noise n.

By substituting equation (1) into equation (2), a value obtained by adding the absolute value of each component h _j ^(k) of the channel vector h ^(k) over all antenna components is obtained as the channel gain. If the transmission power from each antenna element remains the same as when transmitting with one antenna element, if the number of antenna elements is N, the amplitude of the received signal will be N times that when transmitting with one antenna element. Expected. Received signal power is improved proportional to the square of the amplitude to the N ² times. This value is the gain when a plurality of antenna elements are used as an array antenna. Strictly speaking, the gain of the array antenna itself is generally interpreted as N times the received power (which is the result of evaluation under the condition that the total transmitted power is constant). The description will be made on the assumption that the transmission power per element is constant, assuming an environment.

  In general, according to Shannon's theorem, it is known that an increase in transmission capacity with respect to an improvement in SNR (Signal-Noise Ratio) is larger in a lower SNR region and smaller in a higher SNR region. Therefore, rather than aiming to improve the transmission capacity by improving the line gain, the receiving side is often provided with a plurality of antenna elements and spatial multiplexing to improve the transmission capacity. MIMO transmission technology aims to increase the transmission capacity by spatial multiplexing.

FIG. 1 shows an outline of MIMO transmission. Here, as a description focusing on a certain frequency, a suffix “k” representing a subcarrier or a frequency is omitted. In FIG. 1, reference numeral 11 denotes a transmitting station, and reference numeral 12 denotes a receiving station. In this example, both the transmitting station 11 and the receiving station 12 have two antenna elements, and channel information (amplitude and complex phase) between the transmitting antenna # 1 of the transmitting station 11 and the receiving antenna # 1 of the receiving station 12 is provided. The information indicating the amount of rotation) is h ₁₁ , the channel information between the transmitting antenna # 1 of the transmitting station 11 and the receiving antenna # 2 of the receiving station 12 (information indicating the amplitude and the amount of rotation of the complex phase) is h ₂₁ , and transmitted. The channel information (information representing the amplitude and the amount of rotation of the complex phase) between the transmitting antenna # 2 of the station 11 and the receiving antenna # 1 of the receiving station 12 is represented by h ₁₂ , the transmitting antenna # 2 of the transmitting station 11 and the receiving station 12 channel information between the receiving antenna # 2 of expressed (amplitude, information indicating the amount of rotation of the complex phase) as h _22, the signal t _1, t ₂ transmitted from the two transmission antennas of the transmitter station 11 , Two receptions of the receiving station 12 The following equation (3) is used between the signals r ₁ and r ₂ received by the antenna, using the noise signals n ₁ and n ₂ .

  Basically, in MIMO transmission, a signal on the transmission side is estimated based on a reception signal on the reception side and a channel matrix. If the noise term in Expression (3) is sufficiently small, the transmission signal can be estimated from the reception signal by multiplying both sides by the inverse matrix of the channel matrix. The transmission characteristic can be improved by multiplying the transmission side by a predetermined transmission weight matrix, and further multiplying the reception side by a predetermined reception weight, thereby enabling more efficient transmission. For example, when channel information between a plurality of transmitting-side antenna elements and a receiving-side antenna element is known, the channel matrix is subjected to singular value decomposition (SVD), and transmission is performed in an eigenmode. This maximizes transmission capacity.

  More specifically, the channel matrix H is decomposed into a diagonal matrix D having unitary matrices U and V and a singular value λ as a diagonal component, as in the following equation (4).

  At this time, if the unitary matrix V is used as the transmission weight matrix, the reception signal vector Rx is given by the following equation (5) with respect to the transmission signal vector Tx and the noise vector n.

On the receiving side, the following equation (6) is obtained by multiplying the Hermitian conjugate matrix U ^H of the unitary matrix U.

In equation (6), since the off-diagonal components of the diagonal matrix D are zero, the cross term of the transmission signal has already been canceled and the signal is separated. Also, the noise vector is multiplied by the reception weight matrix U ^H and converted to a noise vector n ′ represented by rotating the coordinate axes, but the statistical characteristics of the vector remain equivalent to the original noise vector. FIG. 2 shows a conceptual diagram of eigenmode transmission. 2 (a) shows a basic MIMO channel, FIG. 2 (b) shows a situation where a transmission / reception weight matrix has been multiplied, and FIG. 2 (c) shows a virtual transmission path formed by eigenmode transmission. I have. As shown in equation (4), a channel matrix H between the transmitting and receiving antennas of the transmitting station 11 and the receiving station 12 includes a matrix D having a singular value λ in each diagonal component by singular value decomposition, and two unitary units. matrix U, represented by the product of ^{V H.} Here, as shown in FIG. 2B and Expression (5), when the transmission weight matrix V is used on the transmission side and the reception weight matrix ^UH is used on the reception side, the unit of the unitary matrix of Expression (4) is multiplied to obtain a unit. As a result, as shown in Expression (6), it is possible to represent a matrix D in which the non-diagonal components are zero and only the diagonal components are non-zero. This corresponds to a situation where transmission paths of virtual channels represented by singular values λ ₁ , λ _2, ... Are set in parallel as shown in FIG. At this time, the square value of the absolute value of each singular value λ corresponds to the line gain of an individual signal sequence. Each singular value λ is a different value for each signal system. By optimizing the transmission mode according to the singular value of the eigenmode, the transmission capacity can be maximized. The transmission mode is a specific mode of signal transmission determined by a combination of the M-ary modulation value and the coding rate of error correction.

  Here, if each component of the channel matrix of the MIMO channel is independent and uncorrelated, the absolute value of each singular value is a relatively large value. For example, when there are many reflected waves and the received power of the line-of-sight wave is relatively low, each singular value has a relatively large value as described above. On the other hand, when the transmitting / receiving antenna is in the line-of-sight environment and the reflected wave does not exist much, only the absolute value of the first singular value is extremely large, and the absolute value of the singular value after the second singular value is extremely large. Tends to be smaller. For this reason, it is generally said that MIMO transmission is suitable for a multipath environment, and is not very suitable when the line-of-sight wave is dominant.

  The above is a description of the single-user MIMO transmission technology assuming one base station device and one terminal station device. The same description can be extended to a multi-user MIMO in which one base station apparatus and a plurality of terminal station apparatuses simultaneously communicate on the same frequency axis. In multi-user MIMO, generally, each terminal station device communicates with a smaller number of antenna elements than the total number of signal systems to be spatially multiplexed. Therefore, on the downlink, directivity control for suppressing interference between users is performed in advance on the transmission side. Although the specific formula is slightly different, basically, similarly to the above-described eigenmode transmission, a channel matrix is grasped, and a transmission weight adapted to the channel matrix is used.

  Further, in the above description, the downlink has been mainly described. However, in the uplink, communication using the channel information can be performed after grasping the channel information in advance. For example, in the processing as an array antenna described first, the weight of the in-phase combination given by the equation (1) is used as the reception weight, and the weight of the maximum ratio combination is given by the following equation (7). It is also possible to use one.

The constant C in the equation (7) is a coefficient determined as appropriate. Among the components of the vector, C having a large absolute value of h _j ^(k) is added with a large weight, and C is determined so that a small signal is added with a small weight. As a result, a signal having a large SNR is emphasized, and adjustment is performed so that noise of a signal having a small SNR does not excessively affect the signal.

  It should be noted that downlink channel information is required to calculate the transmission weight, but this can be realized by various forms of channel feedback. In the simplest example, it is possible for the terminal station device to receive the training signal transmitted by the base station device on the downlink, and to report the reception result in the uplink control information. Generally, a channel estimation method for performing such a loopback is called explicit feedback. In addition, for example, a calibration coefficient necessary for conversion between uplink channel information and downlink channel information inside the base station apparatus is acquired in advance, and channel estimation is performed on the uplink. After that, it is also possible to estimate downlink channel information by multiplying by this calibration coefficient. This method is called implicit feedback. In the case of large-scale antennas with a large number of antennas, implicit feedback is generally considered to be advantageous. However, in the present invention, the method of channel feedback is not particularly limited, and a general channel estimation method can be used.

[Future mobile network direction]
As described above, with the explosive spread of smartphones, a further increase in transmission capacity is required. In the current research on wireless communication, technical studies for a fifth-generation mobile phone following a fourth-generation mobile phone are underway. Here, a transmission capacity of 10 times or more that of the fourth generation is realized. Is required. Here, the increase in transmission capacity per unit area as well as the increase in transmission capacity of one base station apparatus and the wireless system under the base station apparatus is also required. Specifically, in places where many people gather, such as Shinjuku, Shibuya, Ginza, and Otemachi, simply increasing the transmission capacity of the wireless system cannot cope with it. The same transmission capacity is realized in a smaller area, and by setting a large number of the small cells, a transmission capacity proportional to the number of the small cells is realized. However, since it is required that this small cell be installed in a place where people gather and further transmission capacity is required, station design cannot be sufficiently performed like a macro cell having a large area area. Originally, if multiple frequency channels were available to introduce a small cell while the frequency resources were depleted, it would not be possible to utilize the resources as frequency repetition (frequency reuse), It is preferable to increase the total transmission capacity by using the frequency channels described above. Therefore, there is a need for a technology that can repeatedly install small cells at a relatively short distance without designing a station even for the same frequency channel.

  Further, as frequency resources in the microwave band are depleted, it is necessary to secure a certain frequency bandwidth in order to realize a transmission capacity of 10 Gbit / s or more, and for that purpose, utilization of a higher frequency band is expected. You. However, when the frequency is increased by a factor of 10, the free space propagation loss increases by 20 dB, so that the propagation distance is reduced to 1/10 in a line-of-sight environment with the same transmission power. Further, it is difficult to increase the output of the high-power amplifier on the transmission side as the frequency increases, and the transmission power per antenna is limited, but the transmission capacity of 10 Gbit / s or more is sufficiently designed in terms of line design. Technologies that can be realized are required.

  From such a viewpoint, a large-scale MIMO (Massive MIMO) transmission technique is currently receiving attention. In the Massive MIMO transmission technology, the number of antennas on the base station device side is increased by at least one digit or more compared to the conventional MIMO transmission which is at most several, and a large number of tens to hundreds of antenna elements are used. Thus, it is possible to improve the line gain to the terminal station apparatus as the destination and reduce the interference given to the terminal station apparatuses other than the destination. There are various variations in the method of realizing and applying Massive MIMO, and for so-called small cells, the reduction of interference given to terminal station devices other than the destination is used for “suppression of inter-cell interference”. In addition, as a Massive MIMO transmission technology, a large-scale MIMO matrix is used as a single-user MIMO in which a single terminal station device is used simply as in the conventional technology, and simultaneous communication is performed with a plurality of terminal station devices. As a multi-user MIMO. Here, in the use as multi-user MIMO, unlike the case of a small cell, it is also possible to utilize a very redundant number of antenna elements for "suppression of interference between users" between terminal station devices in the same area. It is. Hereinafter, a large-scale antenna system (for example, refer to Non-Patent Documents 1 to 4) will be briefly described as an example of a technique relating to these large-scale antennas.

[Overview of large-scale antenna system]
FIG. 3 is a diagram showing an outline of a large-scale antenna system. FIG. 3 shows a base station device 1, a radio station device 2, a line of sight wave 3, a stable reflected wave 4 from a structure, multiple reflected waves 5 to 6 near the ground, and a structure 7. In the large-scale antenna system of FIG. 3, the base station apparatus 1 includes a large number (for example, 100 or more) of antenna elements and is installed at a high place such as a building rooftop or a high steel tower. Similarly, the wireless station device 2 is installed at a high place such as the roof of a building, the roof of a house, a telephone pole or a steel tower. Therefore, the base station apparatus 1 and the wireless station apparatus 2 are generally in a line-of-sight environment. In the meantime, in addition to the path of the line-of-sight wave 3 and the path of the stable reflected wave 4 from the large stable structure 7, Paths of multiple reflected waves 5 and 6 due to a moving body such as a car or a person in the vicinity are mixed. In particular, when a directional antenna is used, the reception levels of the multiple reflection waves 5 and 6 near the ground are lower than those of the line-of-sight wave 3 and the stable reflection wave 4.

  FIG. 4 is a diagram illustrating an impulse response in a line-of-sight environment and a non-line-of-sight environment. FIG. 4A shows an impulse response in a non-line-of-sight environment, and FIG. 4B shows an impulse response in a line-of-sight environment. 4A and 4B, the horizontal axis represents the delay time, and the vertical axis represents the reception level of each delayed wave. In the case of the non-line-of-sight environment shown in FIG. 4A, there is no direct wave component in the line-of-sight section, there are many multiple reflected waves of various paths as components, and each amplitude and complex phase are randomly and strongly increased with time. fluctuate.

  On the other hand, when a line-of-sight environment like the large-scale antenna system shown in FIG. 3 is assumed, the level of the stable path of the line-of-sight wave 3 and the stable reflected wave 4 from the structure 7 is high. The multiple reflection wave of the time-varying path, which generally has a larger delay amount than the line of sight wave 3 and the stable reflection wave 4 from the structure 7, has a relative reflection as shown in FIG. Level becomes smaller. When such channel information is acquired a plurality of times and averaged, the components of the stable path are stable in amplitude and complex phase each time and the same value is obtained. However, the components of the time-varying path are randomly combined in the complex space, averaged, and approach an average value of zero. Therefore, only the stable component can be effectively extracted by the averaging. Note that the absolute channel information depends on the symbol timing, and if the symbol timing is different, the channel information cannot be averaged properly. In order to solve such a problem, in Non-Patent Document 2, it may be considered that relative channel information (or each channel information is divided by channel information of a reference antenna) based on a complex phase of a reference antenna element. Good) technology is introduced. If such averaging is not involved, it is possible to discuss using absolute channel information instead of using relative channel information, but even in such a case, the relative channel information is used in calculating the transmission / reception weight. There is no problem with using it. In the following description, the channel information is basically handled as relative channel information based on the complex phase of the reference antenna, in order to comprehensively handle including the averaging process.

  The base station apparatus 1 (see FIG. 3) calculates the transmission / reception weight based on the channel information of the stable path without time variation obtained in this manner. The base station apparatus 1 performs directivity control for performing in-phase synthesis with a large number of antenna elements using the calculated transmission / reception weights. By using the above transmission / reception weights, the base station apparatus 1 can increase the directivity gain to the wireless station apparatus of the communication partner as the target of the directivity control in proportion to the square of the number N of antennas.

In addition, since the directivity gain of the interference given to the wireless station device other than the destination remains N times, a simple calculation causes an N-times gap between the desired signal and the interference signal. As a result, the expected value of SIR (Signal to Interference Ratio) is ₁₀ Log ₁₀ (N) [dB]. This expected value is 20 dB when N is 100. In the case of selectively spatially multiplexing a wireless station device having a smaller correlation, further improvement in SIR characteristics is expected, and higher spatial multiplexing can be realized.

Non-Patent Literature 3 and Non-Patent Literature 4 select a technique for further suppressing interference that cannot be completely suppressed by the above transmission / reception weights, and a combination of radio station apparatuses having lower correlation of channel information (channel correlation). Technology is introduced. In order to realize ultra-high-order spatial multiplexing, it is important to combine wireless station devices with small correlation of channel information. A channel vector h _j ^(k) (“h _j ^(k) ” is a vector, which has channel information on a k-th subcarrier between a number of antenna elements of the base station apparatus and antenna elements of the j-th wireless station apparatus as components. Yes, the symbol “→” should be given above h to indicate that it is a vector, but this is omitted. Hereinafter, similarly, the description is omitted.) And another i-th wireless station. channel correlation between the channel vector h _{i ^(k)} in the device is given by the following equation (8).

The system assumed a sight environment, assuming a virtual channel model consisting only of sight waves, when the correlation of the channel vector h _i between the antenna and the base station apparatus of the radio station apparatus ^_(k) is small Is suitable for spatial multiplexing, and conversely if the correlation is large, it is not suitable for spatial multiplexing.

[About large-scale MIMO in small cells]
In the description of the above equation (8), it is assumed that the radio station apparatus side is a single antenna, and if the radio station apparatus is different, the correlation of the channel vector may generally be low due to spatial spread. It was assumed. On the other hand, in the fifth-generation mobile phone, a large number of antennas are mounted on the wireless station device side and a large number of antennas are similarly mounted on the base station device side for the purpose of improving the throughput per user. In recent research reports, a study has been made on increasing the capacity of a 256 × 16 large-scale MIMO with 256 antenna elements on the base station apparatus side and 16 antenna elements on the wireless station apparatus side. I have. Here, it is assumed that the wireless station device carried by the user is small in size and portable. Further, for example, in consideration of reducing the mutual interference between adjacent small cells and installing the base station apparatus in a place where people are concentrated, the base station apparatus is installed on the wall surface of an existing building (for example, about 20 m above the ground). It is anticipated that a directivity is given to each antenna element to irradiate a limited area in a manner of looking down from above. In this case, it is not preferable to install a large antenna on a wall surface of a building or the like from the viewpoint of safety and ease of installation. In the case of using a high frequency band such as a millimeter wave or a quasi-millimeter wave, as the wavelength becomes shorter, the miniaturization of the antenna element and the formation of the directivity of the antenna become easier, so that even a base station apparatus has a very narrow area. It is expected that a small antenna set packed with a large number of antennas will be used.

  In this case, for example, if the channel correlation between the j-th antenna and the i-th antenna among the plurality of antennas in the single wireless station device is obtained by the above equation (8), the antenna interval of the user-side wireless station device becomes Under conditions that are very short and that the line of sight between the base station device and the wireless station device can be ensured, a MIMO channel having a very large channel correlation between antenna elements can be seen. As described above, the base station device is installed at a high place such as a wall surface of a building so as to look down, and when a user uses a smartphone or the like in his / her hand, the base station device and the antenna device between the base station device and the radio station device are not used. Is generally expected to be in a line-of-sight environment, and such a situation is expected to be a general use environment.

  In this case, when the singular value decomposition of the MIMO matrix is performed, the absolute value of the first singular value becomes a very high value using the line-of-sight component, but the higher singular value after the second singular value is the first singular value. It tends to be much smaller than the singular value. That is, as spatial multiplexing transmission utilizing the MIMO channel, there is a very large line gain margin for the first path corresponding to the first singular value, but for a higher-order path equal to or higher than the second singular value. Can be said to be relatively inefficient.

  In order to avoid this problem, for example, it is ideal to secure the spatial spread without consolidating the antennas of the base station device into a small-sized housing. For example, if the base station apparatus antennas are arranged in a linear array in which the base station apparatus antennas are evenly arranged in a straight line of about 10 m, even if the line of sight wave is dominant, the antennas of the base station apparatus are spread as a MIMO channel due to the spatial spread. Can be expected to increase the absolute value of a higher-order singular value equal to or larger than the second singular value and contribute to an increase in capacity. However, such a large-scale structure is not preferable in terms of the structure of the antenna. For example, when the number of antenna elements on the base station apparatus side is 256, installing 256 antennas individually on the wall of a building at intervals of about 4 cm increases the burden of installation work. On the other hand, when a structure already assembled in a linear array is installed on the wall surface of a building, it is strict in terms of safety measures to install the structure on the wall surface of the building because the structure becomes large. Further, in order to cooperatively transmit all the antenna elements, it is necessary to distribute a radio frequency signal from a casing of one base station device to an antenna element having a spatial spread of about 10 m using a cable, and in particular, The cable loss of signal transmission in the high frequency band cannot be ignored. For example, assuming a radio frequency of 20 GHz, a cable loss of 10 dB or more at 10 m may cause a loss in the line gain obtained due to the angle and the increase in the number of antenna elements.

  For these reasons, in the operation of large-scale MIMO in which line-of-sight waves are dominant, when it is required to at least reduce the size of the antenna on the radio station side, the capacity is increased by spatial multiplexing transmission using single-user MIMO. It is difficult to plan.

[In case of wireless entrance]
The above situation has been described mainly in a small cell environment. However, under the same condition that the line of sight is dominant, the same problem remains in the wireless entrance line. For example, when a small cell base station for a fifth generation mobile phone is installed on the wall of a building, a large-capacity entrance line is required. Generally, an entrance line to a base station device is generally provided using an optical fiber, but it is not easy to lay the optical fiber on a wall surface of an existing building. For example, drawing optical fiber to the rooftop or somewhere on the ground, and installing cables along the wall of a building may damage the appearance of a building that has already been built, and obtain the consent of the building owner. Hateful. Regarding the work of penetrating a hole in the building wall to provide an optical fiber from the building interior to the building wall, it may be more difficult to obtain the consent of the building owner. In this case, for example, an optical fiber may be drawn to the roof of a building facing the building in which the small cell base station device is installed, and an entrance line may be provided from the roof of the building to the small cell base station device on the building wall surface via a wireless line. Conceivable.

  The problem in this case is that the base station device on the wall surface of the building is assumed to be sufficiently miniaturized in the same manner as described above. Therefore, in a line-of-sight environment, higher-order paths of the second singular value or higher of the MIMO channel are spatially multiplexed. It is considered that the contribution to the transmission is limited and it is effective to utilize the first singular value positively for stable transmission.

[Wireless entrance to moving objects such as trains]
Next, another major problem will be described in consideration of the usage form of the fifth generation mobile phone. As described above, when a large number of users exist in a very small area, it is important to be able to provide a large-capacity wireless line to the large number of users. As the name implies, a small cell constitutes a relatively small service area, and here, a target to provide a large-capacity wireless line is a user who moves at a speed as low as a walking speed at most. Therefore, it is not assumed that the majority of users will switch between small cells (or between small cells and macro cells), and the load of exchanging control information for switching will not be a problem. However, for example, assuming users who move by train or the like, the timings at which each user performs handover are almost the same, and instantaneous mass processing is required for the network side. This signal processing load and transmission capacity compression due to the generation of a large amount of control signals are inefficient.Usually, each user in the train once aggregates traffic in the train, and the aggregated traffic is It is preferable to bundle and secure an entrance line to an external network by a wireless line as viewed from the train. As a result, it is possible to avoid a load caused by a useless enormous amount of handover and a pressure on the transmission capacity due to a tremendous amount of control information. This concept is considered as a moving cell in the sense that a cell, which is a service area of a fifth generation mobile phone, moves with a train.

  The entrance line to the small cell base station device of the 5th generation mobile phone on the wall surface of the building was not impossible to provide by optical fiber if it was impossible, but the moving cell by this train (hereafter, Assuming a “train moving cell”, it is impossible to use an optical fiber for the entrance line, and establishing an efficient construction method of a wireless entrance to this train moving cell is extremely difficult for the fifth generation. It is important in mobile phones.

  Here, for example, a train in an urban area is taken as an example, and the required line capacity is estimated. For example, in a Yamanote line or the like, a situation is assumed in which about 100 passengers are riding per vehicle in an 11-car train. In the case of commuting time or the like, it is assumed that most of the business people and college students carrying a smartphone or the like, and about 550 users in all vehicles, about 550, are trying to access the Internet by wireless communication. In the age of the fifth generation, if one user communicates on average at 10 Mbit / s due to the increase in the capacity of content, a capacity of about 5.5 Gbit / s is required. Considering time fluctuations, if a line capacity about twice as large is secured, a capacity of about 10 Gbit / s is required as a wireless entrance of the train moving cell.

  In order to provide such a large capacity line, it is naturally necessary to provide a line in a high frequency band such as a millimeter wave or a quasi-millimeter wave, and unlike the above-described small cell, a high speed of 100 km / h or more is required. Broadband and large-capacity transmission to a mobile object assuming movement is inevitable. In order to avoid frequent handovers as described above, in the wireless entrance of the train moving cell, it is preferable that the same base station device can provide service for a section of about several hundred meters, and free space propagation loss in a high frequency band. Is large, it is necessary to secure the line gain by the large-scale MIMO.

  Here, when a transmission capacity of about 10 Gbit / s is realized as described above, for example, assuming an efficiency of about 3 bit / Hz · s as a frequency use efficiency, a bandwidth of about 3 to 4 GHz is used without spatial multiplexing. I need. For example, when the e-band of 70 to 80 GHz is used, when one channel occupies a bandwidth of 3 to 4 GHz, only one channel can be secured. However, on the other hand, considering the segregation by the frequency channel for each of a plurality of routes running in parallel, it is necessary to control the bandwidth of one channel to about 1 GHz and operate with a plurality of channels. In this case, it is necessary to secure the insufficient capacity by spatial multiplexing, and in the above example, about four multiplexes are required.

  As explained above, the application of large-scale MIMO is also necessary for a train moving cell. In this case, channel information necessary for directivity formation for gain expansion by large-scale MIMO is accurately obtained. You need to know. However, the wavelength is short in the high frequency band, and the time variation of the channel appears relatively large with respect to the movement of the wireless station device. Further, when the trains passing each other pass each other, the external environment fluctuates more vigorously than the fluctuation of the channel accompanying the normal linear movement, and the reflected wave component is violently disturbed. When transmitting from the base station apparatus to the train, channel information is indispensable for forming directivity, but if a certain degree of estimation accuracy is required for the channel information, the channel information is very frequently used. Need to be exchanged for each other. However, despite the aim of increasing the transmission capacity, the increase in the overhead for control information called channel information feedback is a fall in the end. It is required that a large-capacity spatial multiplexing can be stably realized even in an environment where the time variation is large.

[Example of MIMO transmission device configuration]
(Overall circuit configuration)
FIG. 5 is a schematic block diagram illustrating an example of a configuration of the base station device 80 in the multi-user MIMO system. Here, the description will be made as a multi-user MIMO system, but if it is understood that a plurality of signal sequences are performed by a plurality of antenna elements in the same radio station instead of a terminal station apparatus to be spatially multiplexed, a basic device configuration Is identical to a single-user MIMO system.

  As shown in FIG. 5, the base station device 80 includes a transmitting unit 81, a receiving unit 85, an interface circuit 87, a MAC (Medium Access Control) layer processing circuit 88, and a communication control circuit 820. The MAC layer processing circuit 88 has a scheduling processing circuit 881.

  The base station device 80 inputs and outputs data to and from an external device or a network via the interface circuit 87. The interface circuit 87 detects data to be transferred on the wireless line among the input data, and outputs the detected data to the MAC layer processing circuit 88. The MAC layer processing circuit 88 performs processing related to the MAC layer in accordance with an instruction from the communication control circuit 820 that performs management control of the operation of the entire base station device 80. Here, the processing related to the MAC layer includes conversion of data input / output by the interface circuit 87 and data transmitted / received on the wireless line, addition of MAC layer header information, and the like. In this process, the scheduling processing circuit 881 performs various scheduling processes including a combination of terminal stations that perform spatial multiplexing simultaneously in multi-user MIMO transmission. The scheduling processing circuit 881 outputs a scheduling result to the communication control circuit 820. In the multi-user MIMO, a signal sequence of a plurality of systems is output from the MAC layer processing circuit 88 to the transmission unit 81 in order to transmit a signal to a plurality of terminal station devices at once.

(Circuit Configuration of Transmission Unit 81)
FIG. 6 is a schematic block diagram illustrating an example of the configuration of the transmission unit 81 in the base station device 80 in the multi-user MIMO system. As illustrated in FIG. 6, the transmission unit 81 includes transmission signal processing circuits 811-1 to 811-N _SDM (N _SDM is an integer of 2 or more) and addition and synthesis circuits 812-1 to 812-N _BS-Ant (N _BS-Ant is an integer of 2 or more); IFFT (Inverse Fast Fourier Transform) & GI (Guard Interval: guard interval) providing circuits 813-1 to 813-N _BS-Ant ; and D / A (digital) / Analog) converters 814-1 to 814-NBS- _Ant , local oscillator 815, mixers 816-1 to 816-NBS _-Ant , filters 817-1 to 817-NBS _-Ant , and high power It includes amplifier (HPA) and _{818-1~818-N BS-Ant,} an antenna element _{819-1~819-N BS-Ant,} and a transmission weight processing unit 830 That. A transmission signal processing circuit _{811-1~811-N SDM,} the transmission weight processing unit 830, and a communication control circuit 820 shown in FIG.

The transmission weight processing section 830 includes a channel information acquisition circuit 831, a channel information storage circuit 832, and a multi-user MIMO (MU-MIMO) transmission weight calculation circuit 833. Here, _{N SDM} subscript of the transmission signal processing circuit _{811-1~811-N SDM} in FIG. 6 represents a multiplex number for performing spatial multiplexing simultaneously. The subscript N _BS-Ant of the circuits from the addition / combination circuits 812-1 to 812-N _BS-Ant to the antenna elements 819-1 to 819-N _BS-Ant is the number of antenna elements included in the base station apparatus 80. Represents N _BS-Ant is, for example, 100.

In the multi-user MIMO, a signal is transmitted to a plurality of terminal station devices at one time. Therefore, a plurality of signal sequences are input from the MAC layer processing circuit 88 to the transmission unit 81, and the input plurality of signal sequences are transmitted signals. It is inputted to the processing circuit _{811-1~811-N SDM.} When the transmission signal processing circuits 811-1 to 811-N _SDM receive data (data inputs # 1 to #N _SDM ) to be transmitted to each of the destination terminal station devices from the MAC layer processing circuit 88, the transmission signal processing circuits 811-1 to 811-N _SDM A modulation process is performed by generating a wireless packet to be transmitted. Here, for example, if the OFDM modulation method is used, the signal of each signal sequence is subjected to modulation processing for each subcarrier. Further, the base station multiplies the modulated baseband signal by a transmission weight for each subcarrier. The signal multiplied by the transmission weight corresponding to each of the antenna elements 819-1 to 819 -N _BS-Ant is subjected to the remaining signal processing as necessary, and is added and synthesized as sampling data of the transmission signal in the baseband by the addition and synthesis circuit 812. -1 to 812-N are input to _BS-Ant .

The signals input to the addition / combination circuits 812-1 to 812-NBS- _Ant are combined for each subcarrier. The combined signal is converted from a signal on the frequency axis to a signal on the time axis by IFFT & GI adding circuits 813-1 to 813-NBS- _Ant , and is further inserted into a guard interval or between OFDM symbols (SC-FDE ( In the case of Single-Carrier Frequency Domain Equalization, processing such as waveform shaping (between blocks of block transmission) is performed, and D / A converters 814-1 to 814-1 are provided for each of the antenna elements 819-1 to 819 -NBS _-Ant. The digital sampling data is converted into a baseband analog signal by the 814-N _BS-Ant . Further, each analog signal is multiplied by a local oscillation signal input from the local oscillator 815 by the mixers 816-1 to 816-NBS _-Ant , and is up-converted into a radio frequency signal. Here, since the up-converted signal includes a signal in a region outside the band of the channel to be transmitted, the out _- of-band component is removed by the filters 817-1 to 817-NBS _-Ant , and the signal to be transmitted is removed. Generating a typical signal. The generated signal is amplified by the high-power amplifier _{818-1~818-N BS-Ant,} and transmitted from the antenna elements _{819-1~819-N BS-Ant.}

In FIG. 6, after performing addition and combining of the signals of the respective subcarriers in the addition and combining circuits 812-1 to 812-NBS _-Ant , processes such as IFFT processing, insertion of a guard interval, and waveform shaping are performed. However, the transmission signal processing circuits 811-1 to 811-N _SDM perform these processes, and the IFFT & GI provision circuits 813-1 to 813-N _BS-Ant may be omitted at this position in terms of function distribution. Good. In this case, the rest of the signal processing necessary after transmission weight multiplying in the transmission signal processing circuit _{811-1~811-N SDM} refers IFFT processing, insertion of the guard interval, the process of waveform shaping or the like.

The transmission weight multiplied by the transmission signal processing circuits 811-1 to 811-N _SDM is obtained from the multi-user MIMO transmission weight calculation circuit 833 provided in the transmission weight processing unit 830 at the time of signal transmission processing. In the transmission weight processing section 830, the channel information acquisition circuit 831 separately acquires the channel information acquired by the reception section 85 via the communication control circuit 820, and updates the channel information sequentially to the channel information storage circuit 832. Remember. At the time of signal transmission, the multi-user MIMO transmission weight calculation circuit 833 reads channel information corresponding to the destination terminal station device from the channel information storage circuit 832 in accordance with an instruction from the communication control circuit 820, and based on the read channel information. The transmission weight corresponding to the combination of the terminal station devices as the destination is calculated. Multiuser MIMO transmission weight calculating circuit 833 outputs the calculated transmission weight to the transmission signal processing circuit _{811-1~811-N SDM.}

  Further, the communication control circuit 820 manages control related to the entire communication, such as management of the terminal station apparatus as a destination and overall timing control. The communication control circuit 820 outputs information indicating the destination terminal station device and the like to the transmission weight processing unit 830 that performs the signal processing related to the above-described transmission weight calculation.

(Circuit Configuration of Receiver 85)
FIG. 7 is a schematic block diagram illustrating an example of a configuration of the receiving unit 85 in the base station device 80 in the multi-user MIMO system. As illustrated in FIG. 7, the receiving unit 85 includes antenna elements 851-1 to 851-N _BS-Ant , low noise amplifiers (LNA) 852-1 to 852-N _BS-Ant , a local oscillator 853, and a mixer 854. -1 to 854-N _BS-Ant , filters 855-1 to 855-N _BS-Ant , A / D (analog / digital) converters 856-1 to 856-N _BS-Ant , and FFT (Fast Fourier) transform: it includes fast Fourier transform) circuit _{857-1~857-N BS-Ant,} a reception signal processing circuit _{858-1~858-N SDM,} a reception weight processing unit 860. A reception signal processing circuit _{858-1~858-N SDM,} a reception weight processing unit 860 is connected to the communication control circuit 820 shown in FIG. The reception weight processing unit 860 includes a channel information estimation circuit 861 and a multi-user MIMO (MU-MIMO) reception weight calculation circuit 862.

The signals received by the antenna elements 851-1 to 851-N _BS-Ant are amplified by the low noise amplifiers 852-1 to 852-N _BS-Ant . The amplified signal and the local oscillation signal output from the local oscillator 853 are multiplied by mixers 854-1 to 854-NBS _-Ant , and the amplified signal is down-converted from a radio frequency signal to a baseband signal. You. Since the down-converted signal includes a signal in a region outside the frequency band to be received, the out _- of-band component is removed by the filters 855-1 to 855-NBS _-Ant . The signal from which the out-of-band component has been removed is converted to a digital baseband signal by A / D converters 856-1 to 856-NBS _-Ant . All the digital baseband signals are input to the FFT circuits 857-1 to 857 -NBS- _Ant, and the signal on the time axis is converted into a signal on the frequency axis at a predetermined symbol timing determined by a timing detection circuit which is not described here. It is converted into the above signal (separated into signals of each subcarrier). Each sub-carrier to the separated signals is inputted to the reception signal processing circuit _{858-1~858-N SDM,} is also input to the channel information estimation circuit 861.

In the channel information estimating circuit 861, the antenna element of each terminal station apparatus and the base station apparatus 80 are determined based on a known signal for channel estimation (a preamble signal or the like added to the head of a radio packet) separated into subcarriers. Of each antenna element 851-1 to 851-N _BS-Ant is estimated for each subcarrier, and the estimation result is output to the multi-user MIMO reception weight calculation circuit 862. The multi-user MIMO reception weight calculation circuit 862 calculates a reception weight to be multiplied for each subcarrier based on the input channel information. At this time, the reception weight for combining the signals received by the antenna elements 851-1 to 851-NBS- _Ant differs for each signal sequence, and the reception signal processing circuits 858-1 to 858-1 corresponding to the signal sequence to be extracted. 858- _NSDM .

In the reception signal processing circuits 858-1 to 858 -N _SDM , the signals for each subcarrier input from the FFT circuits 857-1 to 857 -N _{BS-Ant are} input from the multi-user MIMO reception weight calculation circuit 862. The reception weights are multiplied, and the signals received by the antenna elements 851-1 to 851-NBS _-Ant are added and synthesized for each subcarrier. The reception signal processing circuits 858-1 to 858 -N _SDM perform demodulation processing on the signal obtained by the addition and synthesis, and output the reproduced data to the MAC layer processing circuit 88. In the demodulation processing here, for example, the final signal detection may be performed by, for example, performing a soft decision on the received signal once, performing a deinterleaving processing as necessary, and then performing an error correction processing.

Here, the different received signal processing circuit _{858-1~858-N SDM,} the signal processing of the different signal sequences is performed. Further, the MAC layer processing circuit 88 performs processing related to the MAC layer (for example, conversion between data input / output to / from the interface circuit 87 and data transmitted / received on a wireless line, termination of MAC layer header information, and the like). Do. In this process, the scheduling processing circuit 881 performs various scheduling processes including a combination of terminal stations that perform spatial multiplexing simultaneously in multi-user MIMO transmission, and outputs a scheduling result to the communication control circuit 820. The received data processed by the MAC layer processing circuit 88 is output to an external device or a network via the interface circuit 87.

  In addition, the communication control circuit 820 manages control related to the entire communication, such as management of the terminal station device of the transmission source and overall timing control. In addition, information indicating the terminal station device or the like of the transmission source is input from the communication control circuit 820 to the reception weight processing unit 860 that performs signal processing related to the above-described calculation of the reception weight.

As for signal reception, as in the case of transmission, in a wideband system using the OFDM modulation method or the SC-FDE method, the above-described multiplication of the reception weight is performed for each subcarrier. In other words, the FFT circuits 857-1 to 857 -N _BS-Ant perform FFT on the signals output from the A / D converters 856-1 to 856-N _BS-Ant and separate the signals into subcarriers. for each carrier, the signal processing in the channel information estimation circuit 861, and, so that the reception signal processing in the reception signal processing circuit _{858-1~858-N SDM} is performed.

The above is the description of the configurations of the base station device 80, the transmission unit 81, and the reception unit 85 in the multi-user MIMO system. As described above, for example considers the transmission signal processing circuit 811-1～N _SDM and the reception signal processing circuit 858-1～N _SDM respectively a signal processing circuit for the signal sequence of _{N SDM} systems of single terminal station device, further If the multi-user MIMO transmission weight calculation circuit 833 and the multi-user MIMO reception weight calculation circuit 862 are regarded as transmission / reception weight calculation circuits for single-user MIMO, basically, the base station apparatus 80 in the single-user MIMO system will be described in the above description. This shows the configuration of the unit 81 and the receiving unit 85.

What is important here is that the local oscillator 815 in the transmitting unit 81 is shared by the mixers 816-1 to 816-NBS _-Ant in each antenna system of the transmitting unit 81, and the local oscillator 853 in the receiving unit 85 is This is the point that the mixers 854-1 to 854-NBS _-Ant in each of the 85 antenna systems are shared. The phase of the transmission / reception signal is adjusted by each antenna, but the phase relationship between the signals input from the respective local oscillators 815 or 853 is always constant, so that the phase relationship between the transmission / reception signals is constant. It is possible to determine whether to multiply the weight. If a plurality of local oscillators are used in the transmission unit 81 or the reception unit 85, the directivity control of multiplying the transmission weight in at least the transmission unit 81 does not function effectively. Care must be taken in this regard when designing the device.

[Characteristics of MIMO channels dominated by line of sight]
First, the spatial multiplexing characteristics in the propagation path where the line of sight wave is dominant will be summarized. H _LOS is a channel matrix when the transmitting station and the receiving station are in line-of-sight environment, and _{Hi . i. d} . For simplicity, _HLOS is substituted by a matrix in which each component is all “1”, and the following channel matrix is used for the case of 16 transmission antennas and 16 reception antennas, and for the case of 256 transmission antennas and 16 reception antennas. For the two cases, the distribution of the absolute values of the 16 singular values of the channel matrix given by the following equation (9) is evaluated.

FIG. 8 shows the distribution characteristic of the absolute value of the singular value for each channel matrix. FIG. 8A shows _{Hi. i. d.} only (iid channel), and FIG. 8B shows a case of a rice channel (Rician channel) represented by equation (9) when the rice coefficient K = 10 dB. In this case, the right side is a case where the number of transmission sides is 256. When the number of transmission / reception antennas is the same as 16, the distribution of the absolute value of the singular value tends to widen, and the gap between the absolute values of the first singular value and the sixteenth singular value tends to widen. However, when the number of transmitting or receiving antennas becomes redundant, for example, as many as 256 transmitting antennas, the absolute value of the singular value becomes _{Hi. i.} There is almost no difference between the distribution probability of 0% and the 100% value without being affected by the value of the random number for each evaluation of _d . This is common to FIGS. 8 (a) and 8 (b), but _{Hi. i. In} the case of only _d., the gap from the first singular value to the sixteenth singular value is very small, whereas in the case of the rice channel of FIG. The gap between the absolute values is 20 dB, which is 10 dB larger than the Rice coefficient, while the gap between the second singular value and the sixteenth singular value is small. In other words, it can be seen from FIGS. 8A and 8B that when the line of sight wave is dominant, even if the number of antenna elements is increased, the top of FIG. The line gain concentrates too much on the transmission path corresponding to the (λ ₁ ) pipe, and the second (λ ₂ ) and third (λ ₃ ) pipes from the top in FIG. This means that the corresponding transmission line can hardly be used.

  On the other hand, when a millimeter wave or the like is used, the free space propagation loss becomes large depending on the frequency. For example, a gain of about 24 dB must be obtained somewhere at 80 GHz for 5 GHz, for example. For this reason, it is effective to increase the size of the antenna, but it is necessary to implement the enlargement of the antenna for spatial multiplexing and the enlargement of the antenna for gaining the line gain from different viewpoints. .

[Estimation and feedback of channel information]
In large-scale MIMO, the improvement of the line gain by utilizing a large number of antenna elements is only an effect obtained by performing transmission / reception directivity control (beamforming). Information is needed. However, since the signal between one antenna element and one antenna element is a signal at the stage where the line gain has not been improved before the directivity is formed, the estimation accuracy of the acquired channel information is generally low. . Although it is possible to improve the SNR by repeatedly receiving and averaging the same signal, the overhead of channel feedback increases. For example, odd or even subcarriers of all subcarriers are used, and only one of odd and even subcarriers is intermittently allocated to each of the two antenna elements and used. If the channel estimation is performed from the channel information obtained by the subcarriers, the channel estimation of the two antennas can be performed simultaneously, and the SNR of the subcarrier for performing the channel feedback can be improved by about 3 dB. It is. If the period of the intermittent subcarrier is changed from the two subcarrier intervals to the larger subcarrier interval as described above, it is possible in principle to further improve the SNR of the subcarrier on which the channel feedback is performed. In the case of general channel information, there is a problem that the frequency dependency is very large, and the accuracy of channel interpolation between discontinuous subcarriers is greatly deteriorated. Conversely, in order to improve the channel interpolation accuracy between the sub-carrier, to limit the number of subcarriers N _SC to decimation, where improvement amount of the resulting SNR is to be limited. Therefore, there is a need for a technique that realizes highly accurate channel estimation and highly efficient channel feedback in a high SNR environment.

  To deal with such a problem, for example, in Non-Patent Document 5, the base station apparatus combines a large number of antennas with predetermined transmission / reception weights, and forms a highly-gain super-directional beam with a very narrow beam width in the horizontal and vertical directions. Proposals have been made to cope with the problem by forming a large number at predetermined intervals. For example, when the terminal station device is located in an area of ± 60 degrees (120 degrees in total) in the horizontal direction, if the beam is formed at intervals of 5 degrees, the horizontal direction can be covered with 25 directional beams. In the vertical direction, for example, if the user exists only in an area of about 30 degrees, if the beam is formed at intervals of 5 degrees, the vertical direction can be covered with seven directional beams. In order to simultaneously cover the vertical and horizontal directions, a total of 175 fixed directional beams may be used. If the step width is set to 10 degrees, the whole pattern can be similarly covered with 52 patterns of fixed directional beams. If a plurality of such fixed beams are selected and a MIMO channel between the virtual antenna elements corresponding to the number of the fixed beams and the antenna elements of the terminal station apparatus is considered, then even if this virtual antenna element is used. Thus, channel information can be obtained in a situation where the line gain has been sufficiently secured. However, if the training signals are not transmitted by the number of virtual antenna elements, the terminal station apparatus does not know which virtual antenna element to select to secure a favorable communication environment. Overhead required for the transmission of the dynamic antenna element is required. As described above, it is possible to acquire MIMO channel information for realizing high-order spatial multiplexing while securing a sufficient line gain. Further, further ingenuity is required.

[About calibration and implicit feedback]
In an actual wireless communication device, in signal processing on the transmission side, a signal is often amplified by a high power amplifier immediately before transmission. In this case, there is an error in the amplification factor due to the individual difference of the high power amplifier, and the complex phase in the high power amplifier may rotate with a different value for each high power amplifier. Similarly, in signal processing on the receiving side, signal amplification is often performed by a low noise amplifier immediately after reception. In this case, there is an error in the amplification factor due to the individual difference of the low noise amplifier, and the complex phase may rotate with a different value for each low noise amplifier in the low noise amplifier. More strictly, individual differences also occur in transmission and reception circuits including other RF circuits such as filters.

  Generally, the amplification factor and the amount of phase rotation of a high power amplifier and a low noise amplifier have frequency dependence. If the individual difference between the amplification factor and the amount of rotation of the complex phase with frequency dependence is so large that it cannot be ignored, it is necessary to perform a calibration process when estimating downlink channel information from uplink channel information. is there. If the error between the amplification factor and the amount of phase rotation is substantially stable in time, the error between the amplification factor and the amount of phase rotation is measured in advance, and the error is increased using a coefficient for canceling the effect of the error. The link channel information is converted into downlink channel information. In general, a channel estimation method using such a property is called implicit feedback, and is distinguished from explicit feedback in which channel information estimated on the downlink is accommodated in control information as digital data and notified on the uplink. ing.

  Hereinafter, an example of the calibration process will be described. FIG. 9 is a diagram illustrating the asymmetry of channel information between the uplink and the downlink. In FIG. 9, reference numerals 25-0 to 25-2 indicate a wireless module, reference numerals 21-0 to 21-2 indicate a high power amplifier (HPA), and reference numerals 22-0 to 22-2 indicate a low noise amplifier (LNA). Reference numerals 23-0 to 23-2 indicate time division switches (TDD-SW), and reference numerals 24-0 to 24-2 indicate antenna elements.

Here, since only the functions that affect the channel information in the base station apparatus are extracted, configurations other than those illustrated are omitted, but the wireless modules 25-0 to 25-2 include other functions. When the signal passes through each of the high power amplifiers 21-0 to 21-2, the amplitude and the complex phase are Z _{HPA # 0} (f _k ), Z _{HPA # 1} (f _k ), and Z _{HPA # 2} (f _k ). If the amplitude is A and the complex phase is rotated by θ when passing through various amplifiers, the effect can be represented by a coefficient of Z = A · ^ejθ . When a signal passes through each of the low-noise amplifiers 22-0 to 22-2, the coefficient of influence due to the amplitude and the complex phase is Z _{LNA # 0} (f _k ), Z _{LNA # 1} (f _k ), and Z _{LNA # 2} ( f _k ). Here, it is assumed that there is frequency dependency as a general condition, and the frequency “(f _k )” for the k-th subcarrier is indicated.

Here, for example, channel information when a signal is transmitted from the wireless module 25-1 and the wireless module 25-2 to the test wireless module 25-0 will be described. Here, spatial channel information between the antenna element 24-1 of the wireless module 25-1 and the antenna element 24-3 of the wireless module 25-0 is represented by h ₁ (f _k ). channel information on the space between the antenna elements 24-2 and the antenna element 24-0 of the radio module 25-0 -2 is represented by _h 2 _{(f k).}

At this time, the channel information when the signal is actually transmitted from the wireless module 25-1 to the wireless module 25-0 indicates a change in the space h ₁ (f _k ) accompanying the passage of the high power amplifier 21-1. It is observed as a value multiplied by a coefficient Z _{HPA # 1} (f _k ) and a coefficient Z _{LNA # 0} (f _k ) indicating a change accompanying passage through the low noise amplifier 22-0.

Similarly, channel information when a signal is transmitted from the wireless module 25-2 to the wireless module 25-0 includes a coefficient Z ₂ (f _k ) in space indicating a change accompanying the passage of the high power amplifier 21-2. It is observed as a value multiplied by _{HPA # 2} (f _k ) and a coefficient Z _{LNA # 0} (f _k ) indicating a change accompanying passage through the low noise amplifier 22-0.

Therefore, a channel from the wireless module 25-1 to the wireless module 25-0 is represented by Z _{HPA # 1} (f _k ) · h ₁ (f _k ) · Z _{LNA # 0} (f _k ). The channel from the wireless module 25-2 to the wireless module 25-0 is represented by Z _{HPA # 2} (f _k ) · h ₂ (f _k ) · Z _{LNA # 0} (f _k ). Therefore, between the radio module 25-1 and the radio module 25-2, in addition to the difference in the channel information _h 1 _{(f k)} and _h 2 _{(f k),} relatively _{Z HPA} # 2 _{(f k} ) / Z _{HPA # 1} (f _k ) difference occurs.

This situation is the same in the reverse communication. When the signal transmitted from the wireless module 25-0 is received by the wireless module 25-1, the channel information is set to the high power h ₁ (f _k ) on the space. As a value obtained by multiplying a coefficient Z _{HPA # 0} (f _k ) indicating a change caused by passing through the amplifier 21-0 by a coefficient Z _{LNA # 1} (f _k ) indicating a change caused by passing through the low noise amplifier 22-1. Observed.

Similarly, when the signal transmitted from the wireless module 25-0 is received by the wireless module 25-2, the channel information indicates a change in the spatial h ₂ (f _k ) accompanying the passage of the high power amplifier 21-0. The coefficient Z _{HPA # 0} (f _k ) is multiplied by a coefficient Z _{LNA # 2} (f _k ) indicating a change accompanying the passage through the low noise amplifier 22-2.

Therefore, the channel from the wireless module 25-0 to the wireless module 25-1 is represented by Z _{HPA # 0} (f _k ) · h ₁ (f _k ) · Z _{LNA # 1} (f _k ). The channel from the wireless module 25-0 to the wireless module 25-2 is represented by Z _{HPA # 0} (f _k ) · h ₂ (f _k ) · Z _{LNA # 2} (f _k ). Therefore, between the radio module 25-1 and the radio module 25-2, in addition to the difference in the channel information _h 1 _{(f k)} and _h 2 _{(f k),} relatively _{Z LNA} # 2 _{(f k} ) / Z _{LNA # 1} (f _k ).

  As described above, the base station apparatus can acquire channel information including a change caused by the low noise amplifiers 22-1 to 22-2 connected to each antenna element on the uplink, but the base station apparatus does not Cannot be obtained directly. Therefore, downlink channel information is obtained by conversion from uplink channel information. For this conversion, the effects of individual differences between the low noise amplifiers 22-0 to 22-2 and the high power amplifiers 21-0 to 21-2 connected to the antenna elements 24-0 to 24-2 are cancelled. There is a need.

  Therefore, in the manufacturing stage of the base station apparatus, a test wireless module 25-0 serving as a reference is prepared, and the antenna terminals of the test wireless module 25-0 and the antenna terminals of the wireless modules 25-1 and 25-2 are prepared. And are connected directly with a cable. In an environment where channel information on the propagation path has a common value, channel information including changes due to the high power amplifiers 21-0 to 21-2 and the low noise amplifiers 22-0 to 22-2 is measured, and the measured channel information is used. To make corrections.

  FIG. 10 is a diagram showing an outline of the calibration. In FIG. 10, reference numerals 26-0 to 26-2 indicate antenna terminals, and reference numeral 27 indicates a coaxial cable. Note that the same reference numerals are given to the same functional units as those shown in FIG.

  FIG. 10A shows a configuration in which the wireless module 25-0 and the wireless module 25-1 are connected by a coaxial cable. FIG. 10B illustrates a configuration in which the wireless module 25-0 and the wireless module 25-2 are connected by a coaxial cable. FIG. 9 shows a state where a signal propagates in an actual space, whereas FIG. 10 shows a state where a signal propagates on a coaxial cable without passing through an antenna element.

A wireless module 25-1, 25-2, channel information of the coaxial cable 27 as a channel for connecting the wireless module 25-0 _is h 0 _{(f k).}
At this time, channel information from the wireless module 25-1 to the wireless module 25-0 is represented by Z _{HPA # 1} (f _k ) · h ₀ (f _k ) · Z _{LNA # 0} (f _k ). Channel information from the wireless module 25-2 to the wireless module 25-0 is represented by Z _{HPA # 2} (f _k ) · h ₀ (f _k ) · Z _{LNA # 0} (f _k ).

Channel information from the wireless module 25-0 to the wireless module 25-1 is represented by Z _{HPA # 0} (f _k ) · h ₀ (f _k ) · Z _{LNA # 1} (f _k ). Channel information from −0 to the wireless module 25-2 is represented by Z _{HPA # 0} (f _k ) · h ₀ (f _k ) · Z _{LNA # 2} (f _k ).
Therefore, after measuring these channel information, the calibration coefficients C ₁ (f _k ) and C ₂ (f _k ) represented by the following equations (10) and (11) are calculated.

The channel information from the wireless module 25-0 to the wireless module 25-1 is expressed as Z _{HPA # 0} (f _k ) · h ₁ (f _k ) / Z _{LNA # 1} (f _k ). It has been described that channel information from 0 to the wireless module 25-2 is represented by Z _{HPA # 0} · (f _k ) · h ₂ (f _k ) · Z _{LNA # 2} (f _k ). When these are multiplied by the calibration coefficients C ₁ (f _k ) and C ₂ (f _k ) of the equations (10) and (11), the following equations (12) and (13) are obtained.

  The right sides of the equations (12) and (13) are the channel information from the wireless module 25-1 to the wireless module 25-0 and the channel information from the wireless module 25-2 to the wireless module 25-0, as described above. Matches.

  As described above, the calibration coefficients corresponding to Expressions (10) and (11) are acquired at the stage of manufacturing the base station device, and are stored in the base station device. Downlink channel information can be calculated from uplink channel information using the coefficient.

  The following description focuses on the case where these calibration coefficients are obtained in advance and the values are used in digital signal processing. Naturally, on the analog circuit, the calibration coefficients in the base station apparatus and the base station are set such that all of these calibration coefficients have substantially constant values (if the complex phase is a constant value, the absolute value itself may have a difference). If the adjustment is performed in the terminal station device, it is also possible to read the processing assuming that all the calibration coefficients are 1. In particular, since the frequency dependence of the amplification rate of the amplitude is directly linked to waveform distortion, the amplifier is generally industrially designed to have a substantially constant (flat waveform) on the frequency axis as a characteristic of the amplifier.

  Similarly, in the case where the complex phase of the uplink and the downlink is adjusted to be a constant value, the phase is adjusted collectively at the stage after the FFT (at the time of reception) and the stage before the IFFT (at the time of transmission) in digital signal processing. As a result, the complex phase of the calibration coefficient represented by Expressions (10) and (11) becomes substantially constant for all the antenna elements, so that the same effect can be obtained.

  The above is the description regarding the calibration for the one-to-many transmitting / receiving antenna element SIMO channel. However, in the case of the calibration of the channel matrix of the plural antennas to the plural antennas, it is necessary to explain the correction in a slightly corrected form. It is. For example, as shown in FIG. 11, consider the case where there are two transmitting and receiving antenna elements. For the sake of simplicity, the left side is regarded as the base station apparatus 41 and the right side is regarded as the terminal station apparatus 51, and each of the downlink, which is communication from left to right, and the uplink, which is communication from right to left. Organize the relationship between channel matrices.

  First, when considering the downlink, the channel information observed by the antenna element of the terminal station device 51 on the receiving side is obtained by adding the components of the channel matrix in the space between the transmitting antenna end and the receiving antenna end to the base station device. The amplification factor and the rotation amount of the complex phase in the transmission high power amplifier 41 are multiplied by the amplification ratio and the rotation amount of the complex phase in the low noise amplifier of the terminal station device. These amounts of rotation differ for each high-power amplifier corresponding to the plurality of antenna elements of the base station device and for each low-noise amplifier corresponding to the plurality of antenna elements of the terminal station device. On the other hand, in the uplink, the channel information observed by the antenna element of the base station apparatus on the receiving side is added to each component of the channel matrix in the space between the transmitting antenna end and the receiving antenna end, The amplification factor and the rotation amount of the complex phase in the transmission high power amplifier are multiplied by the amplification ratio and the rotation amount of the complex phase in the low noise amplifier of the base station device. These amounts of rotation differ for each high-power amplifier corresponding to the plurality of antenna elements of the terminal station device and for each low-noise amplifier corresponding to the plurality of antenna elements of the base station device. Here, regarding the channel matrix in the space between the transmitting antenna end and the receiving antenna end, the value of each component itself does not change due to spatial symmetry, but in the uplink and downlink, the transmitting antenna and the receiving antenna are different. Since the correspondence between the base station apparatus and the terminal station apparatus is reversed, the transposition of the matrix in the spatial matrix enables the conversion between the uplink and the downlink. In addition to this, if the same processing as the above-described calibration in the SIMO channel is performed, the calibration for converting the channel state between the uplink and the downlink between the transmitting and receiving apparatus for an arbitrary test and the calibration is performed. It is possible to obtain a downlink channel matrix from an uplink channel matrix using the application coefficient.

Here, θ _{BS-HPA, m} ^(k) is the amount of phase rotation of the high-power amplifier corresponding to the m-th antenna element of the base station device, and _{ABS-HPA, m} ^(k) is the m-th antenna of the base station device. The amplification amount of the amplitude of the high-power amplifier corresponding to the element and θ _{BS-LNA, m} ^(k) are converted to the phase rotation amount of the low-noise amplifier corresponding to the m-th antenna element of the base station apparatus, and _{ABS-LNA, m} ^{( Let k) be} the amplitude amplification of the low noise amplifier corresponding to the m th antenna element of the base station device. Further, θ _{MT-HPA, m} ^(k) is used as the phase rotation amount of the high-power amplifier corresponding to the m-th antenna element of the terminal station device, and A _{MT-HPA, m} ^(k) is used as the m-th antenna element of the terminal station device. The amplification amount of the amplitude of the corresponding high-power amplifier and θ _{MT-LNA, m} ^(k) are the phase rotation amounts of the low-noise amplifier corresponding to the m-th antenna element of the terminal station device, and A _{MT-LNA, m} ^(k) Is the amplification amount of the amplitude of the low noise amplifier corresponding to the m-th antenna element of the terminal station device. Further, θ _Test-HPA ^(k) is the phase rotation amount of the high power amplifier of the test station, A _Test-HPA ^(k) is the amplification amount of the amplitude of the high power amplifier of the test station, and θ _Test-LNA ^{( Let k) be} the phase rotation amount of the test station and A _Test-LNA ^{(k) be} the amplification amount of the amplitude of the low noise amplifier of the test station. If the suffixes BS (base station apparatus) and MT (terminal station apparatus) of each coefficient are generalized and indicated as STA (base station apparatus or terminal station apparatus), the calibration coefficient for the m-th antenna element of the STA is expressed by the formula ( 14), and using this, it is possible to obtain the downlink channel matrix from the uplink channel matrix as in equation (15).

In general, a coefficient multiplied equally to all elements of a channel matrix has no effect at all when evaluating the characteristics of the propagation path indicated by the channel matrix or when calculating a transmission / reception weight matrix (vector). It can be treated as a coefficient without any In this sense, in Equation (15), the coefficients θ _Test-HPA ^(k) , A _Test-HPA ^(k) , θ _Test-LNA ^(k) , and A _Test-LNA ^{(k )} Are included, but since these are all coefficients that are multiplied equally to all elements of the channel matrix, this value has no particular effect particularly when calculating the transmission / reception weight matrix (vector). Furthermore, a local oscillator signal input to a mixer when converting a baseband signal to a radio frequency on the transmitting station side, and a local oscillator signal input to a mixer when converting a radio frequency signal to a baseband signal on the receiving station side The state of the complex phase of each of the signals of the signals also actually affects the complex phase of each component of the observed channel matrix, but since all the coefficients are equally multiplied by all the antennas, this is also Also, all components of the channel matrix are multiplied equally as coefficients. Since this coefficient does not affect the transmission / reception weight matrix (vector), it is possible to perform the evaluation ignoring these effects. As described above, if the calibration coefficient is known in advance, it is possible to convert the channel matrix between the uplink and the downlink based on the calibration coefficient. Similarly, if a reception weight vector can be obtained, a transmission weight vector can be calculated by applying a calibration coefficient to the reception weight vector.

[Wireless entrance in mobile fronthaul]
The same applies to the fourth-generation mobile phone. However, assuming that the maintainability of the base station apparatus that needs to be installed in large numbers is improved and that the cooperative transmission from a plurality of base station apparatuses is assumed, most of the base station apparatuses are assumed. Functions (BBU: Base Band Unit) in one place, (Function 1) Generation of analog baseband signal by digital-to-analog conversion from digital sampling data, (Function 2) Radio frequency from analog baseband signal Up-conversion to band signal, (Function 3) Power amplification and transmission from antenna, (Function 4) Reception and power amplification at antenna, (Function 5) Down-conversion from radio frequency band signal to analog baseband signal , (Function 6) A remote radio head having only functions such as generation of digital sampling data from an analog baseband signal by A / D conversion ( A cloud-type radio access network (C-RAN: Centralized Radio Access Network or Cloud Radio Access Network) in which only RRHs (Remote Radio Heads) are distributed is effective. This is a technology called “fronthaul” or “mobile fronthaul” in comparison with a so-called “backhaul line”, and even today, CPRI (Interface Standard) is used as an interface standard for transmitting digital sampling data through an optical fiber. Common Public Radio Interface) has been standardized.

Here, in order to perform joint transmission (JT: Joint Transmit) or joint reception (JR: Joint Reception) with a plurality of RRHs, delay jitter is suppressed with an accuracy of the order of 10 ⁻⁹ and an optical fiber is used. The bit error rate is required to have a high quality of the order of 10 ^{−12 and} a low delay of 100 μs or less. This mobile fronthaul can be provided not only by an optical fiber but also by a wireless line. In this case, the mobile fronthaul has a delay jitter of 10 ⁻⁹ , a bit error rate of 10 ⁻¹² , and a transmission delay of 100 μs or less. High quality must be achieved.

Among the existing wireless entrance devices, a large parabolic antenna forms an ultra-high gain of 40 dBi to 50 dBi and a high directivity, and by opposing this antenna, multipath components are almost eliminated, and stable without nonlinear distortion. Some transmission lines make it possible to achieve a bit error rate of ^10-12 equivalent to that of an optical fiber by one transmission without retransmission control. However, when the directivity gain is ensured by utilizing the large-scale MIMO as described above, the redundancy of the antenna is used, and the multi-path component is used in the same manner as in the normal spatial multiplexing to obtain a plurality of streams. Multiplexing was also possible. However, when such a multipath component is taken in, the effect of nonlinear distortion cannot be ignored, and in general, the error rate characteristic lowers the error floor, and it is difficult to realize a sufficiently low error rate even in a high SNR environment. It was difficult. In this case, it is necessary to improve the error rate characteristics by performing retransmission control. However, generally, retransmission control involves a large transmission processing delay, and it is difficult to complete retransmission control with a low delay of 100 μs or less. Furthermore, in the case of performing high-efficiency retransmission control in ultra-wideband large-capacity transmission, the process of selective retransmission control becomes complicated, and in addition, due to a delay restriction of 100 μs or less, regardless of the success / failure of transmission, two to three times. It is also necessary to terminate the retransmission at the upper limit of the number of transmissions and discard it. When the retransmission is aborted, it may cause inconsistency between the transmitting side and the receiving side, but the inconsistency between the transmitting side and the receiving side leads to a communication deadlock state. However, if negotiation processing or the like for avoiding such a state is performed, the processing is further complicated, which is not preferable. There is a demand for realizing highly efficient retransmission control at an ultra-high speed while avoiding the complication of such signal processing.

  In the mobile fronthaul, as described above, between the BBU and the RRH, digital sampling data of a signal to be transmitted from the antenna element of the RRH is transmitted, and the above (Function 1) to (Function 6) are transmitted by the RRH. ) To transmit and receive the actual radio frequency signal. However, when a large-scale antenna is used in the fifth-generation small cell base station apparatus, if the conventional method of transmitting digital sampling data on an optical fiber is followed, each antenna element has Had to be transmitted. In general, the transmission capacity of digital sampling data is said to be about 16 times as large as the amount of user data to be actually transmitted in a wireless section. This is a case where data transmission of 1 Gbit / s is performed in a wireless section. However, digital sampling data corresponding to 16 Gbit / s must be transmitted in the mobile fronthaul section. However, if a large-scale MIMO technology using 256 antenna elements is adopted in the fifth generation small cell base station apparatus, the transmission capacity required for the mobile fronthaul is 4 Tbit / s, and the transmission medium Is an optical fiber at present. In general, a system that can be used at a relatively low price has a low transmission speed and is widely used, and is a communication system of about 10 Gbit / s at present. Considering future technological innovations, the data capacity to be transmitted on the mobile fronthaul is not allowed to diverge as the number of antenna elements increases, and at least digital sampling for the number of spatially multiplexed signal systems -It is required to fit in the data capacity.

  From such a viewpoint, the review of the function allocation between the BBU and the RRH is currently under consideration, and in addition to the functions of (Function 1) to (Function 6) described above, the function is originally implemented on the BBU side. Consideration has been given to mounting complex MIMO signal processing, FFT / IFFT signal processing, and the like on the RRH side in addition to the physical layer modulation function and demodulation function in power signal processing. As a result, the information capacity to be transmitted on the mobile fronthaul is greatly reduced, but this is far from the original idea of the mobile fronthaul. That is, since functions (function 1) to (function 6) do not include a function specialized for the modulation / demodulation method specific to the radio system to be adopted, only a change on the BBU side at the time of a future system version upgrade. What could be handled without any modification to RRH, if the function allocation was reviewed as described above, even if basic design parameters were changed, for example, the number of subcarriers was changed, the version was upgraded. Therefore, the RRH itself must be remodeled.

  Therefore, there is a need for a device that can support large-scale MIMO but does not increase the transmission capacity of digital sampling data to be transmitted on the mobile fronthaul.

  Hereinafter, description of the mobile fronthaul will be made based on the drawings. FIG. 12 is a diagram showing an outline of the division of functions between a mobile fronthaul and a mobile backhaul in the related art. In particular, FIG. 12A shows a mobile fronthaul, and FIG. 12B shows a mobile backhaul. Here, only functions related to signal transmission in the direction from the network side to the user (in the case of a mobile fronthaul, from the BBU to the RRH direction) are extracted. In the figure, 401 is a MAC layer processing circuit, 402 is a transmission signal processing circuit, 403 is a time axis signal generation circuit, 404 is an optical interface circuit, 405 is an optical fiber, 406 is an optical interface circuit, 407 is a D / A converter, 408 denotes an RF processing circuit, 409 denotes an antenna element, 411-1 denotes a BBU, 412-1 denotes an RRH, 414 denotes an optical interface circuit, 415 denotes an optical fiber, and 416 denotes an optical interface. The MAC layer processing circuit 401, the transmission signal processing circuit 402, and the time axis signal generation circuit 403 constitute regions 400 and 410 in which baseband signal processing relating to wireless is performed as a whole.

  In FIG. 12A, when a signal to be transmitted is input to the BBU 441-1 from the network side, the MAC layer processing circuit 401 performs signal processing of the MAC layer, and a frame format used for transmission / reception in a wireless section and a network format. It converts and terminates the frame format of the data flowing on the side, and inputs a signal in the format of a wireless packet to the transmission signal processing circuit 402. The transmission signal processing circuit 402 performs transmission signal processing of a wireless signal. Here, the transmission method in the wireless section is not particularly limited. For example, if OFDM is used, error correction coding, interleaving, modulation processing for each subcarrier, and the like are performed as necessary. The signal generated in this manner is converted to a time-axis signal by the time-axis signal generation circuit 403. For example, taking the case of OFDM as an example, IFFT is performed to convert a signal on the frequency axis into a signal on the time axis, insert a guard interval, and perform waveform shaping processing between symbols. As a result, each sampling value of the digital baseband signal is converted into a signal that is continuous in time series. The data of these sampling values is converted into a predetermined frame format by the optical interface circuit 404, converted from an electric signal to an optical signal, and output to the optical fiber 405. The signal output to the optical fiber 405 is transmitted to the RRH 442-1, where the optical interface circuit 406 converts the optical signal into an electric signal, terminates a signal of a predetermined format, and converts the signal into a digital baseband signal. Generate an information sequence of sampling values. The D / A converter 407 converts the signal to an analog baseband signal at a predetermined clock rate, the RF processing circuit 408 converts the signal to a radio frequency signal using an up-converter, and removes out-of-band radiation signals using a filter. The signal is amplified by a power amplifier and transmitted to a space through an antenna 409. As described above, the functions corresponding to the radio signal base station apparatus as a whole are accommodated separately in the BBU 441-1 and the RRH 442-1 which are mediated by optical fibers. The feature here is that the radio digital baseband signal processing is integrated in the BBU provided in the station building on the network side, so even if there is any change in the wireless communication system, all of the changes are made by the BBU side. There is a merit that it is done.

  On the other hand, the mobile backhaul has a configuration shown in FIG. 12B as a configuration corresponding to FIG. For example, user data is transmitted as it is via an optical fiber to a base station device installed on the roof of a building or on a telephone pole, and all of the user data is closed within the base station device to perform radio signal processing. Specifically, on the network side, user data to be transmitted by a wireless line is transmitted from the optical interface circuit 414 to the optical interface circuit 416 in the base station device via the optical fiber 415. The optical interface circuit 416 converts the optical signal into an electric signal, terminates a signal of a predetermined format, and outputs user data to the MAC layer processing circuit 401. The MAC layer processing circuit 401 performs signal processing of the MAC layer, converts and terminates a frame format used for transmission / reception in a wireless section and a frame format of data flowing on the network side, and converts a signal in a wireless packet format into a transmission signal processing. Input to the circuit 402. The transmission signal processing circuit 402 performs transmission signal processing of a wireless signal. Here, the transmission method in the wireless section is not particularly limited. For example, if OFDM is used, error correction coding, interleaving, modulation processing for each subcarrier, and the like are performed as necessary. The signal generated in this manner is converted to a time-axis signal by the time-axis signal generation circuit 403. For example, taking the case of OFDM as an example, IFFT is performed to convert a signal on the frequency axis into a signal on the time axis, insert a guard interval, and perform waveform shaping processing between symbols. As a result, each sampling value of the digital baseband signal is converted into a signal that is continuous in time series. The data of these sampled values is input to a D / A converter 407, which converts the data into an analog baseband signal at a predetermined clock rate. After removing the out-of-band radiation signal with a filter, the signal is amplified by a high-power amplifier, and the amplified signal is transmitted from an antenna 409 to the space. The description of the signal processing and the circuit configuration of the base station apparatus described so far is based on this mobile backhaul.

  In the above description of FIG. 12, the number of antenna elements provided in the RRH 442-1 is one. However, in a mobile network in the future, application of the Massive MIMO technology using a multi-element antenna is assumed. Thus, the amount of information to be transmitted increases. FIG. 13 is a diagram showing an outline of the function allocation of the mobile fronthaul and the mobile backhaul in the related art when a multi-element antenna is used. In particular, FIG. 13A shows a mobile fronthaul, and FIG. 13B shows a mobile backhaul. Here, as in FIG. 12, only functions related to signal transmission in the direction from the network side to the user (in the case of a mobile fronthaul, from the BBU to the RRH direction) are extracted. 13, the same components as those in FIG. 12 are denoted by the same reference numerals. 13, reference numeral 412 denotes a transmission signal processing circuit, 424 denotes an optical interface circuit, 425 denotes an optical fiber, 426 denotes an optical interface circuit, 427 denotes a D / A converter, 428 denotes an RF processing circuit, 429 denotes an antenna element, and 430 denotes transmission. A weight multiplication circuit, 431 is a transmission weight processing circuit, 433 is a time axis signal generation circuit, 441-2 is BBU, and 442-2 is RRH. The MAC layer processing circuit 401, the transmission signal processing circuit 412, the transmission weight multiplication circuit 430, the transmission weight processing circuit 431, and the time axis signal generation circuit 433 constitute an area 420 for performing baseband signal processing related to wireless as a whole. Similarly, the MAC layer processing circuit 401, the transmission signal processing circuit 412, the transmission weight multiplication circuit 430, the transmission weight processing circuit 431, and the time axis signal generation circuit 433 constitute a region 420 or 440 in which baseband signal processing related to wireless is performed as a whole. I do. In FIG. 13A, when a signal to be transmitted to the BBU 441-2 is input from the network side, the MAC layer processing circuit 401 performs signal processing of the MAC layer, and a frame format used for transmission and reception in a wireless section and a network format. It converts and terminates the frame format of the data flowing on the side, and inputs a signal in the format of a wireless packet to the transmission signal processing circuit 412. The transmission signal processing circuit 412 performs transmission signal processing of a wireless signal. Here, the transmission method in the wireless section is not particularly limited. For example, if OFDM is used, error correction coding, interleaving, modulation processing for each subcarrier, and the like are performed as necessary. At the same time, for example, reception weight information and the like collected on the receiving side of the BBU not shown here are provided to the transmission weight processing circuit 431 via the transmission signal processing circuit 412. The transmission weight processing circuit 431 performs a calibration process on the reception weight information and the like, calculates a transmission weight, and outputs this to the transmission weight multiplication circuit 430. The transmission weight multiplication circuit 430 multiplies the transmission signal input from the transmission signal processing circuit 412 by the transmission weight. The processing here is performed on the frequency axis when, for example, OFDM is used. That is, a different transmission weight is used for each subcarrier calculated based on a different reception weight for each subcarrier, and a transmission signal different for each subcarrier is multiplied for each subcarrier. The signal generated in this manner is converted to a time-axis signal by the time-axis signal generation circuit 433 for each antenna system. For example, taking the case of OFDM as an example, IFFT is performed to convert a frequency-axis signal into a time-axis signal, insert a guard interval, and perform waveform shaping processing between symbols. It is carried out every time. As a result, each sampling value of the digital baseband signal is converted into a signal for each antenna element that is continuous in time series. The data of these sampling values is converted into a predetermined frame format by the optical interface circuit 424, converted from an electric signal to an optical signal, and output to the optical fiber 425. Here, unlike the case of FIG. 12 described above, it is necessary to simultaneously accommodate signals for the number of antenna elements on the optical fiber, and the capacity can be increased by an arbitrary method such as wavelength multiplexing, TDM, or using a plurality of optical fibers. Is transmitted on the transmission path of the optical fiber, and input to the RRH 442-2. In the RRH 442-2, the optical interface circuit 426 converts an optical signal into an electric signal, terminates a signal of a predetermined format, and generates an information sequence of sampling values as a digital baseband signal. As described above, here, the sampling signals of the systems for the number of antenna elements are reproduced. The D / A converter 427 converts the signal into an analog baseband signal for each antenna system at a predetermined clock rate, and the RF processing circuit 428 uses an up-converter using a common local oscillator for each antenna system to output a radio frequency signal. After individually removing out-of-band radiation signals with a filter, the signals are separately amplified by a high-power amplifier and transmitted to the space from the individual antenna elements 429. As described above, the functions corresponding to the radio signal base station apparatus as a whole are accommodated separately in the BBU 441-2 and the RRH 442-2, which are mediated by optical fibers.

  Although the above description is based on the policy of mobile fronthaul, the number of antenna elements is limited to a very small number, such as a sector antenna with several sectors, considering the transmission capacity of the signal actually flowing on the optical fiber. It is thought that it is possible. It is a general idea that the cost is enormous in consideration of the number of optical fibers to be laid, the capacity of data to be transmitted, and the like for 100 or more antenna elements assumed in Massive MIMO or the like.

  On the other hand, the mobile backhaul adopts the configuration shown in FIG. 13B as a configuration corresponding to FIG. 12B, the area 410 for performing wireless signal processing is replaced with an area 440 for performing wireless signal processing having exactly the same function as the area 420 for performing wireless signal processing in FIG. The processing after the A converter 427 is exactly the same as in FIG. 13A except that it is performed for each antenna system. The feature here is that although the number of antenna elements in the base station device is increasing, the information capacity on the transmission path from the optical interface circuit 414 to the optical interface circuit 416 in the base station device via the optical fiber 415 is increasing. That is not. Therefore, the problem shown in FIG. 13A does not occur. However, originally, the problem in which the mobile fronthaul is useful as shown in FIG. 12A cannot be solved at all. Generally, while suppressing the transmission capacity on the optical fiber, the mobile fronthaul can be reduced. There has been a demand for a configuration that can achieve both advantages.

  In such a study, a review of a function distribution between the BBU and the RRH has been considered as a mobile midhaul intermediate between the fronthaul and the backhaul. FIG. 14 shows a configuration example of a mobile midhaul in the related art. Here, as in FIG. 12, only functions related to signal transmission (from BBU to RRH) from the network side to the user are extracted. 12 and 13 are denoted by the same reference numerals. The MAC layer processing circuit 401, the transmission signal processing circuit 412, the transmission weight multiplication circuit 430, the transmission weight processing circuit 431, and the time axis signal generation circuit 433 form regions 445-1 and 445-2 for performing baseband signal processing related to wireless as a whole. 12 (a) and FIG. 13 (a), the MAC layer processing circuit 401 includes a transmission signal processing circuit 412, a transmission weight multiplication circuit 430, and a transmission weight processing in an area 445-1 on the BBU side. The circuit 431 and the time axis signal generation circuit 433 are separately arranged in a region 445-2 on the RRH side, and the overall function distribution is reviewed.

  By reviewing the above function allocation, in the case of the mobile midhaul of FIG. 14, the transmission capacity on the optical fiber is almost the same as that of FIG. 12 (a), and the amount of information is large as shown in FIG. 13 (a). Without having to increase. In this sense, the effect of suppressing the transmission capacity on the optical fiber is sufficiently obtained. However, from the viewpoint of eliminating the processing depending on the wireless transmission method from the RRH 442-3 side, which is the purpose of the mobile fronthaul in the first place, and consolidating all the processing depending on the wireless method on the BBU 441-3 side, The fact that the processing of the area 445-2 is arranged on the RRH 442-3 side is not sufficiently suitable for the purpose of the front hole as compared with the configuration of the mobile front hole shown in FIG. In this sense, it is also called mid-hole as an intermediate point between the front hole and the backhaul. Ideally, many antenna elements are mounted on the RRH side, but the BBU side depends on the wireless transmission system. It is preferable to be able to consolidate the processes to be performed, and still suppress the information capacity flowing on the optical fiber to the same level as the backhaul.

[Other issues]
It is known that techniques such as OFDM and SC-FDE are generally effective for removing multipath components in a multipath environment. However, in existing systems such as OFDM and SC-FDE, the base station apparatus and the radio station apparatus generally use their own asynchronous clocks and local oscillators, and perform transmission processing in burst mode. In the burst mode, there is no problem because the processing is reset for each burst even if there is a clock error. Further, with respect to the clock error and the frequency error of the local oscillator, if the error does not break the orthogonality of each subcarrier when performing the FFT, further higher-precision synchronization is not required. Therefore, if a certain frequency error is detected, the frequency error is corrected by AFC processing on the transmission side and the reception side, and the remaining frequency error that remains is estimated, for example, in OFDM by estimating the amount of phase rotation using pilot subcarriers. , It was possible to perform the correction processing.

However, as described above, in the case of the mobile fronthaul, the delay jitter is required to be 10 ⁻⁹ or less. could not.

  Further, assuming the use of the e-band of 70 to 80 GHz, the bandwidth in which the phase noise exists becomes wider than the subcarrier interval such as OFDM or SC-FDE, and the bandwidth within the time shorter than the OFDM symbol period is reduced. Due to the phase noise that is an uncertain fluctuation of the complex phase, a component that breaks orthogonality between subcarriers such as OFDM cannot be ignored. This is a problem because the FFT processing is performed only. In the case of single carrier transmission, signal processing of a single carrier is performed in a time period shorter than a block length which is a unit of block transmission such as OFDM or SC-FDE. It was possible to correct the phase fluctuation component before the accumulation of the phase noise became large. However, if spatial multiplexing transmission is performed using the MIMO channel as described above, the transmission and reception weights for separating the signals of each stream have different values for each subcarrier, and the frequency dependence of this weight is different. In order to perform signal processing reflecting the above, it was necessary to temporarily convert the received signal into a signal on the frequency axis by FFT. As described above, the FFT processing is indispensable for the spatial multiplexing signal separation. However, when the FFT is performed, there is a problem that the phase noise cannot be removed. However, before the FFT processing is performed and the spatial separation on the frequency axis is performed, single-carrier signals of a plurality of streams interfere with each other, so that single-carrier signal detection becomes difficult. As described above, if signal separation can be performed on the time axis without performing FFT processing, phase noise compensation can be performed by signal processing of a single carrier signal thereafter, and such a technique is required.

Atsushi Ota, Satoshi Kurosaki, Kazuki Maruta, Takuto Arai, Masataka Iizuka, "Proposal of B-5-175 Large-scale Antenna Wireless Entrance System," Proc. Of the IEICE General Conference, 2013 (Communication_1), 2013 March 5, p. 585 Takuto Arai, Atsushi Ota, Satoshi Kurosaki, Kazuki Maruta, Masataka Iizuka, "B-5-176 Calculation Method of Transmit / Receive Weight in Large-Scale Antenna Wireless Entrance System", Proc. Of the IEICE General Conference, 2013 (Communication_1) March 5, 2013, p. 586 Kazuki Maruta, Atsushi Ota, Satoshi Kurosaki, Takuto Arai, Masataka Iizuka, "B-5-177 Inter-User Interference Suppression Method in Large-Scale Antenna Wireless Entrance System," Proc. Of the IEICE General Conference, 2013 (Communication_1 ), March 5, 2013, p. 587 Satoshi Kurosaki, Atsushi Ota, Kazuki Maruta, Takuto Arai, Masataka Iizuka, "B-5-178 Low Correlation Scheduling Method in Large-Scale Antenna Wireless Entrance System", Proc. Of the IEICE General Conference, 2013 (Communication_1) March 5, 2013, p. 588 Tatsunori Ohara, Satoshi Suyama, Shinkiyun, Yukihiko Okumura, "RCS2013-349 Combination processing of fixed beamforming and eigenmode precoding in ultra-high-speed Massive MIMO transmission using high frequency band", IEICE Technical Report, Wireless Communication System (RCS), Vol. 113, No. 456, February 24, 2014, p. 259-p. 264

  In an environment where a large capacity of a transmission path is required in a mobile network in the future, (Environment 1) In an access system for a mobile user, large capacity transmission from a base station apparatus to a mobile user terminal station apparatus, (Environment 2) Small For example, when a cell radio base station device is installed on a wall of a building or the like, large-capacity transmission at a radio entrance for making an entrance line to the base station device wireless for construction reasons. For example, large-capacity transmission of a wireless entrance as an entrance line that collects data of a large number of mobile users once in a train and collectively transfers the collected data to a terrestrial network can be considered. In the case of (Environment 1), for the purpose of efficient transmission to a user existing in a relatively small area, a service area is formed in a direction looking directly downward from a high place such as a building wall or a roof, and a terminal is formed. The line-of-sight environment is generally ensured due to the physical characteristics of radio waves arriving from above in the station equipment, and the antenna elements mounted in the terminal station equipment have some directivity in consideration of the arrival of radio waves from above. If an antenna with a gain is used, the line of sight becomes a very dominant environment as a result. Similarly, in the case of (Environment 2), since the installation place is fixedly installed, it is expected that the line-of-sight environment is secured and the antenna element uses a high-gain directional antenna. 1) Line of sight is dominant above. In the case of (Environment 3), for example, a train or the like is a moving body, so a directional antenna as high as (Environment 2) cannot be used. However, considering the feature of one-dimensional movement of a train, It is hard to imagine that a directional antenna suitable for the one-dimensional path is used and that an obstacle is interrupted on the track, and it is expected that the line of sight wave is more dominant than (Environment 1).

  As described above, in general, in a spatial multiplexing transmission using a MIMO channel matrix, a multipath environment in which a large number of reflected waves exist is effective, and is not particularly suitable when a direct line-of-sight component is dominant. Not known to. In order to avoid this problem and perform high-order spatial multiplexing transmission in an environment where the line-of-sight wave is dominant, it is necessary to increase the size of the antenna element of the base station apparatus spatially (extend the antenna aperture length). There are two options. In other words, it is possible to pseudo approximate a multipath environment by forming a situation where the path lengths on the transmission path between the antenna elements are different at random.

  On the other hand, as described above, in the future mobile networks are expected to increase the capacity of transmission lines, and considering operation with a wider frequency bandwidth, operation in higher frequency bands such as quasi-millimeter waves and millimeter waves Is forced. It should be noted here that when using a high frequency band, each device or between the device and the antenna is connected with a coaxial cable or the like, and when transmitting and receiving signals and the like over this cable, transmission over the cable is required. The problem is that the loss becomes very large. Specifically, for a signal in the 60 GHz band or the like, a loss of about 15 dB per meter is expected, although it depends on the cable.

  Even when using a millimeter-wave band that is disadvantageous as a gain in circuit design, even if a large number of antennas are installed to secure the line gain, a huge amount of There is no point in having a cable loss. Further, when the number of antenna elements is increased to secure the line gain, a huge number of cables are laid over the spatial spread of the antenna, and the transmission loss during that time is reduced. If a low-loss cable is used to reduce the cable length, the cable management during that time has a low degree of freedom in installation such as bending of the cable, and the restriction on the installation of the base station apparatus becomes very large, which is not practical.

In other words, in the future, in a wireless entrance or access system in a mobile network, even though line-of-sight waves are the dominant environment, high-capacity transmission is realized by high-order spatial multiplexing using high frequency bands such as millimeter waves. It is necessary to construct a wireless system by solving these problems.
In addition, in addition to the above discussion dedicated to wireless links, the provision of mobile fronthaul links using optical fibers to base stations equipped with a large number of antenna elements cannot be ignored in the above wireless trends. The situation. As described above, transmission of sampling data from the mobile fronthaul line originally requires an optical line having a capacity about 16 times the actual amount of user data information. In 5G mobile, it is targeted to increase the capacity of user data to 10 Gbps by utilizing the (quasi) millimeter wave band or the like. If the capacity is increased 16 times, 160 Gbps capacity is required for the mobile fronthaul line, and furthermore, As the number of antenna elements increases, it is theoretically required to increase the capacity up to 160 Gbps × the number of antenna elements. Even if the speed of increasing the capacity of a wireless link is equal to the speed of developing technology for increasing the capacity of an optical link, this increase in the number of antenna elements far exceeds the permissible limit of the optical link. Yes, not very cost-effective.
In this way, even in the future use of mobile fronthaul lines using optical fibers in mobile networks, these problems have been solved in order to economically install wireless base station devices equipped with a large number of antenna elements. It is necessary to build a mobile network that combines light and wireless in the future.

  In view of the above circumstances, the present invention realizes wireless communication between a wireless device and a terminal station device by mounting a large number of antenna elements on the wireless device side while realizing band expansion in a situation where direct waves in a line-of-sight environment are dominant. To provide a wireless communication system and a wireless communication method capable of suppressing an increase in the amount of transmission information between a wireless device and a baseband device while consolidating basic functions of wireless communication in the baseband device while improving characteristics of the wireless communication device It is an object.

  One embodiment of the present invention relates to the eleventh embodiment, a wireless device including a plurality of antenna elements, a baseband device that controls the wireless device connected to the wireless device via a wired line or a wireless line, A terminal station device that wirelessly communicates with the device, a wireless communication system including the baseband device, a transmission signal processing unit that generates a sampling data sequence of a baseband signal from data to be transmitted to the terminal station device, Transmission weight acquisition means for acquiring a transmission weight vector on a time axis used for transmission to the terminal station apparatus, and first transmission means for transmitting the transmission weight vector and the sampling data sequence to the wireless device, The wireless device receives the transmission weight vector and the sampling data sequence from the baseband device First receiving means for sequentially multiplying the sampling data sequence by the transmission weight vector for each sampling data, performing digital / analog conversion on the sampling data corresponding to the vector component of the multiplication result, and obtaining the result by the conversion. Wireless transmission means for transmitting a signal based on the obtained analog signal from the corresponding antenna element as a wireless signal.

  According to one aspect of the present invention, in the above wireless communication system, there are a plurality of the terminal station devices to which data is to be transmitted, and the transmission signal processing unit performs the sampling data sequence for each data to be transmitted to each of the terminal station devices. The transmission weight acquisition means calculates the transmission weight vector for each terminal station device, and the wireless transmission means sets, for each transmission destination, the transmission destination for each sampling data for the sampling data sequence. The corresponding transmission weight vectors are sequentially multiplied, the multiplication results for all transmission destinations are added and synthesized for each sampling data, and the sampling data corresponding to the vector component of the synthesis result is subjected to digital / analog conversion and obtained by the conversion. And transmitting a signal based on the analog signal as a radio signal from the corresponding antenna element. .

  One embodiment of the present invention relates to the eleventh embodiment, a wireless device including a plurality of antenna elements, a baseband device that controls the wireless device connected to the wireless device via a wired line or a wireless line, A terminal station device that wirelessly communicates with the device, wherein the wireless device receives a signal transmitted from the terminal station device via each of the antenna elements, and receives the antenna from the received signal. Wireless receiving means for generating a received sampling signal vector sequence which is a sampling data sequence of a baseband signal corresponding to each element; second receiving means for receiving a reception weight vector on the time axis from the baseband device; The reception weight vector received from the baseband device for a sampling signal vector sequence A multiplying unit for sequentially multiplying each sampling data, and a second transmitting unit for transmitting a series of sampling data sequences generated by the multiplying unit to the baseband device, wherein the baseband device is Receiving weight acquisition means for acquiring the reception weight vector used for reception from the terminal station device, third transmission means for transmitting the reception weight vector to the wireless device, and the one system received from the wireless device. And a received signal processing unit for reproducing data transmitted by the terminal station apparatus from the sampled data sequence.

  One aspect of the present invention is the wireless communication system described above, wherein the wireless device includes a plurality of the antenna elements based on a signal when a known signal transmitted from the terminal station device is received by each of the antenna elements. Is a reference antenna element, and a correlation calculation is performed to calculate a correlation value over a predetermined number of samples between the known signal received by the reference antenna element and the known signal received by another antenna element. Means, and a fourth transmission means for transmitting information for each antenna element regarding the correlation value between the antenna elements calculated by the correlation calculation means to the baseband apparatus, the baseband apparatus, The information for each antenna element regarding the correlation value between the antenna elements calculated by the correlation calculation means is transmitted to the radio. A third receiving means for receiving laid et al, the reception weight acquisition means calculates the receiving weight vector based on the information of each antenna element related to the correlation value between the antenna elements.

  One embodiment of the present invention relates to the eleventh embodiment, which includes a wireless device including a plurality of antenna elements, a baseband device that controls the wireless device, and a terminal station device that wirelessly communicates with the wireless device. In the communication system, the wireless device receives a signal transmitted from the terminal station device via each of the antenna elements, and a sampling data sequence of a baseband signal corresponding to each of the antenna elements from the received signal. A wireless receiving unit that generates a certain received sampling signal vector sequence, and based on a signal when a known signal transmitted from the terminal station device is received by each of the antenna elements, any one of the plurality of antenna elements As a reference antenna element, the known signal received by the reference antenna element and received by the other antenna elements Correlation calculation means for calculating a correlation value over a predetermined number of samples with the known signal in a time domain, and reception weight calculation means for calculating a reception weight vector based on the correlation value calculated by the correlation calculation means; Multiplication means for sequentially multiplying the reception weight vector calculated based on the correlation value calculated by the correlation calculation means with respect to the reception sampling signal vector sequence for each sampling data; Second transmission means for transmitting a sampling data sequence of a system to the baseband apparatus, the baseband apparatus instructing the reception weight calculation means to calculate the correlation value and the reception weight vector. The terminal station device from the one-system sampling data sequence received from the wireless device. Reception signal processing means for reproducing the signal data comprises a wireless communication system.

  In one aspect of the present invention, in the wireless communication system described above, there are a plurality of the terminal station devices serving as data transmission sources, and the multiplying unit includes the reception weight vectors for each of the transmission source terminal station devices and the reception weight vector. Multiplying a sampling signal vector sequence for each sampling data, the second transmission means transmits the one system of sampling data sequences of the number of the terminal station devices of the transmission source to the baseband device, and The processing unit reproduces the data transmitted by each of the terminal station devices from the one-system sampling data sequence for each of the source terminal station devices.

  One embodiment of the present invention relates to the eleventh embodiment, a wireless device including a plurality of antenna elements, a baseband device that controls the wireless device connected to the wireless device via a wired line or a wireless line, A wireless communication method in a wireless communication system, comprising: a terminal station device that wirelessly communicates with a device, wherein the baseband device generates a sampling data sequence of a baseband signal from data to be transmitted to the terminal station device. A processing step, wherein the baseband device obtains a transmission weight vector on a time axis used for transmission to the terminal station device, and the baseband device transmits the transmission weight vector and the A first transmitting step of transmitting a sampling data sequence; A first receiving step of receiving the transmission weight vector and the sampling data sequence from the command device, and the wireless device sequentially multiplies the sampling data sequence by the transmission weight vector for each sampling data, and calculates a multiplication result. Performing a digital / analog conversion of the sampling data corresponding to the vector component, and transmitting a signal based on the analog signal obtained by the conversion as a wireless signal from the corresponding antenna element. .

  One embodiment of the present invention relates to the eleventh embodiment, a wireless device including a plurality of antenna elements, a baseband device that controls the wireless device connected to the wireless device via a wired line or a wireless line, A wireless communication method in a wireless communication system, comprising: a terminal station device that wirelessly communicates with a device, wherein the baseband device obtains a reception weight vector on a time axis used by the wireless device for reception from the terminal station device. A receiving weight acquiring step, a third transmitting step in which the baseband device transmits the receiving weight vector to the wireless device, and a second transmitting step in which the wireless device receives the receiving weight vector from the baseband device. Receiving step, the radio apparatus transmits a signal transmitted from the terminal station apparatus to each of the antenna elements Wireless receiving step of generating a received sampling signal vector sequence that is a sampling data sequence of a baseband signal corresponding to each of the antenna elements from the received signal, and the wireless device, the received sampling signal vector sequence A multiplication step of sequentially multiplying the reception weight vector received from the baseband device by the sampling data for each sampling data, and the wireless device transmits one system of sampling data sequence generated in the multiplication step to the baseband device And a reception signal processing step in which the baseband device reproduces data transmitted by the terminal station device from the one-system sampling data sequence received from the wireless device. It is.

  One embodiment of the present invention relates to the eleventh embodiment, which includes a wireless device including a plurality of antenna elements, a baseband device that controls the wireless device, and a terminal station device that wirelessly communicates with the wireless device. A wireless communication method in a communication system, wherein the baseband device instructs the wireless device to calculate a correlation value and a reception weight vector, and the wireless device receives an instruction from the baseband device. Based on a signal when a known signal transmitted from the terminal station device is received by each of the antenna elements, any one of the plurality of antenna elements is used as a reference antenna element, and received by the reference antenna element. The correlation value over a predetermined number of samples between the known signal obtained and the known signal received by another antenna Calculating a reception weight vector based on the correlation value calculated in the correlation calculation step, a reception weight calculation step of calculating the reception weight vector based on the correlation value, and a signal transmitted from the terminal station device by the wireless device. Receiving via each of the antenna elements, a radio receiving step of generating a received sampling signal vector sequence which is a sampling data sequence of a baseband signal corresponding to each of the antenna elements from the received signal, and the radio apparatus, A multiplication step of sequentially multiplying the reception weight vector calculated based on the correlation value calculated by the correlation calculation step with respect to a reception sampling signal vector sequence for each sampling data; and Sampling of one generated system A second transmission step of transmitting a data sequence to the baseband device; and a reception process in which the baseband device reproduces data transmitted by the terminal station device from the one-system sampling data sequence received from the wireless device. And a signal processing step.

  According to the present invention, while realizing band expansion in a situation where direct waves are dominant in a line-of-sight environment, by mounting a large number of antenna elements on the wireless device side, the characteristics of wireless communication between the wireless device and the terminal station device can be improved. While improving, it is possible to suppress an increase in the amount of transmission information between the wireless device and the baseband device while integrating the basic functions of the wireless communication into the baseband device.

FIG. 3 is a diagram illustrating an outline of an example of MIMO transmission. It is a conceptual diagram of the example of an eigenmode transmission. It is a figure showing the outline of an example of a large-scale antenna system. FIG. 4 is a diagram illustrating an example of an impulse response in a line-of-sight environment and a non-line-of-sight environment. FIG. 2 is a schematic block diagram illustrating an example of a configuration of a base station device in a multi-user MIMO system. FIG. 2 is a schematic block diagram illustrating an example of a configuration of a transmission unit in a base station device in a multi-user MIMO system. FIG. 3 is a schematic block diagram illustrating an example of a configuration of a receiving unit in a base station device in a multi-user MIMO system. FIG. 9 is a diagram illustrating an example of distribution characteristics of absolute values of singular values for each channel matrix. FIG. 3 is a diagram illustrating an example of asymmetry of channel information between an uplink and a downlink. FIG. 4 is a diagram illustrating an outline of an example of calibration. FIG. 7 is a diagram illustrating an example of calibration of a plurality of channel matrices. It is a figure which shows the outline | summary of a function allocation of a mobile fronthaul and a mobile backhaul. FIG. 11 is a diagram showing an outline of a function allocation of a mobile fronthaul and a mobile backhaul in the related art when using a multi-element antenna. FIG. 3 is a diagram illustrating a configuration example of a mobile midhaul. FIG. 2 is a diagram illustrating an example of a linear array in which 100 antenna elements of a base station device are arranged at equal intervals in the first embodiment. FIG. 4 is a diagram illustrating an example of a linear array in which 100 antenna elements of the base station device are arranged in four groups of 25 each in the first embodiment. FIG. 2 is a schematic block diagram illustrating an example of a configuration of a base station device according to the first embodiment. FIG. 2 is a schematic block diagram illustrating an example of a configuration of a first transmission signal processing unit of the base station device according to the first embodiment. FIG. 3 is a schematic block diagram illustrating an example of a configuration of a first reception signal processing unit of the base station device according to the first embodiment. FIG. 2 is a schematic block diagram illustrating an example of a configuration of a terminal station device according to the first embodiment. FIG. 2 is a schematic block diagram illustrating a first example of a configuration of a transmission unit of the terminal station device according to the first embodiment. FIG. 2 is a schematic block diagram illustrating a first example of a configuration of a receiving unit of the terminal station device according to the first embodiment. FIG. 4 is a schematic block diagram illustrating a second example of the configuration of the receiving unit of the terminal station device according to the first embodiment. FIG. 3 is a schematic block diagram illustrating an example of a device configuration of a second reception signal processing unit of the base station device or a second reception signal processing circuit of the terminal station device according to the first embodiment. FIG. 4 is a schematic block diagram illustrating a second example of the configuration of the transmission unit of the terminal station device according to the first embodiment. 5 is a flowchart illustrating an outline of a signal processing flow at the time of signal transmission in the first embodiment. 4 is a flowchart illustrating an outline of a first example of a signal processing flow at the time of signal reception in the first embodiment. 5 is a flowchart illustrating an outline of a second example of a signal processing flow at the time of signal reception in the first embodiment. FIG. 11 is a diagram illustrating a first example of a case in which four parabolic antennas are mounted on a base station device and 16 antenna elements are mounted on a terminal station device in a linear array according to the second embodiment. It is a figure in the 2nd Embodiment which shows the example of the path difference for every antenna element on the terminal station apparatus side from a certain parabolic antenna on the base station apparatus side. FIG. 11 is a diagram illustrating a second example of a case in which four parabolic antennas are mounted on a base station device and 16 antenna elements are mounted on a terminal station device in a linear array according to the second embodiment. It is a figure in the 3rd Embodiment which shows the 1st example which realizes transmission and reception of a signal from a control base station device to a satellite base station device by a radio line. It is a figure in the 3rd Embodiment which shows the 2nd example which realizes transmission and reception of a signal from a control base station device to a satellite base station device by a radio line. It is a figure in the 3rd Embodiment which shows the outline of the signal processing between a control base station apparatus and a satellite base station apparatus, and the signal processing between a satellite base station apparatus and a terminal station apparatus. FIG. 13 is a diagram illustrating a circuit configuration of a satellite base station device according to a third embodiment. 14 shows an outline of a configuration of a wireless communication system according to a fourth embodiment. It is a figure showing the outline of signal processing in the train moving cell using a plurality of virtual transmission paths corresponding to the 1st singular value in a 4th embodiment. FIG. 14 is a diagram illustrating an outline of offline signal processing performed after acquiring coordinate information and channel information in a fourth embodiment. It is a figure showing the outline of another signal processing in the train moving cell using a plurality of virtual transmission paths corresponding to the 1st singular value in a 4th embodiment. It is a figure showing the outline of another signal processing of off-line performed after acquiring coordinate information and channel information in a 4th embodiment. FIG. 14 is a diagram illustrating an example of signal processing when a transmission / reception weight vector is acquired from coordinate information during service operation in the wireless communication system according to the fourth embodiment. It is a block diagram showing the example of composition of the base station device in a 4th embodiment. It is a block diagram showing the example of composition of the terminal station unit in a 4th embodiment. 15 is a graph illustrating a function F (α) according to the fifth embodiment. It is a figure explaining the removal method of the interference signal in a 5th embodiment. It is a flow chart which shows processing of transmission-and-reception weight acquisition of time axis beamforming in the radio communications system of a 5th embodiment. It is a figure showing the example of composition of the 1st transmission signal processing part with which the base station device in a 5th embodiment is provided. It is a block diagram showing the example of composition of the 1st received signal processing part of the base station device in a 5th embodiment. FIG. 21 is a block diagram illustrating an example of a circuit configuration of a transmission unit included in a terminal station device according to the fifth embodiment. It is a block diagram showing the example of composition of the receiving part of the terminal station unit in a 5th embodiment. FIG. 3 is a diagram illustrating an outline of a reception signal received by a plurality of antenna elements. FIG. 3 is a diagram illustrating an outline of channel estimation using limited subcarriers. It is a flow chart which shows processing operation of an approximate solution of channel matrix acquisition in a 6th embodiment. 9 is a flowchart illustrating a processing operation of an approximate solution method of a channel matrix using a plurality of antennas. 9 is a flowchart illustrating a processing operation for obtaining an approximate solution of a channel matrix from a bidirectional channel estimation result. It is a figure showing the result of having evaluated distribution characteristics by simulation. FIG. 9 is a diagram illustrating gain characteristics in an approximate solution of transmission and reception weight vectors using an antenna distant from the center of gravity. FIG. 3 is a diagram illustrating an example of an antenna element used for transmitting a training signal when a linear array is used. FIG. 4 is a diagram illustrating an example of an antenna element used for transmitting a training signal when a two-dimensional antenna arrangement in a close-packed state is used. FIG. 8 is a diagram illustrating an example of a case where an antenna element for transmitting a training signal is added near the center of gravity of a linear array. It is a figure showing the example of linear interpolation of an approximate weight value. 11 is a flowchart illustrating an operation of a process of determining a complex phase offset in linear interpolation of an approximate weight value. It is a flowchart which shows the processing operation of the approximate solution of the weight vector corresponding to several 1st singular values. It is a figure which shows the apparatus structure which performs the simultaneous channel estimation between several small cells which exist in the same area. It is a figure showing the example of the frame composition in an 8th embodiment. It is a flowchart which shows the processing operation | movement for the communication start between the terminal station apparatus and the base station apparatus which newly entered into the area. 6 is a flowchart illustrating a transmission / reception signal processing operation of the base station device. 6 is a flowchart illustrating a transmission / reception signal processing operation of the terminal station device. FIG. 3 is a diagram illustrating a frame configuration serving as a precondition. FIG. 11 is a diagram illustrating an example in which a training signal is omitted and communication is performed using only a data payload. 15 is a flowchart showing another second reception weight matrix calculation processing operation. 9 is a flowchart illustrating another signal processing operation of time-axis beam forming. It is a figure in 10th Embodiment which shows the structure of the base station apparatus at the time of multi-user MIMO application, and a terminal station apparatus. FIG. 21 is a diagram illustrating a circuit configuration of a first transmission signal processing unit of a base station device when a multi-user MIMO is applied in a tenth embodiment. FIG. 25 is a diagram illustrating a circuit configuration of a first reception signal processing unit of a base station device when multi-user MIMO is applied in a tenth embodiment. It is a figure showing the outline of the function allotment of the mobile fronthaul using the virtual transmission way corresponding to the 1st singular value in an 11th embodiment. It is a figure showing the outline (downlink) of the function allotment of the mobile fronthaul using the virtual transmission path corresponding to a plurality of 1st singular values in an 11th embodiment. FIG. 39 is a diagram illustrating an outline (uplink) of function sharing of a mobile fronthaul using virtual transmission paths corresponding to a plurality of first singular values according to the eleventh embodiment. It is a figure showing the outline of the correlation detection circuit in an eleventh embodiment. FIG. 41 is a diagram illustrating an outline (uplink) of another mobile fronthaul function sharing using virtual transmission paths corresponding to a plurality of first singular values according to the eleventh embodiment. FIG. 39 is a diagram illustrating an overview (downlink) of function sharing when multi-user MIMO is applied to a mobile fronthaul using virtual transmission paths corresponding to a plurality of first singular values according to the eleventh embodiment. FIG. 41 is a diagram illustrating an overview (uplink) of function sharing when multi-user MIMO is applied to a mobile fronthaul using virtual transmission paths corresponding to a plurality of first singular values according to the eleventh embodiment. FIG. 4 is a diagram illustrating an example of a configuration in a case where a first signal processing unit and a second signal processing unit are applied to an access system in the embodiment.

An embodiment of the present invention will be described in detail with reference to the drawings.
The symbols in the following embodiments will be described.
N _SDM : number of spatial multiplexing.
N, M, N ', M': natural numbers.
i, j, m, n: serial numbers of antenna elements and the like (general integers).
k: Subcarrier number (frequency component number).
N _BS-Ant : The total number of antenna elements of the base station device.
_N'BS-Ant : Number of antenna elements provided in the first transmission signal processing unit or the first reception signal processing unit of the base station device.
N _MT-Ant : Number of antenna elements of the terminal station device.
N _Ant : Number of antenna elements of the base station device or the terminal station device.
N _SC : number of subcarriers.
N _FFT : number of points in FFT.
L: distance.
K: Rice coefficient.
λ _k : wavelength of the k-th subcarrier.
r _ji : the distance between the i-th antenna element on the transmitting side and the j-th antenna element on the receiving side.
h _ji , h ′ _ji : channel information between the i-th antenna element on the transmitting side and the j-th antenna element on the receiving side (since it has frequency dependency, it is the k-th frequency component if necessary for the description) In some cases).
d: spacing between antenna elements.
Δd _mn : distance between the n-th antenna element and the m-th antenna element.
ΔL _m : path length difference of the m-th antenna element with respect to the first antenna element.
c: Speed of light (3 × 10 ⁸ m / s).
f _c: the center frequency of the radio signal [Hz].
f _k : frequency [Hz] of the k-th subcarrier of the baseband signal.
t: time.
W: bandwidth [Hz].
Δt: sampling period (Δt = 1 / W).
_{ｊ j} (t), Φ _j (t): received signal (sampling value) at the j-th antenna element at time t.
φ _j ^(k) (t): a received signal of the k-th subcarrier at the j-th antenna element at time t (a value focusing on a predetermined subcarrier in the sampling values).
η _k : Offset value of the k-th subcarrier in consideration of the complex phase of 2π period when using the least squares method.
u _m : m-th left singular vector.
v _m : m-th right singular vector.

[First Embodiment]
[Spatial multiplexing transmission using virtual transmission paths corresponding to a plurality of first singular values]
(Overview of Basic Principle According to First Embodiment)
As described with reference to FIG. 8, in an environment where the line-of-sight wave is dominant as shown in FIG. When a transmission path corresponding to a singular value equal to or larger than the value is used, Hi _{. i.} Communication is performed with the slight line gain obtained using the component _d . However, for example, when irradiating a limited small cell area below from a base station device installed on a wall surface of a building, the base station device side mounts an antenna having a high directivity gain. Further, in a situation where it is expected that a small antenna element capable of obtaining a directional gain by utilizing features such as a millimeter wave having a short wavelength is mounted on the terminal station apparatus side, the transmitting side and the receiving side It is expected that the multipath component will be very limited as compared with a microwave band system in which both are equipped with omni-directional antennas. Therefore, the characteristics of MIMO transmission when only the line of sight wave is considered will be summarized.

FIG. 15 is a diagram illustrating an example of a linear array in which 100 antenna elements of the base station device are arranged at equal intervals. In FIG. 15, reference numeral 40 denotes a wireless communication system, reference numeral 301 denotes a base station device, and reference numeral 302 denotes a terminal station device. In FIG. 15, 100 antenna elements of the base station apparatus 301 are mounted in a linear array. 100 antenna elements of the base station apparatus 301 is arranged at regular intervals over the length D _1. Further, 16 of the antenna element of the terminal station device 302 is equally spaced linear array shape over the length D _2.

FIG. 16 is a diagram illustrating an example of a linear array in which 100 antenna elements of the base station apparatus are arranged in groups of four every 25 antenna elements. 16, reference numeral 50 denotes a wireless communication system, reference numeral 303 denotes a base station device, reference numeral 302 denotes a terminal station device, reference numerals 304-1 to 304-4 denote first signal processing units, and reference numeral 305 denotes a second signal. A processing unit (strictly including other base station device functions such as an interface circuit, a MAC layer processing circuit, and a communication control circuit); In FIG. 16, the 100 antenna elements of the base station device 303 are divided into four groups for each of the 25 antenna elements. 25 antenna elements in the same group, in a very narrow interval as compared with the case of FIG. 15, over a short length D ₃ than the length D _1, are arranged in a linear array form.

In FIG. 16, the 100 antenna elements of the base station apparatus 303 are mounted in a linear array for each group (25 antenna elements). That is, the 100 antenna elements of the base station device 303 are mounted in a linear array for each first signal processing unit 304. The first signal processing units 304-1 to 304-4 form one directional beam for each group (25 antenna elements) by signal processing. Further, 16 of the antenna element of the terminal station device 302 is equally spaced linear array shape over the length D _2.

  Here, in each of the two cases shown in FIG. 15 and FIG. 16, the transmission characteristics in the case where four signal systems are spatially multiplexed (four multiplexed) and transmitted are compared. The transmission characteristics can be grasped by the gain of each transmission line shown in FIG. 2C, which corresponds to the square of the absolute value of the singular value obtained by performing the singular value decomposition of the channel matrix. In the case shown in FIG. 15, for example, assuming a downlink and assuming that the base station apparatus 301 is the transmitting station 11 and the terminal station apparatus 302 is the receiving station 12, the size of the channel matrix is 16 × 100. Perform singular value decomposition on this matrix.

On the other hand, in the case shown in FIG. 16, the following procedure is assumed, and its characteristics are grasped. First, the base station apparatus 303 is based on a 16 × 25 channel matrix (in the case of downlink) formed by 25 antenna elements of the base station apparatus 303 and 16 antenna elements of the terminal station apparatus 302. , And transmit using the first right singular vector. Specifically, the base station apparatus 303 includes a portion between each of the 25 antenna elements connected to the first signal processing sections 304-1 to 304-4 and the 16 antenna elements of the terminal station apparatus 302. Singular value decomposition is performed on the channel matrices H _{1 to} H ₄ . Equations (16) show the partial channel matrices H _{1 to} H ₄ .

Here, each of the partial channel matrices H _{1 to} H ₄ is a 16 × 25 matrix. Therefore, v _ij forming each right singular vector is a 25-dimensional vector, and represents the right singular vector corresponding to the j-th singular value in the ith group of the four groups of antennas. Similarly, u _ij forming each left singular vector is a 16-dimensional vector, and represents the left singular vector corresponding to the j-th singular value in the ith group of the four groups of antennas. Equation (17) shows the overall channel matrix between all the antenna elements of the base station apparatus 303 and the terminal station apparatus 302.

The transmission weight matrix _WTx here is shown in equation (18).

Here for convenience of notation, is used a Hermitian conjugate representation of the transmission weight matrix W _Tx, the size of the transmit weight matrix W _Tx itself is 100 × 4. As a result, the product of the overall channel matrix and the transmission weight matrix is shown in equation (19).

_Here, H _{i v i1} is the 16 × 1 matrix (column vector), consistent with the lambda _i u _i1 by equation (16). As a result, the overall size of the product of the overall channel matrix and the transmission weight matrix is 16 × 4. In general, the first left singular vectors of the partial channel matrices H _{1 to} H ₄ are not orthogonal to each other, so that upon reception, a reception weight for signal separation is formed and multiplied. However, when the first left singular vectors of the partial channel matrices H _{1 to} H ₄ are in an environment where they are substantially orthogonal to each other, the singular value decomposition of the matrix represented by the product of the entire channel matrix and the transmission weight matrix is performed. The square values of the absolute values of the four singular values approximately match the line gain of the transmission line in FIG. In this evaluation, it is assumed that each element of the channel matrix is represented by the following equation (20) using a free space propagation model that considers only the line of sight wave.

Here, r _ij represents the distance between the i-th antenna on the transmitting side and the j-th antenna on the receiving side, and λ represents the wavelength. Coefficients that are multiplied as coefficients in order to grasp the overall characteristics are omitted here for simplicity.

Therefore, in FIG. 16, in the case of L = 100 m, D ₁ = 12 m, D ₂ = 10 cm, D ₃ = 30 cm, and the frequency of 80 GHz, the characteristics of FIG. 15 installed with the same antenna aperture length are compared. Here, when the absolute value of the singular value is X as the line gain, the line gain is evaluated as 20 Log (X) [dB]. At this time, the gains of the four lines in FIG. 15 are −56.5 dB, −83.4 dB, −118.3 dB, and −157.2 dB, respectively. Are −62.5 dB, respectively. In the case of FIG. 15, only the line having the maximum gain corresponding to the first singular value in FIG. 2C has a large value, and the gain of the line corresponding to the remaining singular values is relatively small. Although depending on the values of parameters such as the antenna gain, only the line corresponding to the first singular value can be practically used. On the other hand, in the case of FIG. 16, it can be seen that the four transmission lines can be used almost equally. Here, the difference between the gain for the first singular value in FIG. 15 and the gain for the singular value in FIG. 16 is 6 dB, which is 1/100 to 25 in FIG. 4, which is equivalent to 10 Log (1/4) =-6 dB. In other words, the efficiency is reduced to 1/4 by dividing the antenna element group into four parts. However, the channel capacity due to the Shannon limit requires more transmission capacity when multiplexing four signal sequences than when improving SNR by 6 dB. It is overwhelmingly efficient in terms of increase.

  By setting the values of the parameters such as the transmission power and antenna gain, it is different if the received signal power of the reflected wave component is so strong that the line gain of the line corresponding to the singular value after the second singular value can be used sufficiently effectively. However, if the number of antenna elements is generally increased to compensate for a line gain that decreases with the use of a high frequency band such as a millimeter wave, the line gain of a line corresponding to a singular value after the second singular value is sufficient. In general, it is difficult to imagine the situation, and as for data transmission, the transmission capacity is increased in FIG. 16 in which transmission of four lines is performed in parallel, compared with the case of FIG. 15 in which transmission of substantially one line is performed. Can be seen as suitable for. As described above, it is effective to group the antennas and efficiently use the virtual transmission path corresponding to the first singular value in each group.

(Singular value decomposition and calibration)
Here, in the above equation (14), a case is considered where the amplification factor of the low noise amplifier or the high power amplifier does not differ between antenna elements (or can be approximated to be a constant value). In order to simplify the formula, the uplink channel matrix is H _UL , the downlink channel matrix is H _DL , the base station device side calibration matrix is C _BS , and the terminal station side device side calibration matrix is C _MT. , And the singular value decomposition results of the uplink channel matrix H _UL are shown in equations (21) and (22).

Here, it is assumed that the absolute values of the diagonal terms in equation (21) are all equal in each matrix. In this case, both formulas (21) are unitary matrices, and this term behaves as a rotation of the coordinate axes.
Here, the downlink channel matrix _HDL is represented using the calibration matrices C _BS , C _MT and the uplink channel matrix H _UL, and is substituted into the equation (22) and is represented by the equation (23).

Here, equations on both sides of the _{DUL on} the right side are shown in equations (24) and (25).

  If the absolute values of the components of the calibration matrix are approximately equal, the result of performing the singular value decomposition after performing calibration on the uplink channel matrix and calculating the transmission / reception weight and the singular value of the uplink channel matrix Equations (24) and (24) show that the results obtained by multiplying each component of the first right singular vector and the first left singular vector obtained by the decomposition by the calibration coefficient and performing the calibration process match. It can be seen from (25).

  In other words, if singular value decomposition is performed on the uplink channel matrix, calibration processing can be performed on each component of the obtained reception weight vector (or matrix) without performing singular value decomposition on the downlink. Thus, it can be seen that a desired transmission weight vector (or matrix) can be obtained. Therefore, the singular value decomposition needs to be performed only once.

(About the circuit configuration of the base station device in the present invention)
Hereinafter, a circuit configuration of the base station device 303 according to the first embodiment of the present invention will be described with reference to the drawings.
FIG. 17 is a schematic block diagram illustrating an example of the configuration of the base station device 70 in the MIMO system of the present invention. FIG. 16 illustrates a point-to-point type point-to-point type point-to-point communication with one base station apparatus 303 and one terminal station apparatus 302, but a plurality of terminal station apparatuses 302 naturally exist. It does not matter. In the transmission and reception of the signal in FIG. 16, when viewed from the subcarrier of interest, communication with only one terminal station device 302 is performed at the same time, and it is in the form of single-user MIMO transmission. Is selected. If OFDMA is used for access control, a different terminal station device 302 may be assigned to each subcarrier, but if attention is paid to each subcarrier, assignment to one terminal station device 302 is limited. Further, when the terminal is fixed in the point-to-point type communication, the process of selecting the communication partner terminal station device 302 in the scheduling becomes unnecessary. In the first embodiment of the present invention, the present invention can be used in a variation of the form of multi-user MIMO transmission of point-to-multipoint type point-to-multipoint communication. However, in the following description, regardless of such variation, , General single-user MIMO transmission will be described. Also, in the following description, for the sake of simplicity, a wideband system is assumed and signal processing for each subcarrier in the frequency axis such as OFDM or SC-FDE is described. (For example, a narrow band single carrier system).

  As shown in FIG. 17, a base station device 70 corresponding to the base station device 303 includes a first transmission signal processing unit 181-1 to 181-4, a second transmission signal processing unit 71, and a first reception signal processing unit 71. It includes a signal processing unit 185-1 to 185-4, a second received signal processing unit 75, an interface circuit 77, a MAC (Medium Access Control) layer processing circuit 78, and a communication control circuit 120. The MAC layer processing circuit 78 has a scheduling processing circuit 781.

  The base station device 70 inputs and outputs data to and from an external device or a network via the interface circuit 77. The interface circuit 77 detects data to be transferred on the wireless line among the input data, and outputs the detected data to the MAC layer processing circuit 78. The MAC layer processing circuit 78 performs a process related to the MAC layer in accordance with an instruction from the communication control circuit 120 that performs management control of the operation of the entire base station device 70. Here, the processing related to the MAC layer includes conversion between data input / output by the interface circuit 77 and data transmitted / received on the wireless line, addition of header information of the MAC layer, and the like. During this process, the scheduling processing circuit 781 performs various scheduling processes of the terminal station device 302 that performs spatial multiplexing. The scheduling processing circuit 781 outputs a scheduling result to the communication control circuit 120. In the MIMO transmission, signals of a plurality of signal sequences are spatially multiplexed and transmitted at one time. Therefore, a plurality of signal sequences are output from the MAC layer processing circuit 78 to the second transmission signal processing unit 71.

  Although the operation of the second transmission signal processing unit 71 will be described later, basically, a predetermined modulation process is performed on a plurality of series of signals from the MAC layer processing circuit 78, and if necessary, some precoding process (on the transmission side). (E.g., equalization processing and signal separation processing), and outputs the result to the first transmission signal processing units 181-1 to 181-4. At this time, regardless of the use of OFDM or SC-FDE, when the first transmission signal processing units 181-1 to 181-4 perform signal processing on the frequency axis, the second transmission signal processing unit A signal on the frequency axis is generated in 71 and output to the first transmission signal processing units 181-1 to 181-4. When performing signal processing on the time axis in the first transmission signal processing unit as in a fifth embodiment described later, a configuration may be adopted in which a signal on the time axis is output. The first transmission signal processing units 181-1 to 181-4 are each connected to a plurality of antennas as shown in FIG. At this time, a signal obtained by multiplying a transmission weight vector corresponding to the first singular value in the antenna element group grouped for each of the first transmission signal processing units 181-1 to 181-4 (strictly speaking, For example, in the case of OFDM, a signal obtained by combining signals of the respective subcarriers is converted into a time-axis component, and a signal obtained by up-converting the signal into a radio frequency is transmitted from each antenna.

  Next, at the time of reception, signals received by a plurality of antennas connected to each of the first reception signal processing units 185-1 to 185-4 (more precisely, downconvert a received radio frequency signal to a baseband signal) For example, in the case of OFDM, this time-axis signal is converted to a frequency-axis signal by FFT) by a predetermined reception weight vector, and is converted into one complex scalar quantity for each subcarrier. To the received signal processing unit 75. In this example, the second received signal processing unit 75 refers to the four received signal sequences, first performs channel estimation for each subcarrier using a known training singing signal added to the head of the received signal, and A × 4 MIMO channel matrix is obtained for each subcarrier. A reception weight matrix is calculated based on the channel matrix, and a process of detecting a transmitted signal is performed based on the obtained reception weight matrix. For example, a ZF (Zero Forcing) type inverse matrix is used, or an MMSE (Maximum Mean Square Error) type reception weight matrix is used. If there is room in the signal processing, MLD (Maximum Likelihood Detection), simple MLD using QR decomposition (QR-MLD), or the like may be used. The signal detected in the reception signal processing is output to the MAC layer processing circuit 78, performs predetermined MAC layer processing, and is output to the network via the interface circuit 77.

FIG. 18 is a schematic block diagram illustrating an example of a configuration of the first transmission signal processing unit 181 in the base station device 70 according to the first embodiment of the present invention. As shown in FIG. 18, the first transmission signal processing unit 181 includes a first transmission signal processing circuit 111 and an IFFT (Inverse Fast Fourier Transform) & GI (Guard Interval: guard interval) providing circuit 813. −1 to 813-N ′ _BS-Ant , D / A (digital / analog) converters 814-1 to 814- (N ′ _BS-Ant ), local oscillator 815, and mixers 816-1 to 816- ( N ′ _BS-Ant ), filters 817-1 to 817- (N ′ _BS-Ant ), high power amplifiers (HPA) 818-1 to 818- (N ′ _BS-Ant ), and antenna element 819-1 819- (N ′ _BS-Ant ) and a first transmission weight processing unit 130. N ′ _BS-Ant is a value obtained by dividing the total number of antenna elements of the base station device 70 by the number of spatial multiplexing (= N _BS−Ant / N _SDM ). _N'BS-Ant simply represents the number of a plurality of antenna elements provided in one first transmission signal processing unit 181. The first transmission signal processing circuit 111 is connected to the second transmission signal processing section 71 shown in FIG. Also, the first transmission signal processing circuit 111 and the first transmission weight processing unit 130 are connected to the communication control circuit 120 via the second transmission signal processing unit 71 shown in FIG. In the example of FIG. 17, four first transmission signal processing units (181-1 to 181-4) are connected to the base station device 70, and description will be given below focusing on one of them.

The first transmission weight processing unit 130 includes a first channel information acquisition circuit 131, a first channel information storage circuit 132, and a first transmission weight calculation circuit 133. Here, the suffixes (N ′ _BS-Ant ) of the circuits from the IFFT & GI adding circuits 813-1 to 813- (N ′ _BS-Ant ) to the antenna elements 819-1 to 819- (N ′ _BS-Ant ) are as follows: , The number of antenna elements included in the first transmission signal processing unit 181 of the base station apparatus 70.

In the present invention according to the first embodiment, since a plurality of N _SDM (= 4) signals are spatially multiplexed and transmitted to one terminal station device 302, a plurality of signal sequences are transmitted from the MAC layer processing circuit 78. A transmission signal is input to each of the first transmission signal processing units 181-1 to 181-4 via the second transmission signal processing unit 71. When data to be transmitted to the destination terminal station device 302 is input from the MAC layer processing circuit 78, the second transmission signal processing unit 71 generates a wireless packet to be transmitted over a wireless line and performs modulation processing. Here, for example, if the OFDM modulation method is used, the signal of each signal sequence is subjected to modulation processing for each subcarrier. The signal subjected to the modulation processing is subjected to a precoding processing as necessary. Here, the precoding processing may be multiplication of a transmission weight matrix for suppressing signal leakage between virtual transmission paths corresponding to a plurality of first singular values. Alternatively, such precoding processing may not be performed.
Signal generated _{N SDM} system in this manner, is input into the first transmission signal processing unit 181-1～181-4. Each first transmission signal processing unit 181-1～181-4, digital baseband signal input #i which is input (i _{is, 1 to N SDM)} is input to the first transmission signal processing circuit 111-i You. The first transmission signal processing circuit 111 basically performs multiplication of transmission weights and signal processing of the remaining physical layers. For example, if the OFDM modulation method is used, the input modulation-processed baseband signal is multiplied by the transmission weight for each subcarrier. The signal multiplied by the transmission weight corresponding to each of the antenna elements 819-1 to 819- (N'BS _-Ant ) is subjected to the remaining signal processing as necessary, and the IFFT & GI adding circuits 813-1 to 813- ( N ′ _BS-Ant ) to convert the signal on the frequency axis into a signal on the time axis. The converted signal is further subjected to processing such as insertion of guard intervals and waveform shaping between OFDM symbols (or between blocks of block transmission in the case of SC-FDE (Single-Carrier Frequency Domain Equalization)). 'for each _(BS-Ant, D / a converter 814-1~814- (N 1~819- N)' are converted by the _BS-Ant) from digital sampling data into an analog baseband signal. Further, each analog signal is multiplied by the local oscillation signal input from the local oscillator 815 by the mixers 816-1 to 816- (N'BS _-Ant ), and is up-converted into a radio frequency signal. Here, since the up-converted signal includes a signal outside the band of the channel to be transmitted, the signal outside the band is removed by the filters 817-1 to 817- (N'BS _-Ant ) and transmitted. Generate an electrical signal to power. The generated signal is 'is amplified by _(BS-Ant, antenna elements 819-1~819- (N high-power amplifier 818-1~818- N)' is transmitted from the _BS-Ant).

  Note that the transmission weight vector multiplied by the first transmission signal processing circuit 111 is obtained from the first transmission weight calculation circuit 133 provided in the first transmission weight processing unit 130 during signal transmission processing. In the first transmission weight processing section 130, the first channel information acquisition circuit 131 passes the channel information acquired by the first reception signal processing sections 185-1 to 185-4 via the communication control circuit 120 (strictly). Is also separately obtained by the second reception signal processing unit 75 and the second transmission signal processing unit 71), and is stored in the first channel information storage circuit 132 while being sequentially updated. At the time of signal transmission, the first transmission weight calculation circuit 133 reads channel information corresponding to the destination terminal station device from the first channel information storage circuit 132 in accordance with an instruction from the communication control circuit 120, and reads the read channel information. The transmission weight vector is calculated based on

  There are several variations in a channel estimation method and a transmission / reception weight calculation method when a virtual transmission path corresponding to the first singular value is used, and details of a method for efficiently obtaining the weight will be described later. As an example, as the transmission weight vector, for example, singular value decomposition may be performed on the acquired channel matrix, and the resulting first right singular vector may be used.

Or, focusing on one of the central portions of the antennas on the side of the terminal station device 302, the one antenna and the reception signal processing unit (any one of 185-1 to 185-4) of the base station device 70 Each component of the transmission weight vector may be obtained by equation (7) based on the channel vector between the provided antenna elements 819-1 to 819-4 and N ′ _BS-Ant (weight of maximum ratio combining). Alternatively, all the absolute values may be given constant with respect to the value given by Expression (7) (weight of equal gain combining).

  Or, when the transmitting side multiplies a plurality of antenna elements by a predetermined transmitting weight vector and transmits a signal, the signal is effectively transmitted by one transmitting element even though it is actually transmitted from a plurality of transmitting antennas. Is equivalent to the one transmitted from the virtual antenna element, a training signal is transmitted from this one virtual antenna element by multiplying by a predetermined transmission weight vector, and the one virtual antenna element and each A vector of the channel information between the antenna and the receiving antenna may be obtained, and each component of the receiving weight vector may be obtained by the equation (7) based on the vector (weight of the maximum ratio combining) or by the equation (7). All absolute values may be made constant with respect to a given value (weight of equal gain combining).

  If the channel vector at the time of reception is known, it is possible to acquire uplink channel information by an implicit feedback technique. Based on the uplink channel vector obtained in this way, a transmission weight vector May be similarly calculated. Similarly, a transmission weight vector may be calculated based on an uplink reception weight vector by an implicit feedback method of directly performing a calibration process on the received weight vector.

  The first transmission weight calculation circuit 133 outputs the transmission weight vector calculated in this way to the first transmission signal processing circuit 111. Further, the communication control circuit 120 manages control related to the entire communication, such as management of the terminal station apparatus as a destination and overall timing control. The communication control circuit 120 outputs information indicating the destination terminal station device or the like to the first transmission weight processing unit 130 that performs the signal processing related to the above-described transmission weight calculation.

  In the above description, in the first channel information acquisition circuit 131, the channel information acquired by the first received signal processing units 185-1 to 185-4 is acquired via the communication control circuit 120. , The channel information is not required to be updated frequently in a high-altitude environment where channel-time fluctuations can be ignored. The first channel information acquisition circuit 131 acquires channel information in advance, for example, before starting service operation, and further stores a transmission weight vector calculated from the value of the channel information (in FIG. However, in this case, it is realized by a configuration in which a “transmission weight storage circuit” is mounted and recorded), and it may be used repeatedly. As an intermediate between the two, the transmission weight vector is updated based on the sequentially acquired channel information while the transmission weight vector is basically read from the first transmission weight storage circuit. It is also possible to adopt a configuration in which the transmission weight vector is updated at a predetermined time interval based on this.

FIG. 19 is a schematic block diagram illustrating an example of a configuration of the first received signal processing unit 185 in the base station device 70 according to the first embodiment of the present invention. As shown in FIG. 19, the first received signal processing unit 185 includes antenna elements 851-1 to 851- (N ′ _BS-Ant ) and low noise amplifiers (LNA) 852-1 to 852- (N ′ _{BS- Ant} ), a local oscillator 853, mixers 854-1 to 854- (N'BS _-Ant ), filters 855-1 to 855- (N'BS _-Ant ), and A / D (analog / digital) conversion. Units 856-1 to 856- (N'BS _-Ant ), FFT (Fast Fourier Transform) circuits 857-1 to 857- (N'BS _-Ant ), and first reception weight processing unit 160 And a first reception signal processing circuit 158. The first reception signal processing circuits 158-1 to 158-N _SDM (= 4) are connected to the second reception signal processing unit 75 shown in FIG. Further, the first reception signal processing circuits 158-1 to 158-N _SDM (= 4) and the first reception weight processing section 160 are connected via the second reception signal processing section 75 shown in FIG. The communication control circuit 120 is connected. The first reception weight processing section 160 includes a first channel information estimation circuit 161 and a first reception weight calculation circuit 162. As in the description of the first transmission signal processing unit 181, four first reception signal processing units (185-1 to 185-4) are connected to the base station device 70 in the example of FIG. However, an explanation focusing on one of them will be described below.

First, the signals received by the antenna elements 851-1 to 851- (N'BS _-Ant ) are amplified by the low noise amplifiers 852-1 to 852- (N'BS _-Ant ). The amplified signal and the local oscillation signal output from the local oscillator 853 are multiplied by mixers 854-1 to 854- (N'BS _-Ant ), and the amplified signal is converted from a radio frequency signal to a baseband signal. Downconverted. Since the down-converted signal includes a signal outside the frequency band to be received, the out _- of-band component is removed by the filters 855-1 to 855- (N'BS _-Ant ). The signals from which the out-of-band components have been removed are converted to digital baseband signals by A / D converters 856-1 to 856- (N'BS _-Ant ). For example, in the case of OFDM, the digital baseband signal is all input to the FFT circuits 857-1 to 857-(N ′ _BS-Ant ), and a predetermined symbol timing determined by a timing detection circuit not shown here is used. Converts the signal on the time axis into a signal on the frequency axis (separates into signals of each subcarrier). The signal separated into each subcarrier is input to the first reception signal processing circuit 158 and also to the first channel information estimation circuit 161. Although the circuit for OFDM symbol timing detection is omitted in FIG. 19, the symbol timing can be ascertained by any existing method.

The first channel information estimating circuit 161 uses an antenna element of each terminal station apparatus 302 based on a known signal for channel estimation (a preamble signal or the like added to the head of a radio packet) separated into subcarriers, Channel information between each antenna element 851 of the base station device 70 is estimated for each subcarrier, and the estimation result is output to the first reception weight calculation circuit 162. The first reception weight calculation circuit 162 calculates a reception weight vector to be multiplied for each subcarrier based on the input channel information. At this time, the reception weights that combine the signals received by the antenna elements 851-1 to 851- (N ′ _BS-Ant ) are the first reception signal processing units 185-1 to 185-N _SDM (= 4). The first received signal processing units 185-1 to 185-N _SDM are individually calculated.

In the first reception signal processing circuit 158, a signal for each subcarrier input from the FFT circuits 857-1 to 857- (N'BS _-Ant ) (more precisely, signals from a plurality of antenna elements are used as elements). The reception signal vector) is multiplied by a reception weight (accurately, a reception weight vector having reception weights corresponding to a plurality of antenna elements) input from the first reception weight calculation circuit 162, and the result of the multiplication is obtained. The signals received by the antenna elements 851-1 to 851- (N ′ _BS-Ant ) are added and synthesized for each subcarrier. The first reception signal processing circuit 158 outputs the added and combined signal to the second reception signal processing unit 75. Note that the addition synthesis here means addition after multiplication of each component of the vector in the vector product for each subcarrier, and the multiplication of the reception signal and the reception weight and the total addition and synthesis of the result are mathematically vector vectors. Corresponds to product processing.

  Note that the reception weight vector multiplied by the first reception signal processing circuit 158 is obtained from the first reception weight calculation circuit 162 provided in the first reception weight processing unit 160 at the time of signal reception processing. In the first reception weight processing section 160, the first reception weight calculation circuit 162 calculates a reception weight vector using the channel information acquired in the first channel information estimation circuit 161. For example, if the terminal station apparatus 302 is transmitting a signal to the plurality of antenna elements 851 of the first received signal processing unit 185 through a virtual transmission path corresponding to the first singular value, the terminal station apparatus 302 Is a signal transmitted to each first received signal processing unit 185 using one virtual antenna element using a transmission weight for a virtual transmission path corresponding to the first singular value. , Channel information on the receiving side between the one antenna and the plurality of antenna elements 851 of the first received signal processing unit 185 is obtained, and each component of the received weight vector is calculated with respect to this channel vector by Expression (7). It may be obtained (weight of maximum ratio combining), or may be given with all absolute values being constant with respect to the value given by equation (7) (weight of equal gain combining). Alternatively, the channel matrix between the antenna element included in the terminal station device 302 and each of the plurality of antenna elements 851 of the first received signal processing unit 185 included in the base station device 70 is obtained by performing singular value decomposition. Each of the one left singular vectors may be used as a reception weight vector.

  The first reception weight calculation circuit 162 outputs the reception weight vector calculated in this way to the first reception signal processing circuit 158. In addition, the communication control circuit 120 manages control related to overall communication, such as management of a source station and overall timing control.

  In the above description, the reception weights are sequentially calculated using the channel information acquired by the first channel information estimation circuit 161. There is no need to update the channel information. The first channel information estimating circuit 161 acquires channel information in advance, for example, before starting service operation, and stores a reception weight vector calculated from the value of the channel information (not shown in the figure). However, in this case, the "first reception weight storage circuit" is implemented and recorded after the first reception weight calculation circuit 162), and this may be repeatedly used. In this case, the communication control circuit 120 outputs information indicating the terminal station device or the like of the transmission source to the first reception weight storage circuit that outputs the reception weight. As an intermediate between the two, the reception weight vector is updated based on the sequentially acquired channel information while the reception weight vector is basically read out from the first reception weight storage circuit. It is also possible to adopt a configuration in which the reception weight vector is updated at a predetermined time interval based on this.

  Note that the second reception signal processing unit 75 in FIG. 17 multiplies the reception weight vectors from the above-described first reception signal processing units 185-1 to 185-4 by the reception weight vectors and aggregates the reception signals into one system. Although they are input, they are substantially equivalent to a received signal of a 4 × 4 MIMO channel, and can process a signal sequence spatially multiplexed by a received signal detection process of the related art. Specifically, 4 × 4 signals when four sets of virtual antennas composed of a plurality of antennas on the receiving side (base station apparatus 70 side) receive four signal sequences transmitted on the transmitting side. For each of the MIMO channels, a 4 × 4 channel matrix is acquired for each subcarrier using a known training signal for channel estimation added to the head of the received signal. A reception weight matrix is calculated based on the channel matrix, and a process of detecting a transmitted signal is performed based on the obtained reception weight matrix. For example, a ZF (Zero Forcing) type inverse matrix is used, or an MMSE (Maximum Mean Square Error) type reception weight matrix is used. If there is room in the signal processing, MLD (Maximum Likelihood Detection), simple MLD using QR decomposition (QR-MLD), or the like may be used. Further, in the signal detection processing here, for example, a configuration may be adopted in which the final signal detection is performed by, for example, once performing a soft decision on the received signal, performing deinterleaving processing as necessary, and then performing error correction processing. good. The signal detected in the reception signal processing is output to the MAC layer processing circuit 78, performs predetermined MAC layer processing, and is output to the network via the interface circuit 77.

  Further, the MAC layer processing circuit 78 performs processing related to the MAC layer (for example, conversion between data input / output to / from the interface circuit 77 and data transmitted / received on the wireless line, ie, wireless packets, termination of MAC layer header information. Do). In the case of the point-to-point type communication, the scheduling processing circuit 781 is substantially unnecessary, but in the case of performing the point-to-multipoint type communication with the plurality of terminal station devices 302, Various scheduling processes for selecting the terminal station device 302 with which communication is performed are performed, and the scheduling result is output to the communication control circuit 120. The received data processed by the MAC layer processing circuit 78 is output to an external device or a network via the interface circuit 77.

  Further, the communication control circuit 120 manages control related to the entire communication, such as management of the terminal station device 302 of the transmission source and overall timing control. In addition, information indicating the terminal station device or the like of the transmission source is input from the communication control circuit 120 to the first reception weight processing unit 160 that performs the signal processing related to the above-described calculation of the reception weight.

As for signal reception, as in the case of transmission, in a wideband system using the OFDM modulation method or the SC-FDE method, the above-described multiplication of the reception weight is performed for each subcarrier. That is, the FFT circuits 857-1 to 857-(N ′ _BS-Ant ) perform FFT on the signals output from the A / D converters 856-1 to 856-(N ′ _BS-Ant ) and apply the signals to each subcarrier. The signal processing in the first channel information estimation circuit 161 and the reception signal processing in the first reception signal processing circuit 158 are performed for each of the separated subcarriers.

The above is the description of the base station device 70 according to the first embodiment of the present invention. What is important here is that the local oscillator 815 in the first transmission signal processing unit 181 is common to the mixers 816-1 to 816- (N'BS _-Ant ) in each antenna system in the same first transmission signal processing unit 181. On the other hand, the local oscillator 815 is not shared between different first transmission signal processing units 181. Similarly, the local oscillator 853 in the first reception signal processing unit 185 is shared by the mixers 854-1 to 854- (N'BS _-Ant ) in each antenna system in the same first reception signal processing unit 185. It is also important that the local oscillator 853 is not shared between the different first received signal processing units 185. As shown in FIG. 16, generally, the first transmission signal processing units 181-1 to 181-4 (the first signal processing units 304-1 to 304-4 in FIG. 16) are physically separated on the order of several meters. In a high frequency band such as a millimeter wave, the loss at the time of routing with a cable is very large at 10 dB or more per meter. Since a large number of antenna elements are connected to the first transmission signal processing unit 181 and the first reception signal processing unit 185, it is necessary to split the signal into a plurality of systems and input the signals to the mixers 816 and 854. Considering the decrease in the level due to this branching, the loss of the cable length on the order of several meters is not negligible, so it is closed in the individual first transmission signal processing unit 181 and the individual first reception signal processing unit 185 and locally It is effective that the oscillators 815 and 853 are shared and not shared between different first transmission signal processing units 181 and first reception signal processing units 185.

  Here, each antenna adjusts the phase of the transmission / reception signal for directivity control. However, the same first transmission signal processing units 181-1 to 181-4 (the first signal processing units in FIG. 16). Within 304-1 to 304-4), it is easy to always keep the phase relationship between the signals input from the respective local oscillators 815 or 853 constant. It is possible to determine what phase relationship should be multiplied by the transmission / reception weights in a part that does not depend on it. However, when the local oscillator 815 uses a plurality of asynchronous oscillators in the first transmission signal processing unit 181 (or the local oscillator 853 in the first reception signal processing unit 185), at least the first transmission signal When multiplying the transmission weights in the processing unit 181, it is necessary to adjust in consideration of the complex phase relationship between the plurality of local oscillators 815 (or 853). If this adjustment is neglected, the directivity control functions effectively. No longer. Care must be taken in this regard when designing the device. In the first embodiment of the present invention, the same first transmission signal processing unit 181 shares the local oscillator 815 for the above-described reason, and the same first reception signal processing unit 185 uses the local oscillator 853 for the above-described reason. However, since the signal separation between the four spatially multiplexed signal sequences can be performed on the receiving side, it is not necessary to perform complete signal separation on the transmitting side as in multi-user MIMO. .

  Although a plurality of signal sequences can be basically separated by signal processing on the receiving side in this way, for example, the phase between the first transmission signal processing units 181-1 to 181-4 is determined by channel feedback or the like. If the relationship is known, it is possible to multiply (ie, pre-transmit) the transmission weight matrix of the signal separation in advance on the transmission side. In this case, the second transmission signal processing section 71 of the base station device 70 multiplies the transmission weight matrix. In the calculation of the transmission weight matrix, the transmission weight matrix is obtained using a calibration process based on uplink channel information on the four signal sequences after multiplication by the reception weight vector obtained by the second reception signal processing unit 75. It does not matter. However, as described above, even in the downlink, the receiving side terminal station apparatus 302 can acquire the channel matrix from the training signal added to the transmission signal. The multiplication of the transmission weight matrix by the transmission signal processing unit 71 is not necessary.

(Circuit configuration of terminal station device 60 corresponding to terminal station device 302 in the present invention)
FIG. 20 is a schematic block diagram illustrating an example of a configuration of the terminal station device 60 corresponding to the terminal station device 302 according to the first embodiment of the present invention. As shown in FIG. 20, the terminal station device 60 includes a transmission unit 61, a reception unit 65, an interface circuit 67, a MAC (Medium Access Control) layer processing circuit 68, and a communication control circuit 121.

  The terminal station device 60 inputs and outputs data to and from an external device or a network via the interface circuit 67. The interface circuit 67 detects data to be transferred on the wireless line from the input data, and outputs the detected data to the MAC layer processing circuit 68. The MAC layer processing circuit 68 performs processing related to the MAC layer in accordance with an instruction from the communication control circuit 121 which performs management control of the operation of the terminal station device 60 as a whole. Here, the processing related to the MAC layer includes conversion between data input / output by the interface circuit 67 and data transmitted / received on the wireless line, that is, wireless packets, addition of MAC layer header information, and the like. In MIMO transmission, a signal sequence of a plurality of systems is output from the MAC layer processing circuit 68 to the transmission unit 61 in order to spatially multiplex and transmit a signal to one terminal station device 60.

FIG. 21 is a schematic block diagram illustrating an example of a configuration of the transmission unit 61 in the terminal station device 60 according to the first embodiment of the present invention. As illustrated in FIG. 21, the transmission unit 61 includes transmission signal processing circuits 811-1 to 811-N _SDM (N _SDM is an integer of 2 or more) and addition and synthesis circuits 812-1 to 812-N _MT-Ant (N _MT-Ant is an integer of 2 or more, IFFT (Inverse Fast Fourier Transform) & GI (Guard Interval: guard interval) providing circuits 813-1 to 813-N _MT-Ant , and D / A (digital) / Analog) converters 814-1 to 814-N _MT-Ant , local oscillator 815, mixers 816-1 to 816-N _MT-Ant , filters 817-1 to 817-N _MT-Ant , and high power amplifier (HPA) and _{818-1~818-N MT-Ant,} an antenna element _{819-1~819-N MT-Ant,} a first transmission weight processing unit 140 It is provided. A transmission signal processing circuit _{811-1~811-N SDM,} the first transmission weight processing unit 140, and a communication control circuit 121 shown in FIG. 20.

The first transmission weight processing unit 140 includes a channel information acquisition circuit 141, a channel information storage circuit 142, and a first transmission weight calculation circuit 143. Here, _{N SDM} subscripts of the transmission signal processing circuit _{811-1~811-N SDM} in FIG 21 represents a multiplex number for performing spatial multiplexing simultaneously. The subscript N _MT-Ant of the circuits from the addition / combination circuits 812-1 to 812-N _MT-Ant to the antenna elements 819-1 to 819-N _MT-Ant is the number of antenna elements included in the terminal station device 60. Represents N _MT-Ant is 16, for example.

In the first embodiment, since one terminal station device 60 spatially multiplexes and transmits a signal to the base station device 70, a plurality of signal sequences are input from the MAC layer processing circuit 68 to the transmission unit 61, plural systems signal sequences that are are inputted to the transmission signal processing circuit _{811-1~811-N SDM.} The transmission signal processing circuits 811-1 to 811-N _SDM transmit data (data inputs # 1 to #N _SDM ) to be transmitted to the destination base station apparatus 70 from the MAC layer processing circuit 68 via a wireless line (wireless). ), A modulation process is performed on the packet. Here, for example, if the OFDM modulation method is used, the signal of each signal sequence is subjected to modulation processing for each subcarrier. Further, the base station multiplies the modulated baseband signal by a transmission weight for each subcarrier. The signal multiplied by the transmission weight corresponding to each of the antenna elements 819-1 to 819 -N _MT-Ant is subjected to the remaining signal processing as necessary, and each transmission signal processing is performed as sampling data of the transmission signal in the baseband. The circuits 811-1 to 811-N are inputted from the _SDM to the addition / combination circuits 812-1 to 812-N _MT-Ant .

The signals input to the addition / combination circuits 812-1 to 812-NMT- _Ant are combined for each subcarrier. The synthesized signal is converted from a signal on the frequency axis to a signal on the time axis by IFFT & GI adding circuits 813-1 to 812-NMT- _Ant , and furthermore, a guard interval is inserted and an OFDM symbol interval (SC-FDE ( In the case of Single-Carrier Frequency Domain Equalization, processing such as waveform shaping between blocks of block transmission) is performed, and D / A converters 814-1 to 814-1 are provided for each of the antenna elements 819-1 to 819 -N _MT-Ant. The digital sampling data is converted to a baseband analog signal by 814-N _MT-Ant . Further, each analog signal is multiplied by a local oscillation signal input from the local oscillator 815 by the mixers 816-1 to 816-NMT _-Ant , and is up-converted into a radio frequency signal. Here, since the up-converted signal includes a signal in an out-of-band region of a channel to be transmitted, the out _- of-band signal is removed by the filters 817-1 to 817-NMT _-Ant and transmitted. Generate a signal. The generated signal is amplified by the high-power amplifier _{818-1~818-N MT-Ant,} and transmitted from the antenna elements _{819-1~819-N MT-Ant.}

In FIG. 21, after performing addition and synthesis of the signals of the respective subcarriers in the addition and synthesis circuits 812-1 to 812-NMT _-Ant , processes such as IFFT processing, insertion of a guard interval, and waveform shaping are performed. Performs these processes in the transmission signal processing circuits 811-1 to 811-N _SDM , and synthesizes the IFFT sampled signals on the time axis by the addition synthesis circuits 812-1 to 812-N _MT-Ant. , IFFT & GI imparting circuit omitted from _{813-1~813-N MT-Ant} (strictly, these include the transmission signal processing circuit _{811-1~811-N SDM)} may be. In this case, the rest of the signal processing necessary after transmission weight multiplying in the transmission signal processing circuit _{811-1~811-N SDM} refers IFFT processing, insertion of the guard interval, the process of waveform shaping or the like.

The transmission weight multiplied by the transmission signal processing circuits 811-1 to 811-N _SDM is obtained from the first transmission weight calculation circuit 143 provided in the first transmission weight processing unit 140 during the signal transmission processing. I do. In the first transmission weight processing unit 140, the channel information acquisition circuit 141 separately acquires the channel information acquired by the reception unit 65 via the communication control circuit 121, and stores the channel information while sequentially updating the acquired channel information. The information is stored in the circuit 142. At the time of signal transmission, the first transmission weight calculation circuit 143 reads channel information corresponding to the destination station from the channel information storage circuit 142 according to an instruction from the communication control circuit 121, and calculates the transmission weight based on the read channel information. I do. First transmission weight calculating circuit 143 outputs the calculated transmission weight to the transmission signal processing circuit _{811-1~811-N SDM.} It should be noted that, in normal communication, a terminal station device communicates with only a specific base station device. Therefore, in the above description, management regarding a terminal station device as a destination is explicitly shown. Of course, it is also possible to perform processing as one.

  Note that the feature of the present invention according to the first embodiment is that, in the calculation of the transmission weight, between the terminal station device 60 and the first transmission signal processing units 181-1 to 181-4 of the base station device 70, That is, a virtual transmission path corresponding to the first singular value is used. There are several variations in the channel estimation method and the transmission / reception weight calculation method when utilizing the virtual transmission path corresponding to the first singular value, and the method for efficiently obtaining this will be described in detail later. I do. For example, for each channel matrix in the uplink from the terminal station device 60 to the first transmission signal processing units 181-1 to 181-4 of the base station device 70, the first right singularity when the singular value decomposition is performed The vector may be used as the transmission weight vector. In this case, the first transmission weight calculation circuit 143 has a function of calculating the first right singular vector.

  Alternatively, when the base station apparatus 70 transmits a signal by multiplying each of the plurality of antenna elements of the first transmission signal processing units 181-1 to 181-4 by a predetermined transmission weight vector, Despite being transmitted from a plurality of transmission antennas, it is effectively equivalent to each of the first transmission signal processing units 181-1 to 181-4 transmitting from one virtual antenna element. It is. Therefore, a channel information vector between the one virtual antenna element and each of the receiving antennas of the terminal station device 60 is obtained, and the channel vector is subjected to a calibration process by an implicit feedback method. Channel information can be obtained. The first transmission weight calculation circuit 143 calculates a vector obtained by taking a complex conjugate of this channel vector as shown in Expression (7) based on the uplink channel vector obtained in this manner, or each component of the vector. May be used as the transmission weight vector.

  As an alternative to this, the first transmission weight calculation circuit 143 is provided between the terminal station device 60 and one antenna element in the first transmission signal processing units 181-1 to 181-4 of the base station device 70. A vector obtained by obtaining a channel vector of channel information on the receiving side and obtaining a complex conjugate of the channel vector as shown in equation (7), or a vector in which the absolute values of the components of the vector are all constant May be used as a transmission weight vector. Further, the communication control circuit 121 manages control related to the entire communication, such as the entire timing control.

FIG. 22 is a schematic block diagram illustrating an example of a configuration of the receiving unit 65 in the terminal station device 60 according to the first embodiment of the present invention. As illustrated in FIG. 22, the receiving unit 65 includes antenna elements 851-1 to 851-N _MT-Ant , low noise amplifiers (LNA) 852-1 to 852-N _MT-Ant , a local oscillator 853, and a mixer 854. -1 to 854-N _MT-Ant , filters 855-1 to 855-N _MT-Ant , A / D (analog / digital) converters 856-1 to 856-N _MT-Ant , and FFT (Fast Fourier) Transform: Fast Fourier Transform) circuits 857-1 to 857 -N _MT-Ant , reception signal processing circuits 145-1 to 145 -N _SDM, and a first reception weight processing unit 144. A reception signal processing circuit _{145-1~145-N SDM,} the first reception weight processing unit 144, and a communication control circuit 121 shown in FIG. 20. The first reception weight processing section 144 includes a first channel information estimation circuit 146 and a first reception weight calculation circuit 147.

First, signals received by antenna elements _{851-1~851-N MT-Ant} is amplified by the low noise amplifier _{852-1~852-N MT-Ant.} The amplified signal and a local oscillation signal output from the local oscillator 853 are multiplied by mixers 854-1 to 854-NMT _-Ant , and the amplified signal is down-converted from a radio frequency signal to a baseband signal. You. Since the down-converted signal includes a signal outside the frequency band to be received, the out _- of-band component is removed by the filters 855-1 to 855-NMT _-Ant . The signal from which the out-of-band component has been removed is converted into a digital baseband signal by A / D converters 856-1 to 856-NMT _-Ant . For example, in the case of using OFDM, the digital baseband signal is input to the FFT circuits 857-1 to 857 -N _MT-Ant , and the time base is determined at a predetermined symbol timing determined by a timing detection circuit not shown here. The above signal is converted into a signal on the frequency axis (separated into signals of each subcarrier). Each subcarrier is separated signals is inputted to the reception signal processing circuit _{145-1~145-N SDM,} is also input to the first channel information estimation circuit 146.

In the first channel information estimating circuit 146, the first transmission signal of the base station device 70 is based on a known signal for channel estimation (such as a preamble signal added to the head of a radio packet) separated into subcarriers. Channel of channel information between virtual antenna elements formed by transmission weight vectors of processing units 181-1 to 181-4 and antenna elements 851-1 to 851-N _{MT-Ant of} terminal station apparatus 60 The vector is estimated for each subcarrier, and the estimation result is output to first reception weight calculation circuit 147. The first reception weight calculation circuit 147 calculates the reception weight to be multiplied for each subcarrier based on the input channel information. As for the reception weight, for example, as described above, a ZF type pseudo inverse matrix is used, or an MMSE type reception weight matrix is used. At this time, the reception weight vector for synthesizing the signals received by the antenna elements 851-1 to 851-N _MT-Ant differs for each signal sequence, and the above-described ZF-type pseudo inverse matrix or MMSE-type reception is used. corresponds to a row vector, such as weight matrix, are input to the reception signal processing circuit _{145-1~145-N SDM} corresponding to the signal sequence to be extracted.

In the reception signal processing circuits 145-1 to 145-N _SDM , the signals for each subcarrier input from the FFT circuits 857-1 to 857-N _{MT-Ant are} input from the first reception weight calculation circuit 147. The reception weights are multiplied, and the signals received by the antenna elements 851-1 to 851-N _MT-Ant are added and combined for each subcarrier. The received signal processing circuits 145-1 to 145-N _SDM perform demodulation processing on the signals obtained by the addition and synthesis, and output the reproduced data to the MAC layer processing circuit 68.

Here, the different received signal processing circuit _{145-1~145-N SDM,} the signal processing of the different signal sequences is performed. Further, as a received signal processing across multiple reception signal processing circuit _{145-1~145-N SDM,} it may be used a simple MLD or the like using the MLD and QR decomposition. Further, the MAC layer processing circuit 68 performs processing relating to the MAC layer (for example, conversion between data input / output to / from the interface circuit 67 and data transmitted / received on the wireless line, ie, wireless packets, termination of header information of the MAC layer. Do). The received data processed by the MAC layer processing circuit 68 is output to an external device or a network via the interface circuit 67. Further, the communication control circuit 121 manages control related to the entire communication, such as the entire timing control.

  Here, similarly to the transmission side, the terminal station device 60 at the time of reception may perform signal processing that intentionally uses the virtual transmission path corresponding to the first singular value. FIG. 23 illustrates an example of another configuration of the receiving unit 65 in the terminal station device 60 according to the first embodiment.

23, reference numeral 154 denotes a first reception weight processing unit, reference numeral 155 denotes a first reception signal processing circuit, reference numeral 156 denotes a first channel information estimation circuit, reference numeral 157 denotes a first reception weight calculation circuit, and reference numeral 159. Denotes a second reception signal processing circuit, and the other is the same as FIG. In the above description, the first reception weight calculation circuit 157 has been described as calculating the reception weight for directly separating the signal sequence of the _NSDM system signal, but once the virtual weight corresponding to the first singular value is calculated. It is also possible to remove residual interference between signal sequences remaining in two stages while performing signal separation on the transmission path.

  In this case, in the first reception weight calculation circuit 157, for example, a signal from each antenna element of the first transmission signal processing units 181-1 to 181-4 of the base station device 70 to each antenna element of the terminal station device 60 If the channel information can be obtained, the first left singular vector obtained when the singular value decomposition is performed on the channel matrix having the channel information as a component is calculated as the reception weight vector. Alternatively, when the first transmission signal processing units 181-1 to 181-4 perform signal transmission using one virtual antenna element formed by multiplying the transmission weight vectors, A channel vector between the corresponding virtual antenna element and each antenna element of the terminal station device 60 may be obtained, and each component of the reception weight vector may be obtained by Expression (7) based on this vector (maximum ratio combining). Or the absolute value may be given constant with respect to the value given by equation (7) (weight of equal gain combining).

Then, the receiving weight vector obtained in this manner is input to the first received signal processing circuit _{155-1~155-N SDM.} However, in the latter stage, since the second received signal processing circuit 159 performs signal processing for separating residual interference, it is not necessary to frequently update the received weight vector in real time. It is also possible to reuse a common reception weight vector in a time domain in which channel information is not expected to fluctuate rapidly.

The first channel information estimating circuit 156 and the first receiving weight calculating circuit 157 do not sequentially update the receiving weights from such a viewpoint. By averaging the channel estimation accuracy, the first reception weight calculation circuit 157 calculates a first reception weight vector at a predetermined cycle based on the averaged channel information, and calculates the first reception weight vector. it is also possible to adopt a configuration for inputting the reception signal processing circuit _{155-1~155-N SDM.} In this case, when averaging, the channel information may be considered to be relative channel information (or each channel information divided by the channel information of the reference antenna) based on the complex phase of the reference antenna (for example, the first antenna). Good) is preferable.

In the first received signal processing circuit _{155-1~155-N SDM,} inputs a signal from the virtual transmission path corresponding to these first singular value to a second reception signal processing circuit 159. This function sharing of the first received signal processing circuit _{155-1~155-N SDM} and second received signal processing circuit 159, first received signal processing section 185 and the second base station apparatus shown in FIG. 17 Is similar to the relationship of the received signal processing unit 75. That is, a signal obtained by reducing an enormous reception signal corresponding to the number of antenna elements N _MT-Ant to the number of signal sequences N _SDM spatially multiplexed by the first reception signal processing circuits 155-1 to 155-N _SDM , And input to the second reception signal processing circuit 159. The second reception signal processing circuit 159 performs general MIMO signal processing in the space whose dimension is reduced.

Specifically, the second reception signal processing circuit 159 refers to the known training signal added to the head of the reception signal, and obtains an N _SDM × N _SDM channel matrix for the N _SDM system signal sequence. , Based on the channel matrix. As described above, the second reception signal processing circuit 159 multiplies the inverse matrix of the ZF type or the linear reception weight matrix of the MMSE type, or performs non-linear processing such as MLD or simple MLD (QR-MLD, etc.). Can be performed. The second reception signal processing circuit 159 performs demodulation processing on the _NSDM system signal thus separated, and outputs the reproduced data to the MAC layer processing circuit 68. This is equivalent to the signal processing of the second received signal processing unit 75 of the base station device.

Here, an example of a device configuration in the second received signal processing unit 75 of the base station device 70 or the second received signal processing circuit 159 of the terminal station device 60 (basically, processing is common to the base station device and the terminal station device). Is shown in FIG. The basic operation is as described above, and the N _SDM sequence signal sequence is used as the received signal of the virtual transmission line corresponding to the N _SDM first singular values in the second reception signal processing circuit 190 (the second reception signal processing circuit 190 in FIG. 23). 2 (corresponding to the reception signal processing circuit 159 of FIG. 2) (although not explicitly shown here, the signal of each subcarrier is input and the same signal processing is performed). With reference to a known training signal added to the head of a received signal, a channel matrix of N _SDM × N _SDM is obtained for a signal sequence of the N _SDM system. The reception weight matrix calculation circuit 192 calculates the reception weight matrix as an inverse matrix of ZF type or a linear reception weight matrix of MMSE type based on the channel matrix, and inputs this to the reception weight matrix multiplication circuit 193. The reception weight matrix multiplication circuit 193 multiplies subsequent data by a reception weight matrix, and suppresses an interference signal that is a crosstalk component between different virtual transmission paths. The signal detection circuit 194 performs signal detection on each signal whose SINR characteristics are enhanced. Here, signal detection intends a general demodulation process. For example, a soft decision is performed on a received signal, error correction is performed after deinterleaving, and final signal detection is performed. The data that has been developed into a plurality of signal sequences and transmitted in parallel is converted into one-series data by parallel / serial conversion, and these are output to the MAC layer processing circuit. Here, an example of linear signal processing is shown as a typical example, but the signal detection circuit 194 can also perform nonlinear signal processing such as MLD or QR-MLD.

In the description of the configuration of the base station apparatus 70, the second transmission signal processing section 71 in FIG. 17 is explicitly shown, and the terminal station apparatus 60 also has the second transmission signal processing section 71 in FIG. Add the transmission signal processing circuit 148, the transmission signal processing circuit _{811-1~811-N SDM} replaces the first transmission signal processing circuit _{821-1~821-N SDM,} further first transmission weight calculating circuit 143 Can be replaced by the transmission weight calculation circuit 149. In this case, the second transmission signal processing circuit 148 shown in FIG. 21 has the function of the second transmission signal processing unit 71 in FIG. 17, and the second transmission signal processing circuit 148 generates a wireless packet to be transmitted on a wireless line. and performs processing for modulation processing, obtains the transmission weight matrix for compensating the leakage signals between the virtual transmission path corresponding to each first singular value from the transmission weight calculating circuit 149, N _SDM system Is subjected to a process of multiplying the transmission signal vector by a transmission weight matrix (transmission precoding), and output to the first transmission signal processing circuit 821. On the other hand, in the first transmit signal processing circuit _{821-1~821-N SDM,} to the transmission signals of _{N SDM} system input from the second transmission signal processing circuit 148, the virtual corresponding to the first singular value The transmission weight for forming the transmission path is multiplied. Here, the function of the transmission weight calculating circuit 149, calculates a transmission weight for forming a virtual transmission path corresponding to the first singular value in the first transmission signal processing circuit _{821-1~821-N SDM} And a function for calculating a transmission weight matrix for compensating signal leakage between virtual transmission paths corresponding to each first singular value. The transmission weight matrix in this case is obtained based on channel information acquired by a method using, for example, implicit feedback with calibration processing, or a method using direct explicit feedback.

  The feature of the present invention according to the first embodiment is that a virtual station corresponding to a first singular value is provided between the terminal station apparatus 60 and the first transmission signal processing units 181-1 to 181-4 of the base station apparatus 70. Is to use a dynamic transmission path. Therefore, for each channel matrix from the terminal station device 60 to the first transmission signal processing units 181-1 to 181-4 of the base station device 70, the transmission weight is assigned to the first right singular vector when the singular value decomposition is performed. This means that the first transmission weight calculation circuit 143 has a function of calculating the first right singular vector. The details of the approximate solution of the first right singular vector will be described later, but the approximate solution with one antenna element in the first transmission signal processing units 181-1 to 181-4 of the base station device 70 A channel vector is obtained, and either a vector obtained by taking a complex conjugate of the channel vector or a vector in which the absolute values of all components of the vector are made constant may be used as the transmission weight vector. Originally, the antenna element group of the first transmission signal processing units 181-1 to 181-4 of the base station device 70 would form a virtual directional antenna as a whole. A case where a single antenna physically located near the center in the antenna element group is represented as an approximate solution. In this case, the weight of the approximate solution is different from the weight of the exact solution, but the gain obtained as a result of evaluation by simulation does not necessarily have an extremely large deterioration as described later.

The above is the description of the configurations of the terminal station device 60, the transmitting unit 61, and the receiving unit 65 in the first embodiment. What is important here is that the local oscillator 815 in the transmitting unit 61 is shared by the mixers 816-1 to 816-NMT _-Ant in each antenna system of the transmitting unit 61, and the local oscillator 853 in the receiving unit 65 is It is common to the mixers 854-1 to 854-NMT _-Ant in each of the 65 antenna systems. In the directivity control, the phase of the transmission / reception signal is adjusted for each antenna element. By controlling the phase relationship of the signals input from the local oscillators 815 or 853 to be always constant, It is possible to determine whether to multiply transmission / reception weights with such a phase relationship. On the other hand, when the local oscillator 815 or the local oscillator 853 uses a plurality of asynchronous oscillators in the transmission unit 61 or the reception unit 65, the directivity control of multiplying the transmission weight at least in the transmission unit 61 functions effectively. Disappears. Care must be taken in this regard when designing the device. Note that the local oscillator 815 and the local oscillator 853 can be shared.

(Expansion to Point-to-Multipoint)
In the above description, the description has been focused on the point-to-point type wireless entrance line. Therefore, if the arrangement of the first signal processing unit 304 is operated in an optimized state by some method, the first signal Spatial multiplexing transmission by the number of processing units 304 is possible. An optimization method for this purpose may be an antenna arrangement method described later, or may be set at several locations when installing the antenna and search for an arrangement in which virtual transmission paths are substantially orthogonal to each other. However, when one base station apparatus 70 and a plurality of terminal station apparatuses 60 perform point-to-multipoint type communication at the same time, the antenna arrangement on the base station apparatus 70 side is ideal for all the terminal station apparatuses. Since it is difficult to arrange the first signal processing units 304 in a redundant manner, a configuration is employed in which a redundant number of first signal processing units 304 are provided, and a different combination of first signal processing units 304 is selected for each terminal station device 60 with which communication is performed. It does not matter. In this case, the communication control circuit determines the optimal combination of the first signal processing units 304. Therefore, the communication control circuit determines the optimal combination of the first signal processing units 304 for each terminal station device 60. It is necessary to have functions such as a database for information.

(Processing flow of signal processing in the present invention)
Hereinafter, a processing flow of signal processing in the first embodiment of the present invention will be described. Basically, the signal processing flow of the base station device and the terminal station device are common, but since the names and code numbers of the circuits to be quoted are different, the signal processing flow for the base station device will be described here as an example. Further, as described above, the signal processing in the second transmission signal processing unit 71 (corresponding to the second transmission signal processing circuit 148 in FIG. 25 having an explicit description in the case of the terminal station device) It is also possible not to perform weight matrix multiplication (transmission precoding), and in this case, the portion related to signal processing can be omitted.

  FIG. 26 shows an outline of a signal processing flow at the time of signal transmission in the first embodiment. When the transmission process starts, the communication control circuit 120 instructs the first signal processing unit 304 and the second signal processing unit 305 to read the first transmission weight vector and the second transmission weight matrix (step S2601) (Step S2602).

  The reading of the transmission weight vector and the transmission weight matrix here is, for example, a point-to-multipoint type communication and in the case of the base station apparatus 70, the terminal station apparatus 60 which is the communication partner station differs every time communication is performed. Therefore, the reading process is performed every time transmission is performed. On the other hand, in the case of fixed point-to-point type communication, or in the case where the communication partner station is always fixed to the base station apparatus 70 as in the terminal station apparatus 60, the processing may be omitted without reading it every time. Is also possible. However, in the case where the time variation of the channel cannot be ignored and the transmission weight is changed every time, even when the communication partner is fixed, it is possible to read out the data every transmission. Alternatively, a configuration in which the transmission weight updated in a predetermined cycle is read may be adopted. When the matrix multiplication in the second transmission weight multiplication circuit in the second signal processing unit 305 is omitted as described above, it is not necessary to read out the transmission weight matrix in the second signal processing unit 305. .

The second signal processing unit 305 generates transmission signals for the _NSDM system for performing spatial multiplexing (step S2603), and multiplies the transmission signals by a second transmission weight matrix (step S2604). The multiplied transmission data for the _NSDM system is transferred to the corresponding first signal processing unit 304 (step S2605). The second signal processing unit 305 determines whether the transmission of the transmission data has been completed (step S2606). If the transmission of the transmission data has not been completed (step S2606: NO), the second signal processing unit 305 returns the process to step S2603. If the transmission of the transmission data has been completed (step S2606: YES), the second signal processing unit 305 ends the signal processing flow at the time of signal transmission.

On the other hand, the first signal processing unit 304 multiplies the transmission signal (one-dimensional signal) by the transmission weight vector using the transmission weight vector of each first signal processing unit 304 that has been read in advance (step S2607). ), The number of transmission antennas is converted into a N _Ant- dimensional transmission signal vector, and transmission signal processing is performed for each antenna system (step S2608).

  For example, when assuming an OFDM modulation scheme, generation of a transmission signal is performed for each subcarrier and for each OFDM symbol, and multiplication of a transmission weight matrix and multiplication of a transmission weight vector are also performed individually for each subcarrier. Done. As the transmission signal processing here, IFFT processing is performed on the transmission signal for each subcarrier on the frequency axis, a guard interval is added, and waveform shaping between symbols is added as necessary. D / A converting the sampling data to generate an analog baseband signal, upconverting the analog baseband signal with a mixer, removing out-of-band components with a filter, and amplifying the signal with a high power amplifier, and transmitting the amplified signal from an antenna. .

  The above is an example in which the OFDM modulation scheme is used, but the same applies to other schemes such as SC-FDE, and basically various conventional communication schemes can be applied. In addition, the signal processing is not necessarily performed on the frequency axis, and when multiplication of the transmission weight vector is performed on the time axis as described later, sampling is performed using a single transmission weight vector (and matrix). The multiplication of the transmission weight vector (and the matrix) may be performed for each data. In addition, processes such as error correction encoding and interleaving are assumed to be included in the transmission signal generation process, and are omitted in the description here.

  Next, FIG. 27 shows an outline of a first example of a signal processing flow at the time of signal reception in the first embodiment. Upon receiving the signal, the communication control circuit 120 instructs the plurality of first signal processing units 304 to read out the respective first reception weight vectors (step S2701).

  The reading of the reception weight vector here is, for example, a point-to-multipoint type communication as in the case of the transmission side, and in the case of the base station apparatus 70, the terminal station apparatus 60 which is the communication partner station performs each time of communication. Therefore, the reading process is performed every time transmission is performed. On the other hand, in the case of fixed point-to-point type communication, or in the case where the communication partner station is always fixed to the base station apparatus 70 like the terminal station apparatus 60, the processing is omitted without reading every time. It is also possible. However, in the case where the time variation of the channel cannot be neglected and the reception weight is changed every time, even when the communication partner is fixed, the configuration may be read out every time transmission is performed. Alternatively, the reception weight updated at a predetermined cycle may be read.

  Thereafter, when an actual signal is received, a signal reception process is performed for each antenna system (step S2702). Here, the signal receiving process means, for example, amplifying a received signal with a low noise amplifier, performing a down-conversion process with a mixer, removing out-of-band frequency components with a filter, and then using a digital baseband with an A / D converter. Convert to signal sample values. In the reception signal processing, for example, in the case of the OFDM modulation method, the signal is cut out for each OFDM symbol, the guard interval is removed, and the signal is converted into a signal on the frequency axis for each subcarrier component by FFT.

  With respect to the signal on which the above-described reception signal processing has been performed, the reception signal vector having the signal of each antenna element as a vector component is multiplied by the reception weight vector (step S2703), and the result is subjected to the second signal processing. It is transferred to the unit 305 (step S2704). The first signal processing unit 304 determines whether the reception of the reception data has been completed (step S2705). Here, if the reception data is further continued (step S2705: YES), the process returns to the reception signal processing shown in step S2702 and the processing is continued, and if the reception data is completed (step S2705: NO), The processing of the first signal processing unit 304 ends.

On the other hand, the second signal processing unit 305 determines whether or not the signal transferred from the first signal processing unit 304 is, for example, a channel estimation training signal added to the head of a wireless packet (step S2706). ), If it is a training signal (step S2706: YES), channel estimation is performed (step S2707), and a second reception weight matrix is generated based on the obtained channel matrix (step S2708). On the other hand, if the data portion is not a training signal (step S2706: NO), the N _SDM- dimensional reception signal vector of the spatial multiplexing number is multiplied by the second reception weight matrix (step S2709). A signal detection process is performed (step S2710). Here, the signal detection processing means processing for estimating the transmission signal through, for example, soft decision processing, deinterleaving processing, error correction processing, and the like for the transmission signal, and after completion of the signal processing of the physical layer, the MAC layer Output to the processing circuit.

  In the reception signal processing, a general conventional technique such as the SC-FDE scheme can be similarly applied in addition to the OFDM modulation scheme. Furthermore, in the description, signal processing for each subcarrier has been described. However, signal processing is not necessarily performed on the frequency axis. As described later, multiplication of a reception weight vector and a reception weight matrix is performed on the time axis. In some cases, a single reception weight vector and reception weight matrix may be used to multiply the reception weight vector and reception weight matrix for each sampling data. Also, for the MIMO signal processing corresponding to, for example, step S2709 of multiplying the second reception weight matrix (and step S2710 of performing signal detection processing after signal separation), multiplication of the second reception weight matrix as linear signal processing is not necessarily performed. It is not necessary to perform the process, and a general MIMO signal detection process such as QR-MLD or MLD can be applied.

The above description is based on the assumption that the local oscillator 853 used in the first signal processing unit 304 is asynchronous for each first signal processing unit 304 like the base station device 70, and the second reception weight matrix is Has been described for each reception, but in the case of the terminal station device 60, for example, the local oscillator 853 used for the up-conversion processing and / or the down-conversion processing of all antenna elements is shared, so that the reception is not necessarily performed. Does not mean that the second reception weight matrix changes every time. As described above, when the local oscillator 853 is commonly used as a whole and the time variation of the channel can be ignored, the second reception weight matrix acquired in the past can be used. FIG. 28 shows an outline of a second example of the signal processing flow at the time of signal reception in the first embodiment of the present invention. Steps S2801 to S2805 shown in FIG. 28 are the same as steps S2701 to S2705 shown in FIG. The difference from FIG. 27 is that in the signal processing of the second signal processing unit 305, the second signal processing unit 305 determines the second reception weight matrix of the second reception weight matrix from the communication control circuit 120 at the time when the signal reception processing is started. Upon receiving the read instruction, the second reception weight matrix is read (step S2806). Thereafter, when the reception signal is transferred from the first signal processing unit 304, the N _SDM- dimensional reception signal vector of the spatial multiplex number is multiplied by a second reception weight matrix (step S2807), and thereafter, signal detection is performed. The processing is performed (step S2808). Other signal processing is the same as in FIG.

  As described above, the wireless communication system 50 of the first embodiment includes the base station device 70 (first wireless station device) and the terminal station device 60 (second wireless station device). The base station device 70 includes a plurality of first transmission signal processing units 181, a plurality of first reception signal processing units 185, a second transmission signal processing unit 71, and a second reception signal processing unit 75. . The plurality of first transmission signal processing units 181 have a plurality of antenna elements 819. The plurality of first reception signal processing units 185 have a plurality of antenna elements 851. The second transmission signal processing unit 71 executes signal processing of wireless communication associated with the plurality of first transmission signal processing units 181. The second reception signal processing unit 75 executes signal processing of wireless communication associated with the plurality of first reception signal processing units 185. The terminal station device 60 includes a plurality of antenna elements 819, a plurality of antenna elements 851, a transmitting unit 61, and a receiving unit 65. The transmission unit 61 performs wireless communication with the plurality of first reception signal processing units 185 via the plurality of antenna elements 819. The transmission unit 61 communicates with the plurality of first reception signal processing units 185 and the second reception signal processing unit 75 associated with the first reception signal processing unit 185 via the plurality of antenna elements 819. Communication may be performed. The receiving unit 65 performs wireless communication with the plurality of first transmission signal processing units 181 via the plurality of antenna elements 851. The receiving unit 65 is configured to wirelessly connect the plurality of first transmission signal processing units 181 and the second transmission signal processing unit 71 associated with the first transmission signal processing unit 181 via the plurality of antenna elements 851. Communication may be performed. The first transmission signal processing unit 181 and the first reception signal processing unit 185 provide a transmission weight vector and a reception weight for a MIMO channel matrix used for wireless communication between the first antenna element group and the second antenna element group. An approximate solution of at least one of the first right singular vector and the first left singular vector corresponding to the first singular value of the MIMO channel matrix or an approximate solution of the first right singular vector and an approximate solution of the first left singular vector Is calculated based on at least one of the above. The plurality of first transmission signal processing units 181 and the first reception signal processing units 185 include the transmission weight vector and the reception weight vector corresponding to the first transmission signal processing unit 181 or the first reception signal processing unit 185. Spatial multiplex transmission of independent signal sequences is performed using at least one of them.

  This allows the base station device 70, the terminal station device 60, the wireless communication system 50, and the wireless communication method of the first embodiment to increase the transmission capacity by MIMO in an environment where the line-of-sight environment is dominant. For example, the base station device 70, the terminal station device 60, the wireless communication system 50, and the wireless communication method use a high-order spatial multiplexing and a high frequency band in a wireless communication system in a mobile network in the future, while the line-of-sight environment is dominant. As a result, a large capacity can be realized.

  The base station device 70 as the base station device 303 according to the first embodiment includes a first signal processing unit 304 in which antenna elements are bundled in a small area, and a second signal processing unit 305 that aggregates them. The first signal processing unit 304 performs closed beam forming in each signal processing unit, and does not perform beam forming over different first signal processing units 304. The individual first signal processing unit 304 generates transmission / reception weights for the “virtual transmission path corresponding to the first singular value” that makes the best use of the “line of sight wave”, and generates a plurality of first signal processings. Spatial multiplexing transmission is performed using the individual transmission / reception weights in section 304.

  Here, in the above description, it has been described as a typical embodiment that the line-of-sight environment is a dominant environment. Or, when a diffracted wave that is inferior to the line-of-sight wave arrives, the virtual transmission path corresponding to the first singular value is formed in the arrival direction. In this sense, the virtual transmission path corresponding to the first singular value does not necessarily need to be a transmission path constituted by line-of-sight waves, and in the first embodiment of the present invention, the virtual transmission path corresponding to the first singular value. Since spatial multiplexing transmission is performed by positively utilizing a plurality of transmission lines, the first embodiment of the present invention can be applied even in a non-line-of-sight environment. The circuit configuration and processing flow in that case can be applied as they are without any change from the above description.

[Second embodiment]
[Antenna placement conditions for orthogonalization of line-of-sight MIMO transmission]
(Overview of Basic Principle According to Second Embodiment)
Here, conditions for a plurality of virtual transmission paths to have a substantially orthogonal relationship will be summarized. In FIG. 16 described above, since one directional beam is effectively formed for every 25 antenna elements, this is approximately one antenna having a very high directional gain (for example, a parabolic antenna) Is equivalent to using. FIG. 29 shows an example of a case where four parabolic antennas are mounted on a base station apparatus and 16 antenna elements are mounted on a terminal station apparatus in a linear array. In FIG. 29, reference numeral 306 denotes a base station device, reference numeral 302 denotes a terminal station device, and reference numerals 307-1 to 307-4 denote parabolic antennas. The terminal station device 302 corresponds to the terminal station device 60. Base station apparatus 306 is equivalent to base station apparatus 70 equivalently. Basically, transmission equivalent to the parabolic antennas 307-1 to 307-4 shown in FIG. 29 is realized by grouping a large number of antenna elements of the base station device 70.

  Here, the conditions under which the four singular values of the MIMO channel in FIG. 29 in which the line of sight is dominant are almost equally large as in the case described above will be summarized. For example, Patent Documents (Japanese Patent No. 5488894) specify conditions for establishing MIMO transmission in a line-of-sight environment. Are spread as much as possible, and as a result, the distances between the individual transmitting antennas and the receiving antennas are varied, and the relationship between the distances becomes a pseudo-random relationship (or a predetermined relationship). It tries to adjust to.

  On the other hand, for example, in the phased array antenna technology, if the distance between the antenna elements of a one-dimensional linear array is d, setting this value of d to a small value allows the antenna to be viewed from a distant antenna in the direction in which the directivity should be directed. The radio wave arriving at (or being transmitted to) the antenna element of each linear array can be approximated to a plane wave. If the direction of the arriving wave is represented by an angle θ, the path difference between the antenna elements is d · sin θ, which is high. It becomes possible to approximate the accuracy. However, in the above-described basic concept of the related art, in order to set the interval between the antenna elements as wide as possible, such a plane wave approximation cannot be performed, and the approximation needs to be performed under different conditions. Was. As a result, the accuracy of the prior art approximation is reduced. In fact, when performing MIMO communication in a line-of-sight environment, a large parabolic antenna was assumed at both the transmitting and receiving stations. Usually, the size of the parabolic antenna is much larger than the wavelength, and therefore, the above-described path length difference cannot be approximated by a plane wave with negligible accuracy with respect to the wavelength.

However, while increasing the effective aperture length of the antenna only on the base station device 70 side, the interval between the antenna elements on the terminal station device 60 side is set to about several wavelengths (the distance between the base station device 70 and the terminal station device 60 is reduced). If it is set to the order of, for example, 3 wavelengths or less), the path difference between the parabolic antenna on the base station apparatus 306 side and the antenna element on the terminal station apparatus 302 side in FIG. It can be approximated. Therefore, as shown in FIG. 30, the base station apparatus antenna #j (310-j) (1 ≦ j ≦ M) of the base station apparatus 70 corresponding to the base station apparatus 306 and the terminal station corresponding to the terminal station apparatus 302 the distance between the terminal station apparatus antenna # 1 of the device 60 (320-1) and _{L j,} further if the angular difference between the front direction of the terminal station device antenna # 1 (320-1) and theta _j, Using the antenna spacing d of the terminal station apparatus 60, channel information between the channel matrix (the j-th antenna on the base station apparatus 70 side and the i-th (1 ≦ i ≦ N _MT-Ant ) antenna of the terminal station apparatus 60) is _hij. ) Can be approximately expressed by the following equation (26). In equation (26), NMT _-Ant is denoted as N to avoid complexity. Further, in FIG. 29, the number of the parabolic antennas 307-1 to 307-4 included in the base station device 306 is four, but here, the number of antenna elements is M as a general number.

Here, it is displayed in the form of a matrix product. The (j, j) -th component of the second item indicates channel information between the j-th antenna of the base station device 70 and the first antenna of the terminal station device 60. The difference information of each component of the N _MT-Ant antennas of the terminal station device 60 corresponds to the column vector of the j-th column of the first item. Since the (j, j) -th component of the second item is a common term applied to all the components of the column vector, if the column vectors of the first item are orthogonal, each singular value is stably a large value. Become. The condition in which the i-th column vector and the j-th column vector of the first term are orthogonal to each other is expressed by the following equation (27) in which the inner product of the complex vector is zero.

The above equation transformation uses the formula of the sum of geometric series. Since the last condition is that the numerator is zero under the condition that the denominator is not zero, the following conditional expression (28) is obtained from the condition that the term in the parentheses of the numerator exp is an integral multiple of 2πi. Is led. In the following equations, N is described as the original number of antennas N _MT-Ant .

Constant _{K 1} of the formula (28) is an integer not the number of antennas _N a multiple of _MT-Ant terminal station 60. Here, as shown in FIG. 30, it is assumed that the j-th antenna on the base station apparatus 70 side has a displacement of d _j from the front of the antenna on the terminal station apparatus 60 sideways, and the difference between the i-th and j-th antennas is Δd _ij And Let L be the distance between the base station device 70 and the terminal station device 60, and multiply both sides of the first expression of Expression (28) by L / d. The distance between the first antenna of each antenna element and the terminal station device 60 of the base station apparatus 70 slightly different, but the distance L becomes possible approximation with sufficiently large if L ≒ L _j than the displacement d _j.

That is, with respect to the number of antennas _{N MT-Ant} terminal station _{60, N MT-Ant} to integer _{K 2} not be zero at integer multiples, the interval d of the antenna elements of the terminal station 60, terminal station 60 and the base The distance between any two antenna elements may be set to satisfy the formula (29) with respect to the distance L of the station device 70 and the wavelength λ of the signal wave of the wireless communication. If this condition is interpreted geometrically, the condition can be easily realized, for example, under the following conditions. For example, as the simplest conditions, on a line linear array and the distance of the terminal station 60 side L away and directly opposite, allows any offset in the linear direction, N in λL / N _{MT-Ant d} spacing _{MT- The} point of the _Ant point may be set linearly, the M point may be selected from the _NMT-Ant points, and the M antenna elements of the base station apparatus 70 may be arranged. This is because any two integers are selected from consecutive N _MT-Ant integers, and the absolute value of the difference is always (N _MT−Ant −1) or less, and K ₂ becomes N _{MT -It utilizes} the fact that it can be prevented from becoming an integral multiple of _Ant . Note that, by searching and setting an arrangement in which the difference between any two arbitrary integers is not an integral multiple of _NMT-Ant , it is also possible to select the antenna installation location of the base station device from other broader conditions. It is.

As shown in FIG. 29, when the size of the antenna element on the terminal station device 60 side is reduced, the number of antenna elements N _MT-Ant on the terminal station device 60 side is increased accordingly, while the antenna of the base station device 70 is generally increased. Since the number of elements M is set to the upper limit of the assumed number of spatial multiplexing, it is assumed that the value of M is smaller than the number of antenna elements N _MT-Ant on the terminal station device 60 side. That is, when N _MT-Ant > M, the M antennas on the base station device 70 side need not be equally spaced even under the above-described simplest condition, and are intermittently selected from the above-described N _MT-Ant points. The location where the antenna element is arranged may be set. Although the actual number of antenna elements is 100 in the case of FIG. 16, M in FIG. 16 is 4 since the antenna elements can be grouped in units of 25 and regarded as four virtual antenna elements. Should be considered. Although the approximation with very high accuracy is performed until the derivation of Expression (28), the approximation used in Expression (29) has slightly lower accuracy. However, depending on the setting of the parameters, assuming that the distance between the antenna elements of the base station device 70 in the equation (29) is about 1 m, even if there is an error of about ± several centimeters in the base station apparatus 70, they are almost orthogonal. There is no difference. Although the same applies to the first embodiment of the present invention, basically, even when there is an interference component between a plurality of virtual transmission paths on the receiving side, it is possible to suppress the interference by signal processing on the receiving side. If such signal processing is assumed, the loss can be minimized by making the state substantially orthogonal to the last, so that the approximation accuracy of Expression (29) does not significantly affect the system operation.

  Considering such properties, if each grouped antenna element group shown in FIG. 16 is regarded as one virtual antenna, the center point of the physically spread multi-element antenna is assumed to be virtual. If the plurality of antenna element groups are arranged in the above-described antenna arrangement, the weight vector represented by the first singular vector is used, and a plurality of signal sequences with desired characteristics are obtained. Spatial multiplexing and transmission are possible.

In the above description, the antenna element on the base station apparatus side and the antenna element on the terminal apparatus side have been described as an example in a state where they face each other as shown in FIG. 29. As shown in FIG. 31, it is assumed that there is no direct relationship. This is due to restrictions on the installation location of the base station device 306 and the terminal station device 302. In such a case, a desired effect can be obtained by slightly converting each parameter. In FIG. 31, when the terminal station apparatus 302 is viewed from the base station apparatus 306, the terminal station apparatus 302 exists in a direction shifted by an angle θ from the front, and when the base station apparatus 306 is viewed from the terminal station apparatus 302. The base station apparatus 306 exists in a direction shifted from the front by an angle θ ′. In this case, the distance _{D 1} of the parabolic antenna 307-1 and 307-4 of the base station apparatus 306 is equivalently visible state narrowed to _{D 1 '(= D 1 cosθ} ). Similarly, it appears that the width D ₂ of the linear array 312 of the terminal station device 302 is equivalently narrowed to D ₂ ′ (= D ₂ cos θ ′). In other words, it can be understood that the antenna element interval d has been converted to dcos θ × cos θ ′ times. If the distance between the center of gravity of the parabolic antennas 307-1 to 307-4 of the base station device 306 and the distance between the terminal station device 302 is L, these conversions are performed, and the optimum antenna arrangement is obtained using the above-described conditional expression. Can be calculated. That is, the base station apparatus 70 virtually projects each linear array on an axis orthogonal to the line of sight. Similarly, the terminal station device 60 virtually projects each linear array on an axis orthogonal to a line of sight. D in Expression (29) is converted into dcos θ × cos θ ′ based on the interval between the antenna elements on the virtually projected axis. D in Expression (29) is converted by a conversion device or a person based on the distance between the antenna elements on the virtually projected axis.

  29 and 31, the distance between the vicinity of the center of gravity of the parabolic antennas 307-1 to 307-4 of the base station apparatus 306 and the distance between the terminal station apparatus 302 is L, but the exact distance between the individual antenna elements is common. Instead of the distance L, each has an error. However, as can be seen from the use of approximation in the middle of the calculation of the equation (29), a certain error can be tolerated for the distance L here as well, and as shown in FIG. The distance between the elements (the distance between two straight lines of the antenna elements arranged on a parallel straight line) may be used, or the distance approximately connecting the center of gravity of the antenna element as shown in FIG. As described above, the distance L may be handled as an approximate value or an approximate value thereof.

  Note that, in the description of FIGS. 29 and 30 in this description, a case where the terminal station device is configured by a linear array is shown as a typical example. This feature is that the parabolic antennas 307-1 to 307-4 or 310-1 to 310-M (corresponding to the antenna element groups 304-1 to 304-4 in FIG. 16) on the base station device side are arranged on a straight line. Similarly, the antenna elements on the side of the terminal station device are arranged in a linear array on a straight line. In the above-described calculation of the orthogonalization condition, the two straight lines have been described as being in a parallel relationship. Specifically, for example, a horizontal axis parallel to the wall surface of the building is assumed on the wall surface of the building or on the roof of the building, and the plurality of first signal processing units and the antenna element (group) 307 are set along the horizontal axis. When -1 to 307-4 are installed, it is preferable that the linear array of the terminal station device is also arranged on a horizontal axis parallel to the wall surface of the building on the opposite side of the road. Here, a vertical axis orthogonal to the horizontal axis on the terminal station apparatus side is set, and a grid parallel to the horizontal axis and the vertical axis is assumed, and the linear array of the terminal station apparatus described above is vertically shifted by N. 'Consider using a stacked square lattice array. At this time, the total number of antenna elements of the terminal station apparatus is N × N ′. However, when viewing the plurality of antenna elements (groups) 307-1 to 307-4 of the first signal processing unit of the base station apparatus from this terminal station apparatus, although there is an angular difference with respect to the horizontal direction, Since there is no difference in the elevation angle in each of the antenna elements (groups) 307-1 to 307-4, each of the N ′ stages of the above-described square lattice array is not regarded as an independent antenna element, and is substantially vertical. It is understood that N ′ elements arranged in the direction behave equivalently as one virtual antenna element, and that the virtual antenna elements are arranged in a linear array on the horizontal axis. In fact, this effect has been confirmed in the simulation evaluation, and the antenna element of the terminal station apparatus is parallel to the axis connecting the antenna elements (group) 307-1 to 307-4 on the base station apparatus side. When N × N ′ elements are arranged in a lattice formed by an axis and an axis orthogonal to this axis, the N × N ′ elements are parallel to the axis connecting the antenna elements (group) 307-1 to 307-4 on the base station apparatus side. Considering a linear array of N elements on the axis and the element spacing between the N elements, and applying the equation (18) to optimize the element spacing of the antenna element (group) 307-1 to 307-4 on the base station apparatus side good. Of course, when the antenna elements (group) 307-1 to 307-4 on the base station apparatus side are vertically aligned on the wall surface of the building or the like, the antenna elements on the terminal station apparatus side also have N elements in the vertical direction. Equation (18) may be applied by assuming that N ′ elements are arranged in a grid in the horizontal direction and are regarded as an equivalent linear array of N elements arranged in the vertical direction. Also, here, the antenna arrangement on the terminal station apparatus side has been described as a square lattice array, but the arrangement is not necessarily a square lattice, and the element spacing in the horizontal and vertical directions may be rectangular. In this case, the distance between the antenna elements (group) 307-1 to 307-4 on the side of the base station apparatus and the antenna elements arranged on the axis on the side of the terminal station apparatus which is parallel to the axis on which the antenna elements (group) 307-1 to 307-4 are aligned is determined by the antenna element of equation (18). It may be calculated as the interval d.

As described above, the wireless communication system 53 of the second embodiment includes the base station device 306 (first wireless station device) and the terminal station device 302 (second wireless station device). The base station device 306 includes antenna elements (parabolic antennas) 307-1 to 307-4, and the terminal station device 302 includes a terminal station device antenna element group 312. The plurality of parabolic antennas (antenna elements) 307-1 to 307-4 of the base station apparatus 306 include, as shown in Expression (29), the distance Δd _mn between the m-th element and the n-th element, the terminal station apparatus It is arranged based on the distance L between the base station apparatus 302 and the base station apparatus 306, the wavelength λ of the signal wave of wireless communication, the number of antennas N _MT-Ant of the terminal station apparatus 302, and the distance d between the antenna elements of the terminal station apparatus 302. You. Each of the parabolic antennas 307-1 to 307-4 is, for example, a single high-gain antenna element (single antenna element). The high gain antenna element is, for example, a parabolic antenna 307. The parabolic antennas 307-1 to 307-4 may be an antenna element group instead of a single high gain antenna element. When the antenna element group is used as described above, the base station apparatus 306 may be the base station apparatus 303 shown in FIG. 16 and the base station apparatus 17 shown in FIG. The base station device 70 in FIG. 17 includes a plurality of first transmission signal processing units 181, a plurality of first reception signal processing units 185, a second transmission signal processing unit 71, and a second reception signal processing unit 75. And The plurality of first transmission signal processing units 181 have a plurality of antenna elements 819. The plurality of first reception signal processing units 185 have a plurality of antenna elements 851. The second transmission signal processing unit 71 executes signal processing of wireless communication associated with the plurality of first transmission signal processing units 181. The second reception signal processing unit 75 executes signal processing of wireless communication associated with the plurality of first reception signal processing units 185. The terminal station device 60 includes a plurality of antenna elements 819, a plurality of antenna elements 851, a transmitting unit 61, and a receiving unit 65. The transmission unit 61 performs wireless communication with the plurality of first reception signal processing units 185 via the plurality of antenna elements 819. The receiving unit 65 performs wireless communication with the plurality of first transmission signal processing units 181 via the plurality of antenna elements 851.

  That is, the distance between the antenna elements or the distance between the antenna element groups depends on the distance L between the base station apparatus 306 (first wireless station apparatus) and the terminal station apparatus 302 (second wireless station apparatus), and the signal wave of wireless communication. Λ, the number N of linear array-shaped antenna elements constituting the second antenna element group or the number N of any of the vertical or horizontal antenna elements arranged in a lattice, and the second antenna element Each single antenna element forming a plurality of first antenna groups or an integer multiple of a value calculated based on the distance d between any of the vertical or horizontal antenna elements forming the group or Antenna element groups are arranged.

  Accordingly, the base station apparatus 306 or 70, the terminal station apparatus 302 or 60, the wireless communication system 53 or 50, and the antenna element arrangement method of the second embodiment can increase the transmission capacity by MIMO in an environment where the line-of-sight environment is dominant. It is possible to increase. The base station apparatus 306 or 70, the terminal station apparatus 302 or 60, the wireless communication system 53 or 50, and the antenna element arranging method according to the second embodiment transmit a plurality of signal sequences by spatial multiplexing with desired characteristics. Becomes possible. The base station device 306 or 70, the terminal station device 302 or 60, the wireless communication system 53 or 50, and the antenna element arrangement method of the second embodiment can improve the SIR characteristics. The base station device 306 or 70, the terminal station device 302 or 60, the wireless communication system 53 or 50, and the antenna element arrangement method of the second embodiment can realize higher spatial multiplexing.

  The base station device 306 or 70, the terminal station device 302 or 60, the wireless communication system 53 or 50, and the antenna element disposition method of the second embodiment can be used in an environment where the line of sight is dominant and the base station device 306 or 70 In a MIMO channel of a plurality of parabolic antennas 307 and a plurality of element antennas on the terminal station device 302 or 60 side, antenna installation conditions for orthogonalizing each channel are defined. However, here, conditions for mounting a parabolic antenna on the base station side are shown for simplicity of discussion, but naturally a highly directional antenna other than the parabolic antenna is used, and further, the base station side has a plurality of antenna elements. Even when the first signal processing unit as shown in FIG. 16 is used, orthogonalization between transmission paths can be generally achieved under the same conditions. This makes it possible to improve communication quality on the transmission path and increase transmission capacity.

[Third Embodiment]
[Technology applied to access system of multiple virtual transmission paths corresponding to first singular value]
(Basic principle according to the third embodiment)
In the above description, the environment in which the line of sight wave is dominant in the transmission path between the base station apparatus and the terminal station apparatus has been taken up. As an example of such a use, a case is considered in which the terminal station apparatus 60 is regarded as a relay station and the above-described technique is used in an entrance line from the base station apparatus 70 to the relay station. However, the virtual transmission path corresponding to the first singular value is not limited to use in a wireless entrance line. Also in the above description, it is assumed that the terminal station device 60 corresponding to the relay station is small, and that a large number of small antennas are arranged in a very small area. Therefore, for example, if a large number of antennas are mounted on a smartphone or the like, and if it is assumed that reception is performed in a small cell of a fifth generation mobile communication system (5G), the use of a millimeter wave or a quasi-millimeter wave is assumed. Except that the user moves, it can be regarded as an environment in which the antenna is condensed and arranged in a narrow place just like the case of the wireless entrance described above. That is, as an example, when the small cell base station is arranged at a relatively high place such as the wall surface of a building, the base station device 70 and the terminal station device 60 are almost in the line-of-sight environment from almost directly above, and slightly toward the surface of the smartphone. Considering a situation in which an antenna having a directional gain is mounted, it is expected that communication will be performed in an environment where the line of sight wave is dominant. In such a case, it is effective to group the antennas of the base station apparatus into a plurality of groups as in FIG. 16 and actively use the virtual transmission path corresponding to the first singular value in each group. Become.

  Here, considering the difference between the case of FIG. 16 and the case of the access system, in FIG. 16, the straight line in which the antenna elements of the base station device 70 are aligned and the straight line in which the antenna of the terminal station device 60 are aligned are substantially parallel. However, depending on the combination of the arrangement of the antenna of the smartphone, the angle at which the user holds the smartphone, the direction the user is facing, etc., the three-dimensional positional relationship of each antenna is generally It is expected that the relationship is not parallel but twisted. In this case, there is generally no optimum value as shown by the equation (29), but for example, the grouped antenna element groups are provided with some redundancy, and the optimum antenna of a combination having a low correlation in selection diversity is selected. What is necessary is just to select and use an element group. However, in this case, it is necessary to set the interval between the antenna element groups a little larger so that the correlation between the channels can be reduced as much as possible. In that case, it may be effective to install the antenna over a plurality of structures, instead of arranging all the antenna element groups only on the wall surface of the same structure (such as a building). In such a case, in FIG. 16, it is difficult to connect the second signal processing unit 305 and the first signal processing units 304-1 to 304-4 with a wired cable (including an optical fiber).

  Therefore, the antenna element group (first signal processing unit 304) of the base station device 303 is installed as a satellite base station, and the control base station device corresponding to the second signal processing unit 305 of the base station device 303 transmits these antenna elements. It is effective to realize transmission / reception of signals to / from the satellite base station apparatus via a wireless line. FIG. 32 shows a third embodiment of the present invention. In FIG. 32, reference numeral 31 denotes a controlling base station device, reference numerals 32-1 to 32-3 denote satellite base station devices, and reference numeral 33 denotes a terminal station device. The terminal station device 33 corresponds to the terminal station device 60. Reference numeral 37 represents a structure. Reference numeral 38 represents a structure. The control base station device 31 is disposed, for example, on the upper surface of a structure 37 (the roof of a building or the like). The satellite base station device 32 is disposed, for example, on the side surface of a structure 38 (wall surface of a building or the like). The terminal station device 33 is disposed, for example, in a cell 39 between the structure 37 and the structure 38.

  The control base station device 31 and the satellite base station devices 32-1 to 32-3 realize the functions of the base station device 70 as a whole. The control base station apparatus 31 includes a large number of antenna elements, and configures a small antenna as a whole so that the distance between the antenna elements is substantially equal to the wavelength (for example, between several wavelengths to half a wavelength). On the other hand, each of the satellite base station devices 32 is similar to the first signal processing unit 304 of FIG. Antenna. Communication between the control base station device 31 and the satellite base station devices 32-1 to 32-3 is performed using the first singular vectors corresponding to the first singular values of the respective antennas. The terminal station device 33 includes a plurality of antennas. For example, assuming a smartphone or the like, a plurality of antennas will be mounted in a relatively small area. Here, communication is also performed between the terminal station device 33 and the satellite base station devices 32-1 to 32-3 using the first singular vector corresponding to the first singular value of each antenna as a transmission / reception weight vector. Further, communication is performed between the satellite base station devices 32-1 to 32-3 and the control base station device 31 using the first singular vectors corresponding to the first singular values of the respective antennas as transmission / reception weight vectors. Thereby, while utilizing the transmission / reception weight vector for maximizing the gain of the line-of-sight wave, the stable MIMO transmission with a small channel correlation is achieved by utilizing a plurality of different satellite base station devices 32-1 to 32-3. Will be possible.

  Next, FIG. 33 shows another example of the third embodiment of the present invention. In FIG. 33, reference numerals 34-1 to 34-2 denote a general base station device, reference numerals 35-1 to 35-6 denote satellite base station devices, and reference numerals 36-1 to 36-2 denote terminal station devices. FIG. 33 is the same as FIG. 32 described above, but includes a number of satellite base station devices 35-1 to 35-6 arranged on the side surfaces (wall surfaces of the building) and the upper surface of the structure 37 (the roof of the building). 32 is different from FIG. 32 in that the correspondence between the centralized base station devices 34-1 to 34-2 arranged in is not a one-to-one correspondence. The satellite base station devices 35-1 to 35-6 can provide an access link by being flexibly combined with a plurality of control base station devices 34-1 to 34-2. For example, when viewed from the controlling base station device 34-1 near the terminal station device 36-1, the nearest satellite base station devices are the satellite base station devices 35-1 to 35-3. However, assuming that the satellite base station devices having a small correlation from the terminal station device 36-1 are the satellite base station devices 35-1, 35-2, and 35-4, the supervising base station device 34-1 becomes the next supervising base station. It is also possible to utilize the satellite base station device 35-4 closer to the station device 34-2 and provide communication to the terminal station device 36-1 using a virtual transmission path corresponding to the first singular value. become.

  Incidentally, one of the reasons for utilizing the virtual transmission path corresponding to the first singular value in such an access system is that the channel information of the virtual transmission path corresponding to the first singular value is different from that of the other higher order. Compared with the channel information of the virtual transmission path corresponding to the singular value, the time variation is relatively small statistically. In other words, the first right (left) singular vector obtained when the singular value decomposition is performed represents a path corresponding to the line-of-sight wave, and there is no large change in the transmission / reception weight to be applied by a slight movement. However, since the higher-order right (left) singular vector corresponds to a path containing a large amount of reflected wave components, the channel state after the synthesis fluctuates significantly due to slight movement of the terminal, and as a result, the right (left) singular vector Will fluctuate greatly. In other words, if the channel information greatly changes, the transmission / reception weight vector for securing the gain also changes.If the transmission and reception are performed with the same value ignoring the change, a decrease in gain and deterioration of characteristics corresponding to the change will occur. Inevitable. In particular, if the terminal station device 36 is a small portable terminal such as a smartphone, the channel correlation between antennas is large, and it may be difficult to increase the capacity in single-user MIMO. In such a case, as shown in an embodiment to be described later, it is effective to increase the total capacity by multi-user MIMO accommodating a large number of terminals having small transmission capacities at the same time. , The SINR characteristic is degraded, and the transmission capacity is reduced and the communication becomes unstable. However, by actively using the virtual transmission path corresponding to the first singular value whose channel time variation is relatively small, even when multi-user MIMO is used, mutual interference between terminals is significantly reduced. It is also possible to obtain a secondary effect that it can be reduced.

  Note that this multi-user MIMO generally means that one base station apparatus performs spatial multiplexing transmission with a plurality of terminal station apparatuses, but here, as shown in FIG. 34-1 to 34-2 can be understood including the case where services are simultaneously provided to the plurality of terminal station devices 36-1 to 36-2. In this case, the individual control base station devices and the terminal station devices may actually be in one-to-one communication. However, the same frequency channel is used for communication, and the satellite base station devices 35-1 to 35-35 to be further utilized are used. 6, the satellite base station devices 35-1 to 35-6 operate while mutually using resources so as to allow partial duplication, and a certain satellite base station device 32 simultaneously transmits to a plurality of terminal station devices 36. Therefore, such an operation mode can be regarded as a multi-user MIMO in a broad sense, and even in such a case, the effect of reducing the influence of the above-described channel time variation works effectively. Become.

(Outline of non-regenerative re-beamforming relay in satellite base station equipment)
The basic operation when relay transmission is performed via a plurality of satellite base station devices 32 using a virtual transmission path corresponding to the above-described first singular value will be described with reference to the drawings. In the configuration using satellite base station apparatus 32 shown in FIG. 32, second signal processing section 305 and first signal processing sections 304-1 to 304- of base station apparatus 303 shown in FIG. 4 corresponds to the function of the base station apparatus as a whole including the control base station apparatus 31 and the satellite base station apparatuses 32-1 to 32-3 in FIG. 32, and the second signal processing unit 305 in FIG. And the first signal processing units 304-1 to 304-4 are connected by wire, whereas in FIG. 32, the control base station device 31 and the satellite base station devices 32-1 to 32-3 are connected. This corresponds to a configuration in which the connections are made wirelessly. In FIG. 16, the signal transmission between the second signal processing unit 305 and the first signal processing units 304-1 to 304-4 guarantees that there is no signal interference because the transmission medium is wired. Therefore, even in FIG. 32, when the wired portion is completely reworked wirelessly, each satellite base station device 32 is also used in a wireless signal between the control base station device 31 and the satellite base station devices 32-1 to 32-3. This ensures a situation where there is no signal interference between -1 to 32-3. However, in the third embodiment, signal processing for signal separation is performed in the terminal station device 33 (in the case of downlink) or the control base station device 31 (in the case of uplink). It is a point of the present embodiment that it is not necessary to completely separate the signal in the part. However, if signals are transmitted and received between the control base station device 31 and the satellite base station devices 32-1 to 32-3 in a non-directional manner, a line gain corresponding to the first singular value cannot be obtained. Between the base station device 31 and the satellite base station devices 32-1 to 32-3, transmission / reception signal processing is performed using the first singular vector for utilizing the virtual transmission path corresponding to the first singular value as a transmission / reception weight. .

FIG. 34 shows signal processing between the control base station apparatus 31 and the satellite base station apparatus 32 and signal processing between the satellite base station apparatus 32 and the terminal station apparatus 33 in the third embodiment of the present invention. The outline of is shown. In FIG. 34, reference numeral 31 denotes a control base station apparatus, reference numeral 32 denotes a satellite base station apparatus, reference numeral 33 denotes a terminal station apparatus, and reference numeral 52 denotes a wireless communication system, each corresponding to the same number in FIG. There are a plurality of satellite base station devices 32, and a plurality of the satellite base station devices are shown vertically in the center of FIG. In Figure 34, the general base station apparatus 31 and the satellite base station apparatus 32 performs communication using the frequency f _1, the satellite base station apparatus 32 and the terminal station is communicating at the frequency f _2. However, assuming communication in FDD (Frequency Division Duplex), the control base station apparatus 31 and the satellite base station apparatus 32 communicate at frequencies _f1-1 and _f1-2 , and The station device 32 and the terminal station device may be communicating with each other at the frequencies _f2-1 and _f2-2 . Here, for the sake of simplicity, a case will be described in which uplink and downlink of each section use the same frequency, assuming TDD (Time Division Duplex).

Also, as can be seen in Figure 34, except for the differences frequencies _{f 1} and _{f 2,} generally has a folded state symmetric, satellite base station apparatus 32-1 to 32-3 are first in FIG. 16 1 corresponds to the signal processing units 304-1 to 304-4, but if anything, the supervising base station apparatus 31 is similar to the terminal station apparatus 302 of FIG. Has a configuration corresponding to the terminal station device 302. Therefore, in the following description in which the circuit configuration is cited, the description is made with reference to the reference numerals shown in FIG.

  Further, for example, when using OFDM, it is necessary to perform individual signal processing in each subcarrier. However, in order to simplify the description, the notation for identifying the subcarrier in each signal and transmission / reception weight is omitted. I do.

FIG. 34 particularly shows signal processing when transmitting a signal from the control base station apparatus 31 to the terminal station apparatus 33. Although not explicitly shown here, it should be noted that all of these signal processes are performed for each subcarrier as in the above-described processes. However, if the same transmission / reception weight can be used within the entire frequency bandwidth as described later, it is also possible to extend the signal processing on the time axis for each sampling. First, the general base station apparatus 31 _{N SDM} system signal sequences _S 1 to S _N (for convenience of letter notation subscript here, here, the _{N = N SDM)} When sending, the general base station apparatus 31 in the second transmission signal processing circuit 148, in addition to the generation of a signal sequence S ₁ to S _N as a general transmission signal processing as the signal processing illustrated in (i) in FIG. 34, between the satellite base station apparatus 32 _Assuming that a transmission weight matrix for signal separation is W _tx , signal conversion is performed by the following equation (30).

Here, as described in FIG. 25, in the signal processing corresponding to the second transmission signal processing circuit in FIG. 25, it is not always necessary to introduce the transmission weight matrix W _tx , and W _tx is regarded as a unit matrix. , The transmission signal sequences t _{1 to} t _N may remain as the signal sequences S _{1 to} S _N. Next, the directivity formation corresponding to the first singular value for each satellite base station apparatus 32-1 to _{32-N SDM.} Specifically, from each antenna of the general base station apparatus 31, to the channel matrix for each antenna of the k _S satellite base station apparatus 32 of interest, given the first right singular vector at the time of singular value decomposition of the matrix The transmission weight vectors w _{tx, kS, 1 to} w _{tx, kS, Nc} (here, Nc is the number of antenna elements included in the control base station device for convenience of notation) are expressed by Expression (31) ((FIG. 34) ii) Multiply the transmission signal _tkS as shown in (ii)).

As shown in Expression (32) ((iii) in FIG. 34), the _NSDM system signal addressed to each satellite base station device 32 is added and synthesized for each antenna element, and a time axis signal is generated.

Further, the IFFT & GI adding circuit 813 of the control base station apparatus 31 performs an IFFT conversion on the signal on the frequency axis shown in Expression (32) and converts it into a signal on the time axis (strictly, for example, OFDM). For example, signal processing such as insertion of a guard interval and a wave forming system is also included, but the description is omitted here). The D / A converter 814 of the control base station apparatus 31 performs D / A conversion on a result of the IFFT conversion to generate an analog baseband signal. The mixer 816 of the general base station apparatus 31, an analog baseband signal, multiplies the signal from the local oscillator 815 of the frequency of f _1, upconverts the analog baseband signal to a frequency band of the signal f _1. Filter 817 of general base station apparatus 31 removes out-of-band signals from the signal in the frequency band of f _1. HPA818 of general base station apparatus 31 amplifies the signal in the frequency band of _{f 1,} and transmits the amplified signal from the antenna element 819.

Antenna elements of the satellite base station apparatus 32 receives a signal in the frequency band of f _1. Low-noise amplifier of the satellite base station apparatus 32 (LNA) amplifies the signal in the frequency band of _{f 1.} Mixers satellite base station apparatus 32 down-converts the signal in the frequency band of f ₁ into an analog baseband signal. The filter of the satellite base station device 32 removes out-of-band signals from the analog baseband signal. The A / D converter of the satellite base station device 32 converts an analog baseband signal into a digital baseband signal. The FFT circuit of the satellite base station device 32 converts the digital baseband signal into a subcarrier reception signal based on the symbol timing of the sampled signal.

Here, in the satellite base station device 32, the first left singular vector when the matrix is singular value-decomposed from each antenna of the control base station device 31 to the channel matrix for each antenna of the satellite base station device 32 of interest. The reception weight vectors w ′ _{rx, kS, 1 to} w ′ _{rx, kS, M given} by the following equations are converted into reception signal vectors r ′ _{kS, 1 to} r ′ as shown in Expression (33) ((iv) in FIG. 34). Multiply _{kS, M.} Here, M is the number of antenna elements of the satellite base station apparatus 32, and represents the number of antenna elements that perform communication with the control base station apparatus 31.

The signal R ′ _kS of the scalar amount obtained here does not estimate the transmission signal of the control base station apparatus 31, but is merely an intermediate one-dimensional signal for extracting a virtual transmission path of the first singular value. And is different from regenerative relay. However, unlike ordinary non-regenerative relay, it corresponds to generating the intermediate signal R ′ _kS and performing reception directional beam formation and transmission directional beam formation individually. In the transmission from the satellite base station apparatus 32 corresponding to the last access system to the terminal station apparatus 33 with respect to the intermediate signal R ′ _kS obtained in this manner, the last one-hop channel Signal transmission is performed using a virtual transmission path corresponding to the first singular value. Specifically, a transmission weight vector given by a first right singular vector when the matrix is singular-value-decomposed from each antenna of the satellite base station apparatus 32 of interest to a channel matrix for each antenna of the terminal station apparatus 33 The received signal vector R ′ _kS is multiplied by w ′ _{tx, kS, 1 to} w ′ _{tx, kS, M} as shown in Expression (34) ((v) in FIG. 34).

Here, the number of antenna elements for the terminal station apparatus 33 in the satellite base station apparatus 32 has been described as M. However, this does not necessarily need to match the number of antenna elements on the control base station apparatus 31 side, and if it is different, it is generally used. In this case, the number M of antenna elements may be understood as the number m of antenna elements. The transmission signal vector of each subcarrier obtained in this manner is converted into a time axis signal by IFFT (strictly, for example, in the case of OFDM, signal processing such as insertion of a guard interval or a wave forming system). but also includes, here will not be described), D / a conversion to generate an analog baseband signal, which is multiplied with the signal from the local oscillator frequency f ₂ frequency band f ₂ by mixer After up-converting the signal to a signal of the same type and removing the out-of-band signal with a filter, the signal is amplified by a high power amplifier and transmitted.

Here, the equations (33) and (34) are obtained by multiplying the reception signal vector by the reception weight vector, and then multiplying the obtained R ' _kS by the transmission weight vector. _{May be} configured to multiply the transmission weight vector and the reception weight vector in advance to generate a transmission weight matrix and directly multiply the transmission weight matrix by the received signal vectors r ′ _{kS, 1 to} r ′ _{kS, M.} I do not care. However, the amount of computation at this time requires 2 × M complex multiplications when Equations (33) and (34) are performed separately, whereas the transmission weight matrix generation and matrix The complex multiplication of 2 × M ² times in total is required for both the multiplication of the vector and the multiplication, and the efficiency is better when divided into individual operations.

Next, signal processing for a signal received by the terminal station device 33 will be described. Antenna elements of the terminal station 33 851, receives a signal in the frequency band of _{f 2.} LNA852 amplifies the signal in the frequency band of _{f 2.} The mixer 854 down-converts the signal in the frequency band of f ₂ to the analog baseband signal. The filter 855 removes out-of-band signals from the analog baseband signal. A / D converter 856 converts an analog baseband signal into a digital baseband signal. The FFT circuit 857 converts the digital baseband signal into a subcarrier reception signal based on the symbol timing of the sampled signal.

Here, in the terminal station device 33, first, the first channel information estimating circuit 146 or the first channel information estimating circuit 156 determines a virtual antenna element formed by each antenna element of each satellite base station device 32 by a transmission weight vector. Channel information between the antennas of the terminal station device 33 and each antenna is obtained from the training signal added to the received signal. The first reception weight calculation circuit 147 or the first reception weight calculation circuit 157 generates a reception weight from the acquired channel information. The reception weight here may be a ZF-type or MMSE-type reception weight matrix for directly separating the signal sequence of the _NSDM system in one stage (corresponding to FIG. 22), or corresponds to the first singular value. It is also possible to use a two-stage reception weight multiplication process (corresponding to FIG. 23) in which signals are once separated into virtual transmission paths and then the mutual interference between the virtual transmission paths is removed. In the latter case, a vector vector is defined based on a channel vector between a virtual antenna element formed by each antenna element of each satellite base station apparatus 32 by a transmission weight vector and each antenna of the terminal station apparatus 33. Each of the components of the reception weight vector may be obtained by the equation (7) (weight of the maximum ratio combination), or all the absolute values may be fixed with respect to the value given by the equation (7). (Weight of equal gain combining). The reception weight is assumed to be w _{rx, kS, 1} to w _{rx, kS, Nm} (here, N _m is the number of antenna elements N _MT-Ant provided in the terminal station device for the sake of suffix notation). Assuming that the signal vector received by each antenna element of the device 33 is Rx, the signal is once separated into a virtual transmission path by the expression given by Expression (35) ((vi) in FIG. 34).

Here r _kS is the received signal of the virtual transmission channel via the first k _S satellite base station apparatus 32. At this point, since the signal is an N _SDM × N _SDM MIMO transmission signal, the second reception signal processing circuit 159 of the terminal station device 33 then performs general MIMO signal processing. More specifically, an N _SDM × N _SDM channel matrix is obtained for the N _SDM system signal sequence with reference to a known training signal added to the head of the received signal, and the received signal is detected based on the channel matrix. Perform processing. As described above, it is also possible to multiply by an inverse matrix of ZF type or a linear reception weight matrix of MMSE type, or to perform nonlinear signal processing such as MLD. Demodulation processing is performed on the N-system signals separated in this way, and the reproduced data is output to the MAC layer processing circuit 68. When a ZF-type or MMSE-type linear reception weight matrix W _rx is used as an example, the calculation shown in Expression (36) ((vii) in FIG. 34) is performed. Here are the _{N SDM} in Formula (36) and (37) in order to avoid complication represents a N.

Note that, as described above, instead of the two-stage signal processing of Expressions (35) and (36), direct signal separation using the reception signals of the antenna elements of the terminal station device 33 may be performed in one stage. It is possible. This is expressed by equation (37) using a ZF-type or MMSE-type linear reception weight matrix ~ W _rx (~ is expressed above the matrix W _rx ) which directly separates the received signal vector Rx. This corresponds to performing signal processing.

  The feature of the above-described embodiment of the present invention is that the satellite base station apparatus 32 does not have the regenerative relay function in the related art, and is not limited to the function of only normal non-regenerative relay, Intermediate processing of multiplying the reception weight vector and further multiplying the signal by the transmission weight vector is performed, and the transmission and reception weight vector basically utilizes a virtual transmission path corresponding to a plurality of first singular values effectively. This is the point that the directivity control is performed.

  FIG. 35 shows a circuit configuration of the satellite base station device according to the embodiment of the present invention. In FIG. 35, reference numeral 700 denotes an antenna element. Reference numeral 701 indicates a TDD-SW. Reference numeral 702 indicates a low noise amplifier (LNA). Reference numeral 703 indicates a mixer. Reference numeral 704 indicates a filter. Reference numeral 705 indicates an A / D converter. Reference numeral 706 indicates an FFT circuit. Reference numeral 707 indicates a reception weight multiplication circuit. Reference numeral 708 indicates a transmission weight multiplication circuit. Reference numeral 709 denotes an IFFT & GI adding circuit. Reference numeral 710 indicates a D / A converter. Reference numeral 711 indicates a mixer. Reference numeral 712 denotes a filter. Reference numeral 713 indicates a high power amplifier. Reference numeral 714 indicates a TDD-SW. Reference numeral 721 indicates an antenna element. Reference numeral 722 indicates a low noise amplifier (LNA). Reference numeral 723 indicates a mixer. Reference numeral 724 indicates a filter. Reference numeral 725 indicates an A / D converter. Reference numeral 726 indicates an FFT circuit. Reference numeral 727 indicates a reception weight multiplication circuit. Reference numeral 728 denotes a transmission weight multiplication circuit. Reference numeral 729 indicates an IFFT & GI adding circuit. Reference numeral 730 indicates a D / A converter. Reference numeral 731 denotes a mixer. Reference numeral 732 indicates a filter. Reference numeral 733 indicates a high power amplifier. Reference numeral 740 indicates a transmission weight processing unit. Reference numeral 741 denotes a local oscillator. Reference numeral 742 indicates a local oscillator. Reference numeral 743 indicates a communication control circuit. Hereinafter, for convenience of explanation, the left side will be described as performing communication with the centralized base station apparatus 31, and the right side will be described as performing communication with the terminal station apparatus 33.

  First, the downlink direction (from the control base station device 31 to the satellite base station device 32 to the terminal station device 33) will be described. The signal received by each antenna element 700 is input to the low noise amplifier 702 via the TDD-SW 701. Here, the switching of the TDD-SW 701 is performed by the communication control circuit 743 in accordance with the transmission / reception timing. The weak signal input to the low noise amplifier 702 is amplified, the amplified signal is multiplied by the local oscillation signal output from the local oscillator 741 in the mixer 703, and the amplified signal is converted from a radio frequency signal to a baseband signal. Down-converted to Since the down-converted signal includes frequency components outside the frequency band to be received, the filter 704 removes the out-of-band components. The signal from which the out-of-band component has been removed is converted into a digital baseband signal by the A / D converter 705. All the digital baseband signals are input to the FFT circuit 706, and a signal on the time axis is converted into a signal on the frequency axis at a predetermined symbol timing determined by a timing detection circuit (not shown). Signal). The signal separated into each frequency component is input to the reception weight multiplication circuit 707 and also to the transmission / reception weight processing unit 740-1.

  The reception weight multiplying circuit 707 multiplies the reception weight vector of the first singular value in correspondence with the virtual transmission line carried by the satellite base station device 32, and obtains a single-system scalar-like signal from a vector-like signal by a plurality of antenna elements. Converted to a signal. This signal is input to the transmission weight multiplication circuit 708, and the transmission weight multiplication circuit 708 associates the transmission weight vector with the virtual transmission path corresponding to the first singular value from the satellite base station apparatus 32 to the terminal station apparatus 33 of interest. Are multiplied to generate signals corresponding to the respective antenna elements 721. These signals are input to the IFFT & GI adding circuits 709 of the respective antenna element systems. And processing such as insertion of guard intervals and waveform shaping between OFDM symbols (or between blocks of block transmission in the case of SC-FDE (Single-Carrier Frequency Domain Equalization)) is performed. D / A converter 710 converts digital sampling data to baseband analog It is converted to issue. Further, each analog signal is multiplied by the local oscillation signal input from the local oscillator 742 by the mixer 711, and is up-converted into a radio frequency signal. Here, since the up-converted signal includes a signal in a frequency component outside the band of the channel to be transmitted, the filter 712 removes the frequency component outside the band and generates an electrical signal to be transmitted. . The generated signal is amplified by the high power amplifier 713 and transmitted from the antenna element 721 via the TDD-SW 714.

  Next, the uplink direction (from the terminal station device 33 to the satellite base station device 32 to the control base station device 31) will be described. The signal received by each antenna element 721 is input to the low noise amplifier 722 via the TDD-SW 714. Here, the switching of the TDD-SW 714 is performed by the communication control circuit 743 in accordance with the transmission / reception timing. The weak signal input to the low noise amplifier 722 is amplified, and the amplified signal is multiplied by a local oscillation signal output from the local oscillator 742 by a mixer 723. The amplified signal is converted from a radio frequency signal to a baseband signal. Down-converted to Since the down-converted signal includes frequency components outside the frequency band to be received, the filter 724 removes the out-of-band components. The signal from which the out-of-band component has been removed is converted into a digital baseband signal by the A / D converter 725. All the digital baseband signals are input to the FFT circuit 726, and a signal on the time axis is converted into a signal on the frequency axis at a predetermined symbol timing determined by a timing detection circuit (not shown). Signal). The signals separated into the respective frequency components are input to the reception weight multiplying circuit 727 and also to the transmission / reception weight processing unit 740-2.

  The reception weight multiplying circuit 727 multiplies the reception weight vector of the first singular value in correspondence with the virtual transmission line carried by the satellite base station device 32, and obtains a single-system scalar-like signal from a vector-like signal by a plurality of antenna elements. Converted to a signal. This signal is input to the transmission weight multiplying circuit 728, and the transmission weight multiplying circuit 728 sets the transmission weight corresponding to the virtual transmission path corresponding to the first singular value from the satellite base station apparatus 32 of interest to the control base station apparatus 31. Vectors are multiplied to generate signals corresponding to the respective antenna elements 700. These signals are input to the IFFT & GI adding circuits 729 of the respective antenna element systems. The signal is converted into a signal, and processing such as insertion of guard intervals and waveform shaping between OFDM symbols (or between blocks of block transmission in the case of SC-FDE (Single-Carrier Frequency Domain Equalization)) is performed. , D / A converter 730 converts the digital sampling data to a baseband analog signal. It is converted into grayed signal. Further, each analog signal is multiplied by a local oscillation signal input from a local oscillator 741 by a mixer 731 and up-converted into a radio frequency signal. Here, since the up-converted signal includes a signal in a frequency component outside the band of the channel to be transmitted, the filter 732 removes the frequency component outside the band and generates an electrical signal to be transmitted. . The generated signal is amplified by the high power amplifier 733 and transmitted from the antenna element 700 via the TDD-SW 701.

  The transmission / reception weight vector multiplied by the reception weight multiplication circuit 707 and the transmission weight multiplication circuit 728 is calculated by the transmission / reception weight processing unit 740-1. The transmission / reception weight vector multiplied by the reception weight multiplication circuit 727 and the transmission weight multiplication circuit 708 is calculated by the transmission / reception weight processing unit 740-2. The receiving weight vector is calculated based on the channel information acquired using the downlink and uplink FFT circuits, and the transmission weight vector in the reverse direction is calculated by performing a calibration process. In the transmission / reception to / from the control base station apparatus 31, once the channel information is obtained because the propagation path is fixed and the line of sight is at a high place, the value may be continuously used thereafter. It is possible.

  Therefore, it is possible to repeatedly perform channel estimation before the start of service operation, and to use a highly accurate channel estimation result (in this case, relative channel information is used) by averaging and a transmission / reception weight vector calculated therefrom. It is possible. On the other hand, since channel information between the satellite base station device 32 and the terminal station device 33 changes due to movement around the terminal station device 33 and surroundings and environmental fluctuations, it is necessary to obtain channel information and update weight vectors in real time. Becomes This channel information acquisition (channel feedback) method may use any means. The transmission / reception weight processing units 740-1 and 740-2 calculate the transmission / reception weight vector of the virtual transmission path corresponding to the first singular value by using a method such as singular value decomposition, for the channel information thus obtained. The calculated values are input to the reception weight multiplication circuits 707 and 727 and the transmission weight multiplication circuits 708 and 728. The communication control circuit 743 grasps the transmission / reception timing and the symbol timing, and issues various instructions to the TDD-SWs 701 and 714 and various circuits. The reason why the communication control circuit 743 grasps such a timing can also be grasped by synchronizing with the control base station device 31 in communication with the control base station device 31, and can be determined by using the GPS or other wireless system (for example, This may be performed using information obtained from a signal from a macro cell base station).

  As described above, the wireless communication system 52 according to the third embodiment includes an integrated base station device 31 (first wireless station device), a plurality of satellite base station devices 32 (second wireless station device), and a terminal. And a station device 33 (third wireless station device). The control base station device 31 includes a first wireless station signal processing unit that performs wireless communication with a plurality of satellite base station devices 32 via the first antenna element group. The satellite base station apparatus 32 executes wireless communication with the control base station apparatus 31 via the second antenna element group, and executes wireless communication with the terminal station apparatus 33 via the third antenna element group. It has two radio station signal processing units.

  The second wireless station signal processing unit of the satellite base station device 32 (second wireless station device) of the third embodiment is used for wireless communication between the first antenna element group and the second antenna element group. The reception weight vector for the MIMO channel matrix is defined as at least one of a first right singular vector and a first left singular vector corresponding to a first singular value of the MIMO channel matrix, or an approximate solution of the first right singular vector and the first left It is calculated based on at least one of the approximate solutions of the singular vectors. The second radio station signal processing unit generates a transmission signal sequence by multiplying a signal received via the second antenna element group by a reception weight vector. The second wireless station signal processing unit transmits a signal based on the generated signal sequence to the terminal station device 33 (third wireless station device) via the third antenna element group.

  The second radio station signal processing unit of the satellite base station device 32 (second radio station device) according to the third embodiment includes a fourth antenna element group of the terminal station device 33 (third radio station device) and a satellite. The transmission weight vector for the MIMO channel matrix used for wireless communication with the third antenna element group of the base station device 32 (second wireless station device) is represented by the first right value corresponding to the first singular value of the MIMO channel matrix. Calculated based on at least one of the singular vector and the first left singular vector, or at least one of the approximate solution of the first right singular vector and the approximate solution of the first left singular vector, via the third antenna element group A plurality of signal sequences are generated by multiplying the received signal by a transmission weight vector, and a signal based on the plurality of signal sequences is transmitted through a second antenna element group. And transmits to end station device 33 (third radio station).

  The first wireless station signal processing unit of the wireless communication system 52 according to the third embodiment calculates a transmission weight vector for a MIMO channel matrix used for wireless communication between the first antenna element group and the second antenna element group, At least one of a first right singular vector and a first left singular vector corresponding to a first singular value of the MIMO channel matrix, or at least one of an approximate solution of the first right singular vector and an approximate solution of the first left singular vector , A signal sequence is generated for each second wireless station apparatus, the generated signal sequence is multiplied by a transmission weight vector corresponding to each signal sequence, and the multiplication result is added and synthesized for all signal systems. Then, a signal based on the added and combined signal sequence is transmitted to the satellite base station device 32 (second wireless station device) via the first antenna element group.

  The first wireless station signal processing unit of the wireless communication system 52 according to the third embodiment calculates a reception weight vector for a MIMO channel matrix used for wireless communication between the first antenna element group and the second antenna element group, At least one of a first right singular vector and a first left singular vector corresponding to a first singular value of the MIMO channel matrix, or at least one of an approximate solution of the first right singular vector and an approximate solution of the first left singular vector Is calculated for each satellite base station device 32 (second wireless station device) based on the above equation, and a signal received via the first antenna element group is multiplied by a reception weight vector for each second wireless station device. A signal sequence of a plurality of systems is generated, and a residual interference component is removed from a signal based on the signal sequences of the plurality of systems.

  Accordingly, the control base station device 31, the satellite base station device 32, the terminal station device 33, the wireless communication system 52, and the wireless communication method of the third embodiment increase the transmission capacity by MIMO in an environment where the line-of-sight environment is dominant. It is possible to do. The control base station device 31, the satellite base station device 32, the terminal station device 33, the wireless communication system 52, and the wireless communication method according to the third embodiment can increase the transmission capacity by non-regenerative re-beamforming relay. Becomes In the control base station device 31, the satellite base station device 32, the terminal station device 33, the wireless communication system 52, and the wireless communication method according to the third embodiment, a plurality of signal sequences can be spatially multiplexed and transmitted with desired characteristics. Will be possible.

  In the first embodiment, the base station device 70, the terminal station device 60, the wireless communication system 50, and the wireless communication method use a plurality of virtual transmission paths corresponding to a first singular value for an access system, and perform spatial multiplexing. It is also possible to do. However, since the coordinates of the terminal station device 60 cannot be fixedly designed, it is preferable to spatially multiplex a plurality of first signal processing units 304 on the base station device 70 side at distant locations. For this reason, when a function corresponding to the first signal processing unit 304 is arranged as the satellite base station device 32, it is preferable to relay wirelessly rather than routing the space between them. In realizing this usage form, signal processing in which only beamforming is performed in a relay station while performing non-regenerative relay is effective, and a method for realizing this is defined.

[Fourth embodiment]
[Train moving cell using a plurality of virtual transmission paths corresponding to the first singular value]
(Outline of train moving cell and basic principle)
As described in the third embodiment, in the future wireless access for 5G (fifth generation), instead of accommodating traffic of a large number of users in the train in the macro cell, these traffics are transmitted in the train. Aggregating these and collectively detouring the user data to the wired network side at the wireless entrance and offloading is effective in effectively utilizing the frequency resources of the macro cell and improving the total transmission capacity of the entire system. is there. Here, no distinction is made between trains and trains, and all vehicles such as conventional lines and Shinkansen are referred to as trains.

  However, in the case of a train, the traveling speed is generally high, and in the case of a Shinkansen, the speed exceeds 300 [km / h]. Must. In particular, the use of high frequency bands such as millimeter waves is inevitable in order to provide large capacity lines. Considering that the output is very small compared to the microwave band, and that high output is expensive, it is necessary to secure the line gain by adopting a large-scale antenna using a large number of low output antenna elements. In particular, in consideration of the traveling speed of the train, in order to reduce the frequency of handovers and reduce the control load, an area of about 500 [m] intervals must be covered by the entrance-side base station device. In securing the gain of a large-scale antenna, it is necessary to perform accurate feedback of channel information and perform high-accuracy directivity control.However, the time variation of the channel is generally large due to the fast moving speed of the train, At this time, it is necessary to follow the fluctuation with high accuracy and to form the directivity at high speed.

  As mentioned above, generally, the time variation of the MIMO channel tends to be relatively small in the virtual transmission path corresponding to the larger singular value obtained when the singular value decomposition of the channel matrix is performed. Conversely, in a virtual transmission path corresponding to a small singular value, channel time variation tends to be very large. Here, since it is assumed that the transmitting and receiving stations are in line-of-sight environments, the virtual transmission line corresponding to the first singular value is a transmission line corresponding to the line-of-sight wave. In this case, considering that the train line is generally linear, a highly directional antenna is installed near the upper part of the line (such as an overhead pole) with directivity oriented in a direction parallel to the line. If the wireless signals are transmitted and received in a manner facing each other in the front direction, the line-of-sight environment can be stably secured for a long time. The channel information between the antenna elements changes very rapidly because the moving direction of the train and the arrival direction of the radio wave are substantially parallel. On the other hand, the relative relationship (complex phase difference) of the channel information of each path between the antenna elements can be almost approximated to a plane wave, and thus hardly changes with time. In other words, the channel itself fluctuates at a high speed, but the transmission / reception weight of the virtual transmission line corresponding to the first singular value itself changes extremely slowly, and even if a constant transmission / reception weight value is kept for a certain period of time, it is almost ideal It is possible to maintain a proper directivity.

  On the other hand, the virtual transmission path corresponding to the second and subsequent singular values uses many reflected waves positively, and incident waves from various directions that are not parallel to the moving direction are dominant. Become a target. At this time, if waves from various angles are combined, the situation is clearly different from the plane wave, and the relative relationship of the channel information corresponding to each path between each antenna element fluctuates significantly with time. I do. Therefore, it is necessary to update the transmission / reception weights at high speed in order to follow the fluctuation of the relative relationship of the channel information. If the update cannot be performed at high speed, an appropriate directivity cannot be formed. Here, even if implicit feedback is introduced to reduce the overhead of feedback of channel information, training signals for channel estimation must be exchanged very frequently in order to follow this rapid change in channel information. And it is difficult to achieve the required accuracy in practice.

  Here, various transmission / reception circuits (high-power amplifier, low-noise amplifier, filter, mixer, etc.) provided in the terminal station device mounted on the train and the base station device of the wireless entrance are complex circuits that change due to temperature characteristics and the like. It is assumed that the phase rotation amount can be always treated as substantially constant without time variation by a highly accurate calibration process. In this case, paying attention to a certain train, if the line to be passed is determined, the movement route of the train (strictly speaking, each antenna element of the terminal station device mounted on the train) shows one-dimensional displacement with extremely high accuracy. If the coordinates on the one-dimensional movement route can be grasped, the channel information when there is a train at the same place is almost always the same every time.

  Of course, for example, assuming 80 GHz or the like, the wavelength is 3.75 mm, and a deviation of coordinates of only 1 mm results in a coordinate error of 25% or more of the wavelength, and the absolute channel information greatly changes. This is relative channel information required when calculating the transmission / reception weight. In the case of a line-of-sight wave arriving as a plane wave, as long as the plane and the plane on which the antenna elements are two-dimensionally arranged are substantially parallel, the relative channel information is not significantly affected by such a coordinate error. Generally, regarding the fluctuation of the transmission / reception weight, the transition in the direction perpendicular to the line-of-sight direction (the direction parallel to the straight line connecting the transmission / reception antenna) and the transition in the direction parallel to the line-of-sight direction are more vertical. In this case, the fluctuation amount of the transmission / reception weight becomes larger. Of course, the play between the track and the width of the train wheels may be several millimeters or more, so they do not pass exactly the same coordinates, and there is a slight error in the direction perpendicular to this line of sight. Become. However, in the case of a virtual transmission path corresponding to the first singular value as described above, it can be approximated by a plane wave, and a change in transmission / reception weight is within an allowable range if the error is of such a level. On the other hand, when transiting in a direction parallel to the line-of-sight direction, the accuracy of determining the position of the train is low due to the high moving speed. However, in the case of a plane wave, the transmission / reception weight is completely insensitive compared to the case where the train transitions in the vertical direction, and can sufficiently cope with it.

  Although the transmission / reception weight itself may have a small time variation, the complex phase of a signal synthesized using the transmission / reception weight fluctuates with the coordinates, which is observed as a Doppler shift. However, since the Doppler shift itself does not essentially differ from the frequency error, even if, for example, OFDM is used, if it is smaller than the subcarrier interval, it does not break the orthogonality of OFDM. Therefore, it is possible to cope with this by executing the compensation of the residual frequency error by a tracking process or the like.

  Therefore, first, a mechanism for determining the position (coordinates) of the train in one-dimensional train movement with high accuracy is introduced. For example, a mark for identifying some coordinates is arranged at a predetermined interval beside the track, the mark is photographed by a camera or the like, and the coordinates are measured by performing image processing on the video. For example, a mark of barcode-like geometric coordinate information is arranged at an interval of 10 [m], an image captured by a camera is analyzed, and the coordinates of the mark are specified from the barcode. There is a time lag between the time of actually passing the mark and the time at which the coordinates of the mark can be specified by image analysis, but it is possible to obtain in advance how long this time lag will be. On the other hand, since this time lag is estimated to be substantially the same for all landmarks, if the landmarks are set at intervals of 10 [m], the difference between the time when the landmarks are recognized by image analysis and From the relationship of the moving distance of 10 [m], it is possible to estimate the moving speed at that moment with high accuracy. Furthermore, from the traveling speed and the time of the above-mentioned time lag, it is possible to estimate with high accuracy where the train is actually located at the moment when the coordinates of the landmark are determined. Can be estimated.

  In this way, it is possible to accurately acquire instantaneous coordinates in one-dimensional movement. Here, a barcode is shown as an example, but an optical signal from an LED or the like may be used, or the coordinates may be specified using an electromagnetic signal such as a wireless tag. Although the current GPS coordinate estimation accuracy is not very high, when GPS correction value information is obtained via a wireless line (different frequency band or through a different system) separately set as Assisted GPS, It is known that the positioning accuracy of GPS can be suppressed to about 5 to 10 [m]. Further, in the case of a train, the traveling route is one-dimensional in order to travel on the track, and even when using a GPS or the like, the accuracy is further improved if the location is limited to a location on the one-dimensional route (on the track). . It is said that it will be possible to estimate the coordinates with an accuracy of about several centimeters in the coordinate specification using the quasi-zenith satellite in the future, and it is naturally possible to use such a future GPS function. Furthermore, like a speedometer or distance meter of a car, it is also possible to obtain the travel distance from the number of rotations of the wheels, perform high-precision coordinate correction when stopping at a station platform, etc., and from that point If the combination is used, such as using the moving distance of the GPS information and other coordinate acquisition means together, the measurement accuracy is dramatically improved. With this configuration, highly accurate coordinate estimation can be performed without using a camera and an image recognition function as described above. The present embodiment can be applied similarly even if coordinate information is acquired by any of these means.

  In the above description, the case where the coordinates are estimated on the train side has been described, but similarly, it is necessary to specify the coordinates of the train as the communication partner on the base station apparatus side. Basically, the base station device acquires the coordinates of the train by using the coordinate specifying means of the running train connected to the base station device via the wired network. When the coordinates of the train are specified on the wired network side in this way, for example, a configuration may be employed in which a light emitter and its light receiver are arranged beside the track, and that light is blocked when passing through the train. Alternatively, a configuration may be adopted in which a light emitter is installed on the train side, and the light from the light emitter is detected by a light receiver beside the track. Although it is impractical in terms of cost to install a large number of cameras beside the track, it is possible to obtain the coordinates with high accuracy by other methods. Alternatively, if the coordinate information obtained by the train can be notified to the base station apparatus side through a low-delay wireless line (different frequency band or through a different system) which is separately set, a wired network can be used. It is also possible to specify the coordinates of the train on the base station apparatus side without using the coordinate information acquisition means connected in this way.

  Next, in such a state that the coordinates of the train can be specified, for example, during a time zone outside normal service such as midnight, or before starting service operation, while running an actual vehicle, the channel matrix at that time is calculated. The information is acquired in advance in association with the coordinate information and stored in the base station device or the terminal station device. If the average value is obtained by performing measurements several times, the correspondence between the channel matrix and the train coordinates with higher accuracy can be obtained. When calculating the average value, instead of the absolute channel information as measured, the relative channel information based on the complex phase of the reference antenna is used, and the transmission / reception weight is calculated for the relative channel information. I do. The measurement of the channel matrix is performed not only on the train side but also on the base station device on the wired network side. Alternatively, the information of the uplink channel may be obtained from the information of the downlink channel by a method such as implicit feedback.

  As described above, before the actual service operation starts, a database of the relationship between train coordinates and uplink and downlink channel matrices is constructed in advance. To be precise, based on the channel matrix, the transmission / reception weight of the virtual transmission path corresponding to the first singular value is calculated in advance, and the transmission / reception weight is stored in the database. For example, on the train side, a transmission weight vector used for uplink and a reception weight vector used for downlink of each virtual transmission path are stored in a database in association with discrete coordinate information. Similarly, on the base station apparatus side, the reception weight vector used in the uplink of each virtual transmission path and the transmission weight vector in the downlink are stored in a database in association with discrete coordinate information.

  FIG. 36 shows an outline of the configuration of the wireless communication system according to the fourth embodiment. In the figure, reference numeral 600 denotes a base station device, reference numerals 601-1 to 601-8 denote first signal processing units of the base station device, reference numerals 602-1 to 602-8 denote geometric coordinate information, and reference numeral 603 denotes a train. Reference numeral 604 denotes a terminal station device, reference numeral 605 denotes a camera, reference numeral 606 denotes a train-side weight matrix / coordinate database, and reference numeral 607 denotes a base station-side weight vector / coordinate database. Note that, when any of the first signal processing units 601-1 to 601-8 is indicated, the first signal processing unit 601 will be described.

  The train 603 has a terminal station device 604 provided with a large number of antenna elements above the head of the leading vehicle in the traveling direction or above the rear of the last vehicle. The installation location here may be configured to transmit and receive signals from the inside of the train 603 through a window, or may be installed outside the train 603 and further covered with a cover (radome) for reducing the effects of rain and wind. good. In addition, the base station device 600 includes a plurality of first signal processing units 601-1 to 60-8 such as the first signal processing units 304-1 to 304-4 illustrated in FIG. 16 in the first embodiment. The first signal processing units 601-1 to 60-8 are physically installed at a large distance. For example, a large number of the first signal processing units 601-1 to 60-8 are provided at intervals of 1 m or 2 m. In addition to these first signal processing units 601-1 to 601-8, base station apparatus 600 also includes second signal processing unit 305 shown in FIG. The first signal processing units 601-1 to 601-8 included in the base station device 600 are installed on, for example, overhead wire poles provided to cross the line of the train 603 in a direction perpendicular to the line. . The first signal processing units 601-1 to 601-8 include a large number of antenna elements and circuits for signal processing, similarly to the first signal processing units 304-1 to 304-4 shown in FIG.

  The terminal station device 604 also includes a large number of antenna elements. The antenna elements of the terminal station apparatus 604 and the antenna elements of the first signal processing units 601-1 to 601-1-8 each have a configuration in which directivity is enhanced in a forward direction as viewed from the antenna element in order to secure a line gain. Since the movement of the train 603 may be accompanied by a curve instead of a perfect straight line, it is not necessarily sufficient that the directivity gain is excessively high. When viewed from the terminal station device 604 installed on the train 603, the base station device 600 It is ideal that the antenna element has a high directivity gain within a range where the directivity gain does not decrease with respect to the direction in which the terminal station device 604 of the train 603 exists as viewed from the base station device 600. is there.

  In the example shown in FIG. 36, a camera 605 is mounted on the train 603, and a device that captures the geometric coordinate information 602 on the side of the track and grasps the coordinates of the train 603 by an image analysis function not specified in the drawing is used as a train. 603 is provided. Further, the base station device 600 includes a weight vector / coordinate database 607, and the terminal station device 604 of the train 603 includes a weight matrix / coordinate database 606. The geometric coordinate information 602-1 to 602-8 are installed on the side of the track at substantially equal intervals, and are arranged at positions where images can be taken by the camera 605 mounted on the train 603. The geometric coordinate information 602-1 to 602-8 includes coordinate information indicating the coordinates where the geometric coordinate information 602-1 to 602-8 is set or geometric coordinate information 602-1 to 602-8. Is recorded in a bar code or the like.

  For example, when the train 603 captures geometric coordinate information 602-1 to 602-8 with the camera 605 while traveling, and recognizes the image of the geometric coordinate information 602-7, the geometric coordinate information The coordinates of the geometric coordinate information 602-7 are specified from the information of 602-7. Actually, if there is a time lag in the image recognition, the time ΔT of the time lag is grasped at the time of factory shipment or product design. Since the geometric coordinate information 602-1 to 602-8 are sequentially image-recognized by the camera 605, the geometric coordinate information 602-8 is recognized at a certain time t1 [s] and another time t2 [s]. Assuming that the geometric coordinate information 602-7 is recognized by the following equation, v = x / for the distance x [m] between the geometric coordinate information 602-8 and the geometric coordinate information 602-7. At (t2−t1), the second speed [m / s] is grasped, and at time t2 + Δt at which a minute time Δt has elapsed from time t2, the coordinates ahead of the coordinates of the geometric coordinate information 602-7 by v × ΔT [m]. It is determined that it is located in.

  On the other hand, before the service operation starts, the train 603 runs on a predetermined track while transmitting the training signal transmitted from each of the first signal processing units 601-1 to 601-8 of the base station device 600 to the terminal station. The signal is received by each antenna element of the device 604, and channel information of the MIMO channel is obtained. Based on the obtained channel information, the reception weight vector and the transmission weight vector used in the communication using the virtual transmission path corresponding to the first singular value are converted into the first signal processing units 601-1 to 601-8. For each time, and this is recorded in the weight matrix / coordinate database 606. When the reception weight vector and the transmission weight vector are acquired a plurality of times, it is possible to improve the accuracy by obtaining an average value or the like. In addition, at the time of this averaging, for example, the transmission weight vector of the antenna element included in the terminal station apparatus 604 is determined based on the complex phase of the transmission / reception weight vector of a predetermined antenna element among the antenna elements included in the terminal station apparatus 604 It is preferable to perform processing capable of averaging even information acquired at different times, such as using each component as a relative transmission / reception weight.

  A similar database is also created on the base station apparatus 600 side, and stored in the weight vector / coordinate database 607. In the above description, the case where the training signal is transmitted from the base station apparatus 600 side has been described, but the coordinates and time information of the train 603 are recorded on the train 603 (or the terminal station apparatus 604) side by any method described later, A training signal is transmitted from the terminal station apparatus 604 side, received by each antenna element of the first signal processing units 601-1 to 601-8 of the base station apparatus 600, and channel information of the MIMO channel is received together with reception time information. Record. Based on the recorded time information and the coordinate information of the train, and the relationship between the time information and the channel information, the reception for the predetermined train coordinates used in the communication using the virtual transmission path corresponding to the first singular value is performed. The weight vector and the transmission weight vector are obtained for each of the first signal processing units 601-1 to 601-8, and are recorded in the weight vector / coordinate database 607. Alternatively, as long as the train 603 side can acquire the downlink channel information, it is also possible to acquire the uplink channel information using the calibration technique, so that the training signal transmitted from the base station apparatus 600 side can be acquired. After acquiring one set of channel information on the train 603 side, the channel information of the uplink is acquired by a calibration process offline, and the first signal processing units 601-1 to 601- The transmission weight vector and the reception weight vector may be calculated for every 8, and may be stored in the weight vector / coordinate database 607 as a database.

  In the above processing, information obtained on the base station apparatus 600 side and information obtained on the terminal station apparatus 604 side are comprehensively processed and recorded in the weight matrix / coordinate database 606 and the weight vector / coordinate database 607. In order to generate the information, necessary information is once aggregated in an environment such as another computer, and the information processed there is separately stored in the weight vector / coordinate database 607 of the base station apparatus 600 and the terminal station apparatus 604. A configuration in which the data is recorded in the weight matrix / coordinate database 606 may be used.

  Here, the intervals between the coordinates corresponding to the transmission / reception weight vectors in the weight matrix / coordinate database 606 and the weight vector / coordinate database 607 match the intervals between the positions where the geometric coordinate information 602-1 to 602-8 are installed. There is no necessity. As described above, at any position between the geometric coordinate information 602-1 to 602-8, the coordinate information can be obtained from the moving speed and the elapsed time of the train, and the channel information obtained at the arbitrary position It is also possible to calculate the transmission / reception weight vector at that arbitrary position based on the following equation. Can be set to

  In the weight matrix / coordinate database 606 of the terminal station apparatus 604 and the weight vector / coordinate database 607 of the base station apparatus 600, transmission / reception weight vectors for discrete coordinates are recorded. Needs to acquire the transmission / reception weight vector between the coordinates corresponding to the discrete transmission / reception weight vectors. However, if the step size of the coordinates corresponding to the transmission / reception weight vector / matrix recorded in the weight matrix / coordinate database 606 and the weight vector / coordinate database 607 is sufficiently short, the transmission / reception weight vector / matrix corresponding to each of the continuous coordinates is used. If the difference is sufficiently small, it does not matter much if the same transmission / reception weight vector / matrix is continuously used until the coordinates where the next transmission / reception weight vector / matrix is recorded. If there is no restriction on the storage capacity of the weight vector / coordinate database 606 and the weight vector / coordinate database 607, the same transmission / reception weight vector / matrix can be continuously used up to the newly recorded coordinate point. It is preferable to record the transmission / reception weight vector / matrix of the coordinate information in the weight vector / coordinate database 606 and the weight vector / coordinate database 607. However, when the storage capacity is limited, for example, interpolation interpolation is performed in a section between the coordinates of two points by using a transmission / reception weight vector / matrix corresponding to each of the coordinates of two consecutive points, and approximation is performed. Alternatively, the transmission / reception weight vector / matrix in the section may be calculated.

  Note that when the train 603 is moving at high speed, a Doppler shift occurs in the frequency of the received signal observed by the base station device 600 and the terminal station device 604. Therefore, in the above description, the channel information that is measured in advance before the service operation starts includes the influence of the Doppler shift that depends on the traveling speed of the train 603, as a matter of course. In this Doppler shift, a wavelength shift occurs due to the frequency offset, and thus the value of the channel information itself varies depending on the speed of the train 603. In the operation of the ordinary train 603, it is expected that the vehicle will run at the same speed at the same place. However, in practice, the possibility of decelerating due to some operational trouble cannot be denied. Deterioration of the communication quality of the entrance line that would hinder the user in 603 from accessing the Internet must be avoided.

  However, each of the antenna elements of each of the first signal processing units 601-1 to 601-8 of the base station apparatus 600 and each of the antenna elements of the terminal station apparatus 604 correspond to the first singular value. When using a virtual transmission path, it is assumed that the antenna elements are concentrated in a very narrow area. In this case, the difference in path length between the antenna elements of the first signal processing units 601-1 to 601-8 of the base station apparatus 600 and the antenna elements of the terminal station apparatus 604 is small. Are set, and the viewing directions of the respective antenna elements are in the same direction with very high accuracy. In this case, the amount of Doppler shift generated in each antenna element is almost the same value, and the complex phase difference of the relative channel state between each antenna element is a value that does not depend on the moving speed of the train 603 with high accuracy. . That is, the channel information (and the channel vector having the channel information as a component) has different values due to the moving speed of the train 603, but the transmission / reception weight vector itself corresponding to the channel vector does not depend on the moving speed of the train 603. Will be. Therefore, it is possible to calculate the moving speed of the train 603 by the above-described method, and also to estimate the Doppler shift amount from the moving speed and the relationship of the coordinates, and to correct the channel vector for an arbitrary moving speed. Although it is possible to do so, the wireless communication system according to the fourth embodiment in which the virtual transmission path corresponding to the first singular value is positively used without performing the correction processing up to that point, the Doppler shift It is possible to use the transmission / reception weight vectors stored in the database without correction.

(Overview of Signal Processing in Wireless Communication System of Fourth Embodiment)
Hereinafter, an overview of signal processing performed in the wireless communication system according to the fourth embodiment will be described with reference to the drawings. The training signal used here may be, for example, a training signal of an OFDM signal from which channel information of each frequency component can be obtained, or a training signal having a small autocorrelation for calculating a time correlation. Even when an OFDM signal is used as a training signal, it is not always necessary to provide a guard interval as in a general OFDM signal, and a training signal in which each frequency component continues without a guard interval can be used. is there. This is because absolute channel information (or a channel matrix or a channel vector) is not always necessary for calculating the transmission / reception weight vector, and it is sufficient to obtain only the relative relationship of the complex phases between the antenna elements. This is because if the phase relationship with any one reference antenna element can be obtained, it is not necessary to detect the symbol timing of the OFDM signal with high accuracy. Also, in the case of an OFDM signal, it is not necessary to be a training signal using all the subcarriers used as described later. The training signal may be used in a state where the SNR on the receiving side is increased.

<When transmitting a training signal from the base station device side>
FIG. 37 is a diagram illustrating an outline of signal processing in a train moving cell using a plurality of virtual transmission paths corresponding to a first singular value in the fourth embodiment. Here, an example is shown in which a training signal is transmitted from the base station apparatus to the terminal station apparatus side, and the terminal station apparatus side records coordinate information and channel information. Although not explicitly described in the following description, in the case of a wideband system such as when using OFDM, it is necessary to acquire channel information for each subcarrier and calculate transmission / reception weights. However, for the sake of simplicity, the description and detailed description of subscripts and the like regarding subcarriers and the like are omitted.

  FIG. 37A is a flowchart illustrating a process performed by the base station device 600. Upon starting the process, the base station device 600 continuously transmits a training signal (step S3701). Base station apparatus 600 ends the process after continuously transmitting the training signal for a predetermined period. The base station device 600 detects, for example, that the train 603 exists in the area of the base station device 600, and continuously transmits a training signal over a period in which the train 603 exists in the area. The terminal station device 604 provided in the train 603 receives the training signal transmitted from the base station device 600. The detection of the train 603 by the base station device 600 is performed based on operation information of the train 603 or the like, or performed by some other method.

  FIG. 37B is a flowchart illustrating a process performed by the terminal station device 604 in the train 603. Upon starting the process, the terminal station device 604 acquires coordinate information by an arbitrary method such as analysis of a captured image captured by the GPS or the camera 605 while the vehicle is traveling (step S3711). Time information is also obtained by the technique (step S3712). The terminal station device 604 stores the coordinate information acquired in step S3711 in association with the time information acquired in step S3712 corresponding to each coordinate information (step S3713), and ends the process. As a method of acquiring the time in this process, for example, information from a GPS or another wireless communication system of a macro cell may be used, or a clock whose time has been adjusted in advance may be mounted and the time stamped by the clock may be set. You may record it as it is. Here, when the coordinate information and the time information are recorded in association with each other in step S3713, not the time information after the time lag for analyzing the coordinate information, but the raw information including the coordinate information before the time lag occurs (for example, camera information). Time information at the time when the image or the GPS signal is acquired. When the time information and the coordinate information after the analysis are recorded together, the time lag is related to the information processing in the offline processing described later. No correction is required. It is not always necessary to record the correspondence between the time information and the coordinate information by the terminal station device 604 of the wireless communication system, as long as the time information and the coordinate information can be recorded even in a completely independent system. Any device may be used.

  FIG. 37C is also a flowchart illustrating the processing performed by the terminal station device 604 in the train 603. The terminal station device 604 performs the process shown in FIG. 37C in addition to the process shown in FIG. Upon starting the process, the terminal station device 604 receives a training signal (Step S3721), and acquires time information in accordance with the reception of the training signal (Step S3722). The terminal station device 604 performs received signal processing on the received training signal to perform channel estimation (step S3723). As the received signal processing here, for example, a received signal is amplified by a low noise amplifier, converted from a radio frequency to a baseband signal via a mixer, a sampling signal on a time axis is generated by A / D conversion, and OFDM is generated. In the case of the modulation method, FFT processing is performed at a predetermined symbol timing, and channel estimation is performed based on comparison with a known training signal. If the training signal is a continuous signal not including the guard interval, the FFT processing may be performed at an arbitrary timing. Further, the channel estimation result may be divided by the channel estimation result of the reference antenna element or its complex phase and converted into relative channel information actually required for calculating the transmission / reception weight vector.

  The terminal station device 604 stores the channel information acquired in step S3723 in association with the time information acquired in step S3722 (step S3724), and ends the process. Here, the timing at which the channel information can be acquired in step S3723 differs from the time at which the training signal was received at step S3721, but there is a time lag, but the time information acquired at step S3722 is the time at which the training signal was received at step S3721. With such management, even if a time lag occurs before the channel information can be acquired in step S3723, it is possible to eliminate the time lag from the time information recorded in association with the channel information in step S3724. In the description, it is assumed that the acquisition time of the channel information does not include a time lag. Note that the above-described signal processing and channel estimation can be separately performed as offline processing at different times, so that raw sampling data may be recorded as it is. However, in the following description, it is assumed that the channel estimation result (relative channel information) is recorded for convenience.

  FIG. 38 is a diagram illustrating an outline of off-line signal processing performed after acquiring coordinate information and channel information according to the fourth embodiment. Here, as in FIG. 37, an example of the offline processing when a training signal is transmitted from the base station apparatus 600 to the terminal station apparatus 604 and coordinate information and channel information are recorded on the terminal station apparatus 604 side is shown. I have. This off-line signal processing is performed by a signal processing device. The signal processing device may be provided in the terminal station device 604, or may be provided as an independent device that can acquire the coordinate information, the time information, and the channel information acquired by the terminal station device 604.

  When the off-line signal processing is started, the signal processing device reads the data of the combination of the recorded time information and coordinate information (step S3801), and chronologically calculates the difference between the coordinates and the coordinate information corresponding to the previous time. (Moving distance) is calculated (step S3802). The signal processing device also calculates the elapsed time as the difference between the time information and the time information corresponding to the previous time (step S3803). The signal processing device calculates the moving speed by dividing these differences (step S3804), and calculates the distance traveled by the train 603 during the time lag from the time lag required to actually acquire the coordinate information which is separately measured. The coordinate information is calculated by adding only the calculated moving distance (step S3805). Here, the time lag means, for example, when image processing is performed on an image captured by a camera to acquire coordinate information, means a time required for the image processing, and the actual image acquisition and the coordinate information are recorded. Equivalent to time. Since the acquisition time of the channel information and the acquisition time of the coordinate information do not necessarily have to coincide, the signal processing device converts the correspondence between the time and the coordinates from the relationship between the coordinate information, the time information, and the moving speed. Expressions are extracted and managed (step S3806). However, when the time information and the coordinate information are recorded in association with each other in a form excluding the time lag in the above-described preliminary processing, the correction processing regarding the time lag in step S3805 can be omitted.

  On the other hand, the signal processing device reads the combination of the channel information and the time information in parallel with the processing from step S3801 to step S3806 (step S3807), and estimates the coordinate information from the time information by the above conversion formula (step S3807). S3808). The signal processing device calculates a channel matrix from the channel information (step S3809), and performs singular value decomposition on the channel matrix (step S3810). The signal processing device records the first left singular vector obtained by the singular value decomposition in step S3810 as the reception weight vector of the terminal station device 604 in association with the coordinate information (step S3811), and records the first left singular vector obtained by the singular value decomposition. The right singular vector is recorded as the transmission weight vector of the base station device 600 in association with the coordinate information (step S3812).

  Further, the signal processing device performs a calibration process on the first left singular vector (reception weight vector) and the first right singular vector (transmission weight vector) calculated in step S3810 (step S3813). The signal processing device records the vector obtained by the calibration process on the reception weight vector as the transmission weight vector of the terminal station device 604 in association with the coordinate information (step S3814), and obtains the vector by the calibration process on the transmission weight vector. The resulting vector is recorded as a reception weight vector of the base station apparatus 600 in association with the coordinate information (step S3815). The above processing is performed for each virtual transmission path corresponding to each first singular value according to the first signal processing units 601-1 to 601-8 on the base station apparatus 600 side, and information is recorded for each of them. Is performed.

  In the processing so far, the processing for each virtual transmission path corresponding to each first singular value according to all the first signal processing units 601-1 to 601-8 on the base station apparatus 600 side has been described. In a situation where the distance between the base station apparatus 600 and the terminal station apparatus 604 changes every moment due to the movement of the first signal processing unit 601, the first signal processing unit 601 is adjusted to a specific distance as shown in the second embodiment. The spacing between 1 and 601-8 cannot be optimized, and as a result, not all virtual transmission paths are approximately orthogonal. Therefore, the signal processing device selects an efficient partial virtual transmission path from among the slightly redundant number of first signal processing units 601-1 to 601-8 on the base station device 600 side (step S3816). ). Data transmission between the base station apparatus 600 and the terminal station apparatus 604 is performed using the transmission / reception weight vector corresponding to the virtual transmission path selected in step S3816.

For example, when the target number of spatial multiplexing is N _SDM , the correlation among the reception weight vectors of the terminal station device 604 corresponding to the virtual transmission path related to the first signal processing unit 601 of the base station device 600 is Selects a combination of the N _SDM reception weight vectors from the plurality of recorded reception weight vectors. This selection is performed by searching the combination of N _SDM number of receiving weight vector correlation value is minimized from a plurality of receiving weight vectors for example recorded. As an example of this search method, for example, when N _SDM reception weight vectors are selected, among the correlation values calculated for each combination of two reception weight vectors included in the N _SDM reception weight vectors, maximum correlation value, the correlation value in the combination of N _SDM number of receiving weight vectors, may be as to select a combination of receiving weight vector to which the correlation value is minimized. The correlation value calculated for the combination of the two reception weight vectors here corresponds to the inner product of the normalized vectors. Note Besides, for example, the correlation values in N _SDM number of receiving weight vectors, Shun power sum of the absolute value of the correlation value calculated for each combination of the two receiving weight vectors included in N _SDM number of receiving weight vector , A combination that minimizes the power sum may be selected.

  In this way, a combination of reception weight vectors having a small correlation is selected based on some criterion, and the combination information (that is, this is a combination of all the first signal processing units 601-1 to 601-8 on the base station apparatus 600 side) , Which corresponds to the combination of which first signal processing unit is selected and communicated) also corresponds to the transmission / reception weight vector described above, and the transmission / reception weight vector for the terminal station device 604 on the train 603 side and the base station device 600 side And the combination information in the first signal processing units 601-1 to 601-8 to be used are recorded and managed. The recording management interval is a constant value between minute coordinate differences where the time variation of the transmission / reception weight vector due to the time variation of the channel is negligible, and each time a significant difference appears, coordinate information and transmission / reception weight vector information are set. May be recorded and managed.

In the above description, the number of virtual transmission paths to be used, that is, the number of spatial multiplexing is fixed to _NSDM , but this number is made variable, and the correlation value (standard The maximum number of virtual transmission paths may be utilized in a range in which the maximum value of the inner product value of the converted vector falls below a predetermined threshold. Similarly, the maximum number of virtual transmission paths may be used in a range where the sum of powers of the absolute values of the correlation values is below a predetermined threshold. In the case of a wideband system, the number of multiplexes for each subcarrier does not need to be constant, so that a different number of multiplexes can be managed for each subcarrier. In these cases, together with the information of the transmission / reception weight vectors stored in the database, the number of these multiplexes and which virtual transmission path of the first signal processing unit 601 to use is managed.

  In this case, the communication control circuit provided in the base station apparatus 600 refers to the weight vector / coordinate database 607, and instructs the first signal processing unit used for communication at the coordinates for each coordinate to transmit and receive signals. It is good also as a structure which performs. The coordinate information used in the database here does not need to represent the two-dimensional or three-dimensional geographical coordinates, but may be one-dimensional such as a distance from an arbitrary starting point on the track. However, any information may be used as long as it can correspond to the channel information.

  Further, although the previous coordinate information and the previous time information are required here, the moving speed information cannot be obtained for the data at the head of the database because there is no previous coordinate and time information. However, there is an overlap between the area covered by one base station apparatus 600 and the area covered by the adjacent base station apparatus 600, and if data acquisition is actually performed before the service area is switched, There is no problem if the information at the head of the recorded data is ignored. For example, in the case of the base station apparatus 600 including the station platform in the service area, if the data is acquired while the station platform is stopped, there is no problem even if the first data is discarded. Alternatively, the moving speed and the like can be directly obtained by measuring the number of rotations of the train wheels, and in that case, the moving speed can be obtained directly without obtaining the moving speed from the difference between the coordinates and the time. It is also possible to record the obtained value as the moving speed together with the measurement of the channel information, and use this value. In this way, by analyzing the acquired data off-line, necessary information can be processed and prepared.

<When transmitting a training signal from the terminal station device>
FIG. 39 is a diagram illustrating an outline of another signal processing in a train moving cell using a plurality of virtual transmission paths corresponding to the first singular value in the fourth embodiment. The signal processing illustrated in FIG. 39 is different from the signal processing illustrated in FIG. 37 in that a training signal is transmitted from the terminal station device 604 to the base station device 600, the terminal station device 604 acquires coordinate information, and the base station device 600 4 shows signal processing when recording channel information.

  FIG. 39A is a flowchart of a process performed by the terminal station device 604. When the terminal station device 604 starts the processing together with the operation of the train 603, it continuously transmits a training signal (step S3901). The terminal station device 604 terminates the process after continuously transmitting a training signal while collecting channel information between the terminal station device 604 and the base station device 600 before starting service operation.

  The terminal station device 604 performs the processing of FIG. 39B in parallel with the processing of FIG. Upon starting the channel information collection process before the service operation starts, the terminal station device 604 obtains coordinate information and time information by some means (step S3911, step S3912). The terminal station device 604 records data in which the acquired coordinate information is associated with the time information (step S3913), and ends the process.

  On the other hand, for example, upon detecting that the train 603 is present in the area of the base station device 600, the base station device 600 starts the processing of FIG. (Step S3921). The base station device 600 acquires time information together with the reception of the training signal (step S3922). The base station apparatus 600 performs channel estimation based on the training signal after the same reception signal processing as the above-described reception signal processing in the terminal station apparatus 604 (step S3923). Base station apparatus 600 records the obtained channel information and time information in association with each other (step S3924), and ends the process. The detection of the existence of the train 603 in the area of the base station device 600 is performed using any method such as a method using operation information.

  These processes are performed simultaneously and concurrently in all first signal processing units 601-1 to 601-8 on the base station apparatus 600 side. The time information may be acquired by each of the first signal processing units 601-1 to 601-8, or the entire base station device 600 including the plurality of first signal processing units 601-1 to 601-8. May be provided with a function of acquiring time information in common, and the time information may be notified to all the first signal processing units connected by wire. Alternatively, the channel information acquired by the plurality of first signal processing units 601-1 to 601-8 is aggregated into the base station device 600 via a wire, and the time information acquired by the base station device 600 and the plurality of first signal processing are combined. The channel information from the units 601-1 to 601-8 may be recorded and managed in association with each other.

  FIG. 40 is a diagram illustrating an outline of another off-line signal processing performed after acquiring coordinate information and channel information in the fourth embodiment. Here, similarly to FIG. 39, a case where a training signal is transmitted from the terminal station apparatus 604 to the base station apparatus 600 side, coordinate information is acquired on the terminal station apparatus 604 side, and channel information is recorded on the base station apparatus 600 side 4 shows an example of off-line signal processing. The processing shown in FIG. 40 is basically the same as the signal processing shown in FIG. 38, and the processing from step S4001 to step S4010, the processing in step S4013, and the processing in step S4016 are respectively the same as those in FIG. Are the same as steps S3801 to S3810, step S3813, and step S3816. Here, the duplicate description will be omitted, and steps S4011, S4012, S4014, and S4015 will be described.

  There is a difference that the process illustrated in FIG. 40 is a process based on the training signal in the uplink, and the process illustrated in FIG. 38 is a process based on the training signal in the downlink. Due to this difference, the first left singular vector is used as the reception weight vector of the terminal station device 604 on the train 603 side in step S3811 in FIG. The vector is a reception weight vector of the base station device 600. Also, in step S3812, the first right singular vector is used as the transmission weight vector of base station apparatus 600, whereas in step S4012 of FIG. And

  In addition, the handling of the vector obtained by the calibration processing on the first right singular vector and the first left singular vector obtained by the singular value decomposition in step S4010 in FIG. 40 is similarly different. In step S4014 of FIG. 40, the first right singular vector, that is, the vector obtained by performing the calibration process on the reception weight vector of base station apparatus 600 is set as the transmission weight vector of base station apparatus 600. In step S4015 of FIG. 40, the first left singular vector, that is, the vector obtained by performing the calibration process on the transmission weight vector of the terminal station device 604 is set as the reception weight vector of the terminal station device 604.

  In the above description of FIG. 38 and FIG. 40, the transmission weight vector and the reception weight vector on the base station apparatus 600 side and the transmission weight vector and the reception weight vector on the terminal station apparatus 604 side in each figure by the calibration process. Although it has been calculated, it is natural that the calculation is performed in duplicate, and therefore, for example, only FIG. 38 or only FIG. 40 can be used instead. For example, the processing after step S3813 in FIG. 38 and the processing in FIG. In the processing after step S4013, the processing of obtaining the remaining transmission / reception weight vectors by calibration may be omitted.

  In the above description, since the first signal processing units 601-1 to 601-8 are physically independent on the base station apparatus 600 side, the description has been made as transmission / reception weight vectors. On the side, the independent first signal processing units 601-1 to 601-8 do not exist as in the base station apparatus 600, so they are expressed as transmission / reception weight vectors in the description. It is correctly expressed as a transmission / reception weight matrix composed of vectors, and is actually expressed as a weight matrix / coordinate database 606 of the terminal station device 604. However, this point is not essential for understanding the present embodiment, and this point is expressed in accordance with the description before and after.

<Overview of signal processing during train operation>
FIG. 41 is a diagram illustrating an example of signal processing when a transmission / reception weight vector is acquired from coordinate information during service operation in the wireless communication system according to the fourth embodiment. The transmission / reception weight vector here requires both the transmission weight vector and the reception weight vector, and both the base station apparatus 600 and the terminal station apparatus 604 need to acquire the transmission / reception weight vector. This shows a process common to both. Of course, the weight vector / coordinate database 607 of the base station device 600 and the weight matrix / coordinate database 606 of the terminal station device 604 used in this case are different from each other. And the base station device 600 is different. For example, when acquiring the coordinate information of the train 603 on the base station apparatus 600 side, a plurality of sensors may be provided along the track as described above to detect the position, or the coordinate information acquired by the train 603 side may be obtained. The transmission may be performed by a separately set wireless channel (which may be in a different frequency band like a macro cell or through a different system). Even if the coordinate information of the train 603 is acquired by any of these methods, for example, if there is a difference in the time lag of the information acquisition, any time that the coordinate information at the time of information acquisition can be accurately estimated by adjusting the time lag. Even if a method is used, it is essentially possible to use the present embodiment similarly.

  First, in any of the terminal station device 604 and the base station device 600 on the train side 603, the current coordinate information of the train 603 is acquired by some method (step S4101). Further, time information at that time is also acquired (step S4102). The difference between the previous coordinate information and the previous time information is obtained (steps S4103, S4104), the moving speed is calculated by dividing the difference (step S4105), and the current coordinate is obtained by performing correction in consideration of the processing time lag. Is calculated (step S4106), and the corresponding transmission / reception weight vector is read from the weight matrix / coordinate database 606 or the weight vector / coordinate database 607 using the coordinate information as an argument (step S4107). The above-described processing in steps S4101 to S4107 is performed every time the train coordinate information is obtained, or repeatedly at a predetermined cycle, and the transmission / reception weight vector / matrix at that moment is always obtained.

  In parallel with the above processing, when receiving a signal, a virtual transmission path corresponding to a plurality of first singular values (a plurality of first signal processing units 601-1 to 601-8 of the base station device 600) Are sequentially obtained in step S4107 (step S4111), and the received signal processing is performed in a symbol unit at the time of signal reception (step S4112). For example, in the case of an OFDM signal, a signal on the frequency axis separated into frequency components is multiplied by a reception weight vector for each frequency component (step S4113), and a reception signal on each virtual transmission path is individually extracted. . Here, signals from a plurality of virtual transmission paths are extracted, but if these signal separations are necessary, the reception signals on each virtual transmission path can be extracted using a training signal added to the head of a series of reception signals. , And signal detection is performed by an arbitrary signal detection process using the channel matrix of the MIMO channel obtained by the channel estimation (step S4114), and if there is subsequent data, the above process is repeated (step S4114). If the data ends (step S4115: NO), the receiving process ends.

  Similarly, when transmitting a signal in parallel with the above processing, a virtual transmission path corresponding to a plurality of first singular values (corresponding to a plurality of first signal processing units of the base station apparatus 600). ), The transmission weight vectors read out in step S4107 are sequentially obtained (step S4121), and the transmission weight vectors are generated for each virtual transmission path together with the generation of the transmission signal to be transmitted on each virtual transmission path (step S4122). The multiplication is performed (step S4123), and the transmission signal processing of the signal is further performed (step S4124). If transmission data remains, the processing is continued (step S4125: NO), and the processing ends when the data ends. Step S4125: YES).

  In the above description, the processing performed by the base station apparatus 600 and the processing performed by the terminal station apparatus 604 are collectively described. For example, the base station apparatus 600 performs signal processing on one virtual transmission path for each of the plurality of first signal processing units 601-1 to 601-8, but the terminal station apparatus 604 performs processing on all virtual transmission paths. Signal processing for signals is performed in parallel in the terminal station device. For example, at the time of reception signal processing, each reception signal received by each antenna element is multiplied by a different reception weight vector, and a signal detection process of MIMO transmission is performed on a plurality of signal sequences obtained as a result. At the time of transmission signal processing, an individual transmission weight vector is multiplied for each signal sequence of each virtual transmission path. However, in the terminal station apparatus 604, the signal obtained by multiplying the transmission weight vector in the actual transmission signal processing is added and combined with the signal components related to all the virtual transmission paths for each antenna element, and the transmission signal processing is performed on the combined result. Will be applied. Also, in the base station apparatus 600, the first signal processing unit 601-1 to 601-8, which is provided redundantly compared to the actual number of spatial multiplexing, uses the first signal used for transmitting and receiving the actual signal. The processing unit manages which one of the first signal processing units 601-1 to 601-8, and in both transmission and reception, the communication control circuit or the like refers to the database and the corresponding first signal processing unit. , And instruct each of the first signal processing units to perform various signal processing. Excluding such a slight difference, both the base station apparatus 600 and the terminal station apparatus 604 can explain the signal processing flow in this processing flow.

(Selection of the first signal processing unit of the train moving cell using a plurality of virtual transmission paths corresponding to the first singular value)
Here, in the configuration of the wireless communication system according to the present embodiment shown in FIG. 36, eight first signal processing units 601-1 to 601-8 are shown. Does not necessarily mean that For example, when the transmission weight vectors corresponding to the first signal processing units 601-1 to 601-8 recorded in the weight vector / coordinate database 606 on the train 603 side and the reception weight vectors show a high correlation, Means that communication between a certain first signal processing unit 601 and a terminal station device 604 causes interference with communication between another first signal processing unit 601 and the terminal station device 604. In the example shown in FIG. 29 in the second embodiment, the optimum condition of the interval between the antenna elements (corresponding to the first signal processing unit) on the base station apparatus side is shown by design. The distance L between the station device and the terminal station device is included. That is, in the wireless communication system of the present embodiment in which the distance between the base station device and the terminal station device changes with the movement of the train 603, the first signal processing unit always satisfies the relationship of Expression (29) It is impossible to install 601.

Therefore, a situation may occur in which the correlation between the transmission / reception weight vectors corresponding to the individual first signal processing units 601 becomes high. In such a case, since efficient spatial multiplexing cannot be performed, the spatial multiplexing is used when simultaneously transmitting a plurality of first signal processing units 601 using a combination of transmission weight vectors or reception weight vectors having low correlation with each other. A predetermined number of the first signal processing units 601 are selected. This selection also can be implemented in advance before starting the service operation, e.g., if the spatial multiplexing number of the target is 4, in the 70 patterns of combinations of ₈ C _4, each transceiver weights or a combination pattern the maximum value of the absolute value of the inner product between vectors normalized vector ₍₄ absolute value of the six inner product C ₂ is calculated) is minimized, minimum power sum of the absolute value of the inner product The spatial multiplexing transmission is performed by using the transmission / reception weight vector of the combination pattern having the following relationship or the combination having a relatively low correlation. The combination pattern of the first signal processing unit 601 used in the combination is also recorded in the weight vector / coordinate database 607 of the base station apparatus 600, and the combination pattern of the first signal processing unit 601 is recorded. Communication may be performed.

  In addition, since communication is normally performed over a wide band, the optimal combination pattern of the first signal processing unit 601 differs for each frequency component to be used. May be used. Alternatively, the same combination pattern of the first signal processing unit 601 may be used for all subcarriers. In this case, for example, the absolute value of the above inner product for each subcarrier is compared with the absolute value of the inner product for all subcarriers. A transmission / reception weight vector combination pattern that minimizes the maximum absolute value may be selected, or a transmission / reception weight vector combination pattern that minimizes the sum of power values of absolute values of all inner products for all frequency components may be selected. May be.

(Base station apparatus configuration example of a train moving cell using a plurality of virtual transmission paths corresponding to the first singular value)
FIG. 42 is a block diagram illustrating a configuration example of a base station device 600 according to the fourth embodiment. Base station apparatus 600 includes interface circuit 77, MAC layer processing circuit 78, second transmission signal processing section 71, first transmission signal processing sections 181-1 to 181-4, and first reception signal processing. It includes units 185-1 to 185-4, a second reception signal processing unit 75, a communication control circuit 614, a weight vector / coordinate database 607, a coordinate information acquisition circuit 612, and a time information acquisition circuit 613.

  Also in the wireless communication system according to the fourth embodiment, the base station apparatus 600 uses the spatial multiplexing using the virtual transmission paths corresponding to the plurality of first singular values. It has the same configuration as the base station device 70 shown, but differs in the following configuration. The base station apparatus 600 according to the fourth embodiment includes a weight vector / coordinate database 607, a coordinate information acquisition circuit 612, and a time information acquisition circuit 613, and includes a communication control circuit 614 instead of the communication control circuit 120. Are different from the base station device 70.

  The weight vector / coordinate database 607 stores the transmission / reception weight vectors obtained as described above and associated with the coordinate information. When some of the plurality of first signal processing units 601 provided in the base station apparatus 600 are selected and used, information indicating the first signal processing unit 601 to be used is added to the transmission / reception weight vector. Are stored in the weight vector / coordinate database 607 in association with each other. Each of the plurality of first signal processing units 601 includes one first transmission signal processing unit 181 and one first reception signal processing unit 185.

  The coordinate information acquisition circuit 612 acquires the coordinate information indicating the position of the running train 603 by the means described above. The time information acquisition circuit 613 acquires time information indicating the current time. The communication control circuit 614 reads out a transmission / reception weight vector corresponding to the coordinate information acquired by the coordinate information acquisition circuit 612 from the weight vector / coordinate database 607. The communication control circuit 614 outputs the transmission weight vector among the read transmission / reception weight vectors to the first transmission signal processing units 181-1 to 181-4 via the second transmission signal processing unit 71. The communication control circuit 614 instructs the first transmission signal processing units 181-1 to 181-4 to transmit using the transmission weight vector. Note that the communication control circuit 614 may directly output the transmission weight vector to the first transmission signal processing units 181-1 to 181-4 without passing through the second transmission signal processing unit 71. Further, the communication control circuit 614 outputs the reception weight vector among the read transmission / reception weight vectors to the first reception signal processing units 185-1 to 185-4 via the second reception signal processing unit 75. The communication control circuit 614 instructs the first reception signal processing units 185-1 to 185-4 to receive using the reception weight vector. Note that the communication control circuit 614 may directly output the reception weight vector to the first reception signal processing units 185-1 to 185-4 without passing through the second reception signal processing unit 75.

  Therefore, the first transmission weight processing section 130 provided in the first transmission signal processing sections 181-1 to 181-4 in the fourth embodiment is different from the first transmission signal processing section 130 shown in FIG. Like the first transmission weight processing unit 130 included in the signal processing unit 181, it is not necessary to include the first channel information acquisition circuit 131, the first channel information storage circuit 132, and the first transmission weight calculation circuit 133. Instead, the transmission weight vector specified by the communication control circuit 614 is output to the first transmission signal processing circuit 111. Similarly, the first reception weight processing section 160 provided in the first reception signal processing sections 185-1 to 185-4 in the fourth embodiment is different from the first reception signal processing section 160 shown in FIG. Unlike the first reception weight processing section 160 provided in the reception signal processing section 185, it is not necessary to include the first channel information estimation circuit 161 and the first reception weight calculation circuit 162. The received weight vector is output to the first received signal processing circuit 158.

  However, if the channel information is acquired on the side of the base station apparatus 600 before the service operation starts (corresponding to FIG. 39), it is necessary to perform channel estimation on the received signal. Unit 160 may have a configuration including first channel information estimation circuit 161. Similarly, the first reception weight processing section 160 may have a configuration including a first reception weight calculation circuit 162 for calculating the reception weight vector. However, the channel estimation and the calculation of the reception weight vector can be performed in an off-line process as a pre-process before the start of the service operation, and therefore, it is not always necessary to implement them. As described above, if the first reception weight processing unit 160 implements only the first channel information estimation circuit 161, the first reception weight processing unit 160 The estimated channel information is output to the communication control circuit 614. The communication control circuit 614 records the channel information acquired from the first reception weight processing unit 160 in association with the time information acquired in the processing of step S3922 shown in FIG. In FIG. 42, it is assumed that this recording function is also mounted in the communication control circuit 614, and is not explicitly shown.

  In the case where the first reception weight processing section 160 does not include the first channel information estimation circuit 161, the first reception weight processing section 160 converts the acquired sampling value of the channel estimation training signal to Output to the communication control circuit 614. The communication control circuit 614 may record the sampling information acquired from the first reception weight processing unit 160 in association with the time information acquired in the process of step S3922 shown in FIG. . In this case, for example, instead of the process of step S4007 in FIG. 40, a process of reading sampling data and performing a channel estimation process may be performed.

  Note that the time information acquisition circuit 613 is necessary when acquiring channel information in advance before starting service operation, and therefore, it is possible to omit this function during actual service operation. In addition, since a point-to-point type communication is basically performed for a train moving cell, the scheduling processing circuit 781 can be omitted. However, since the terminal station device 604 is mounted on a plurality of trains with respect to the fixedly installed base station device 600, information recorded in the weight vector / coordinate database 607 may differ slightly for each train. For this reason, in that case, a database relating to all trains that may pass through the place is recorded in the weight vector / coordinate database 607, and the scheduling processing circuit 781 refers to operation information and the like to determine which train It may be provided with a function of managing whether or not it is time to pass.

  In addition, when acquiring channel information and the like before the start of service operation, various information acquired in the communication control circuit 614 is recorded. However, such information is naturally recorded in other circuits. Alternatively, a configuration in which information can be read through the communication control circuit 614 may be employed.

(Example of terminal station device configuration of train moving cell using multiple virtual transmission paths corresponding to first singular value)
FIG. 43 is a block diagram illustrating a configuration example of a terminal station device 604 according to the fourth embodiment. The terminal station device 604 includes an interface circuit 67, a MAC layer processing circuit 68, a transmission unit 61, a reception unit 65, a communication control circuit 624, a weight matrix / coordinate database 606, a coordinate information acquisition circuit 622, And an information acquisition circuit 623.

  In the wireless communication system of the fourth embodiment, too, since the spatial multiplexing using the virtual transmission paths corresponding to the plurality of first singular values is used, the terminal station device 604 is configured as shown in FIG. The configuration is the same as that of the terminal station device 60, but the following configuration is different. The terminal station device 604 according to the fourth embodiment includes a weight matrix / coordinate database 606, a coordinate information acquisition circuit 622, and a time information acquisition circuit 623, and includes a communication control circuit 624 instead of the communication control circuit 121. Are different from the terminal station device 60.

  The weight matrix / coordinate database 606 stores the transmission / reception weight matrix (vector) associated with the coordinate information acquired as described above. In addition, as for the base station apparatus 600, when selecting and using some from the plurality of first signal processing units 601 provided in the base station apparatus 600, in addition to the transmission / reception weight vector, the first Although the information indicating the signal processing unit 601 is stored in the weight matrix / coordinate database 606 in association with the coordinate information, the terminal station apparatus 604 does not select the physically different first signal processing unit 601. Since there is no information, the information of the vector forming the transmission / reception weight matrix includes which first signal processing unit 601 in the base station apparatus 600 is used, and explicitly indicates which first signal processing unit 601 is used. It is not necessary to record whether to use the unit 601. However, the combination of the first signal processing unit 601 assumed to be used by the base station device 600 and the first signal processing unit 601 actually used in the transmission / reception weight matrix assumed to be used by the terminal station device 604 Must be consistent with each other.

  The coordinate information acquisition circuit 622 acquires the coordinate information indicating the position of the running train 603 by the means described above. The time information acquisition circuit 623 acquires time information indicating the current time. The communication control circuit 624 reads out a transmission / reception weight matrix associated with the coordinate information acquired by the coordinate information acquisition circuit 622 from the weight matrix / coordinate database 606. Communication control circuit 624 outputs the read transmission weight matrix to transmitting section 61, and causes transmitting section 61 to perform transmission using the transmission weight matrix. The communication control circuit 624 outputs the read reception weight matrix to the reception unit 65 at the time of reception, and causes the reception unit 65 to perform reception using the reception weight matrix.

Therefore, the first transmission weight processing section 140 provided in the transmission section 61 in the fourth embodiment is different from the first transmission weight processing section 140 provided in the transmission section 61 shown in FIG. 21 in the first embodiment. As described above, it is not always necessary to include the channel information acquisition circuit 141, the channel information storage circuit 142, and the first transmission weight calculation circuit 143. not required during normal operation), it outputs the transmission weight vector designated by the communication control circuit 624 (column vectors constituting the transmission weight matrix) to the transmission signal processing circuit _{811-1~811-N SDM.} Similarly, the first reception weight processing unit 44 provided in the reception unit 65 in the fourth embodiment is different from the first reception weight processing unit 144 provided in the reception unit 65 shown in FIG. 22 in the first embodiment. During the service operation, the reception weight vector specified by the communication control circuit 624 is output to the reception signal processing circuit 145.

  However, when the terminal station device 604 on the train 603 performs acquisition of channel information before starting service operation (when the process illustrated in FIG. 37 is performed), the terminal station device 604 performs channel acquisition based on the received training signal. Since the estimation needs to be performed, a configuration in which the first reception weight processing unit 144 includes the first channel information estimation circuit 146 may be adopted. However, since the channel estimation and the calculation of the reception weight vector (matrix) can be performed by off-line processing, it is not always necessary to implement them. As described above, when the first reception weight processing section 144 includes only the first channel information estimation circuit 146, the first reception weight processing section 144 is estimated by the first channel information estimation circuit 146. The channel information is output to the communication control circuit 624. The communication control circuit 624 records the channel information acquired from the first reception weight processing unit 144 in association with the time information acquired in the process of step S3722 shown in FIG. In FIG. 43, it is assumed that this recording function is also mounted in the communication control circuit 624, and is not explicitly shown. In the first embodiment, another configuration example of the reception unit 65 of the terminal station device 60 shown in FIG. 23 and another configuration example of the transmission unit 61 of the terminal station device 60 shown in FIG. Except for the difference, the same configuration can be used.

  As described above, when the traffic of the information terminal or the like used by the user who gets on the train 603 is aggregated and the offload from the macro cell is attempted at the wireless entrance, the terminal station device 604 provided in the train 603 moving at high speed Channel variation between the base station apparatus 600 and the base station apparatus 600 becomes a problem. Therefore, in the wireless communication system according to the fourth embodiment, in an environment where the arrival direction of the radio wave and the moving direction of the train 603 are substantially aligned, the time variation of the transmission path corresponding to the first singular value tends to decrease. Use for Specifically, a plurality of first signal processing units 601 are installed on the overhead wire column, while the relation between the coordinate information and the channel information is acquired in advance and made into a database, and the weight matrix / coordinate database 606 and the weight vector / The information is recorded in the coordinate database 607 and used. The terminal station device 604 provided in the train 603 sequentially obtains its own coordinates, and reads out the transmission / reception weight matrix from the weight matrix / coordinate database 606 using the coordinates as an argument, thereby enabling stable communication to enable stable communication. The capacity can be increased. Similarly, the base station apparatus 600 sequentially obtains the coordinate information of the train 603, and reads out the transmission / reception weight vector used by the first signal processing unit 601 from the weight vector / coordinate database 607 using this as an argument, thereby stabilizing the operation. Communication can be performed, and the capacity of a communication line can be increased.

  In the wireless communication system of the present embodiment, each of the first transmission signal processing units 181 multiplies a baseband signal obtained by performing a modulation process or the like in the second transmission signal processing unit 71 by a transmission weight vector. To generate signal vector components. A signal vector composed of a signal component generated in each of the first transmission signal processing units 181 is transmitted from an antenna element provided in the first transmission signal processing unit 181. The component of the signal vector transmitted from each antenna element is a component of the signal vector obtained by multiplication with the transmission weight vector corresponding to the antenna element.

  In the wireless communication system of the present embodiment, each of the first received signal processing units 185 acquires a baseband signal vector from a received signal received by a plurality of antenna elements provided, and acquires the acquired baseband signal vector and reception weight. By multiplying by a vector, one signal sequence is obtained. The second received signal processing unit 75 reproduces data transmitted from the terminal station device from each of the one signal sequence acquired by each first received signal processing unit 185.

  In the wireless communication system according to the fourth embodiment, the base station apparatus 600 performs spatial multiplex transmission of a plurality of data sequences # 1 to #L with the terminal station device 604. May be transmitted to and received from the terminal station device 604.

  When the antenna element provided in first transmission signal processing section 181 is a single-polarization antenna element, first transmission signal processing section 181 may transmit one signal sequence. . When the antenna elements provided in the first transmission signal processing unit 181 are a plurality of types of polarization antenna elements and a plurality of types of polarization antenna elements are used at the same time, the first transmission signal processing unit 181 may be a single system. Only or a signal sequence of up to two systems may be transmitted. When the antenna element provided in first reception signal processing section 185 is a single polarization antenna element, first reception signal processing section 185 acquires only one baseband signal from the reception signal. You may do. When the antenna elements provided in the first reception signal processing unit 185 are a plurality of types of polarization antenna elements and a plurality of types of polarization antenna elements are used at the same time, the first reception signal processing unit 185 includes one system. Alternatively, baseband signals of up to two systems may be obtained.

[Fifth Embodiment]
[About time axis beamforming]
(Overview of basic principles)
First, two antenna elements (a first antenna element and a second antenna element) having an interval of d [m] are considered. Assuming that a transmission signal from a communication partner station has a very strong directivity and is in a line-of-sight environment, signals arriving at the two first and second antenna elements can be approximated by plane waves, and The received signal at the time t of the second antenna element is represented by φ ₁ (t) and φ ₂ (t). When the signal component at time t k-th frequency component and S _{k (t),} the distance between the transmitting station and the first antenna element L, and the center frequency f _c, the frequency bandwidth W, around the f _c - Assuming that the frequency of the k-th frequency component between W / 2 and + W / 2 is f _k and the wavelength of the k-th frequency component in the radio frequency is λ _k , the received signal φ ₁ (t) is expressed by Expression (38). be able to.

Here, the direction of arrival and an angle theta, (the c here the speed of light) the received signal phi _{2 (t)} is the received signal phi _{1 (t)} is (dsin) / c in the case where it can approximate plane wave only delayed Therefore, the received signal φ ₂ (t) can be approximated as in Expression (39).

  Here, Δt represents a sampling interval Δt [s] (= 1 / W). The coefficient α is expressed by Expression (40) using the signal bandwidth W [Hz], the arrival direction angle θ, the antenna element interval d [m], and the light speed c [m / s].

Comparing Expression (38) with Expression (39), each frequency component k in Expression (39 ⁾ is multiplied by a weight (coefficient) w _k ^{(freq-domain)} on the frequency axis shown in Expression (41). I have.

Since this coefficient is a coefficient on the frequency axis, if it is converted into a coefficient on the time axis by IFFT, the weight (coefficient) of the n-th (0 ≦ n ≦ _NFFT- 1, _NFFT is the number of FFT points) sample on the time axis w _n ^{(time-domain)} is represented by equation (42).

Here, the value expressed in [dB] is the ratio of the sum of the square of the absolute value of the time axis weight of the leading n = 0 and the square of the absolute value of the weight from n = 1 to _NFFT -1. Is expressed by the following equation (43) as a function F (α) of the coefficient α.

  FIG. 44 is a graph illustrating a function F (α) according to the fifth embodiment. In the figure, the vertical axis indicates the value of the function F (α) and the relative power ratio, and the horizontal axis indicates the value of α. The function F (α) corresponds to the ratio between the received power of the preceding wave and the received power of all delayed waves other than the preceding wave. It can be seen from the figure that when α is sufficiently small, almost one component of the preceding wave occupies most of the entire power, and conversely, as α increases, the delayed wave component increases.

Also, as can be seen from equation (40), although it depends on the bandwidth W, in general, if the antenna element spacing d is sufficiently small, α will be a small value. The approximate value of φ ₂ (t) can be obtained by multiplying the value of the received signal φ ₁ (t) by the coefficient of the preceding wave component. That is, since the weight w _k ^{(freq-domain)} on the frequency axis differs for each frequency component, conventionally, the signal on the time axis is converted into a signal on the frequency axis by FFT, and then the weight is multiplied. Was. However, in FIG. 44, if the function F (α) is designed to reduce the coefficient α by narrowing the antenna element interval so as to be equal to or more than a predetermined value (for example, 30 [dB]), the predetermined coefficient is added to the signal on the time axis. it is possible to calculate as only the receiving signal phi _{1 (t)} from the received signal phi _{2 (t)} the equation multiplied (44).

Here, the relationship between the received signal φ ₁ (t) and the received signal φ ₂ (t) has been shown. The received signal φ _m (t) can be approximated by Expression (45) using the received signal φ _m-1 (t) of the adjacent (m−1) th antenna element.

That is, the reception signal φ _m (t) of the _m- th antenna element can be approximated by Expression (46) using the reception signal φ ₁ (t) of the first antenna element.

Here, _cm in Expression (46) is given by Expression (47).

  Here, the mathematical meaning of the above description will be organized using Equation (41) and Equation (43). First, the situation in which energy is concentrated on one time-axis component when IFFT, that is, Fourier inverse transform is performed, means that a behavior like a delta function is seen on the time axis. It is known that the Fourier transform of a delta function becomes a constant on the frequency axis. Conversely, if IFFT is applied to a component that looks like a constant on the frequency axis, mathematical description can be made with high accuracy on the time axis using a single time axis component.

The first half of the right side of Expression (41) is a constant that does not depend on the frequency component, and when focusing on the exponent of the natural logarithm of the second half, it is -2πjf _k αΔt. The frequency f _k of the k-th frequency component takes a value between −W / 2 and + W / 2, and the sampling interval Δt is 1 / W, so that f _k Δt takes a value from − か ら to １ ／. take. The complex phase of Exp (−2πjf _k αΔt) is from −απ to + απ. If α is sufficiently small, the complex phase is approximately 0, and Exp (−2πjf _k αΔt) can be approximated by a constant of 1. That is, if α is sufficiently small, the frequency dependence of w _k ^{(freq-domain)} can be ignored. If α is large to some extent, the frequency dependence cannot be ignored, and the frequency component obtained after Fourier inverse transformation deviates from the behavior of the delta function type, and the delayed wave component cannot be ignored.

Here, in order to quantitatively discuss the value of the coefficient α, a frequency bandwidth of 1 GHz is assumed in the 80 GHz band. The wavelength is 3.75 mm, the antenna elements are arranged at 5 mm intervals, and d = 0.005. Assuming that the angle range of the arriving wave is ± 10 degrees, the value of α is 0.005 × sin (10 °) × 10 ⁹ / (3 × 10 ⁸ ) = 0.002894 from equation (40). This is 1.04 degrees in angle, and it can be seen that this is a constant having almost no frequency dependence. That is, if the frequency dependence is small, the value of the function F (α) becomes a very large value in the equation (43), and it can be seen that the function F (α) is more like a delta function.

In this manner, in a situation where the coefficient α is sufficiently small and Exp (−2πjf _k αΔt) can be approximated by 1, Equation (42) becomes a non-zero value only when n = 0, and Equation (48) Can be approximated as follows.

  When this is substituted into Expression (47), the coefficient of the m-th antenna element is expressed by Expression (49).

  In this way, if the complex phase relationship in the received signals of each of the N antenna elements can be obtained, the reception weight when combining the received signals of each antenna element can be determined as in equation (50).

  Although the reception weight vector of this combination is the weight of the maximum ratio combination, it is expected that the reception levels are approximately the same in a line-of-sight environment. May be.

That is, the signal from the target transmitting antenna element is extracted by multiplying the received signal vector of each antenna element by the received weight vector (〜w ₁ , ｗw ₀ ,..., Ｗw _N ). Becomes possible.

  Here, α in equation (40) is a coefficient depending on the path length difference, and can be expressed in the form of equation (49) if it is a linear array. If the geometric feature is utilized, such a reception weight can be described in a similar procedure.

Note that the coefficient _cm can be directly obtained in addition to being obtained from the equation (47). In particular, in cases other than the linear array, the periodic training signal transmitted from the transmitting station is acquired over one period without reflecting the characteristic of the geometrical arrangement in the mathematical expression, and the training signal of the training signal is acquired. Based on the cross-correlation, it is also possible to calculate as in equation (52). Here, j = m (j and m are values indicating the number of the antenna element, and are written as j instead of m).

Here, S _j (n) represents the n-th sampling value of the j-th antenna element. Here, since signal processing is performed on a signal of one antenna element versus one antenna element, the value of one symbol should not be used as it is as a line gain, but an average value of many times should be used. It is desirable to increase the SNR value. For example, averaging 100 symbols can improve the SNR characteristic by 20 dB. Therefore, averaging may be performed according to the required channel estimation accuracy. In averaging here, it is also preferable to use relative channel information with reference to the complex phase of the reference antenna element. By the way, the effect of phase noise is a problem in high frequency bands such as millimeter waves, but since the degree of phase fluctuation can be expected to some degree of randomness, the effect of phase noise was also suppressed by averaging here. A correlation value in the form can be obtained.

  Here, the training signal may be a signal with an OFDM guard interval or a continuous signal without a guard interval. This is because, in order to obtain absolute channel information, it is necessary to accurately acquire symbol timing and measure a channel. This is because it is only necessary to relatively understand the relationship. Further, other training signals can be similarly used as long as the signals have low autocorrelation in the time direction.

  The value of the reception weight may be obtained by the equation (50) or the equation (51) based on the coefficient thus obtained. With respect to the reception weight obtained in this manner, for example, in reception signal processing, a signal sequence 軸 S (n) on one time axis may be calculated by equation (53).

Here, it to w ₁ is 1 in the formula (53), is added to the formula to the (53) _S 1 (n) from j = 2 N 'to _BS-Ant of _{to w} j _S j (n) process It is carried out. One signal sequence ＾ S (n) obtained in this way is regarded as a signal received by one virtual receiving antenna for receiving a signal on a virtual transmission path corresponding to the first singular value. , Reception signal processing may be performed.

  In general, if a narrow-band signal is combined with a signal having a complex phase difference of 2π (d · sin θ / c) / λ with respect to the direction of the angle θ, directivity can be directed in the direction of the angle θ. However, for a wideband signal having a bandwidth of 1 GHz, it is necessary to change the complex phase difference for each frequency component. Further, in order to separate signals with sharp directivity having a narrow beam width, it is preferable to widen the antenna element interval. In this case, the complex phase difference for each frequency component is increased (that is, the frequency dependency is increased). It was effective. For this reason, in general, communication is performed by a method in which the directivity is stably formed over a wide band with different weights for each frequency component while narrowing the directional beam while widening the antenna aperture length by widening the interval between antenna elements. I was aiming for that. However, if the directivity can be narrowed by increasing the number of antenna elements to an enormous number, the antenna aperture length is rather narrowed, and the evaluation function F (α) shown in equation (43) is used to determine the delay other than the preceding wave. If the range of the antenna element interval and the direction θ of the arriving wave is limited (that is, the value of α is limited) so that the contribution of the wave component can be determined to be a negligible level, different antenna elements can be used even for a wideband signal. A signal arriving from a predetermined direction is efficiently extracted by multiplying the received signal by a predetermined coefficient, and signal separation at the time of spatial multiplexing can be realized on the time axis by using the coefficient.

  For example, when spatial multiplexing is performed using a plurality of virtual transmission paths corresponding to a plurality of first singular values as shown in FIG. 16 in the first embodiment, Equations (49) to (52) are used. Is calculated corresponding to each virtual transmission path, and a weight vector is calculated using Equations (50) to (51), whereby a plurality of signal sequences can be spatially multiplexed. It is possible to calculate the reception weight vector on the time axis for the first-stage signal separation.

  Also, the same calibration processing as the calibration processing on the frequency axis can be performed on the time axis. It is also possible to obtain a transmission weight vector.

(Outline of the method for removing residual interference components)
As described above, it is possible to extract the signal of the virtual transmission path corresponding to the target first singular value. However, between virtual transmission paths that are not completely orthogonal, mutual Although weak, the interference signal leaks. Hereinafter, a method of removing the interference signal will be described. The following interference signal removal method corresponds to signal processing in the second received signal processing unit 75 in the base station device. Also in the above description of the first embodiment, the interference signal that cannot be completely removed by the first received signal processing unit 185 is removed in two stages by using the second received signal processing unit 75, but it is essentially required. Is equivalent to the signal processing.

  In the following description, for example, as shown in FIG. 16 in the first embodiment, a plurality of first signal processing units each including a plurality of antenna element groups are mounted on the base station device side, and Supposes a case where there is one antenna element group composed of a plurality of antenna elements. FIG. 45 is a diagram illustrating a method of removing an interference signal according to the fifth embodiment. As shown in the figure, the wireless communication system according to the fifth embodiment includes a base station device 204 and a terminal station device 206. The base station device 204 includes a plurality of first signal processing units, and the first signal processing unit includes a plurality of antenna elements. The plurality of antenna elements provided in each first signal processing unit form virtual antenna elements 201, 202, and 203, respectively. The terminal station device 206 includes an antenna element group 205 formed by a plurality of antenna elements. In the wireless communication system according to the fifth embodiment, the communication path between the base station apparatus 204 and the terminal station apparatus 206 is such that communication is performed between the virtual antenna elements 201, 202, and 203 and the antenna element group 205. Can be regarded as a system that is

FIG. 45A is a diagram illustrating an interference component generated in a downlink in the wireless communication system according to the fifth embodiment. The directional beam (transmission beam) directed to the terminal station apparatus 206 formed by the virtual antenna element 201 of the base station apparatus 204 is directed by the antenna element group 205 of the terminal station apparatus 206 to the virtual antenna element 201. Consider the case of receiving with a directional beam (reception beam). In this case, the receive beam towards a virtual antenna elements 201 of the antenna element group 205, the signal transmitted from the virtual antenna elements 202 in addition to the signal [psi ₁ transmitted from virtual antenna elements 201 [psi _When a part of the signal ₂ leaks, consider removing a part of the signal ψ2 as an interference component. This is equivalent to removing the interference component in the uplink shown in FIG. 45B, and can be processed in the same way in terms of signal processing. In the uplink, when transmitting the signal _{１ 1} with a transmission beam formed in the antenna element group 205 of the terminal station device 206 and directed to the virtual antenna element 201, the virtual antenna element 201 A part of the signal ψ ₂ transmitted by the antenna element group 205 toward the virtual antenna element 202 leaks into the reception beam directed to the antenna element group 205 of the terminal station apparatus 206, and the leaked signal ψ ₂ Is equivalent to removing a part of as an interference component.

First, taking the downlink as an example, in the antenna element group 205, for the arriving wave from the angle θ direction heading toward the virtual antenna element 201, the signal of the signal is represented by the reception weight vector represented by Expressions (48) to (52). Perform a standby. Here is awaiting a signal [psi ₁ from virtual antenna elements 201, here the angle theta 'of direction of the virtual signal [psi ₂ from the antenna element 202 comes leaks. At this time, if a value obtained by substituting θ ′ into θ in Expression (40) is a coefficient β, the coefficient α in Expression (49) is replaced with a coefficient β in each antenna element of the antenna element group 205 of the terminal station apparatus 206. signal [psi ₂ is received by the coefficients. However, each element of the reception weight vector to be multiplied when the antenna element group 205 of the terminal station apparatus 206 is waiting with the directional beam directed to the virtual antenna element 201 is obtained by the equation in which the coefficient α is not replaced with the coefficient β. This is the reception weight represented by the equation (50) that takes the complex conjugate of (49). Therefore, the complex phase is not originally canceled to 0 by the multiplication of the expression (49) and the expression (50) in the multiplication of the reception weight. When taking the summation first m coefficients in the antenna element Exp multiplied by _{{2πjf c (α-β)} × (m-1) Δt} constituting the antenna element group 205 is synthesized at each complex phase different complex phase Will be done. Here, m = 1, 2,..., K, and K is the number N _MT-Ant of the antenna elements constituting the antenna element group 205 provided in the terminal station device 206.

However, although it is described here that different complex phases are combined, since the combination is actually regular, the “antenna arrangement condition for orthogonalization of line-of-sight MIMO transmission” described in the second embodiment is satisfied. In circumstances, this summed value converges to approximately zero. Incidentally, applying the received weights shown in equation (50) or formula (51) with respect to the signal [psi ₁ from virtual antenna elements 201, the signals received by all the antenna elements constituting the antenna element group 205 The complex phase has a constant value. As a result, it is possible to add and combine signals in the same phase, the amplitude of the signal becomes N _MT-Ant (the number of antenna elements) times, and a large gain is obtained.

According to such a procedure, when the receiving side directs the directivity to the j-th virtual antenna element of the base station apparatus 204, how the signal from the i-th virtual antenna element of the base station apparatus 204 is received. You only have to figure out. Signal indicating whether virtual signal [psi ₂ transmitted from the antenna element 202 from leaking to what kind in the state of waiting in the directional beam directed to virtual antenna elements 201 in the above example, the formula (49) Is replaced by (α-β) and the sum of all m is obtained. Since this sum is given as the sum of geometric series as shown on the left side of the equation (54), it is expressed by the following equation by the formula of the sum of geometric series.

Here, when f _c (α-β) N _MT-Ant Δt is an integer, the sum of geometric series becomes zero and no residual interference exists, but generally, residual interference remains. Therefore, it is necessary to suppress the interference component.

On the other hand, ψ ₁ (t) is transmitted from the virtual antenna element 201 with a directional beam directed to the antenna element group 205, and waits with a directional beam directed from the antenna element group 205 to the virtual antenna element 201. The signal _{１ 1} (t) received in the state includes the relationship between the signal ψ ₁ (t) transmitted from the antenna element group 205 to the first virtual antenna element and the first virtual antenna element. using the coefficients _{^ h 11} shown, to be included as a desired signal component _{^ h 11 × ψ 1 (t} ). Similarly, a signal ψ _j (t) transmitted from the j-th virtual antenna element and a signal _{ｉ i} (t) received by a reception beam from the i-th virtual antenna element toward the antenna element group 205 are generated. Using a coefficient ＾ h _ij indicating the relationship between the signals, it can be expressed that the components of the signal _{ｉ i} (t) include only ＾ h _ij × ψ ₁ (t).

However, it should be noted here that when the weights on the frequency axis of the equation (41) do not have frequency dependence, only the time axis weights of the equation (42) where n = 0 are non-zero and all others are non-zero. Since the value becomes zero, the above description is possible. However, if the weight of the equation (41) has frequency dependency, the term n = 1 of the time axis weight of the equation (42) and the subsequent terms are also used. It has a value and consequently cannot be expressed in such a simple expression. Therefore, if this signal is converted into a signal on the frequency axis and understood, generally, both the coefficient ＾ h _{ij, the} signal _{１ 1} (t) and the signal ＾ ψ _i (t) are decomposed into frequency components, For the k-th subcarrier, the signal _{ｉ i} ^(k) (t) is calculated using the coefficient ＾ _ij ^(k) , the signal ψ ₁ ^(k) (t), and the signal ＾ ψ _i ^(k) (t). Must be treated as including the components of the signal _{１ 1} ^(k) (t) ＾ h _ij ^(k) × ψ ₁ ^(k) (t).

However, conversely, when the weight on the frequency axis in equation (41) has almost no frequency dependence, n = 1 of the time axis weight in equation (42) and the subsequent terms are approximated to 0 with high precision. In this case, the signal processing on the time axis becomes possible for the subsequent processing. As described above, whether the signal processing on the frequency axis is necessary or can be dealt with by simple signal processing on the time axis depends on the frequency dependence of the weight on the frequency axis in equation (41). Yes, depending on various parameters of the target system. Accordingly, in the case where the frequency dependence of all the coefficients 可能 h _ij is negligible in the system design and the separation of each signal is possible by signal processing between each sampling value on the time axis, FIG. The communication state shown can be represented by equation (55). Here, M represents the number of spatial multiplexing (the number of virtual transmission paths) for convenience of notation.

Since this is the same format as that of the conventional MIMO signal processing on the frequency axis, a signal vector (＾ ψ _{1) in} which a slight interference signal leaks and is observed by, for example, an inverse matrix of this channel matrix or linear processing by MMSE. _{(t), ^ ψ 2 (} t), ···, ^ ψ M (t)) from the cancellation signal vector interference components _{(ψ 1 (t), ψ} 2 (t), ···, ψ M (T)). In this manner, signals between a plurality of virtual transmission paths can be separated.

  What is important here is the intention of this equation. The time t in the equation (55) can be applied to any time, and if this relational expression is understood at any time as an instantaneous value, this becomes all This means that this relationship is established between the sampling values at the same time with respect to the sampling values. That is, it means that the signal can be separated without changing the sampling value without performing FFT or the like after sampling.

  For this reason, in the above description, the description has been made mainly on the assumption that the OFDM modulation method is applied. If the corresponding channel matrix can be obtained in advance, it is possible to separate the signals on the time axis without performing FFT. It is possible to perform signal processing of single carrier transmission.

  Generally, in the case of a high frequency band such as a millimeter wave, phase noise cannot be ignored, and when a wideband OFDM signal or the like is used, the phase noise component breaks orthogonality between OFDM subcarriers by FFT processing, and the entire band is It is expected that noise and crosstalk between subcarriers will be dispersed in frequency components, making it difficult to correct phase noise. With single carrier transmission, it is possible to estimate and compensate for the phase noise component from the relationship between the signal detected and the received signal for each symbol, and it is possible to deal with relatively large phase noise. However, in the related art, when signals of a single-carrier transmission of a plurality of sequences are combined in a spatial MIMO channel, the signals cannot be separated unless they are once converted into signals on a frequency. Implementation of the FFT was essential. However, since the phase noise is converted into an uncompensable state by performing the FFT, a signal separation technique that can be realized without performing the FFT is required for the phase noise compensation.

  The MIMO channel on the time axis in Expression (55) is obtained by a theoretical analysis operation as shown in Expression (54), and a correlation operation is performed by using a training signal for one cycle corresponding to one OFDM symbol, for example. Each component of the channel matrix can be directly obtained. In the signal processing for multiplying the reception signal vector received by a plurality of antenna elements by the reception weight vector, the SNR of the original signal was low, so that some measure such as averaging was required. However, since the signal obtained by multiplying the reception weight vector for extracting the signal of the virtual transmission path corresponding to the first singular value has already been improved in SNR by the directivity gain, the reception weight vector When performing the operation of calculating by a correlation operation, it is not always necessary to perform the averaging process a plurality of times. For example, there is no problem if the averaging process is performed in a single process using a training signal for one cycle corresponding to one OFDM symbol. .

In this equation, N _FFT sample points are used. However, since it is not necessary to use FFT in particular, N _{FFT in} equation (56) does not necessarily have to be the number of OFDM FFT points, but is an arbitrary value. It is also possible.

  The approximation conditions here are based on the assumption that the virtual transmission path corresponding to the first singular value is substantially suppressed in the interference component by the directivity control of both transmission and reception. In other words, the pair of the MIMO channel matrix When the absolute value of the angle component is relatively large compared to the absolute value of the off-diagonal component, approximation with high accuracy is possible.

By using the coefficient ＾ h _ij obtained in this way and performing an operation such as multiplying both sides by a linear weight by the ZF method or the MMSE method on the channel matrix of Expression (55), the signal separation of the time axis signal is performed. If the resulting time axis signal is subjected to, for example, single-carrier signal processing, even in cases where phase noise is a problem, the existing single-carrier phase noise countermeasures can be used to achieve high accuracy. Signal detection processing can be performed.

  In the above description, when the frequency dependence of the weight on the frequency axis of the equation (41) is almost negligible, that is, n = 1 of the time axis weight in the equation (42) and the subsequent terms are 0 with high precision. The above description focuses on the case where approximation can be made. However, if such a condition is not satisfied, the FFT processing is performed straightforwardly, and processing for signal separation is individually performed on the frequency axis.

(Time axis beam forming process flow)
FIG. 46 is a flowchart illustrating a process of acquiring transmission / reception weights for time-axis beamforming in the wireless communication system according to the fifth embodiment. The flowchart shown in FIG. 46A is a flowchart of the process of acquiring the transmission / reception weight of the first stage of the time axis beamforming. Here, for example, a case will be described in which a correlation coefficient as shown in Expression (52) is used instead of a method of obtaining a coefficient on the time axis from various parameters as shown in Expression (49). However, these values differ only in the method of calculating the weight, and an analytical method can be similarly applied by replacing the calculation method. Also, the acquisition of the transmission / reception weight vector is required for both the base station device side and the terminal station device side, and the processing described here is performed bidirectionally. In the following, processing performed on the downlink will be described.

  First, the base station device transmits a training signal for channel estimation (step S4601). The training signal here is a signal for performing channel estimation and does not depend on a transmission method used in data communication after channel estimation. A training signal having a predetermined periodicity may be used. Of course, it is also possible to use the training signal used in the transmission method used in the data communication as it is.

Upon receiving the training signal transmitted from the base station apparatus by each antenna element (step S4611), the terminal station apparatus individually performs a down-conversion process from a radio frequency to a baseband on each received signal, and performs A / D conversion. After that, record the sampling value. This training signal is acquired in a predetermined cycle (for example, 100 cycles), and the signals of each cycle are added and synthesized according to the periodicity and averaged (step S4612). Assuming that the averaged sampling value of the n-th sample of the j-th antenna element is S _j (n), the correlation coefficient c _j of the j-th antenna element with respect to the first antenna element is obtained by Expression (52) (step S4613) ).

The reception weight of the j-th antenna element is calculated by taking the complex conjugate of the correlation coefficient c _j obtained in step S4613 (or further dividing the value by its absolute value), and combining them to obtain the reception weight vector. Is obtained (step S4614). In particular, when spatial multiplexing is not performed, the reception weight vector is simply calculated on the time axis of directivity formation. However, when spatially multiplexing a plurality of systems of signals as shown in FIG. 16 shown in the first embodiment, the above-described reception weight vector for each of the first signal processing units 304-1 to 304-4 in FIG. Is calculated.

  If the calibration coefficient at the time of performing implicit feedback does not have frequency dependency, a calibration process is performed on the reception weight vector on the time axis using a common calibration coefficient in the frequency direction, A transmission weight vector is obtained (step S4615). The obtained transmission / reception weight vector is recorded and managed (step S4616), and the transmission / reception weight vector is used for subsequent signal transmission / reception. If the training signal can be received in a state where the SNR is high by any method, the averaging process can be omitted. Further, even when the calibration coefficient has frequency dependence, if the deviation for each frequency is relatively small, the average value of the calibration coefficients for all subcarriers is regarded as a calibration coefficient common to all subcarriers. Processing may be performed. In this way, the first-stage reception weight vector and transmission weight vector for extracting the virtual transmission path corresponding to each first singular value can be obtained.

  FIG. 46 (b) is for acquiring a second-stage (transmission) reception weight matrix for removing the residual interference with respect to the signal of each virtual transmission path received by increasing the directivity gain as described above. 6 is a flowchart showing the processing of FIG. First, the base station device transmits a training signal using the transmission weight vector obtained by the processing shown in FIG. 46A (step S4621). The training signal here is a signal for performing channel estimation, and may be a signal that does not depend on a transmission method used in data communication after channel estimation.

  Upon receiving the training signal transmitted from the base station apparatus by each antenna element (step S4631), the terminal station apparatus individually performs a down-conversion process from a radio frequency to a baseband on each received signal, and performs A / D conversion. To obtain the sampling value. The reception signal vector at each of these antenna elements is multiplied by the reception weight vector (step S4632). The cross-correlation of the training signal for each reception weight vector thus obtained (that is, for each virtual transmission path corresponding to the first singular value) is obtained by equation (56) (step S4633), and a channel matrix is obtained. (Step S4634). The terminal station device calculates a reception weight matrix based on the acquired channel matrix (step S4635), and records and manages the calculated reception weight matrix as a second-stage time axis weight matrix (step S4636).

  Note that in the case of a wireless entrance where both the base station device and the terminal station device are fixedly installed, the weights of the first and second stages hardly change over time, and therefore, before the start of service operation. If you obtain it in advance, you can continue to use it. On the other hand, for example, in a train moving cell, the channel fluctuates at a high speed. There is no need to calculate the eye transmit / receive weight vector. However, the second-stage reception weight matrix (vector) is for removing an interference signal remaining in a different state depending on the imperfection of the first-stage reception weight vector, and is generally used. These need to be updated sequentially. In this case, the processing from step S4631 to step S4636 may be performed each time a wireless packet signal is received.

  When a plurality of virtual transmission paths corresponding to the first singular value are used in the access system, not only the second-stage reception weight matrix (vector) but also the first-stage transmission / reception weight vector is sequentially updated. There is a need to. In the frame configuration shown in FIG. 65 described in an eighth embodiment described later, a slot for accommodating a training signal for acquiring a transmission / reception weight vector for that purpose is provided in an example described in an eighth embodiment described later. The first slot of the frame and the last four slots of the frame are used. However, the training signal for calculating the reception weight matrix of the second stage can be arranged in the above-mentioned slots, but the sub-slots of the subdivided sub-slots in intermediate slots different from these slots can be arranged. It is also possible to arrange at the head (that is, at the head of the wireless packet) and calculate the second reception weight matrix based on this.

(Circuit configuration of time axis beam forming)
FIG. 47 is a diagram illustrating a configuration example of the first transmission signal processing unit 381-j included in the base station device according to the fifth embodiment. Note that j indicates a serial number in the plurality of first transmission signal processing units 381 provided in the base station apparatus. As shown in the figure, the first transmission signal processing unit 381-j includes an IFFT & GI addition circuit 813-j, a first transmission signal processing circuit 311-j, and D / A converters 814-1 to 814-. N ' _BS-Ant , local oscillator 815, mixers 816-1 to 816-N' _BS-Ant , filters 817-1 to 817-N ' _BS-Ant , and high power amplifier (HPA) 818-1 to 818-1 818-N ′ _BS-Ant , antenna elements 819-1 to 819-N ′ _BS-Ant, and a first transmission weight processing unit 330. The first transmission weight processing unit 330 includes a first channel information acquisition circuit 332, a first channel information storage circuit 333, and a first transmission weight calculation circuit 334. The configuration of the base station apparatus according to the fifth embodiment has the same configuration as the base station apparatus 70 (FIG. 17) according to the first embodiment, but is shown in FIG. And a first transmission signal processing unit 381.

  The first transmission signal processing unit 381-j in the fifth embodiment applies time-axis beamforming to the first transmission signal processing unit 181-j shown in FIG. 18 in the first embodiment. Circuit configuration. The difference between the first transmission signal processing unit 381-j and the first transmission signal processing unit 181-j shown in FIG. 18 is that a large number of IFFT & GI adding circuits 813 are integrated into one. In that a first transmission signal processing circuit 311 and a first transmission weight processing section 330 are provided in place of the transmission signal processing circuit 111-j and the first transmission weight processing section 130, and the IFFT & GI provision circuit 813 This is a point arranged before the transmission signal processing circuit 311.

  Also, in the first transmission signal processing unit 181-j shown in FIG. 18, since multiplication of the transmission weight vector on the frequency axis is indispensable, regardless of OFDM or SC-FDE, IFFT processing was essential in any arrangement as transmission signal processing. On the other hand, the first transmission signal processing unit 381-j in the fifth embodiment can perform the directional beam forming signal processing in units of sampling data on the time axis. May also be omitted, and pure single-carrier signal processing may be performed. In this sense, the IFFT & GI providing circuit 813 shown in FIG. 47 is represented by a dotted square.

An operation of the first transmission signal processing unit 381-j provided in the base station device according to the fifth embodiment will be described. When the digital baseband signal is input from the second transmission signal processing unit 71 to the first transmission signal processing unit 381-j, if the digital baseband signal is a signal on the frequency axis, the IFFT & GI addition circuit 813-j determines the frequency. The signal on the axis is converted into a signal on the time axis. In the case of a system in which sampling data on the time axis is input like a signal of a pure single carrier, a digital signal is output without passing through the IFFT & GI adding circuit 813-j (the IFFT & GI adding circuit 813-j is omitted). The baseband signal is input to a first transmission signal processing circuit 311-j. The first transmission signal processing circuit 311-j multiplies one system signal sequence by a transmission weight vector on the time axis for each sampling value, and applies each component of the resulting transmission signal vector to a D / D signal for each antenna element system. A converters 841-1 to 814-N 'are output to _BS-Ant . The subsequent processing is the same as the processing performed in the first transmission signal processing unit 181-j shown in FIG.

  Note that the transmission weight vector multiplied by the signal sequence in the first transmission signal processing circuit 311-j is the same as the first transmission signal processing unit 181-j shown in FIG. It is obtained from the first transmission weight calculation circuit 334 provided in the transmission weight processing unit 330. In the first transmission weight processing unit 330, the first channel information acquisition circuit 332 acquires the channel information acquired by the reception unit via the communication control circuit 120, and stores the acquired channel information in the first channel information storage circuit 333. It is stored and updated sequentially. At the time of transmission, in accordance with an instruction from the communication control circuit 120, the first transmission weight calculation circuit 334 reads channel information corresponding to the destination terminal station device from the first channel information storage circuit 333, and based on the read channel information. Calculates the transmission weight vector. The method for calculating the transmission weight vector here can be any method as described above.

  Note that the communication control circuit 120 shown in FIG. 17 and the communication control circuit 120 shown in FIG. 47 are slightly different in details such as channel information transferred via the communication control circuit 120 strictly. For example, in FIG. 17, different channel information needs to be transferred for each subcarrier because a general communication method is targeted, but in FIG. And only the common channel information for all subcarriers need to be transferred. However, since other functions are equivalent except for such a difference in specific information, description will be made using the same reference numerals.

  Further, the first transmission signal processing unit 181-j shown in FIG. 18 calculates an individual transmission weight vector for each frequency component, whereas the first transmission signal processing unit 381-j in the fifth embodiment simply obtains a transmission weight vector. The point that only the transmission weight vector in one time axis component is obtained is the difference from the first transmission signal processing unit 181-j in FIG. When the wireless station device of the communication partner is fixedly installed, the transmission weight vector itself is calculated and recorded in advance, and the transmission weight vector is simply read out to transmit to the first transmission signal processing circuit 311. The configuration may be such that a weight vector is notified, and further, in the case of a point-to-point type point-to-point communication, the transmission weight vector may be directly stored in the first transmission signal processing circuit 311. good.

FIG. 48 is a block diagram illustrating a configuration example of the first reception signal processing unit 385-j of the base station device according to the fifth embodiment. As shown in the figure, the first received signal processing unit 385-j includes antenna elements 851-1 to 851-N ' _BS-Ant and low noise amplifiers (LNA) 852-1 to 852-N' _BS-Ant. , A local oscillator 853, mixers 854-1 to 854-N ' _BS-Ant , filters 855-1 to 855-N' _BS-Ant , and A / D converters 856-1 to 856-N ' _{BS-. Ant} , a first reception signal processing circuit 358-j, an FFT circuit 857-j, and a first reception weight processing unit 360. The first reception weight processing section 360 includes a first channel information estimation circuit 362 and a first reception weight calculation circuit 363.

  The first reception signal processing unit 385-j according to the fifth embodiment applies time-axis beamforming to the first reception signal processing unit 185-j illustrated in FIG. 19 in the first embodiment. Circuit configuration. The difference from the first reception signal processing unit 185-j shown in FIG. 19 is that the large number of FFT circuits 857 are integrated into one, Instead, a first reception signal processing circuit 358 and a first reception weight processing section 360 are provided, and an FFT circuit 857 is arranged at a subsequent stage of the first reception signal processing circuit 358.

  Further, in the first reception signal processing unit 185-j shown in FIG. 19, since multiplication of the reception weight vector on the frequency axis is indispensable, regardless of OFDM or SC-FDE, FFT processing was indispensable in any arrangement as reception signal processing. On the other hand, the first received signal processing unit 385-j in the fifth embodiment can perform directional beam forming signal processing in units of sampling data on the time axis, so that the first received signal It is also possible to omit the FFT circuit 857-j from the processing unit 385-j and perform pure single-carrier signal processing. More specifically, as shown in FIG. 46B, the FFT circuit 857-j is omitted here even when the reception weight matrix on the time axis is used in the weights after the two-stage weight multiplication, Furthermore, it is also possible to adopt a configuration in which signal processing on the time axis is continued. In this sense, the FFT circuit 857-j shown in FIG. 48 is represented by a dotted square.

The operation of the first received signal processing unit 385-j provided in the base station device according to the fifth embodiment will be described. Among the processing on the signals received by the antenna elements 851-1 to 851-N ' _BS-Ant , the low-noise amplifiers 852-1 to 852-N' _BS-Ant to A / D converters 856-1 to 856-N '. _The processing performed up to _BS-Ant is the same processing as the processing in the first received signal processing unit 185-j of the first embodiment. By this processing, the received signal is converted into a digital baseband signal (sampling data). The A / D converters 856-1 to 856-N'BS- _Ant input a received signal vector including a digital baseband signal in each sampling to the first received signal processing circuit 358-j. According to the instruction from the communication control circuit 120, the first reception signal processing circuit 358-j multiplies the reception signal vector on the time axis input from the first reception weight processing unit 360 by the sampling signal value for each sampling value. Then, it is converted into a single signal sequence.

  When signal processing on the frequency axis is required thereafter, as in OFDM or SC-FDE, and when processing using a reception weight matrix on the frequency axis is performed at a subsequent stage, the first reception signal processing is performed. The circuit 358-j outputs one signal sequence obtained by the conversion to the FFT circuit 857-j, and the FFT circuit 857-j converts a signal on the time axis into a signal on the frequency axis. In the case of a system that performs all signal processing on the time axis like a pure single-carrier signal, or in the subsequent stage where the reception weight matrix on the time axis is continuously multiplied, the signal must be passed through the FFT circuit 857-j. (The FFT circuit 857-j is omitted), and this signal is output. The FFT circuit 857-j converts the signal input from the first received signal processing circuit 358-j into a signal on the time axis by FFT at a predetermined symbol timing determined by a timing detection circuit not shown. The signal is converted into a signal on the frequency axis (digital baseband signal for each subcarrier). FFT circuit 857-j outputs a digital baseband signal for each subcarrier to second received signal processing section 75. When the first reception signal processing unit 385-j does not include the FFT circuit 857-j as described above, the first reception signal processing circuit 358-j converts the reception weight vector on the time axis for each sampling value. And outputs a signal for one system obtained by multiplying the received signal vector to the second received signal processing unit 75.

When each sampled received signal vector obtained by each of the A / D converters 856-1 to 856-N'BS- _Ant is based on a training signal for channel estimation, the received signal vector Is input to the first channel information estimation circuit 362. The first channel information estimating circuit 362 calculates the antenna elements 819-1 to 819 -N _MT-Ant of each terminal station apparatus and the antenna elements 851-1 to 851 -N of the base station apparatus 70 by some method as described above. 'Estimate the channel information between the virtual antenna element constituted by _BS-Ant on the time axis (corresponding to, for example, the correlation operation shown in equation (52)), and report the estimation result as the first reception weight. Output to the calculation circuit 363. Based on the input channel information on the time axis, the first reception weight calculating circuit 363 calculates the complex conjugate or the complex conjugate value obtained by dividing each component by its absolute value. Calculate the reception weight vector. In the first embodiment, the first reception signal processing unit 185 shown in FIG. 19 obtains an individual reception weight vector of each frequency component, whereas the first reception signal processing unit 185 shown in FIG. The difference between the processing unit 385 and the first reception signal processing unit 185 is that the processing unit 385 calculates only the reception weight vector of a single time axis component. The first reception weight calculation circuit 363 outputs the reception weight vector calculated in this way to the first reception signal processing circuit 358-j. The calculation of the reception weight vector by the first reception weight calculation circuit 363 corresponds to the processing described with reference to FIG.

  In addition, the communication control circuit 120 manages control related to overall communication, such as management of a source station and overall timing control. Here, when the terminal station device of the communication partner is fixedly installed, the reception weight vector itself is calculated and recorded in advance, and this is simply read out to the first reception signal processing circuit 358-j. Alternatively, the reception weight vector may be notified, or in the case of a point-to-point type point-to-point communication, the reception weight vector is directly stored in the first reception signal processing circuit 358. May be.

  In the above description, the signal processing is performed only on the virtual transmission line corresponding to the first singular value. However, while the virtual transmission line corresponding to the first singular value mainly considering the line of sight is effectively utilized. For example, when relatively clean reflected wave components such as large structures are also used in an auxiliary manner, in addition to addition and synthesis in which coefficients between sampling values at the same time between multiple antenna elements are multiplied, several samples of A configuration including a delay component may be used. For example, in the case where the reference antenna is the first antenna and a coefficient that causes a delayed wave of one sampling clock to leak into the j-th antenna is obtained, the following equation (57) obtained by expanding the equation (52) may be used.

Here, S _j (n) represents the n-th sampling value of the j-th antenna element, as in Expression (52). Here, since signal processing is performed on a signal of one antenna element versus one antenna element, the SNR is not obtained by using the value of one symbol as it is, but by using the average value of many times as the line gain. It is desirable to increase the value. Based on this coefficient, the value of the reception weight may be obtained in the same manner as in Expression (50) or Expression (51). With respect to the reception weight obtained in this manner, for example, in the reception signal processing, a signal sequence on one time axis may be calculated by the following equation (58).

Here, ｗw ′ _j is a reception weight calculated using the equation (50) or the equation (51) for the coefficient c ′ _j . In this manner, it is also possible to perform processing in consideration of a delay wave for one sampling clock. Similarly, when considering a delayed wave corresponding to two sampling clocks, the following equation (59) may be applied.

Here, ｗw ″ _j is a reception weight calculated using the equation (50) or the equation (51) for the coefficient c ″ _j . By performing a similar extension, it is possible to consider a further delayed wave component. However, both of Expressions (58) and (60) indicate that the respective antenna elements at the same time shown in Expression (53) that contains components for adding combines the signals obtained by multiplying the reception weight to w _j to the signal sampling values is a common feature, the essence of this embodiment is here.

  In the present embodiment, in the synthesis of transmission / reception signals of a large number of antenna elements, transmission / reception weights having frequency dependence are used, and the conventional directivity control of multiplying different transmission / reception weights for each individual subcarrier on the frequency axis. Instead, by using a common transmission / reception weight for all subcarriers, directivity control on the time axis can be performed for each sampling value, and this can be similarly used for both the base station apparatus and the terminal station apparatus. It is possible, or it can be implemented only in the base station device or only in the terminal station device. Although the above description is for a base station device, a description for a terminal station device will be given below. Here, similarly to the case of the base station apparatus, it is possible to apply time-axis beamforming on both the transmitting side and the receiving side.

FIG. 49 is a block diagram illustrating an example of a circuit configuration of a transmission unit 361 included in a terminal station device according to the fifth embodiment. As shown in the figure, the transmission unit 361, a second transmission signal processing circuit 348, IFFT & GI imparting circuit _{813-1~813-N SDM} and the first transmission signal processing circuit _{331-1~331-N SDM} , Addition / synthesizing circuits 812-1 to 812-N _MT-Ant , D / A converters 814-1 to 814-N _MT-Ant , local oscillator 815, and mixers 816-1 to 816-N _MT-Ant. A filter 817-1 to 817-N _MT-Ant , a high power amplifier (HPA) 818-1 to 818-N _MT-Ant , an antenna element 819-1 to 819-N _MT-Ant, and a first A transmission weight processing unit 340. The first transmission weight processing section 340 includes a first channel information acquisition circuit 341, a first channel information storage circuit 342, and a first transmission weight calculation circuit 343. The configuration of the terminal station device in the fifth embodiment has the same configuration as the terminal station device 60 (FIG. 20) in the first embodiment, but differs in that a transmitting unit 361 is provided instead of the transmitting unit 61.

  The transmission unit 361 according to the fifth embodiment has a circuit configuration when time-axis beamforming is applied to the transmission unit 61 illustrated in FIG. 25 in the first embodiment. The difference between the transmission unit 361 and the transmission unit 61 shown in FIG. The point is that the processing unit 340 is provided, and the IFFT & GI adding circuit 813 is arranged at a stage before the first transmission signal processing circuit 331.

  Also, in the transmitting section 61 shown in FIG. 25, since multiplication of the transmission weight vector on the frequency axis is essential, IFFT processing is performed as transmission signal processing on the frequency axis regardless of OFDM or SC-FDE. Required in either arrangement. On the other hand, the transmitting unit 361 in the fifth embodiment can perform directional beam forming signal processing in units of sampling data on the time axis, so that the IFFT & GI adding circuit 813 is also omitted, and pure single It is also possible to perform carrier signal processing. In this sense, the IFFT & GI providing circuit 813 in the figure is represented by a dotted square.

The operation of the transmitting unit 361 provided in the terminal station device according to the fifth embodiment will be described. When data (data inputs # 1 to #N _SDM ) corresponding to the number of spatial multiplexing N _SDM systems are input from the MAC layer processing circuit 68 to the second transmission signal processing circuit 348, the second transmission signal processing circuit 348 The transmission signal processing is performed on each input data system to generate a digital baseband signal for each data system. This signal processing is general, and may be OFDM, single carrier, SC-FDE, or any type of wireless signal. In addition, when performing arbitrary precoding on the transmission signal, the second transmission signal processing circuit 348 performs precoding signal processing. If the signal output from the second transmission signal processing circuit 348 is a signal on the frequency axis, the IFFT & GI adding circuit 813 converts the signal on the frequency axis into a signal on the time axis for each signal system. In the case of a system in which sampling data on the time axis is input like a pure single-carrier signal, the second transmission is performed without passing through the IFFT & GI adding circuit 813 (that is, by omitting the IFFT & GI adding circuit 813). the signal processing circuit 348 inputs the signal of each data lines obtained by the signal processing to the first transmission signal processing circuit _{331-1~331-N SDM.}

The first transmission signal processing unit 381-j included in the base station apparatus illustrated in FIG. 47 performs signal processing on one system of signal sequence. In contrast, for processing a plurality of signals lines in the terminal station apparatus, mobile station apparatus has a configuration comprising a plurality of first transmission signal processing circuit 331-1~331-N _SDM. First transmission signal processing circuit 331-1~331-N _SDM, each multiplied by the transmission weight vector on the time signal sequence of the corresponding data line axis for each sampling value, each of the transmission signal vector of the results The components are output to the additive synthesis circuits 812-1 to 812-NMT _-Ant . Additively combining circuit _{812-1~812-N MT-Ant} is a signal of all signal sequences which are respectively input from the first transmission signal processing circuit _{331-1~331-N SDM} adds synthesized, the same strain To the D / A converters 814-1 to 814 -N _SDM . The subsequent processing is the same as the processing performed in the transmitting unit 61 shown in FIG. The transmission weight vector on the time axis that is multiplied by the signal sequence of each data system in the first transmission signal processing circuits 331-1 to 331-N _SDM is similar to the transmission unit 61 during signal transmission processing. It is obtained from the first transmission weight calculation circuit 343 provided in the first transmission weight processing section 340. Alternatively, the transmission weight vector on the time axis acquired in advance as described above is stored in the first transmission weight processing unit 340, and the first transmission signal processing circuit 331-1 is used therefrom. it may be configured to instruct the _{~331-N SDM.}

In the first transmission weight processing unit 340, the first channel information acquisition circuit 341 separately acquires the channel information acquired by the reception unit 365 via the communication control circuit 121, and sequentially updates the acquired channel information. It is stored in the first channel information storage circuit 342. At the time of signal transmission, the first transmission weight calculation circuit 343 reads channel information corresponding to the destination terminal station device from the first channel information storage circuit 342 in accordance with an instruction from the communication control circuit 120, and reads the read channel information. The transmission weight vector on the time axis is calculated based on An arbitrary method similar to that of the above-described base station apparatus can be used for the method of calculating the transmission weight vector here. Further, the transmission unit 61 shown in FIG. 25 obtains an individual transmission weight vector for each frequency component, whereas the first transmission weight calculation circuit 343 in the fifth embodiment employs a transmission weight for a single time axis component. The point that only the vector is obtained is the difference from the transmission unit 61. When the terminal station apparatus is fixedly installed, the transmission weight vector itself is calculated and recorded in advance, and the transmission weight vector itself is simply read to obtain the first transmission signal processing circuits 331-1 to 331-N _SDM. transmission weight to the vector may be configured to notify, if more Point-to-Point-type one-to-one communication with respect to the first transmission signal processing circuit _331-1~331-N in _SDM May be directly stored in the transmission weight vector.

FIG. 50 is a block diagram illustrating a configuration example of the receiving unit 365 of the terminal station device according to the fifth embodiment. As shown in the figure, the receiving unit 365 includes antenna elements 851-1 to 851-N _MT-Ant , low noise amplifiers 852-1 to 852-N _MT-Ant , a local oscillator 853, and mixers 854-1 to 854-1. 854-N, filters 855-1 to 855-N, A / D converters 856-1 to 856-N, first reception signal processing circuits 355-1 to 355-N _SDM , and FFT circuit 857- comprising a one to eight hundred _{fifty-seven-N SDM,} the second received signal processing circuit 359, and a first reception weight processing unit 354. The first reception weight processing section 354 includes a first channel information estimation circuit 356 and a first reception weight calculation circuit 357.

The receiving unit 365 in the fifth embodiment has a circuit configuration in the case where time-axis beamforming is applied to the receiving unit 65 shown in FIG. 23 in the first embodiment. The difference from the receiving unit 65 shown in FIG. 23 is that a large number of FFT circuits 857 are integrated into one, the first received signal processing circuit 155, the second that provided in place of the part 154 and the first received signal processing circuit 355 and the second received signal processing circuit 359 and the first reception weight processing unit 354, a further FFT circuit _{857-1~857-N SDM,} Is disposed after the first reception signal processing circuit 355.

In addition, in the receiving unit 65 shown in FIG. 23, since multiplication of the reception weight vector on the frequency axis is essential, FFT processing is performed as reception signal processing on the frequency axis regardless of OFDM or SC-FDE. Required in either arrangement. In contrast, the reception section 365 in the fifth embodiment, since it is possible to signal processing beamforming at the sampling data unit on the time axis, also the FFT circuit _{857-1~857-N SDM} It is also possible to omit this and perform pure single-carrier signal processing. In this sense, FFT circuit _{857-1~857-N SDM} shown in FIG. 50 is expressed by a dotted line box.

An operation of the receiving unit 365 provided in the terminal station device according to the fifth embodiment will be described. Among the processing on the signals received by the antenna elements 851-1 to 851-N _MT-Ant , the low-noise amplifiers 852-1 to 852-N _MT-Ant to the A / D converters 856-1 to 856-N _MT-Ant The processing performed up to is the same as the processing in the receiving unit 65 of the first embodiment. A / D converter _{856-1~856-N MT-Ant} inputs the received signal vector consisting of the digital baseband signal at each sampled first received signal processing circuit _{355-1~355-N SDM} . In the base station apparatus, as shown in FIG. 48, the first received signal processing unit 385-j performs processing of one signal sequence. In contrast, for processing a plurality of signals lines in the terminal station apparatus, the reception unit 365 has a structure comprising a plurality of first received signal processing circuit _{355-1~355-N SDM.} First received signal processing circuit 355-1~355-N _SDM in accordance with an instruction from the communication control circuit 121, a receiving weight vector on the time axis corresponding to each signal series by multiplying the received signal vector for each sampled value Are converted into signals of each signal series. A reception weight vector corresponding to each signal sequence is input from first reception weight processing section 354. The first received signal processing circuits 355-1 to 355-N _SDM output the obtained signals of the respective signal sequences to the FFT circuits 857-1 to 857-N _SDM corresponding to the signal sequences.

  It should be noted that the communication control circuit 121 shown in FIG. 20 and the communication control circuit 121 shown in FIG. 50 are slightly different in details such as channel information transferred via the communication control circuit 121 strictly. For example, in FIG. 20, since a general communication scheme is targeted, different channel information needs to be transferred for each subcarrier, but in FIG. 50, a single channel information on the time axis (that is, And only the common channel information for all subcarriers need to be transferred. However, since other functions are equivalent except for such a difference in specific information, description will be made using the same reference numerals.

As the OFDM and SC-FDE, which if necessary signal processing on the frequency axis in the following, the frequency axis signal of each signal sequence on the time axis at the FFT circuit _{857-1~857-N SDM} Convert to the above signal. In the case of a system that performs all signal processing on the time axis like a signal of a pure single carrier, or in a case where signal processing is performed using a reception weight matrix on the time axis even in the subsequent stage, the FFT circuit 857-1 is used. not through the _{~857-N SDM} (i.e. omitted from the FFT circuit _{857-1~857-N SDM),} the signal of each signal sequence is output to the second reception signal processing circuit 359.

For example, when signal processing on the time axis is performed by the second reception signal processing circuit 359, the first reception signal processing circuit 355-1 to 355-N _{SDM determines the} spatial multiplexing number N _SDM on the time axis. The signal of each signal series is input, and this is multiplied by the reception weight matrix on the time axis, and the signal separation of the spatially multiplexed signal series is performed on the time axis. And performs a signal detection process, and outputs the reproduced data sequence to the MAC layer processing circuit 68.

Here, the reception weight matrix by which the second reception signal processing circuit 359 multiplies the signal of each signal sequence is a reception weight matrix obtained by the processing shown in FIG. Here, it is possible to use the previously obtained reception weight matrix, each time receiving a wireless packet, the beginning of a signal input from the first received signal processing circuit 355-1~355-N _SDM (wireless packet) , A channel matrix related to channel information is obtained by the operation shown in equation (56), and a general MIMO signal process is performed using the ZF method or the MMSE method. A linear weight may be calculated and used as a reception weight matrix. The signal of the _NSDM system separated in this manner may be subjected to single-carrier signal processing if it is a single-carrier signal, or timing detection omitted here for OFDM or SC-FDE. A signal on the time axis is converted into a signal on the frequency axis (digital baseband signal for each subcarrier) by FFT at a predetermined symbol timing determined by the circuit for normal demodulation, and the demodulated data is reproduced. The sequence is output to the MAC layer processing circuit 68.

On the other hand, when performing signal processing on the frequency axis in the second received signal processing circuit 359, the FFT circuit _{857-1~857-N SDM} to a signal on the frequency axis second received signal processing circuit 359 is input, may be used pre-obtained reception weight matrix, each time receiving a wireless packet, the beginning of a signal input from the first received signal processing circuit 355-1~355-N _SDM (wireless packet) , A channel matrix is obtained by performing channel estimation based on the relationship between the information of the training signal portion for channel estimation and the known training signal given to ZF and MMSE as general MIMO signal processing. The weight may be calculated as a linear weight such as a reception weight matrix, or the signal detection processing may be performed by non-linear processing such as MLD or simplified QR-MLD. The data sequence reproduced by performing such signal detection processing is output to the MAC layer processing circuit 68. Note that the signal detection processing or demodulation processing in the above description may be configured to include deinterleaving, error correction processing, and the like as necessary.

If the received signal vector having each sampling value as a component obtained by each of the A / D converters 856-1 to 856-N _MT-Ant corresponds to a training signal for channel estimation, this The received signal vector is input to first channel information estimation circuit 356. The first channel information estimating circuit 356 calculates the antenna elements 851-1 to 851-N _MT-Ant of each terminal station apparatus and the virtual antennas of the base station apparatus 70 by some method similar to that of the base station apparatus 70 described above. Channel information between the device and the element is estimated on the time axis, and the estimation result is output to the first reception weight calculation circuit 357. The first reception weight calculation circuit 357 calculates a reception weight vector to be multiplied based on the input channel information on the time axis. In the first embodiment, the receiving unit 65 shown in FIG. 23 obtains an individual transmission weight vector for each frequency component, whereas the receiving unit 365 of the terminal station device according to the fifth embodiment uses a single time axis. The difference from the receiving unit 65 is that only the transmission weight vector of the component is obtained. First reception weight calculating circuit 357 outputs the reception weight vector calculated in this way in the first received signal processing circuit _{355-1~355-N SDM.} Alternatively, the reception weight vector on the time axis acquired in advance as described above is stored in the first reception weight processing unit 354, and the first reception signal processing circuit 355-1 is used therefrom. it may be configured to instruct the _{~355-N SDM.}

Further, the communication control circuit 121 manages control related to the entire communication, such as management of the source station and overall timing control. Here, when the terminal station apparatus is fixedly installed, the reception weight vector itself is calculated and recorded in advance, and the reception weight vector is simply read out to the first reception signal processing circuit 355 to receive the reception weight vector. may be configured to notify, if even a one-to-one communication Point-to-Point type, the direct receiving weight vector to the first reception signal processing circuit _{355-1~355-N SDM} You may memorize it. The above is the description of the configuration example of the receiving unit 365 of the terminal station device according to the fifth embodiment.

  The point of the fifth embodiment is that, when a plurality of antenna elements are provided, the difference between the path lengths of the line-of-sight waves is made sufficiently small by reducing the interval between the antenna elements, and the reception weight in the frequency domain is reduced. The frequency dependence tends to be small. At this time, if the reception weight is converted into the time domain by IFFT or the like, the first wave component (line-of-sight wave component) almost occupies most of the entire reception power. In such a situation, MIMO signal separation performed in the frequency domain can be performed in the time domain. Also, MIMO signal separation that can be performed in the time domain using only the first wave component can be performed for each sampling value.

  Therefore, in the wireless communication system according to the fifth embodiment, for example, at the time of signal reception, the first reception signal processing unit 385 provided in the base station device and the first reception signal processing in the reception unit 365 provided in the terminal station device are provided. The circuit 355 performs a first-stage signal separation for separating a signal on a virtual transmission path corresponding to a first singular value obtained by singular value decomposition of a channel matrix from a received signal for each sampling value on a time axis. I do. Thereafter, the second reception signal processing unit 75 provided in the subsequent stage of the first reception signal processing unit 385 by the base station apparatus and the second reception signal processing circuit 359 in the reception unit 365 that the terminal station apparatus can include are virtually connected. A second-stage signal separation for separating signals between transmission paths is performed. In the device on the receiving side of the wireless communication system according to the fifth embodiment, when the second-stage signal separation is performed on the time axis, the reception weight matrix used in the signal separation is the signal received for each antenna element. Can be calculated based on the channel matrix obtained by obtaining the correlation value of.

  By performing the MIMO signal separation in the time domain, it is possible to omit the FFT required for the signal separation, and to reduce the operation load and the operation circuit. In addition, when performing spatially multiplexed single-carrier transmission, it is not necessary to convert signals on the frequency axis. It becomes possible to use the phase noise compensation technique in carrier transmission. In addition, as in the first to fourth embodiments, the bandwidth can be expanded in a situation where direct waves are dominant in a line-of-sight environment, while reducing the calculation load for MIMO signal separation in the base station apparatus and the terminal station apparatus. be able to.

  In the wireless communication system according to the fifth embodiment, the configuration has been described in which the reception weight / transmission weight is obtained from the correlation coefficient. However, channel information may be fed back to some subcarriers, and the reception weight / transmission weight of the entire frequency band may be calculated based on the average value.

[Sixth Embodiment]
[Approximate solution for channel matrix acquisition when line of sight is dominant]
(Overview of Basic Principle According to Sixth Embodiment)
In the above description, spatial multiplexing transmission that actively utilizes virtual transmission paths corresponding to a plurality of first singular values has been described. In this spatial multiplexing transmission, it is necessary to perform singular value decomposition of a channel matrix and calculate a first right singular vector or a first left singular vector. Also, in order to obtain the original channel matrix used for this singular value decomposition, channel estimation and feedback of channel information are required. Here, in order to improve the gain by using a large number of antenna elements, it is necessary to find a component of a channel matrix corresponding to a product of the number of each of a large number of transmitting antennas and a large number of receiving antennas. In general, the overhead associated with the transmission of a training signal for channel estimation increases in proportion to the number of components of the channel matrix, and becomes an insignificant amount.

  The implicit feedback is, for example, when the uplink channel matrix can be obtained, the downlink channel matrix is estimated by the calibration process, and the overhead associated with the increase in the number of antenna elements of the base station device can be reduced. it can. On the other hand, if there are a plurality of antenna elements on the terminal side, unless a training signal is individually transmitted from each antenna element, an uplink channel matrix cannot be obtained, and the overhead due to an increase in the number of antenna elements on the terminal side is Cannot be reduced.

  When the number of antenna elements on the terminal side is relatively small, for example, if the number of antenna elements is two, each antenna transmits a training signal on even-numbered subcarriers and odd-numbered subcarriers. Channel estimation is possible. In this case, the channel information of the subcarrier for which channel estimation has not been performed is estimated by interpolation. In addition, it is possible to efficiently perform channel estimation by using a plurality of orthogonalized preamble signals. However, in the case where a multi-element antenna element such as 16 or 32 elements is provided, there is a problem that the subcarrier interval is too wide and the accuracy of channel estimation by interpolation is extremely reduced. There are limitations to channel estimation and its feedback.

  Furthermore, even when a large-scale antenna is applied to improve the line gain, a single-element antenna cannot expect a high SNR (signal-to-noise ratio) enough to enable effective channel estimation in line design. For this reason, for example, long-term measurement and averaging processing on the assumption that there is no channel time variation are required. However, in a case involving time fluctuations, for example, in the case of a train moving cell, long-term averaging cannot be employed, and it is necessary to acquire channel information of all necessary paths instantaneously. In a general MIMO system, for example, when N antenna elements are provided on the transmission side, channel estimation is performed using N symbols of a training signal of one OFDM symbol. However, as described above, when both the transmitting and receiving stations are provided with a large number of antenna elements, there is a possibility that the channel may fluctuate during N channel estimation. For this reason, a channel estimation method is required in which necessary channel information is obtained in as short a time as possible.

  Here, when the virtual transmission path for the first singular value is positively used, the antenna group of the terminal device and the antenna group of each first signal processing unit of the base station device are located in a very narrow area. They are concentrated. For this reason, the channel information changes minutely due to the subtle displacement. However, if attention is paid only to the line-of-sight component, the change is expected to change in a form that maintains the regularity between the components of the channel vector. For example, when a signal transmitted from an antenna element on the transmission side is received by an antenna group on the reception side, channel information changes depending on a difference in path length of the signal. For this reason, it is necessary to obtain the type of this regularity on a propagation model that considers only the “line of sight wave”, and to efficiently use the regularity to efficiently perform channel estimation. Can be. Hereinafter, a method of efficiently performing the channel estimation will be described.

FIG. 51 is a diagram illustrating an outline of a reception signal received by a plurality of antenna elements. In this figure, reference numerals 221-1 to 221-1 denote antenna elements, and another antenna element in which the antenna elements 221-1 to 5 are located at a distance L from the antenna element 221-1 in the angle θ direction (FIG. (Not shown). Here, the received signal at the time t at the m-th antenna element 221-m is represented as Φ _m (t). When the plane wave approximation is performed by setting the distance of the first antenna element 221-1 of the arriving wave from the angle θ direction from the transmission antenna to L, the wavefront becomes a line indicated by a dotted line.

Therefore, the difference between the arrival distances of the first antenna element 221-1 and the m-th antenna element 221-m is ΔL _m as shown in the figure. If the antenna element interval is d, the path difference between the mth and (m + 1) th antenna elements is d · sin θ. However, in this case, the path length difference is defined based on the first antenna element 221-1 using ΔL _m for generalization including an arrangement other than the linear array. Here, if the received signal of the m-th antenna element 221-m is divided into k-th subcarriers φ _m ^(k) (t) and expressed, the received signal Φ _m (t) of the m-th antenna element 221-m at time t t) can be expressed by the following equation (61) as the sum of the subcarrier components.

Here, if indicated a transmission signal component at time t in _S k (t), the received radio frequency signal _f c + _{f k} with respect to the center frequency _{f c} can be expressed by the following equation (62).

Here, _cm ^(k) is a coefficient (relative channel information) of the reception signal at the m-th antenna element 221-m when the reception signal at the first antenna element 221-1 is used as a reference in the k-th subcarrier. And is given by the following equation (63).

Here, α _m is a constant related to the m-th antenna element 221-m, and is defined by the following relational expression (Expression (64)).

The right side of equation (63) is expressed as the product of two natural logarithmic powers. The former is a constant without frequency dependence for each subcarrier and takes a different value for each antenna element, while the latter is a frequency Show dependencies. Considering that W is a frequency bandwidth, assuming that x = f _k / W, when α _m is smaller than 1, when x changes in a value between − ／ and + ／, a complex phase linearly changes with 2πα _{m x.} If the feature that the complex phase changes linearly is used, even if the reception weight is not calculated for all subcarriers, if some subcarriers are determined and linear interpolation of complex phase components is performed, all subcarriers can be used. Can be calculated.

In the above description, the right side of Expression (63) is a complex number having an absolute value of 1, but the actual measurement result is not strictly a complex number having an absolute value of 1 due to the influence of noise and reflected waves. In such cases, observed _c ^m a ^{_{(k) c m (k)}} / | c m (k) | with substituted on normalized value and performing the following processes. When the reception weight is calculated by using the complex conjugate value of the channel information thus standardized, the weight is obtained not as the maximum ratio combination but as the equal gain combination weight. Since the environment where the line of sight wave is dominant is assumed, it is expected that there is no large difference in characteristics between equal gain combining and maximum ratio combining.

FIG. 52 is a diagram illustrating an outline of channel estimation using limited subcarriers. In this figure, performs channel estimation for the m th antenna element in the frequency bandwidth W with respect to four sub-carriers f _A ~f _D, the relative channel information c _m which is normalized obtained by measuring A value obtained by dividing a value obtained by taking a natural logarithm (a logarithm using a base e of the natural logarithm) with respect to ^(fk) by 2πj is defined as Y _m (f _k ) (Equation (65)).

Referring to the equation (65), if f _k / W is x and Y _m (f _k ) is y, y is plotted on the graph. linear y = ax + b a straight line near the observed Y _m of _{(f k)} is to be plotted as a. Therefore, if this straight line is fitted by the least square method, the frequency dependence of the channel information can be obtained. Conventionally, there has been a technique for performing channel estimation by skipping subcarriers for transmitting a training signal. However, since there was no information on the frequency dependence of the channel information in the entire band, the linear interpolation between the two observed subcarriers and the frequency dependence before and after the two points were considered. In consideration of this, the processing has been limited to processing such as performing nonlinear interpolation. This embodiment utilizes linear frequency dependency in terms of complex phase in all frequency bands when focusing only on the line-of-sight component, and linear interpolation utilizing the frequency dependency over the entire band. Is what you do. However, it should be noted here that when taking the natural logarithm of a complex number, Y _m (f _k ) has the same value even if an integer multiple of 2πj is added to the exponent part of e. The point is that correction must be made.

For example, in the example shown in FIG. 52, the least squares method uses a regression line and four points (f _A / W, Y _m (f _A )), (f _B / W, Y _m (f _B )), (f _C / W, Y _m (f _C )) and (f _D / W, Y _m (f _D )). In addition to this, for example, with respect to η _A , η _B , η _C , and η _D given by any one of the offset values ± 1 and 0, respectively, a regression line and four points (f _A / W, Y _m (F _A ) + η _A ), (f _B / W, Y _m (f _B ) + η _B ), (f _C / W, Y _m (f _C ) + η _C ), (f _D / W, Y _m (f _D ) + η _D ), a and b are calculated for each combination of offset values, and the combination of a and b that minimizes the following equation (66) is calculated. It can be regarded as a true regression line.

In this case, since the offset value can take a value of −1, 0, and +1 in each of the four subcarriers, 3 ⁴ = 81 kinds of least squares methods are performed as a combination of all of them. Channel information of the following equation (67) may be calculated for an arbitrary frequency _fk using a and b obtained in this manner, and the reception weight may be calculated as the complex conjugate.

Supplementing a little, the coefficient a, which is the slope of this regression line, is a physical quantity corresponding to α _m in equation (64). If this is a value sufficiently smaller than 1, for example, the straight line of y = ax + b is such that when x changes between -1/2 and 1/2, the change of y is sufficiently smaller than 1 and noise and reflected waves It is considered that the offset amount within the range of ± 1 is sufficient even if the influence of the above is considered. However, when the value of ΔL _m becomes large to some extent, the change of y exceeds 1, and in such a case, it is possible to cope by expanding the selection range of, for example, 0, ± 1, ± 2. Becomes

If the value of [Delta] L _m is small to some extent, on the other hand, for example, _f B / and _Y m _{(f B)} observed in W, it is observed at the front and rear of _f A / W and _f C / W _Y m ( f _A ) and Y _m (f _C ) do not differ so much. In this case, _{| Y m (f A) -Y} m (f B) + η B | and _{| Y m (f C) -Y} m (f B) + η B | of both values is a predetermined value (e.g., 0 The offset amount η _{B that} is larger than (a value smaller than 1 such as 0.5 or 0.3) can be excluded from consideration. Here, for example, Y _m (f _A ), Y _m (f _D ), and the like are observation points at both ends. Therefore, instead of comparison with the observation points at both ends, comparison may be made with only one of the observation points. By doing so, it is possible to limit the search range while considering the offset amount, and it is possible to suppress an increase in the amount of calculation due to performing the individual least squares method.

  In the above channel estimation, it is only necessary to transmit a training signal with a limited number of subcarriers from a certain transmission antenna. Will be able to do it. For example, if the total number of subcarriers is 1000, when focusing on four subcarriers, it is possible to transmit at 250 times the transmission power as compared with the case of transmitting to all subcarriers. You can earn line gain. Therefore, the channel estimation accuracy is improved accordingly.

Furthermore, the use of the four sub-carriers _{'f D from'} f _A different subcarriers and f _D subcarriers f _A described above, it can be simultaneously carried out in the same manner as the channel estimation for the different antenna elements It is. For example, even if the transmitting side is provided with an antenna of 250 elements, if each of them transmits with only four subcarriers, it is sufficient if there are a total of 1000 subcarriers. That is, even if a MIMO channel is composed of 250 antenna elements and a large number (for example, 100) of antenna elements, channel information of a 250 × 100 MIMO channel matrix of all subcarriers can be obtained by one channel estimation. Can be obtained.

  Here, if the number of effective subcarriers (subcarriers used in data communication) is not a multiple of 4, at least a measure for adjusting the fraction is necessary. For example, with respect to a certain antenna element, a training signal to which only a slightly smaller number of 3 subcarriers are assigned may be partially used, or adjustment may be performed without assigning a training signal to some of the effective subcarriers. good. This neighborhood depends on system parameters.

  Here, for example, as shown in FIG. 16, consider a case where a plurality of first signal processing units 304 include a plurality of antenna elements. Assuming that each of the first signal processing units 304 includes 50 antenna elements and further includes five first signal processing units 304-1 to 304-5, the total number of antenna elements is 250. . When channel information in the downlink is acquired using these antennas as described above, they can all be acquired collectively. Also, when spatial multiplexing is performed using a plurality of virtual transmission paths corresponding to the first singular value, transmission directivity control over different first signal processing units 304 is not performed. When ten signal processing units 304 are provided, the same processing is performed by the first to fifth first signal processing units 304 and the sixth to tenth first signal processing units 304. Channel estimation may be performed twice.

  What is important is that the channel estimation relating to the antenna element for the same first signal processing unit 304 is simultaneously performed, and the channel estimation is performed collectively for the antenna elements belonging to the different first signal processing units 304. do not have to. By the way, when the number of subcarriers is 2,000, 4 subcarriers × 50 antennas × (5 sets + 5 sets) becomes 2000, so that channel estimation is possible by dividing into two. As described above, the total number of subcarriers, the number of subcarriers used for one-time channel estimation, and the like are parameters in the system design, and the present embodiment is similarly applicable to any other values.

Here, points to be noted when the transmitting side performs channel estimation using a plurality of antenna elements will be described. First, in the above description, as shown in Expression (63), the reception signals of the plurality of antenna elements on the reception side are divided by the reception signals of the reference antenna element (here, the first antenna element is assumed). That is, the relative channel information is used as a relative relationship of the channel information based on the first antenna element instead of the absolute channel information. If this is explicitly shown, the relative channel matrix H _relative obtained by the above-described procedure is such that the component of the channel information between the j-th transmitting antenna and the i-th receiving antenna of the k-th subcarrier is h _ij ^(k) . Then, it is given by the following equation (68).

A feature of the relative channel matrix H _relative is that each component of the row vector of the first row is 1, and all information is based on the first antenna element on the receiving side. The relative channel information for each subcarrier obtained by the above-mentioned least square method is all about the receiving antennas after the second antenna, and the channel information about the first antenna element on the receiving side is out of the above discussion. Of course, since the sampling information regarding Φ ₁ (t) has been obtained, absolute channel information can be obtained through FFT processing or the like based on this information over one OFDM symbol. However, the signal to be transmitted for example from the first antenna of the transmitting side as described above in the description only signals in the f _D subcarriers f _A, the signal at the _{'f D from'} example, another subcarrier f _A on the transmitting side The signal is transmitted from the second antenna.

That is, for the same sub-carrier f _A to only be acquired channel information of the first transmission antenna, not only can obtain channel information of the second transmission antenna with respect to the subcarrier f _{A '.} For example, when the antenna element is changed between the even subcarrier and the odd subcarrier, the channel information of the subcarrier to which the training signal is not transmitted can also be predicted by linear interpolation. If the training signal is limited to the carrier, linear interpolation between them becomes impossible.

  However, as will be described later, when this matrix is subjected to singular value decomposition, the left singular matrix is affected by any diagonal component of the second term of the product of the right side of equation (68). Will result in a similar left singular matrix.

The reason is as follows. First, when all the elements of a matrix are multiplied by a predetermined coefficient, when the matrix is singular value-decomposed, all the singular values are only multiplied by the predetermined coefficient. Becomes Next, 1 / h _1j ^(k) of each diagonal component of the second term on the right side of Expression (68) is a line-of-sight wave and corresponds to the amplitude of the received signal | h _1j because the antenna elements are near each other. ^(K) | take substantially equal values.

That is, when this matrix is multiplied by the coefficient | h ₁₁ ^(k) |, each diagonal component of the second term on the right side of Expression (68) becomes | h ₁₁ ^(k) | / h _1j ^(k) . All diagonal terms have absolute values of 1 and differ only in the complex phase. In this matrix, arbitrary column vectors are orthogonal to each other, and the diagonal term of the matrix that takes the complex conjugate of this matrix is the reciprocal of the original diagonal term, and the product of multiplying the complex conjugate matrix and the original matrix is It is a unit matrix. That is, since this is a unitary matrix, when the matrix of the first term on the right side of the equation (68) is singular value-decomposed, the left singular matrix and the matrix having singular values as diagonal terms are common, and the right singular matrix Only the matrix converted to the right singular matrix rotated by the unitary matrix described above is obtained.

Even if the entire channel matrix, which is the matrix of the first term on the right side of Expression (68), is not known, it is sufficient to know the relative channel matrix H _{relative for} calculating the reception weight vector for reception. By the way, if the reception weight vector is known, the transmission weight vector can also be obtained by the calibration process.As a result, if the channel estimation is performed by the above-described method, the reception weight vector and the transmission weight vector are not obtained on the side receiving the training signal. , Which means that both can be obtained.

  Thus, in the above-described method, for example, by performing channel estimation on the downlink, it is possible to obtain the reception weight vector and the transmission weight vector (or the weight matrix formed by combining the vectors) on the terminal station apparatus side. However, in order to obtain a reception weight vector and a transmission weight vector (or a weight matrix formed by combining the vectors) on the base station apparatus side, a training signal is transmitted from the terminal station apparatus side in the same manner, and the uplink signal is transmitted. And the same processing may be performed.

(Optional function for improving channel estimation accuracy)
Since a system to which the above method is applied targets a case where the direction of arrival of a radio wave is in a relatively narrow range, it is assumed that an antenna element having a high directivity gain is used. In this case, the line-of-sight wave is dominant in the line-of-sight environment, so the channel information is expected to show a linear behavior on the frequency axis as described above. Is expected to be slightly, and the effect may deviate from a straight line due to frequency-selective distortion. Although the degree of this shift depends on the ratio of the power of the reflected wave, for example, if regression calculation is performed using only four subcarriers, the accuracy may be low. Therefore, a comprehensive regression calculation is considered for the entire channel information of a plurality of antenna elements.

For example, considering a linear array as shown in FIG. 51, it is assumed that the path length difference ΔL _m between the first antenna element and the m-th antenna element is represented by the following relational expression (69).

That is, in Expression (64), α _m is given by the following Expression (70).

In this case, it suffices to acquire the entire channel information with the constraint condition of equation (70). It should be noted that there are some variations in how to apply the constraint. For example, a, and b determined by the least squares method described above individually for all antennas, the gradient a of the straight line by calculating a / (m-1) to two or more m because matching alpha _m its an average value, an average value ^ alpha ₂ of alpha ₂ of the formula (70) with respect to two or more m. Next, with respect to the m antenna (m-1) by substituting × ^ alpha ₂ in a, may be obtained Y-intercept in the slope for each antenna element by the least square method.

As another method, for example, there is the following method. If the arrival direction of the angle θ and the antenna element spacing d of the radio wave is obtained, alpha ₂ is because it can estimate the approximate value d · sinθ × W / c, α 2 value for testing a plurality of values in the vicinity A = α _m (ie, the slope of the regression line) is given by (m−1) × α ₂ , and the a is used to introduce the offset value η _k for each antenna as described above using the least squares method. The value of the Y-intercept b _{m of} the m-th antenna and the offset value group {η _k } used at that time are determined together, and using this α ₂ and the corresponding {b _m {η _k }}, the following equation ( F (α ₂ , {b _m }, {η _k }) given by (71) is calculated.

Here, Σ of the sum with respect to _k means the sum with respect to only the subcarriers used in each antenna element, and η _k means the offset value of the k-th subcarrier used with the m-th antenna. _{F (α 2, {b m} }, {η k}) for such alpha ₂ in a, _{F (α 2, {b m} }, {η k}) and alpha ₂ when is minimum, Using the set of Y intercepts {b _m } at that time, channel information of each subcarrier of each antenna element can be obtained.

Is a where supplemental originally value of the Y-intercept is because given by f _c ΔL m _{/ c,} should have a periodicity of formula (69) with respect to antenna number m. However, for example, if the center frequency is considered the case of 80 GHz, even if divided by the speed of light _{^{3 × 10 8, f c /}} c is the value of [Delta] L _m becomes about 267 is 3.35mm complex phase resulting in Tsu orbiting 1 Will be. will be f _c ΔL m _{/ c} + wrapping offset by exceeded 2π as the η is observed in a state of being applied, and since its offset value η can become a relatively large value, the formula (70) Such a proportional relationship cannot be grasped.

  Therefore, the constraint condition such as the equation (70) set to the value of b of the regression line for each antenna element cannot be expressed in a simple form, and each has a relationship to be evaluated as an independent Y-intercept value. This is why it is difficult to calculate an exact solution to this problem. However, if the reception weight vector is calculated using the channel information obtained by taking measures to increase the accuracy of the above-described least square method in consideration of such effects, the present embodiment can be applied with high accuracy. It is possible.

  In the example described above, an example is shown in which four subcarriers are used, but the number of subcarriers other than four may naturally be used. If the number of subcarriers used is small, the accuracy of the least-squares method will decrease, but if the number of subcarriers is increased, the transmission power per subcarrier will decrease. The estimation accuracy deteriorates. The accuracy of the slope a of the regression line of the least-squares method can be improved by performing the correction using a plurality of receiving antennas in combination, but the accuracy of the above-described method cannot be improved for the Y intercept b. . For this reason, in an actual system design, the number of subcarriers used for channel estimation is optimized in a form that balances these.

In the above description, the channel estimation by one transmission of the training signal has been described. However, as is apparent from equation (63), when acquiring the relative channel information with the reference antenna element, It is not necessary that channel estimation on the carrier be performed at the same time. For example, after the channel estimation for only f _D subcarriers f _A, by performing the channel estimation applies _{'f D from'} another subcarrier f _A de changing the time, with respect to eight subcarriers It is possible to obtain information by using By repeating these processes, it is possible to increase the accuracy of fitting a regression line by using more subcarriers.

(Supplement on how to obtain the complete channel matrix)
In the above description, each component of the matrix of the second term on the right side of equation (68) is assumed, for example, in the case of a downlink, assuming that the terminal station apparatus alone calculates the reception weight and the transmission weight on the terminal station apparatus side. Was indeterminate. However, if it is applied to a fixed-installed wireless entrance or train moving cell, it is possible to aggregate the information acquired on the downlink and the information acquired on the uplink, and perform processing after grasping both. It is.

For example, if the equation (68) is downlink information, the component of the first column vector of the relative channel matrix H _relative obtained in the uplink is subjected to calibration processing. Assuming that the component of the channel information between the j-th transmitting antenna and the i-th receiving antenna of the k-th subcarrier in the downlink converted by this calibration process is h ′ _ij ^(k) , the above-described process for the uplink is performed. h ′ ₁₂ ^(k) / h ′ ₁₁ ^(k) , h ′ ₁₃ ^(k) / h ′ ₁₁ ^(k) ,..., h ′ _1M ^(k) / h ′ ₁₁ ^(k) . When each component of the downlink relative channel matrix _Hrelative is multiplied by the coefficient h ′ ₁₁ ^(k) , the (j, j) diagonal component of the matrix of the second term on the right side is h ′ ₁₁ ^(k) / h _1j ^(K) . _Assuming that this can be approximated by h ′ ₁₁ ^(k) / h ′ _1j ^(k), it can be _{calculated as} h ′ ₁₂ ^(k) / h ′ ₁₁ ^(k) and h ′ ₁₃ ^(k) / h obtained for the uplink. ^{_{'11 (k), ···,}} h' corresponds to the reciprocal of the ^{_{1M (k) / h '11}} (k). Therefore, it is possible to obtain all channel information by using this information.

  Note that, as a supplement, the above-described approximate solution of channel estimation is a channel estimation method using the condition that only the line-of-sight wave is dominant. Naturally, in the single MIMO channel matrix, the absolute It is expected that the absolute value of a singular value equal to or smaller than the second singular value will be a very small value. Therefore, it is not suitable for spatially multiplexing a plurality of signal sequences using the MIMO channel itself. However, if a plurality of virtual transmission paths corresponding to the first singular value are used in parallel using a plurality of different antenna groups, efficient spatial multiplexing transmission is possible. If used in such an application, such an approximate solution for channel estimation will function effectively.

Furthermore, regarding the "optional function for improving the channel estimation accuracy", the accuracy in consideration of the regularity of the path length difference for each antenna element is utilized by using the case where the antenna elements on the receiving side are arranged in a linear array. The signal processing for improvement has been described. Regarding the preceding technology, the path length difference does not impose any restrictions on the arrangement of the antenna elements as described as a general ΔL _m . Since this discussion is based on the premise that the line of sight component is dominant, this technique can be applied even if the distance between the antenna elements is slightly widened if the range of the offset value η _k is allowed to be slightly expanded. It is possible. In particular, when the antenna element spacing is widened, the column vectors or row vectors of the relative channel matrix H _relative have relatively high absolute values of inner product values, but their components change almost irregularly. However, the present embodiment is applicable to a wide range of conditions if the line of sight wave is dominant, since restrictions on the antenna configuration and arrangement are not particularly strongly limited as applicable conditions.

(Processing flow of approximate solution for channel matrix acquisition when line of sight is dominant)
Hereinafter, the processing operation of the approximate solution method for acquiring the channel matrix will be described with reference to the drawings. Here, no distinction is made between the uplink and the downlink, and one of the base station apparatus and the terminal station apparatus transmits a training signal for channel estimation, and the other receives the training signal and acquires a channel matrix. If a similar process is performed in the opposite direction after acquiring the channel matrix in one direction, it is possible to acquire a bidirectional channel matrix. When performing transmission and reception on the virtual transmission path corresponding to the first singular value, a complete channel matrix is not necessarily required, and it is sufficient to obtain a relative channel matrix H _relative as in Expression (68). A method for obtaining a channel matrix will be described. Further, for convenience of explanation, a case where a training signal is transmitted on the downlink will be described as an example.

  Next, with reference to FIG. 53, a description will be given of a processing operation of an approximate solution method for obtaining a channel matrix in the present embodiment. FIG. 53 is a flowchart showing the processing operation of the approximate solution method for acquiring the channel matrix in the present embodiment. First, when the base station device transmits a training signal (step S5301), the terminal station device receives the training signal with each antenna element (step S5302). In this reception, the signal is amplified by a low noise amplifier, down-converted from a radio frequency to a baseband signal by a mixer, a signal component outside the band is removed by a filter, and sampling is performed by A / D conversion. This signal is subjected to FFT processing and converted into a signal on the frequency axis.

The training signal here is a training signal in which the transmission signal exists only on a specific subcarrier on the frequency axis in order to increase the transmission power per subcarrier, and the terminal station apparatus uses the discontinuous subcarrier. A subcarrier including a signal component is extracted from among the subcarriers, and is used as a channel estimation result of the subcarrier (step S5303). Then, the terminal station device divides the result by the same channel estimation result of the reference antenna to obtain _cm ( _k ) of Expression (63) (step S5304).

Next, the terminal station device obtains Y _m (f _k ) using equation (65) (step S5305), and calculates a and b that minimize equation (66) by the least square method in consideration of offset value η _k. (Step S5306). Subsequently, the terminal station device performs channel estimation for each subcarrier by substituting into Equation (67) based on a and b obtained here (step S5307), and further performs calibration to perform bidirectional communication. Is obtained (step S5308), and this is recorded and managed as a relative channel matrix (step S5309).

  On the other hand, it is more important to effectively manage the transmission and reception weight vectors than to record and manage the channel matrix. In this case, the terminal station apparatus obtains the channel estimation result of each subcarrier by equation (67), The value decomposition is performed, and the reception weight vector is acquired as the first left singular vector (step S5310). Next, the terminal station device performs a calibration process on the reception weight vector (step S5311) to obtain a transmission weight vector. Then, the terminal station device records and manages the transmission / reception weight vector thus obtained (step S5312).

The above is a case in which channel estimation is performed with a single antenna element closed. However, as described above, it is also possible to improve estimation accuracy in consideration of a plurality of antenna elements. FIG. 54 is a flowchart showing a processing operation of an approximate solution method of a channel matrix using a plurality of antennas. As in the processing operation shown in FIG. 53, when receiving a training signal at each antenna element in the terminal station apparatus, a predetermined reception process is performed (steps S5401-1 to S5401-3), and a signal component is generated in the discontinuous subcarriers. Are extracted and used as channel estimation results for the subcarriers (steps S5402-1 to S5402-3). Subsequently, the terminal station device divides the result by the same channel estimation result of the reference antenna to obtain _cm ( _k ) of Expression (63) (Steps S5403-1 to S5403-2), Y _m (f _k ) is obtained for each antenna element according to equation (65) (steps S5404-1 to S5404-2).

Here, the terminal station apparatus, for example, alpha value of ₂ is set at a certain interval, the slope of the regression line relative to the Y _m of the m antenna elements _{(f k)} as a = (m-1) α 2 are fixed and determine the b by the least square method in consideration of the offset _{value, F (α 2, {b} m}, {η k}) of the formula (66) determine the alpha ₂ that minimizes (step S5405) Using the α ₂ , the least squares method in consideration of the offset value in each antenna element is performed using Equation (71) (Step S5406), and based on a and b obtained here, Equation (67) is used. Substitution is performed for the channel estimation value of each subcarrier (step S5407), and calibration is further performed (step S5408) to obtain a bidirectional relative channel matrix, which is recorded and managed (step S5409).

  On the other hand, it is more important to effectively manage the transmission and reception weight vectors than to record and manage the channel matrix. In this case, the terminal station apparatus obtains the channel estimation result of each subcarrier by the equation (67), Value decomposition is performed, and a reception weight vector is acquired as a first left singular vector (step S5410). Next, the terminal station device performs a calibration process on the reception weight vector (step S5411) to obtain a transmission weight vector. Then, the terminal station device records and manages the transmission / reception weight vector thus obtained (step S5412).

  Next, a processing operation for obtaining an approximate solution of a channel matrix from a bidirectional channel estimation result will be described with reference to FIG. FIG. 55 is a flowchart showing a processing operation for obtaining an approximate solution of a channel matrix from a bidirectional channel estimation result. First, the downlink will be described. When the base station device transmits the training signal (step S5501), the terminal station device receives the training signal at each antenna element (step S5502).

The training signal here is a training signal in which the transmission signal exists only on a specific subcarrier on the frequency axis in order to increase the transmission power per subcarrier, and the terminal station apparatus uses the discontinuous subcarrier. A subcarrier including a signal component is extracted from among the subcarriers, and is used as a channel estimation result of the subcarrier (step S5503). Then, the terminal station device divides the result by the same channel estimation result of the reference antenna to obtain _cm ( _k ) of Expression (63) (step S5504).

Next, the terminal station device obtains Y _m (f _k ) using equation (65) (step S5505), and calculates a and b that minimize equation (66) by the least squares method in consideration of offset value η _k. (Step S5506). Subsequently, the terminal station device performs channel estimation for each subcarrier by substituting the obtained a and b into equation (67) based on the obtained a and b (step S5507).

  Next, the uplink will be described. When the terminal station device transmits the training signal (step S5508), the base station device receives the training signal at each antenna element (step S5509).

The training signal here is a training signal that is missing on the frequency axis in order to increase the transmission power per subcarrier, and the base station apparatus includes a signal component in the discontinuous subcarrier. The subcarrier is extracted and used as the channel estimation result of the subcarrier (step S5510). Then, the base station apparatus divides the result by the same channel estimation result of the reference antenna to obtain _cm ( _k ) in Expression (63) (Step S5511).

Next, the base station apparatus obtains Y _m (f _k ) from equation (65) (step S5512), and calculates a and b that minimize equation (66) by the least squares method in consideration of offset value η _k. (Step S5513). Subsequently, the base station apparatus performs channel estimation for each subcarrier by substituting the obtained a and b into equation (67) based on the obtained a and b (step S5514).

  Next, the absolute channel matrix is calculated from the relative channel matrix of the uplink and the downlink by equation (68) (step S5515). Then, calibration is performed (step S5516), and this is recorded and managed as a bidirectional channel matrix (step S5517).

  Here, the calibration processing in step S5516 is performed. Basically, the processing in steps S5502 to S5507 and the individual processing in steps S5509 to S5516 are performed to calculate the row vector component of the first row in equation (68). Other than indefiniteness (relative relation of channel information) can be obtained. Therefore, if the information is matched in step S5515 to compensate for the uncertainty of the row vector component of the first row in both the uplink and downlink directions, calibration is not performed (ie, step S5516 is omitted). It is also possible to obtain a bidirectional channel matrix.

As described above, the above description is a procedure for obtaining the relative channel matrix. However, if information on both the uplink and the downlink is known, not only the relative channel matrix but also the channel matrix as an absolute value can be obtained. You can ask. After the training signal is received by each antenna element as in the above description, the channel estimation of each subcarrier is performed by equation (67) by a predetermined process. This is obtained separately on the downlink terminal station device side and on the uplink base station device side. Collected both obtained _{^{results, (h 11 (k) /}} h 11 (k), h 12 (k) / h 11 (k), h 13 (k) / h 11 (k), ···, h _{1M ^(k) / h} ^{11 (k))} to the first item of the matrix of the right side of the equation (68) by multiplying the matrix with diagonal elements right from the relative channel matrix _{1 / h} ¹¹ a ^(k) The absolute multiplied channel matrix can be obtained. If this is subjected to a calibration process, a bidirectional channel matrix can be obtained, and this is recorded and managed.

  As described above, in a line-of-sight environment and when the antenna element spacing is narrow, even if the channel information has frequency dependence, it is expected that the relative channel information will show a linear frequency dependence in terms of complex phase. it can. In addition, by limiting the subcarriers used for the training signal and increasing the transmission power per subcarrier for transmission, it is possible to improve the channel estimation accuracy before forming directivity. By using these features and performing channel estimation with a small number of subcarriers, it is possible to provide a method of estimating while maintaining accuracy by estimating channel information of subcarriers that have not been obtained by the least square method. . Furthermore, if a different subcarrier is used for each antenna element, it is possible to simultaneously transmit a training signal from a plurality of antenna elements, and it is possible to support a plurality of transmission antennas that cannot be solved by implicit feedback, Channel information can be efficiently and simultaneously fed back to a plurality of antenna elements or antenna element groups.

  Since the actual channel contains the reflected wave, the channel information does not become a straight line on the frequency axis but includes many errors.However, if the least square method is performed at a larger number of samples, the line-of-sight wave component It is expected that features can be extracted.

[Seventh Embodiment]
[Approximate solution of weight vectors corresponding to a plurality of first singular values]
(Overview of Basic Principle According to Seventh Embodiment)
In the case of normal MIMO transmission, it has been considered ideal to keep the antenna element spacing as large as possible for the purpose of reducing the correlation between antenna elements. However, in order to actively use the virtual transmission path corresponding to the first singular value as described above, it is better to narrow the antenna element interval and increase the correlation. Of course, if the distance between the antenna elements is less than 1/2 wavelength, mutual coupling between the antenna elements cannot be ignored. Therefore, it is necessary to set a distance of about 1/2 wavelength. If so, it is ideal to reduce the element spacing of the antenna to increase the correlation. In the above-described approximate solution of channel estimation, a method which can be applied without much restriction on the antenna arrangement has been described. The transmission / reception weight can be calculated by a simpler method.

  Here, although the channel correlation itself of the channel matrix is large, for example, when the channel information between the first transmitting antenna and the first receiving antenna is compared with the channel information of the second antenna receiving with the same first antenna for transmission, In most cases, the values indicate completely different values due to the difference in path length. In order to perform singular value decomposition, it is necessary to perform exactly the same operation irrespective of whether the channel correlation is strong or weak. The load required for the calculation is also not insignificant. Particularly, since the number of antenna elements is enormous, the matrix size becomes enormous, and it is necessary to calculate the right singular vector and the left singular vector corresponding to the first singular value by a simple operation.

  Here, considering the high correlation of the antennas, an approximate solution of the right singular vector and the left singular vector corresponding to the first singular value is obtained by the following method. First, when the first left singular vector is used as the reception weight vector, one virtual antenna element is formed by multiplying the reception signal of each reception antenna element by the weight vector. Since this forms a beam tuned to the line of sight wave, it is expected that the beam is directed from the receiving antenna group to the transmitting antenna group side. Since the transmitting-side antenna elements are arranged at intervals of at least about 波長 wavelength, the first right singular vector of the singular value decomposition is formed strictly in a form that takes its spread into account. However, since it is close to a state in which one virtual antenna element is arranged in a very small area, reception toward a specific one antenna installed near the center of gravity of these transmission antenna groups is considered. It is expected that the approximation can be made approximately by the weight vector.

Therefore, a reception weight vector directed to one specific transmission antenna element near the center of gravity of the transmission antenna group is determined, and a gain at the time of transmission using the first right singular vector determined by singular value decomposition on the transmission side, Let us consider a case where the right and left singular vectors corresponding to the first singular value are used as the transmission and reception weight vectors. First, as an example, a channel matrix is assumed in which the (i, j) -th component of the matrix is given by the line-of-sight wave model of Expression (20) using 31 antennas for transmission and reception. Let the first right singular vector be v ₁ and the first left singular vector be u ₁ when the singular value decomposition is performed as in equation (4). The gain when the transmission / reception weight vector is formed based on these is given by the following equation (72).

On the other hand, a SIMO channel vector (h _1,16 , h _2,16 , h _3,16 ,..., H _31,16 ) ^T between the 16th element at the center of the 31 elements of the transmission antenna group and the reception antenna group. , The vector obtained by normalizing this Hermite conjugate vector is approximately equivalent to the reception weight vector _wrx-app. (Equation (73)).

  The gain in this case is as shown in the following equation (74).

  Further, a configuration in which a training signal is transmitted in a limited manner from the antenna element near the center of gravity of the antenna group not only in one direction but also in both directions to obtain both reception weight vectors, and a transmission weight vector is obtained therefrom by a calibration process. Then, the transmission weight vector and the reception weight vector can be extremely simplified from the same channel information feedback to the transmission / reception weight vector calculation processing. The gain at this time is given by the following equation (75).

Here, _Grx-app. -G _ideal and G _app. FIG. 56 shows the result of evaluating the distribution characteristics of -G _ideal . The evaluation conditions were as follows. The antenna 31 elements were arranged as a linear array on an xy two-dimensional plane, which is one wavelength apart, parallel to the y-axis, and ± 5 [ m] and installation points were determined by uniform random numbers in the range of ± 5 [m] in the y-axis direction, and the CDF distribution of the gain difference was evaluated. As parameters, the case where the frequency is 20 GHz and the distance L in the x direction is 20 m, 30 m, and 100 m are evaluated.

Looking at these results, there is almost no difference between the case of using the approximate weight vector and the case of using the ideal weight vector by singular value decomposition, and the closer the distance between the transmitting and receiving stations, the larger the gain gap becomes. It can be seen that the gain difference is within 0.25 dB or less even at 20 m. That is, in the present embodiment, when performing signal transmission using the virtual transmission path corresponding to the first singular value, the reception weight vector w _rx-app. And a calibration process is performed on the transmission weight vector w _tx-app. By performing signal transmission / reception with the above, communication can be performed which is almost equal to the case where the optimum transmission / reception weight vector obtained by singular value decomposition is used. Here, the description has been given by taking the 20 GHz band as an example, but in the case of a higher frequency such as the 80 GHz band, the antenna element interval can be shortened by shortening the wavelength, so that it is physically almost one point. Since the antenna is arranged near, the approximation accuracy is further improved and the gain gap can be ignored.

(Option for approximate solution of weight vectors corresponding to multiple first singular values)
The above is the basic principle of the simple weight vector calculation method, but some problems remain as it is. For example, when a large antenna is used to gain a gain in line design as described above, the accuracy of channel estimation for one antenna to one antenna is not very high. Therefore, similarly to the technique used in “approximate solution for channel matrix acquisition when line-of-sight waves are dominant” (sixth embodiment), the number of subcarriers used for channel estimation is greatly limited, and Is applied to improve the channel estimation accuracy for the subcarrier by increasing the transmission power of the subcarrier. Furthermore, by utilizing the fact that the amount of space between antenna elements is small, it is possible to maintain channel estimation with fairly high accuracy even when the transmitting antenna that transmits the training signal in channel estimation is slightly away from the vicinity of the center of gravity of the antenna group. Become.

  For example, when performing the same evaluation as in FIG. 56, it is considered that the transmission / reception weight vector is obtained using an antenna element slightly away from the center of gravity without using an antenna near the center of gravity. As an example, when 61 elements are arranged in a linear array in the same manner as in the evaluation of FIG. 56 at a half wavelength interval, and a transmission / reception weight vector using the 31st antenna element corresponding to the center of gravity is used, the 46th antenna separated by 15 elements is used. When the element is used, the gain when the 61st antenna element at the end of the linear array is used is evaluated as a deterioration amount from the case where an ideal transmission / reception weight using singular value decomposition is used for transmission and reception.

  FIG. 57 is a diagram illustrating a gain characteristic in an approximate solution of transmission and reception weight vectors using an antenna distant from the center of gravity. As can be seen from the figure, the median gain gap (50% CDF value) is 0.5 dB for the 46th antenna element, and even for the 61st antenna element, even under severe conditions such as a frequency of 20 GHz and a distance of 20 m. 0.8 dB. If this degree of gain reduction is allowed, the training signals of different subcarriers are transmitted using all the 61 antenna elements, and the training signals of all the frequency components for which the training signals have been transmitted by any of the antenna elements are used. The transmission / reception weight vector at the receiving station can be easily obtained.

  For example, if a training signal of 4 subcarriers is transmitted from each antenna element and the number of antenna elements is, for example, 64, it is possible to simultaneously perform channel estimation for 256 subcarriers. In the above example, a linear array was described for the sake of simplicity. However, if a 16 × 16 square array antenna (256 elements in total) of a square lattice with a half wavelength interval is used, the above-described FIG. Since similar characteristics are expected, it is possible to estimate 1024 subcarriers at once by using 256 elements at the same time.

  Here, assuming an uplink, there may be cases where not so many antenna elements can be arranged on the terminal station apparatus side, but even in that case, it was described in "Approximate solution for channel matrix acquisition when line of sight is dominant" Similarly, it is known that the weight value between the subcarriers of each component of the transmission / reception weight vector (corresponding to an individual antenna element) shows a gradual variation in complex phase. In the graph of the complex phase, if the offset value is corrected in consideration of the return of the 2π period, the weight value is calculated assuming that the weight value falls within a linear gradual change in the entire band. It is also possible to acquire the weight value of a subcarrier that has not been acquired by performing linear interpolation while extending the complex phase of the weight value outside the range of -π to + π. In this case, it is preferable to use relative channel information based on the complex phase of the reference antenna as the channel information, and obtain the reception weight from the relative channel information.

  Here, an example of using an antenna element near the center of gravity in the entire antenna will be described with reference to FIG. FIG. 58 is a diagram illustrating an example of an antenna element used for transmitting a training signal when a linear array is used. In this figure, the case of a linear array of 31 elements is taken as an example, and the center of gravity antenna is the 16th antenna in the middle (indicated by a double circle in the figure). Basically, as shown in FIG. 58 (a), only the sixteenth element, which is the antenna element at the center of gravity, is used for transmitting the training signal. When the number of subcarriers used for the channel information is limited, the number of subcarriers from which channel information can be obtained decreases, and the accuracy of interpolation of channel information during that period decreases. Therefore, for example, even if the training signal is transmitted within a range of ± 5 elements as shown in FIG. 58 (b), or the training signal is transmitted within a range of ± 10 elements as shown in FIG. 58 (c). Good. As shown in FIG. 57, if there is no problem in channel estimation accuracy, it is also possible to use all elements for transmitting the training signal as shown in FIG. 58 (d).

  Next, FIG. 59 is a diagram illustrating an example of an antenna element used for transmitting a training signal when a close-packed two-dimensional antenna arrangement is used. In this figure, the case of a two-dimensional array of 37 elements is taken as an example, and the symbols a to z and A to K are given for convenience of explanation. The center-of-gravity antenna is the a-th antenna in the middle (indicated by a double circle in FIG. 59). Since the basis is the antenna element of the center of gravity as shown in FIG. 59 (a), only the a-th element is used for transmitting the training signal. However, in this case, in order to improve the transmission power per subcarrier, the training signal is not transmitted. If the subcarriers used for transmission are limited, the number of subcarriers from which channel information can be obtained decreases, and the accuracy of interpolation of channel information during that period decreases. Therefore, for example, as shown in FIG. 59 (b), the training signal is transmitted in the range of the bth to sth elements (shown by black circles in FIG. 59) in addition to the ath element, and the remaining tth to tth elements are transmitted. The K element antenna is used for communication but not for channel feedback. Alternatively, the configuration may be such that all of these antenna elements are used for both transmission of the training signal and data communication.

  The same concept can be applied when the number of antenna elements available for transmitting the training signal is small. For example, when a one-dimensional linear array is used for transmission / reception for some reason, only antenna elements near the center of gravity as shown in FIG. 58B can be used for channel estimation accuracy for transmitting a training signal. Separately, it is possible to add and operate an antenna element used only for training signal transmission. FIG. 60 is a diagram showing an example in which an antenna element for transmitting a training signal is added near the center of gravity of the linear array. As shown in this figure, the antenna elements E, D, C, B, A, z, r, g, a, d, l, t, u, v, w, x, and y constitute a linear array. The center of gravity is the element a, and the elements a to s are arranged at the center of this element in the closest packing manner. The antenna elements b, c, e, f, h, i, j, k, m, n, o, p, q, and s that are not in the linear array are not used for data communication. A training signal is transmitted using the a to s elements.

  This is merely an example. For example, even when a close-packed two-dimensional antenna arrangement is used as shown in FIG. 59, the training signal is basically placed around the a-th element while performing close-packing at intervals of 5 wavelengths. A configuration may be used in which antenna elements are intensively arranged and used in a close-packed manner at an interval of 1/2 wavelength. As described above, the density of the antenna elements may be increased near the center of gravity, and the training signal may be transmitted using only the antenna elements near the center of gravity. Also, in the description of FIG. 60, the antenna elements b, c, e, f, h, i, j, k, m, n, o, p, q, and s deviating from the linear array are not used for data communication. There is no problem in using these antenna elements for data communication.

FIG. 61 is a diagram illustrating a specific example of linear interpolation of approximate weight values. The horizontal axis of the graph represents the subcarrier, and the vertical axis represents the complex phase of the weight value. For example, it is assumed that the weight values of f ₁ , f ₉ , f ₁₆ , and f ₂₅ are obtained at plot intervals 291 to 294 at eight subcarrier intervals. This because complex phase are given so as to be a range of [pi from + [pi, linear interpolation based on example plot points 291-295, the complex phase of the weight value of f ₂₅ is exceeds the + [pi, actual observations The plotted point 294 has a value near -π. However, since the subcarriers are complex phase when changes f ₂₅ from f ₁₆ can not cause any change near -2.pi., inevitably plotted points 295 plus the offset of the plot points 294 + 2 [pi actual complex It can be considered a phase.

By using the complex phase value corrected by adding the offset in this way, it is possible to obtain the complex phase therebetween by linear interpolation. For example, for sub-carrier f ₂ ~f _8, it is also possible to between interpolation in tie between the plot points 291 and plot points 292 in a straight line. Similarly, if the least squares method is applied using the plot points 291 to 293 and the plot point 295, a linear regression line can be obtained, and the weight value of each subcarrier may be given by the regression line. At this time, it is possible to use the weight values of the acquired subcarriers f ₁ , f ₉ , f ₁₆ , and f ₂₅ as they are. When a weight value of a carrier is given, an estimation error component such as noise may be suppressed in some cases. Furthermore, in addition to the other plot points, an extremely large change point (for example, a change amount of ± π / 2 or more) in consideration of the offset of 2π from the weight value of the neighboring subcarrier is estimated. It is also possible to increase the accuracy by excluding the low accuracy from the linear interpolation.

  Next, with reference to FIG. 62, an operation of a complex phase offset determination process in linear interpolation of approximate weight values performed by the base station device or the terminal station device will be described. FIG. 62 is a flowchart showing the operation of the process of determining the complex phase offset in the linear interpolation of the approximate weight value. Here, description will be made assuming that the processing operation shown in FIG. 62 is performed by the base station apparatus. If the number of subcarriers used for channel estimation can be secured to some extent (including a case where training signals are transmitted from a plurality of antenna elements), the interval between subcarriers on which channel estimation is performed can be relatively narrow. It is considered that the difference between the complex phases of the channel estimation results of adjacent subcarriers is sufficiently small.

  Therefore, after performing channel estimation using discrete subcarriers, the base station apparatus reads out each component of the transmission / reception weight vector (or channel vector or channel matrix) for complex phase information interpolation (step S6201). Then, the base station apparatus reads weight values Wk and Wk 'of the continuous subcarriers (step S6202). At this time, for each vector or matrix, the acquired channel information is read out, for example, in ascending order of the subcarrier number. Then, the base station apparatus calculates the complex phases θ (Wk) of Wk and Wk ′ of the continuous subcarriers that have been acquired (since they are discrete, k and k ′ are not necessarily adjacent). θ (Wk ′) is obtained (step S6203). Subsequently, the base station apparatus adds + 2π when the difference between the complex phases θ (Wk ′) and θ (Wk) is equal to or less than −π (steps S6204 and S6205), and conversely, when the difference is equal to or more than π. By subtracting 2π (steps S6206 and S6207), the absolute value of the difference between the corrected complex phase θ (Wk) and θ (Wk ′) becomes equal to or smaller than π.

  Basically, this alone is sufficient. However, when it is determined that the information of the complex phase difference of ± π / 2 or more has low credibility, the corrected complex phases θ (Wk) and θ (Wk ′) Are compared again (steps S6208 and S6209). If the difference is equal to or less than -π / 2 or equal to or more than π / 2, the channel information is determined to be unreliable, and the weight value Wk ′ is discarded (step S6208). S6210). If all subcarriers that have been acquired have been inspected, the processing ends.If not, the next subcarrier is read out, and the same processing is performed between the values of the subcarriers that have been processed immediately before. This is performed (step S6211).

  Although the correction processing is performed for the complex phase, since the actual weight value is given by Exp (jθ (k), the correction of the complex phase 2π does not cause any difference as a weight. It is possible to apply an arbitrary interpolation process on the corrected complex phase in this manner, and to perform linear interpolation from two adjacent complex phases. It is also possible to perform linear interpolation in a least-squares manner from the complex phases of a plurality of points, or to perform other non-linear interpolation. Extrapolation may be used.

  In the various descriptions above, the “offset value” was introduced by linear interpolation (regression calculation by the least squares method) of the channel vector, the channel matrix, the transmission / reception weight vector, etc. The correction of the complex phase performed without involving a rapid change in the complex phase between the subcarriers is basically equivalent to the introduction of the “offset value”. Introduction of “offset value” (processing such as performing regression calculation by a least squares method on a plurality of offset value candidates) is not always necessary and can be omitted. Note that the above-described correction processing can be similarly used in the processing in the sixth embodiment.

(Processing of approximate solution of weight vector corresponding to a plurality of first singular values)
Next, the processing operation of the approximate solution method of the weight vector corresponding to the plurality of first singular values will be described with reference to FIG. FIG. 63 is a flowchart showing a processing operation of an approximate solution method of a weight vector corresponding to a plurality of first singular values. Here, regardless of the uplink and downlink, either the base station apparatus or the terminal station apparatus transmits a training signal for channel estimation, and the other receives the training signal and obtains a channel vector. Is used to calculate the reception weight vector. Further, a transmission weight vector is also obtained by a calibration process. By obtaining the transmission / reception weight vector in one direction and then performing the same operation in the opposite direction, it is possible to obtain a bidirectional transmission / reception weight vector. Also, for convenience of explanation, the following description will be made taking a case where a training signal is transmitted on the downlink as an example.

  The base station apparatus transmits a training signal using some subcarriers in the whole in order to increase transmission power per subcarrier (step S6301). At this time, the training signal is transmitted by the antenna elements around the center of gravity of the antenna group. The terminal station device receives the training signal and performs a predetermined receiving process (step S6302). In the reception process, the signal is amplified by a low-noise amplifier, down-converted from a radio frequency to a baseband signal by a mixer, a signal component outside the band is removed by a filter, and sampled by A / D conversion. Processing such as performing FFT processing on this signal to convert it into a signal on the frequency axis is performed. The terminal station apparatus calculates a reception weight vector by taking a complex conjugate (and, if necessary, including a normalization process) for the channel vector obtained in this way (step S6303).

  Subsequently, the terminal station device performs the complex phase correction process (the process shown in FIG. 62) (step S6304). Then, the terminal station device acquires channel vectors of all subcarriers by linear interpolation or the like (step S6305). Subsequently, the terminal station device predicts the reception weight vector of the subcarrier that has not been obtained by various interpolation processes such as linear interpolation and nonlinear interpolation, and calculates the transmission weight vector by the calibration process after obtaining all the reception weight vectors. (Step S6306). Finally, the terminal station device records and manages the transmission weight vector (step S6307).

  As described above, when the transmission / reception weight for using the virtual transmission path corresponding to the first singular value is obtained in the line-of-sight environment, the fact that the antenna element group is limited to a very narrow place is used. Then, a reception signal vector is obtained using a single antenna element near the center of gravity of the antenna group, and the reception weight can be calculated extremely easily by taking its complex conjugate. If the calibration process is performed, the transmission weight can also be obtained. If the process is performed in both directions, the base station device and the terminal station device can easily approximate the transmission and reception weight of the virtual transmission path corresponding to the first singular value. Can be sought. In addition, while only the antenna near the center of gravity is used for training signal transmission, data transmission / reception can use the entire antenna element with an increased aperture length to improve line gain and spatial multiplexing characteristics.

  If the transmission weight vector is known on the transmitting side, it is not necessary to transmit a training signal by using only the antenna near the center of gravity and limiting the subcarriers to be used. It is also possible to transmit a training signal by increasing the gain by forming a directivity using transmission weight vectors in all antenna elements using a carrier. For this reason, the present embodiment is used as a method for acquiring the initial value of the transmission weight vector, and after acquiring the transmission weight vector, forming a directivity using the transmission weight vector, transmitting the training signal, and Channel estimation is performed with the line gain increased, the reception weight vector is obtained with high accuracy by the complex conjugate of the channel vector, and the calibration process is performed on the high-precision reception weight vector to achieve high accuracy. A procedure such as obtaining a transmission weight vector may be performed. By repeating such processing sequentially, communication can be performed in a more stable state in a steady communication state.

[Eighth Embodiment]
[Simultaneous channel estimation in multiple wireless stations]
(Overview of Basic Principle According to Eighth Embodiment)
In the above “approximate solution for obtaining a channel matrix when line of sight waves are dominant” and “approximate solution for weight vector corresponding to a plurality of first singular values”, 1 It has been shown that the channel estimation accuracy of one antenna to one antenna can be improved. Further, in these techniques, training signals with different subcarriers are simultaneously transmitted from a plurality of antennas, so that channel information on different antenna elements can be obtained.

  In the description of these techniques, for example, a training signal is transmitted from a plurality of antenna elements of the same terminal station apparatus using different subcarriers, or a plurality of first signals are transmitted on the base station apparatus 303 side as shown in FIG. The case where the processing unit 304 is included has been described. In this case, the transmission of the training signal from the antenna element of the first signal processing unit 304 is adjusted by a communication control circuit or the like inside the base station device 303, and the training signals are transmitted at the same time with the timing adjusted. Was controlled to do so. On the other hand, it may be effective to simultaneously perform channel estimation for a plurality of antenna elements under other conditions. As an example, in addition to applying the present embodiment to the wireless entrance line as described above, it can be used for the access system of the small cell base station apparatus.

  FIG. 64 is a diagram illustrating an apparatus configuration for performing simultaneous channel estimation between a plurality of small cells existing in the same area. In FIG. 64, the left diagram shows, for example, an environment in which a plurality of small cell base station devices 231 to 234 are installed on a building wall of a building street, and the respective service areas spread around the base station devices 231 to 234. ing. In this service area, there are terminal station devices 235 to 239 that perform wireless communication with the base station devices 231 to 234. The terminal station device 235 is under the control of the small cell base station device 231, the terminal station device 236 is under the control of the small cell base station device 232, and the terminal station device 237 is under the control of the small cell base station device 233. The device 238 and the terminal station device 239 are under the control of the small cell base station device 234.

  As described in the background art, a small cell is installed to offload traffic in an area where traffic is locally concentrated from a macro cell. A small cell in a narrow area is set for an area where traffic is concentrated, and the small cell base station apparatus performs communication only for terminal stations in the small cell. In the case of a macro cell, it is general to perform station placement design strictly, manage the level of mutual interference between macro cells, and reuse frequency resources. However, small cells are installed in a spot manner in accordance with required conditions such as traffic concentration, and station design is not always performed systematically.

  In such an environment, a small cell configuration technology using Massive MIMO technology has been attracting attention for the purpose of reducing mutual interference in accordance with a requirement for securing line gain. In this small cell, the transmission power per element is not high in order to reduce mutual interference, but a huge line gain is secured by the directional gain formed by a large number of antenna elements. However, to secure the channel gain, the MIMO channel matrix between each transmitting antenna element and receiving antenna element needs to be known, and the channel feedback technique for this is important.

  Furthermore, the accuracy of channel estimation is also important in order to obtain this line gain efficiently, but when a training signal is transmitted for channel estimation, interference waves from neighboring small cells on the same subcarrier are generated. In the case of reception, the channel estimation accuracy is greatly degraded due to the interference signal, and not only sufficient line gain cannot be secured, but also radiation in directions other than the desired direction increases, so that the risk of mutual interference increasing. There is also. This problem is called Pilot Contamination, and the problem of contamination of the training signal due to mutual interference is an important problem.

  To simply avoid this problem, shift the transmission timing of the training signal for each small cell (when the training signal is transmitted in some small cells in the vicinity, refrain from communication in other small cells) Was considered valid. However, in this method, the overhead due to the training signal for channel estimation increases, and the transmission efficiency in the MAC layer decreases. In particular, in a small cell in 5G, since it is assumed that one base station device collectively accommodates an extremely large number of terminal station devices, those terminal station devices frequently transmit a training signal and each small cell has If the transmission timing is separated, the MAC efficiency will be greatly reduced.

  Therefore, as shown in the right diagram of FIG. 64, among a number of subcarriers in the entire frequency bandwidth W, each of the base station devices 231 to 234 separates subcarriers and performs the same time in downlink. And send the training signal. Similarly, the terminal station devices 235 to 238 of each small cell also segregate subcarriers among a large number of subcarriers in the entire frequency bandwidth W and align training signals at the same time on the uplink. Send. For example, the base station device 231 and the terminal station device 235 use a plurality of subcarrier groups 241 in the right diagram, the base station device 232 and the terminal station device 236 use a plurality of subcarrier groups 242, and the base station device 233 and the The terminal station device 237 uses a plurality of subcarrier groups 243 in the right diagram, and the base station device 234 and the terminal station device 238 use a plurality of subcarrier groups 244 in the right diagram. Since they do not overlap on the frequency axis, there is no interference of the same frequency (same subcarrier) at the same time, and the problem of pilot containment does not occur.

  For the uplink, for example, the terminal station device 238 and the terminal station device 239 under the control of the base station device 234, for example, while the terminal station device 238 uses the subcarrier group 244, the terminal station device 239 It is assumed that the training signal is transmitted using the subcarrier group 243 and that the channel information of the terminal station device 238 and the terminal station device 239 are simultaneously estimated (assuming that the terminal station device 233 and the terminal station device 237 are not communicating with each other). It is also possible to configure.

  In general, the base station device is allowed to increase the device scale, while the terminal station device side cannot increase the device scale so much. Also, as shown in FIG. 16, when a plurality of first signal processing units 304 are provided, and each of the plurality of first signal processing units 304 is provided with a large number of antenna elements, the number of antennas of the terminal station It is expected that the total number of antennas will be small. In this case, it is reasonable to simultaneously perform channel estimation with many terminal station devices in the same cell (or may include different cells) as described above.

  Specifically, for example, it is assumed that the base station apparatus includes five first signal processing units 304, each of which includes 50 antenna elements. When a training signal is transmitted on four subcarriers per antenna element, channel estimation of 1000 subcarriers can be performed simultaneously with 4 × 50 × 5 = 1000. Further, if the total number of subcarriers is 2000, for example, the base station apparatus 231 and the base station apparatus 233 perform channel estimation using even subcarriers, and the base station apparatus 232 and the base station apparatus 234 use odd subcarriers. Is possible. Depending on the directivity characteristics of the antenna, etc., assuming that each antenna element on the base station apparatus side has a certain degree of directivity, adjacent small cells may interfere with each other, In some cases, the interference may be negligible with such a small cell, and in such a case, the above-described setting is established.

  On the other hand, assuming that the number of antenna elements provided in the terminal station apparatus is 25, four subcarriers are used per element, so that one terminal station apparatus uses 100 subcarriers in total. However, since the number of subcarriers used by one base station apparatus for channel estimation is 1000, even if ten terminal station apparatuses simultaneously transmit uplink channel estimation training signals at the same time, the same frequency ( There is no interference on the same subcarrier), and channel estimation can be performed on these at the same time. Regarding the downlink, when the base station device transmits the training signal once, all the terminal station devices that have received the training signal can simultaneously perform downlink channel estimation for the own station. However, for the uplink, each terminal station device must individually transmit a training signal, and how to prevent a decrease in MAC efficiency due to overhead associated with the transmission of the training signal, and how to increase channel estimation accuracy Is an issue.

  Here, in order to simultaneously transmit and receive the above-described channel estimation between different small cells or between different terminal station apparatuses in the same small cell while changing subcarriers, (1) training to be transmitted by each terminal station apparatus It is important to be able to grasp the subcarriers of the signal, and (2) to be able to grasp the timing at which each terminal station device should transmit a training signal.

FIG. 65 is a diagram illustrating an example of a frame configuration according to the present embodiment. In FIG. 65, reference numerals 251-1 to 251-3 denote slots for the base station device to transmit training signals on the downlink, and reference numerals 252-1 to 252-3 denote data communication between the base station device and the terminal station devices. It is a slot for performing. Reference numerals 253-1 to 253-3, 254-1 to 254-3, 255-1 to 255-3, and 256-1 to 256-3 allow the terminal station device to transmit a training signal on the uplink. Represents a slot. The time length from slot 251-1 to slot 256-1, the time length from slot 251-2 to slot 256-2, and the time length from slot 251-3 to slot 256-3 are each the basic frame period. _Tb-frame . Strictly, between the end of the slot 256-1 and the slot 251-2, between the end of the slot 256-2 and the slot 251-3, etc., the switching from the uplink to the downlink is performed. May be set as a predetermined time.

  In the example shown in FIG. 65, a superframe structure having a long cycle in which a basic frame is repeated N times is adopted, and slots 253-1 to 253-3, 254-1 to 254-3, and 254-1 to 253-3 in different basic frames in the superframe are taken. 255-1 to 255-3 and 256-1 to 256-3 are allocated to different terminal station devices to transmit uplink training signals. Focusing on any one of the slots 253-1 to 253-3, 254-1 to 254-3, 255-1 to 255-3, and 256-1 to 256-3, each is divided into subcarrier groups. Have been. In the above-described example, one terminal station apparatus transmits training signals on four subcarriers in each of 25 antenna elements for a total of 1000 subcarriers. For this reason, for example, for the subcarriers of the j-th group (j is an integer of 1 to 10), the (10 × m + j) -th subcarrier is used for the integer m of 0 to 99, and the n′th subframe of the nth basic frame is used. (Since four slots from 253 to 256 are allocated in the figure, n 'is an integer of 1 to 4.) The sub-carriers of the j-th group are allocated to the slots, and the allocation is performed to the terminal station apparatus.

  In FIG. 65, slots 251-1 to 251-3 are assigned to each frame at the beginning of each basic frame, but it is sufficient that at least one slot is included in a long-period superframe structure. If it is arranged at the head of a predetermined basic frame in the above, it is possible to handle even if it is not arranged at the head of all basic frames. Similarly, the slots 253-1 to 253-3, 254-1 to 254-3, 255-1 to 255-3, and 256-1 to 256-3 are described as the same number of slots in each basic frame. However, the number of frames does not necessarily have to be the same, and in some cases, there may be a basic frame that does not include any basic frame.

  The data communication slots 252-1 to 252-3 may be provided with a bandwidth request from a terminal station device or a base station from the network side if time division TDD is assumed on the assumption that implicit feedback is performed. Bands in the data communication slots 252-1 to 252-3 are adaptively allocated according to data arriving at the station device. At this time, the uplink and downlink time lengths may be fixedly set, or the allocation may be variable according to the traffic situation. In any case, the present embodiment is applicable. Further, the data communication slots 252-1 to 252-3 are further subdivided to perform transmission and reception in units of fine slots.

  It is assumed that each base station device and terminal station device have a known start timing of a super frame and a basic frame by, for example, GPS (Global Positioning System) or some other means. Other than the GPS, the signal of the macro cell may be used, or the timing information may be notified by another wireless system. In addition, a terminal station device newly entering the area needs to first access the base station for belonging processing, but for that access, performs downlink channel estimation, and based on the estimation result, Unless the obtained reception weight vector and uplink transmission weight vector are obtained, it is not possible to secure the line gain by using a large number of antennas.

  In an ordinary wireless system, a signal usable for timing detection is generally added to the head of a wireless packet. In the present embodiment, signals in slots 256-1 to 256-3 are omitted as subcarriers. Since the signal is an imperfect signal and has no directivity, it is a signal that is generally not very suitable for timing detection. Further, even if the base station apparatus transmits another signal for timing detection, if the terminal station apparatus does not form a directional beam using the reception weight vector, the base station apparatus cannot transmit signals due to insufficient line gain. The signal for timing detection cannot be received. Therefore, it is preferable that some mechanism for assuming that the heads of the basic frame and the superframe are known is implemented in the present embodiment, and in the following description, it is assumed that such timing information is known. This timing information may be grasped by utilizing another radio system such as GPS, or may be grasped by utilizing information of a macrocell (that is, synchronized with the periodicity of communication of the macrocell).

(Processing flow of simultaneous channel estimation by multiple radio stations)
Here, a processing operation for starting communication between the terminal station device and the base station device newly entering the area will be described. FIG. 66 is a flowchart showing a processing operation for starting communication between the terminal station device and the base station device newly entering the area. When the terminal station apparatus starts communication (for example, when the user turns on the power of the terminal station apparatus), the terminal station apparatus obtains the start timing of the basic frame and the superframe by a predetermined means (step S6601), and Is received (step S6602). Then, the terminal station device acquires a reception weight vector on the downlink and a transmission weight vector on the uplink from the received slots 251-1 to 251-3 (step S6603).

  Next, the terminal station apparatus determines any one of the slots 253-1 to 253-3, 254-1 to 254-3, 255-1 to 255-3, and 256-1 to 256-3, and any one of the slots. In a group of subcarriers, a combination of subcarriers with conditions assigned to the system as slots for random access and subcarriers with conditions similarly assigned to the system as subcarriers for random access (Step S6604), and a training signal for channel estimation is transmitted by random access thereto (step S6605). For example, predetermined subcarriers in the slots 256-1 to 256-3 at the end of the basic frame (in the above example, divided into ten groups, all of the groups or some of the groups) may be randomly assigned. A slot for access may be used, or a slot for random access may be arranged only in a predetermined group of subcarriers in the last slot 256-1 in the first basic frame in the superframe.

  Each time the base station apparatus receives the random access slot (step S6621), it performs FFT processing and acquires the reception level of a subcarrier that may include a training signal as a random access slot (step S6622). ). Then, if the subcarrier of the received signal is at a predetermined level (step S6623), the base station apparatus regards this as initial access from a new terminal station apparatus, and performs uplink channel estimation from the training signal. (Step S6624). Subsequently, the base station apparatus calculates the downlink reception weight vector and acquires the downlink transmission weight vector by calibration (step S6625). Using the transmission weight vector obtained here, the base station apparatus transmits an uplink signal addressed to the terminal station apparatus to any of the subdivided slots among the data communication slots 252-1 to 252-3. Signal transmission of a slot assignment instruction for transmission is performed (step S6626). Here, since the base station apparatus does not know the identity of the terminal station apparatus, it is not possible to use the identification number of the terminal station apparatus or the like for the assignment instruction. For example, the information such as the slot and subcarrier at which the terminal station apparatus transmitted the training signal by random access is specified, and the terminal station apparatus is informed that the assignment is to the own station.

  In response to this, after transmitting the training signal by the random access described above, the terminal station device forms a directional beam with the obtained reception weight vector, and determines a timeout (step S6606) and transmits a signal from the base station device. It waits (step S6607). If the slot assignment instruction signal transmitted by the base station apparatus can be received using the above-described reception weight vector (step S6608), control information containing information of the own station is transmitted using the designated slot. (Step S6609), and a request for belonging to the base station is made.

  The base station apparatus waits for a signal from the terminal station apparatus while determining a timeout in the slot assigned by the base station using the reception weight vector acquired in the previous random access slot (step S6627). The signal is received with high quality by utilizing the line gain secured by forming a directional beam in both the base station apparatus and the base station apparatus (step S6628). After exchanging the information, the base station device manages the belonging of the device on its own database (step S6629).

  In addition, the base station apparatus is configured to use any one of the slots 253-1 to 253-3, 254-1 to 254-3, 255-1 to 255-3, and 256-1 to 256-3, and any one of the slots. The group of carriers is allocated as a slot for channel feedback with the terminal station device (step S6630), and the contents are similarly notified by subdivided slots among data communication slots 252-1 to 252-3. I do. In response to this, the terminal station device grasps a channel estimation slot for periodically transmitting a training signal to the base station device on the uplink (step S6610). Thereafter, band allocation and band request are performed in the subdivided slots among the data communication slots 252-1 to 252-3, and the base station apparatus and the terminal station apparatus appropriately communicate while updating the transmission / reception weight vector. Continue.

  Although detailed description has been omitted in the above processing, in order to avoid stagnation of processing due to incorrect information exchange due to code errors, a general technique of setting a timer is used to determine whether there is a response by timeout. It is also possible to apply a combination of management technologies. In addition, when the base station device implements a plurality of first signal processing units as shown in the first embodiment, the amount of control information such as an association request is limited, so that a plurality of first There is no necessity to perform communication while performing spatial multiplexing by utilizing the signal processing unit. Therefore, one specific first signal processing unit is selected, and only the transmission weight vector directed to the first signal processing unit is used. Alternatively, the control information may be transmitted. Similarly, in transmitting control information from the base station apparatus, a configuration may be adopted in which one specific first signal processing unit is selected and control information is transmitted without performing spatial multiplexing.

  Next, a transmission / reception signal processing operation of the base station device according to the present embodiment will be described with reference to FIG. FIG. 67 is a flowchart showing the transmission / reception signal processing operation of the base station apparatus. In FIG. 67, (a) shows a reception process, and (b) shows a transmission process. First, the receiving process will be described. Upon acquiring the frame timing (step S6701), the base station apparatus recognizes the reception timing of the wireless packet from the terminal station, which is grasped due to the centralized base station control, in the data communication slot subsequent to the first slot of the basic frame. (Step S6702). Then, the base station apparatus reads out the reception weight vectors of the plurality of first signal processing units 304 of the own station for the terminal station apparatus from the memory (step S6703), and receives the reception weight vectors of the respective first signal processing units. The weight vector is multiplied (step S6704).

  Next, the base station apparatus aggregates one signal for each virtual transmission path, and based on the channel estimation result for the training signal received in the head area of each wireless packet (step S6705), A second reception weight matrix is calculated (step S6706), and a signal detection process is performed using the matrix (step S6707). Then, the base station apparatus repeats the signal detection processing in the same manner as long as the data covers a plurality of symbols (step S6708), and determines whether or not the frame is completed after the data is completed (step S6709). , The above-described processing (signal reception from another terminal station apparatus) is repeated. At the end of the frame, the base station apparatus receives a slot for transmitting a training signal from the terminal station apparatus at the end of the frame (step S6710), and calculates a transmission / reception weight vector for the terminal station apparatus based on the received signal. Then, this is recorded and managed, and the process ends (step S6711).

  Next, transmission processing will be described. When the base station device acquires the frame timing (step S6721), the terminal station device transmits a training signal for calculating the reception weight vector used in the first signal processing unit in the first slot of the basic frame (step S6722). . Then, the base station apparatus recognizes the transmission timing of the wireless packet to the terminal station apparatus that has been grasped due to the base station centralized control (step S6723), and stores a plurality of first stations of the own station for the terminal station apparatus from the memory. The transmission weight vector in the signal processing unit 304 is read (step S6724).

  Subsequently, after generating the transmission signal (step S6725), the base station apparatus multiplies the first transmission weight vector for each first signal processing unit (step S6726), and performs signal transmission (step S6727). The base station apparatus repeats the above processing if the data continues (step S6728), and determines whether or not the frame has ended if the data has ended (step S6729), and the frame has ended. If not, the next transmission processing or the reception processing of FIG. 67 (a) is repeated. If the frame has ended, the process ends.

  Next, a transmission / reception signal processing operation of the terminal station device according to the present embodiment will be described with reference to FIG. FIG. 68 is a flowchart showing the transmission / reception signal processing operation of the terminal station device. In FIG. 68, (a) shows a reception process, and (b) shows a transmission process. First, the receiving process will be described. Upon acquiring the frame timing (step S6801), the terminal station apparatus receives the training signal transmitted by the base station apparatus in the first slot of the basic frame (step S6802), and based on this, sets the first reception weight vector to a virtual value. It is calculated for each dynamic transmission path (for each of the plurality of first signal processing units) (step S6803). Upon receiving the wireless packet from the base station apparatus (step S6804), the terminal station apparatus reads out the reception weight vectors in the plurality of first signal processing units 304 of the terminal station apparatus from the memory, and converts each of the virtual weights into a reception signal vector. The receiving weight vector corresponding to the transmission path is multiplied (step S6805).

  Next, the terminal station device aggregates one signal for each virtual transmission path, and based on the channel estimation result (step S6806) for the training signal received in the head area of each wireless packet, obtains a linear signal. A second reception weight matrix is calculated (step S6807), and a signal detection process is performed using this (step S6808). The terminal station device repeats the signal detection process similarly while the data covers a plurality of symbols (step S6809). After the end of the data, the terminal station device determines whether or not the frame has ended (step S6810), and if the frame continues, if there is an assignment to its own station, repeats the above-described reception processing. At the end of the frame, the process ends.

  Next, transmission processing will be described. Upon acquiring the frame timing (step S6821), the terminal station device performs a calibration process on the reception weight vector obtained in the above-described reception process, and calculates a first transmission weight vector (step S6822). When recognizing the transmission timing of the own station (step S6823), the terminal station device generates a transmission signal (step S6824), and then multiplies the signal for each virtual transmission path by the first transmission weight vector. (Step S6825), the result of the multiplication is added and synthesized for each antenna element, and signal transmission is performed (Step S6826).

  Next, the terminal station device repeats the above processing when the data is continued (step S6827), and determines whether or not the frame is completed when the data is completed (step S6828). If it has not been completed, the next transmission processing is repeated if there is an assignment to the own station. If the frame has ended, the terminal station device transmits a training signal in the slot and subcarrier allocated to the own station for the training signal at the end of the frame (step S6829), and ends the process.

  Although the details are omitted here, the above processing is basically performed for each subcarrier on the frequency axis. The above processing has been described in the case of the configuration shown in FIG. 69 as a precondition. FIG. 69 is a diagram showing a frame configuration as a precondition. In this figure, reference numerals 257-1 to 257-3 indicate training signals, and reference numerals 258-1 to 258-3 indicate data payloads. In the above description of FIG. 67 and FIG. 68, first, channel estimation for signal separation of MIMO transmission is performed on signal sequences for each of a plurality of virtual transmission paths that are aggregated by multiplying by the first reception weight vector. As described above, the training signals 257-1 to 257-3 are used for this. For the training signals 257-1 to 257-3 and the data payloads 258-1 to 258-3, it is possible to adaptively use the uplink and the downlink in a time-division manner so that it is not specified whether the uplink or the downlink is used. It is possible. For example, the training signals 257-1 to 257-2 and the data payloads 258-1 to 258-2 are downlink signals (transmitted by the base station), and the training signal 257-3 and the data payload 258-3 are uplink signals. (Transmitted by the terminal station), all may be downlink, and all may be uplink. In any case, the training signals 257-1 to 257-3 are signals having a sufficient line gain multiplied by the first transmission weight vector irrespective of the uplink or the downlink, and therefore have a high reception level at a sufficient level. Channel estimation is possible with high accuracy.

  On the other hand, since the training signals 251-1 to 251-3 are all signals in which directivity is not formed by the first transmission weight vector, the transmission power is concentrated on a small number of subcarriers. The estimation accuracy is relatively low. However, if it is possible to secure channel estimation accuracy that does not cause a problem in operation, it is also possible to calculate the second reception weight matrix using the training signals 251-1 to 251-3. In this case, the training signals 257-1 to 257-3 are omitted as shown in FIG. 70 instead of FIG. 69, and communication is performed using only the data payloads 258-1 to 258-3. FIG. 70 is a diagram illustrating an example in which the training signal is omitted and communication is performed using only the data payload.

  The training signals 257-1 to 257-3 shown in FIG. 69 are for performing channel estimation for signal separation of MIMO transmission. For example, if the number of spatial multiplexing is large, orthogonal training signals are necessary for the number. Yes, and a corresponding symbol number overhead is required. However, in FIG. 70, if it is assumed that the channel estimation for signal separation in MIMO transmission is performed by making full use of the training signals 251-1 to 251-3 at the head of the frame, the spatially multiplexed signal sequence is clearly separated. It is only necessary to perform channel estimation necessary for signal detection and demodulation processing of the independent signal sequence thus performed. All training signals are transmitted at the same time (for example, one symbol in OFDM), and the receiving side is as if after signal separation. The received signal processing can be performed as if the signal was a SISO (Single Input Single Output) signal. In this sense, FIG. 70 may include a training signal for signal detection / demodulation of the SISO signal.

  Next, another processing operation of calculating the second reception weight matrix in the present embodiment will be described with reference to FIG. FIG. 71 is a flowchart illustrating another processing operation of calculating the second reception weight matrix. This processing operation can be applied to both processing of the base station apparatus in the uplink and processing of the terminal station apparatus in the downlink. In the case of the uplink, the training is performed on the training signal transmitted from the terminal station device at the end of the frame, and in the case of the downlink, the training is performed on the training signal transmitted from the base station device at the beginning of the frame. Alternatively, the processing may be performed on the training signals 257-1 to 251-3 shown in FIG. Here, the operation of the base station apparatus will be described.

  First, upon receiving a training signal (step S7101), the base station apparatus calculates a first reception weight vector for each virtual transmission path (step S7102). In addition, this first reception weight vector is any one of the slots 253-1 to 253-3, 254-1 to 254-3, 255-1 to 255-3, and 256-1 to 256-3, and its slot. A reception weight vector calculated using any one of the subcarrier groups may be used instead. The base station apparatus then multiplies the training signal used to calculate the first reception weight vector by the first reception weight vector for each virtual transmission path (step S7103). By this multiplication, signal sequences of the number of virtual transmission paths are obtained. In the reception of the training signal for each virtual transmission path, since subcarriers are separated for each virtual transmission path, the crosstalk component between each virtual transmission path can also be individually extracted, and individually. The extracted channel estimation result corresponds to each component of the channel matrix of Expression (55). A second reception weight matrix is calculated based on the channel matrix thus obtained (step S7104).

For example, the case where the number of virtual transmission paths is 3 will be described as an example. As an example, the first virtual transmission path has a multiple of 3 subcarriers, the second virtual transmission path has a multiple of 3 + 1 subcarriers, and the third virtual transmission path has a multiple of 3 + 2 subcarriers. It is assumed that a training signal is transmitted using a carrier. At this time, attention is paid to a signal obtained by multiplying each subcarrier component of the received signal by the reception weight of the first virtual transmission path. Focusing on 3 multiple subcarriers, since the first receiving weight vector may perform channel estimation of the desired signal after the multiplication, h ₁₁ components of the channel matrix of 3 × 3 can be obtained.

On the other hand, when attention is paid to subcarriers of a multiple of 3 + 1, a signal component of the signal of the second virtual transmission path leaking into the first virtual transmission path can be extracted. ₁₂ components can be obtained. Similarly, focusing on a subcarrier of a multiple of 3 + 2, it is possible to extract a signal component of a signal on the third virtual transmission path leaking into the first virtual transmission path, so that h ₁₃ of the 3 × 3 channel matrix can be extracted. Components can be obtained.

Similarly, attention is paid to a signal obtained by multiplying the reception weight of the second virtual transmission path. Focusing on subcarriers 3 multiples +1, since the second receiving weight vector may perform channel estimation of the desired signal after the multiplication, h ₂₂ components of the channel matrix of 3 × 3 can be obtained. On the other hand, when attention is paid to subcarriers that are multiples of 3, a signal component leaking from the first virtual transmission path to the second virtual transmission path can be extracted, so that h ₂₁ of the 3 × 3 channel matrix can be extracted. Components can be obtained. Similarly, paying attention to subcarriers of a multiple of 3 + 2, a signal component leaking from the third virtual transmission path into the second virtual transmission path can be extracted, so that h ₂₃ of the 3 × 3 channel matrix can be extracted. Components can be obtained.

Further similarly, attention is paid to a signal obtained by multiplying the reception weight of the third virtual transmission path. Focusing on multiples of 3 + 2 subcarrier, because the third receiving weight vector may perform channel estimation of the desired signal after the multiplication, h ₃₃ components of the channel matrix of 3 × 3 can be obtained. On the other hand, when attention is paid to subcarriers that are multiples of 3, a signal component of the signal on the first virtual transmission path leaking into the third virtual transmission path can be extracted, so that h ₃₁ of the 3 × 3 channel matrix can be extracted. Components can be obtained. Similarly, focusing on a subcarrier of a multiple of 3 + 1, a signal component of the signal on the second virtual transmission path leaking into the third virtual transmission path can be extracted, and therefore, h ₃₂ of the 3 × 3 channel matrix can be extracted. Components can be obtained.

  In this way, all 3 × 3 matrix elements can be obtained. Here, the case where the number of virtual transmission paths is 3 has been described as an example, but a channel matrix can be obtained for an arbitrary number of virtual transmission paths. If these channel matrices can be obtained, the subsequent processing is the same as that of spatial multiplexing signal processing using an ordinary MIMO channel. -Thousands of signals detection can be performed by arbitrary signal processing including nonlinear signal processing such as MLD. In the above description of FIG. 67 and FIG. 68, an example in which a linear reception weight matrix is used has been described. However, as described above, signal detection processing of an arbitrary MIMO channel can be applied as a matter of course.

  Here, an example is shown in which orthogonalization is performed by dividing subcarriers used for a training signal for each virtual transmission path, but if other orthogonalized training signals can be added to each signal sequence to be spatially multiplexed, Using the orthogonality, similar signal processing can be performed using the training signals 257-1 to 257-3 shown in FIG. For example, in the case of OFDM, if 3 OFDM symbols are used, and if a training signal for each signal sequence is used while being shifted in time, it is naturally possible to perform channel estimation using all subcarriers.

(Utilization of time axis beam forming)
As described in the above-mentioned section on time-axis beamforming, in general, when an antenna element exists in a very small area, the frequency dependence of the reception weight (that is, each component of the reception weight vector) for each antenna element is described. Is expected to be relatively small. In order to increase the degree of signal separation between each virtual transmission path, it is possible to add and combine the complex phases of the received signals of each antenna element with their respective subcarriers by using the best possible reception weight. preferable. However, if the crosstalk component between the virtual transmission paths can be separated with high accuracy using the reception weight matrix of the second stage at the subsequent stage, the first reception weight vector is not a serious problem even if the accuracy is low to some extent. No.

  For example, consider the addition of Sin (θ) and Sin (θ + δ). From the trigonometric function formula, the following relational expression (76) is obtained.

  This equation means that if the efficiency at the time of addition is incomplete with in-phase synthesis and there is a phase error δ, the amplitude will be 2 × Cos (δ / 2) times instead of 2 times by the in-phase synthesis. I do. Taking the case where δ is 30 degrees as an example, the amplitude becomes Cos (30/2) = 0.6959..., and the gain loss is about 3.4%. That is, if the error is within 30 degrees for all antennas, the gain loss due to this error is only about 0.3 dB. Even when δ is 60 degrees, it is only about 1.2 dB with a gain loss of about 13.4%.

  Therefore, even in a case where F (α) in FIG. 44 is not an area exceeding 30 dB, the first signal processing unit 304 executes the processing if it is assumed that signal separation processing on the frequency axis is performed in the subsequent stage. Even if beamforming on the time axis is performed with low accuracy, it means that there is no problem if a limited decrease in gain can be tolerated. That is, after the reception weight vector on the frequency axis is obtained from the training signals 251-1 to 251-3 in FIG. 65, the reception weight for each frequency is subjected to the IFFT processing for each element of the reception weight vector, so that the reception weight vector on the time axis is obtained. Convert to weight. Then, a reception weight vector on the time axis is formed using only the value of the head component (corresponding to the line-of-sight component). At each sampling value, a reception signal vector of a sampling value constituted by a sampling value of each antenna element is multiplied by a reception weight vector on a time axis, and a sampling signal in a state where a line gain is increased by directivity control is virtually transmitted. It can be obtained for each road. In this way, it is possible to obtain a received signal in a state where the timing can be detected.

  If the speed of the channel time variation is moderate to some extent with respect to the basic frame period, when receiving the training signals 251-1 to 251-3, the relative channel information extracted from the signal a plurality of times (ie, By averaging channel information using a relative complex phase based on the complex phase of the reference antenna, noise components can be suppressed and channel estimation accuracy can be improved. If averaging is not performed, it is also possible to use absolute channel information instead of relative channel information.

  Next, with reference to FIG. 72, another signal processing operation when time-axis beamforming is applied to the present embodiment will be described. FIG. 72 is a flowchart showing another signal processing operation of time-axis beamforming. In the description of the fifth embodiment, considering that the SNR between one antenna element on the transmission side and one antenna element on the reception side is significantly short in line design, the equation (52) is used. ) Is described, but one antenna element on the transmitting side and one antenna element on the receiving side are obtained by the techniques shown in the sixth, seventh, and eighth embodiments. Channel information between the antenna elements can be obtained with high accuracy on the frequency axis, and the channel information on the time axis used for time-axis beamforming by IFFT processing using the channel information on the frequency axis It becomes possible to obtain. Here, the processing of the terminal station apparatus in the downlink is shown as an example, but the same processing is also possible in the uplink. First, the base station apparatus transmits a training signal for channel estimation in the frame head area (step S7201). The terminal station device receives the training signal and performs a received signal process (step S7211). As described above, in the present signal processing, a training signal having a length slightly longer than the OFDM symbol period that does not include the guard interval is assumed, and the sampling signal cut out by an appropriate FFT window even with a rough frame timing as a reference. To perform the FFT processing.

  Next, the terminal station apparatus performs channel estimation on the signal subjected to the FFT processing with reference to a known training signal, and performs channel information for each virtual transmission path (or for each virtual transmission path for the reference antenna). Relative channel information). Since the channel information obtained here is in a state where information exists only in a specific subcarrier for each virtual transmission path, using an arbitrary method such as linear interpolation, even for a subcarrier component that has not been obtained. Similarly, a channel estimation result is obtained (step S7212). Further, based on the acquired channel information, a reception weight vector on the frequency axis is calculated (step S7213).

  Next, the terminal station device performs an IFFT process on the subcarrier of the weight information of each antenna element (step S7214). Then, the terminal station apparatus extracts the value of the leading component (corresponding to the line-of-sight component) from the weight information on the time axis obtained as a result, calculates the reception weight on the time axis of each antenna element, A reception weight vector having these as vector components is obtained for each virtual transmission path (step S7215). Calibration processing is performed on the reception weight vector on this time axis (step S7216). Subsequently, the terminal station device calculates a transmission weight vector, and records and manages the transmission weight vector on the time axis of the first stage (step S7217).

  In the above description, the time axis weight calculation operation of the first stage is performed, but the frequency axis weight calculation of the subsequent stage is performed as follows. At the stage of calculating the reception weight vector on the time axis (step S7215), the terminal station device uses the sampling value of each antenna element as an element with respect to each sampling value of the reception training signal used for the calculation of the reception weight. The received signal vector (sampling value) is multiplied by the reception weight vector for each virtual transmission line, and separated for each signal sequence of each virtual transmission line (step S7221). Subsequently, the terminal station device individually performs FFT processing on the signal sequences of the sampling values corresponding to the number of the obtained virtual transmission paths, and performs channel estimation with reference to the known training signal (step S7222). ). Here, the channel estimation result is such that only the subcarriers to which the training signal is allocated for each virtual transmission path are channel estimation results of the desired virtual transmission path, and the subcarriers for which no training signal exists are transmitted from different virtual transmission paths. Represents a crosstalk component.

  Here, the case where the value of the leading component (corresponding to the line-of-sight component) is extracted from the weight information on the time axis and used is introduced, but the time axis of the delay component of one sample shown in Expression (57) is used. Can be calculated based on the IFFT second component, and the reception weight on the time axis of the two-sample delay component shown in equation (59) can be calculated based on the IFFT third component. Similarly, it is possible to use a delayed wave component larger than that.

Next, the terminal station apparatus determines that the channel information leaked to the j-th virtual transmission path by the signal transmitted to the i-th virtual transmission path in the k-th subcarrier is h _ji ^(k) . Since information is present only in the subcarriers, interpolation processing of the subcarriers that have been dropped by linear interpolation or the like is performed, and a complete channel matrix including crosstalk components from different virtual transmission paths is obtained. (Step S7223). Then, the terminal station apparatus calculates a reception weight matrix for signal separation based on the reception weight matrix obtained in this manner (step S7224), and uses this as a reception weight matrix on the frequency axis of the second stage. Is recorded and managed (step S7225).

  The first-stage time-axis transmission / reception weight vector and the second-stage frequency-axis reception weight matrix acquired as described above can be continuously used in a subsequent frame.

  As described above, in obtaining transmission / reception weights for using the virtual transmission path corresponding to the first singular value in the line-of-sight environment, channel estimation is performed using a small number of subcarriers, the antenna element interval is small, and the line-of-sight wave is small. If it is dominant, it is possible to estimate the transmission / reception weight of the remaining subcarriers by linear interpolation or the like, utilizing the characteristic that the frequency dependence of the channel information is reduced. Therefore, it is possible to perform channel estimation simultaneously and in parallel in a form in which subcarriers do not overlap in a plurality of wireless stations. In order to perform this efficiently, a training signal for channel estimation by the base station apparatus and the individual terminal station apparatuses in a frame is set as a fixed slot in a frame configuration having periodicity, and is transmitted in the same subcarrier. In order to avoid mutual interference, the timing and subcarriers are segregated, slots are allocated, and channel feedback is periodically performed in the slots.

  Note that a training signal for performing feedback of channel information in the related art generally adds an identification number of a transmitting station to a payload portion subsequent to the training signal. In addition, since the frequency dependence of the channel information is generally large, it has been designed so that the estimation of the channel information for each antenna can be performed for all subcarriers. These overheads were very large, causing a significant loss in MAC efficiency. However, in the eighth embodiment, the training signal is assigned to a known fixed location (slot and subcarrier) for each base station device or terminal station device. If it is not necessary to add, it is possible to simultaneously perform a large number of channel estimation and channel feedback while separating subcarriers in one slot.

[Ninth embodiment]
In the above description of the eighth embodiment, description has been made mainly on the case where the channel information or the transmission / reception weight has frequency dependency. However, as shown in the fifth embodiment, the present invention can be extended to a case where the channel information or the transmission / reception weight has no frequency dependency.

In the fifth embodiment described above, channel information is obtained from the correlation value evaluation on the time axis of the received training signal by using, for example, Expression (52). Equation (52) means that if the channel information has no frequency dependence, S _j (n) S ₁ (n) ^* at each sampling time in Σ in equation (52) excludes noise components. For example, since the values are all constant, the noise components are averaged by the N _FFT sampling values, and as a result, relative channel information (correlation value for each antenna) can be obtained with high channel estimation accuracy.

  However, in the eighth embodiment, since signals from different antenna elements are mixed on the frequency axis, if a correlation is made between received signals in which signals from different terminal station apparatuses and different antenna elements are mixed, a specific transmission antenna Rather than the correlation of the signal from each receiving antenna, the correlation value in a state where signals from different terminal station devices and different antenna elements are mixed is obtained, and the specific terminal station device and the specific The correlation value for the antenna element cannot be extracted. In a state where signals from different terminal station apparatuses and different antenna elements are mixed, it is possible to easily perform signal separation by performing FFT. In this case, the obtained channel estimation result of each subcarrier is , It is required that the channel estimation accuracy be sufficiently high (that is, sufficient SNR characteristics).

  In the above-described sixth, seventh, and eighth embodiments, the number of subcarriers to be used is limited to four as an example to increase the channel estimation accuracy. If there is no frequency dependency, it is possible to further average these channel estimation results and acquire the common channel information for all subcarriers based on the average value.

Here, c _j (f _k ) represents channel information on the frequency axis in the subcarrier f _k . Alternatively, relative channel information based on the complex phase of a reference antenna may be used.

Here, taking the case of using four subcarriers as an example, a coefficient of 1/4 is specified for the purpose of averaging the four used subcarriers _fk . Averaging is performed according to the number. The channel information obtained as a result is equivalent to that obtained by equation (52), and the reception weight is calculated using equation (50) or equation (51).

  Note that, although the same applies to the seventh embodiment in addition to the eighth embodiment, a case where a training signal including four subcarriers is used as a training signal for acquiring channel information has been described as an example. However, the number of subcarriers is 4 and there is no probability, and any other number of subcarriers may be used. As the ultimate condition, the number of used subcarriers can be set to one.

  In the fifth embodiment described above, the antenna element spacing of each first signal processing unit on the base station apparatus side is narrowed to create a situation in which the frequency dependence of channel information or transmission / reception weight is extremely small. Common carrier channel information or transmission / reception weights are used for subcarriers. Although it is possible to expect the suppression of the measurement error by using a plurality of subcarriers, there is a certain merit in using a single subcarrier as the ultimate condition.

  If the purpose of this training signal using only a single subcarrier is to obtain relative channel information for each antenna element (that is, if there is no purpose such as timing detection), a case where OFDM is used Even in this case, it is not necessary to add a guard interval, and continuous transmission of a simple sine wave (that is, an unmodulated signal) of the frequency of the allocated subcarrier is used. The characteristic of this signal is that since the amplitude of the signal is completely constant, the ratio PAPR (Peak to Average Power Ratio) of the temporal variation between the average transmission power and the instantaneous transmission power is 1. .

  In general, when OFDM is used, even when a modulation method having a constant amplitude such as QPSK is used, the amplitude of a signal fluctuates with time in order to synthesize an independent signal for each subcarrier. For this reason, the PAPR has a relatively large value, and the transmission power at the peak has a very large value compared to the average transmission power. Since the linearity of the high power amplifier is limited, if the power at the peak exceeds the linear region of the high power amplifier, non-linear distortion occurs and communication characteristics deteriorate. For this reason, in OFDM, a transmission power margin called so-called back-off is expected, and signal transmission is performed by lowering the transmission power by the margin.

  From the viewpoint of circuit design, for example, in a system using a 1024-point FFT (using about 1000 as an effective subcarrier), it is necessary to expect a backoff of at least about 10 dB. However, in the case of a sine wave having a completely constant amplitude, it is not necessary to expect such a back-off, and it is possible to increase the effective transmission power by the back-off value and transmit the signal. It should be noted that increasing the transmission power for this back-off value can be used even when a single subcarrier is not used. For example, even when using four subcarriers, the initial phase of each subcarrier is adjusted to an arbitrary value. By configuring the training signal so that the PAPR in one symbol period is minimized, operation with a lower back-off value according to the PAPR value at that time becomes possible.

  When the minimum back-off value used in other data communication is xdB when transmitting the training signal, it is desirable to set the back-off value at the time of transmitting the training signal to (x / 2) dB or less. Specifically, it is desirable to set the back-off value at the time of transmitting the training signal to 0 dB.

As a result, for example, if the transmission power for 1000 subcarriers is focused on one subcarrier, a line gain of 30 dB (= ₁₀ Log ₁₀ 1000) can be obtained, and a line of about 10 dB for the back-off can be obtained. Channel estimation in a good state with an added gain is possible. That is, an improvement of about 40 dB can be expected. Originally, in high frequency bands such as the millimeter wave band, circuit design was strict due to the high frequency. For example, if the frequency is increased by a factor of 10, the increase in the loss due to the frequency dependent term of the free space propagation loss is 20 dB. Further, the output of the high power amplifier also relatively decreases.

  Further, in the Massive MIMO, since it is necessary to mount 100 or more RF devices, it is expected that the price of each device is reduced, and the linearity of the amplifier is reduced by using inexpensive component materials. As a result, there is a problem that the channel estimation accuracy is reduced unless a line gain of 30 dB or more is additionally secured. However, a channel that focuses on one or four subcarriers and suppresses the PAPR value to increase transmission power Estimation can solve these problems.

  When channel estimation is performed using only one subcarrier per antenna element, the effect of various reflected wave components actually affects the channel estimation result of that subcarrier. May indicate different values. However, as shown in the seventh embodiment, a radio station transmitting a training signal is provided with a plurality of antenna elements, and transmits unmodulated continuous signals of different subcarriers from the plurality of antenna elements. Then, the wireless station receiving the training signal will receive training signals of a plurality of subcarriers. If relative channel information is obtained from the obtained channel information, the influence of different antenna elements on the transmitting side is relatively small, and the relative channel information obtained with these multiple antenna elements is substantially constant on the frequency axis. Should be considered. If averaging is performed using this feature, the reflected wave components that are affected differently for each subcarrier are relatively suppressed, and the relative channel information on the time axis in the form of effectively extracting the line-of-sight component Can be obtained, and the transmission / reception weight on the time axis can be calculated based on this.

  Note that the averaging process here is basically intended to average the complex phase. However, since the amplitude is expected to be substantially constant and the variation width of the complex phase is expected to be small. Alternatively, the channel information itself given by a complex number may be simply averaged on a complex plane.

[Tenth embodiment]
[Multi-user MIMO transmission using virtual transmission path corresponding to first singular value]
(Overview of Basic Principle According to Tenth Embodiment)
In the above description, for example, as shown in FIG. 16, each first signal processing unit 304 transmits and receives a single signal sequence between a single terminal station device 302 and base station device 303, The spatial multiplexing transmission is performed as a whole using the signal processing unit 304. This is equivalent to that communication between the base station apparatus 303 and the terminal station apparatus 302 is performed by single-user MIMO transmission. However, it is needless to say that the present invention can be applied to a case where spatial multiplexing transmission (that is, multi-user MIMO) is performed between a plurality of terminal station apparatuses 302 and base station apparatuses 303.

  Normally, in multi-user MIMO, when signals are simultaneously transmitted on the same frequency to a plurality of terminal station apparatuses, null formation is performed in the downlink so that mutual signals do not cause mutual interference between the terminal station apparatuses. Signal transmission was performed while performing. However, the null formation can be broken if there is a time variation of the channel, resulting in residual interference. This is because the transmission weight used for transmission directivity control is calculated based on the result of channel feedback performed at a predetermined cycle, but in order to suppress the overhead associated with the feedback of this channel, the feedback of the channel is performed so frequently. This cannot be performed, and as a result, the transmission weight must be calculated based on the channel information acquired in the past. Therefore, no matter how highly accurate the transmission weight is calculated, it is meaningless if the channel changes over time, and conversely, a transmission system that is less susceptible to the time fluctuation of the channel is required.

  When the virtual transmission line corresponding to the first singular value shown in the first embodiment is positively used, the value of the line gain obtained on the transmission line is different from that on the other transmission line using the reflected wave. Considering that when a large number of antenna elements are used on both the base station apparatus side and the terminal apparatus side, a line gain corresponding to the product of the number of antenna elements can be additionally obtained. Then, a desired SIR characteristic can be obtained without intentionally forming complete null control.

  FIG. 73 is a diagram illustrating a configuration of a base station device and a terminal station device when multiuser MIMO is applied in the tenth embodiment. 16 is different from FIG. 16 in that the terminal station apparatus is only the terminal station apparatus 302 in FIG. 16, but in FIG. 73, the terminal station apparatuses are added to the terminal station apparatus 302 and the terminal station apparatuses 308 and 309 are added. In each of the first signal processing units 304-1 to 304-4 of the base station apparatus 303, a virtual transmission path corresponding to the first singular value is formed toward each of the terminal station apparatuses 302, 308, and 309. Multi-user MIMO transmission.

  Conventionally, a plurality of first signal processing units 304-1 to 304-4 have not been implemented, so when a plurality of signal sequences are simultaneously spatially multiplexed for a single terminal station device, One transmission / reception signal processing circuit performs multi-user MIMO signal processing using different transmission / reception weight vectors. Here, it is necessary to perform complete null control on the transmitting side so that mutual interference does not occur between the terminal station devices.

On the other hand, according to the embodiment of the present invention, unlike the conventional multi-user MIMO, each of the first signal processing units 304-1 to 304-4 uses one transmission / reception weight vector for one terminal station apparatus. Only signal processing is performed. For example, one first signal processing unit 304-1 transmits signals to the terminal station devices 302, 308, and 309, one system each, for a total of three systems. At this time, the transmission weight vector for the virtual transmission path corresponding to the first singular value for each of the terminal stations 302,308,309 and _{v 11, v 12, v 13,} respectively. Since these are transmission weight vectors for the individual single-user MIMO described above, they are not orthogonal to each other. On the other hand, transmission weight vectors v ′ ₁₁ , v ′ ₁₂ , and v ′ _{13 that} are actually used for multi-user MIMO are as follows: vector v ′ ₁₁ is orthogonal to v ₁₂ and v ₁₃ , and vector v ′ ₁₂ is v ₁₁ and _v so as to orthogonal to _13, the vector v _'13 to set so that orthogonal to _{v 11} and _{v 12.} Specifically, the transmission weight vector v ′ ₁₁ for the terminal station device 302 is expressed by the following equation (78) using Gram-Schmidt orthogonalization.

This process is similarly performed on the terminal station devices 308 and 309 to obtain v ′ ₁₂ and v ′ ₁₃ . The above is the description of the first signal processing unit 304-1, but the same processing may be performed on the first signal processing units 304-2 to 304-3. In this manner, the transmission weight vectors v ′ _{11 to} v ′ ₁₃ for the first signal processing unit 304-1, the transmission weight vectors v ′ _{21 to} v ′ ₂₃ for the first signal processing unit 304-2, and the first The transmission weight vectors v ′ _{31 to} v ′ ₃₃ for the signal processing unit 304-3 and the transmission weight vectors v ′ _{41 to} v ′ ₄₃ for the first signal processing unit 304-4 may be obtained. According to the above processing, the first signal processing units 304-1 to 304-4 transmit three signal sequences, respectively, and can perform spatial multiplexing transmission of 12 systems in total. As a supplement, the transmission weight vectors v ′ ₁₁ and v ′ _2j are not orthogonal to j, for example, j = 1, unlike the case of the conventional multi-user MIMO, resulting in complete null formation. Not. However, since these are different combinations of the first signal processing unit 304 of the transmission source and the terminal station device of the reception, the mutual interference is relatively small because of the virtual transmission path having a small correlation. This basis is due to the fact that the absolute value of the first singular value shows a remarkably high gain as described above, and the high directivity gain at the pinpoint resulting from the product of a large number of antenna elements of the base station device and the terminal station device. I do. Therefore, the orthogonality of the collateral transmission weight vector v _'11 and v' _2j are become unnecessary.

Note that the same processing may be performed similarly to the processing on the base station apparatus side in the uplink. That is, although the above equation (78) shows the orthogonalization process related to the transmission weight vector, the same orthogonal relationship is applied to the reception weight, and the first singularity for each of the terminal station devices 302, 308, and 309 is obtained. The reception weight vectors directed to the virtual transmission path corresponding to the values are read as v ₁₁ , v ₁₂ , and v ₁₃ , respectively, and the reception weight vectors v ′ ₁₁ , v ′ ₁₂ , and v ′ ₁₃ used for multi-user MIMO are replaced with the vector v _'11 so as to orthogonal to the _{v 12} and _{v 13,} the vector v' as the ₁₂ orthogonal to the _{v 11} and _{v 13,} it Rure be set so that orthogonal to the vector v _{'13 v 11} and _{v 12} . Expression (78) can be applied as it is to the reception weight calculation at this time. Further, the radio station apparatus side may have exactly the same processing and circuit configuration as the radio station apparatus according to the above-described first embodiment without being conscious of multi-user MIMO.

  In this case, the circuit configuration of the base station device 303 needs to be changed as compared with FIG. FIG. 74 is a diagram illustrating first transmission corresponding to first transmission signal processing units 181-1 to 181-4 of base station apparatus 70 corresponding to base station apparatus 303 when multi-user MIMO is applied in the embodiment of the present invention. FIG. 3 is a diagram illustrating a circuit configuration of a signal processing unit 182. In FIG. 74, as shown in FIG. 73, the number of terminal station apparatuses to be simultaneously spatially multiplexed for multi-user MIMO transmission is three. The configuration of FIG. 74 is similar to the configuration of FIG. 6, except that the first transmission signal processing circuits 113-1 to 113-3 correspond to three terminal station apparatuses that perform spatial multiplexing simultaneously. The transmission weight processing unit 150 generates and manages a transmission weight vector for multi-user MIMO used for transmission on a virtual transmission path corresponding to the first singular value for three terminal stations that perform spatial multiplexing at the same time. Specifically, in the transmission weight processing unit 150, the channel information acquisition circuit 151 separately acquires the channel information acquired by the reception unit via the communication control circuit 120, and sequentially updates the acquired channel information while sequentially updating the channel information. 152. At the time of signal transmission, in accordance with an instruction from the communication control circuit 120, the multi-user MIMO (MU-MIMO) transmission weight calculation circuit 153 reads channel information corresponding to the destination station from the channel information storage circuit 152, and based on the read channel information. Then, the transmission weight vector is calculated as in the above equation (78). It should be noted here that in the case of single-user MIMO, the first right singular vector is calculated and used as it is as the transmission weight vector. In order to orthogonalize the transmission weight vector between the terminal station devices, for example, the first right singular vector component for the terminal station device 308 and the first right singular vector component for the terminal station device 308 and the first right singular vector for the terminal station device 309 are obtained. The components of the right singular vector are subtracted, and signal transmission is performed in a mutually orthogonal manner. The multi-user MIMO transmission weight calculation circuit 153 outputs the transmission weight calculated in this way to the first transmission signal processing circuits 113-1 to 113-3.

Next, FIG. 75 illustrates a first embodiment corresponding to the first reception signal processing units 185-1 to 185-4 of the base station device 70 corresponding to the base station device 303 when multi-user MIMO is applied in the tenth embodiment. 3 shows a circuit configuration of the received signal processing unit 186. Also in FIG. 75, as shown in FIG. 73, the number of terminal station apparatuses to be spatially multiplexed simultaneously for multi-user MIMO transmission is three. The configuration of FIG. 73 is similar to the configuration of FIG. 7, but the first received signal processing circuits 114-1 to 114-3 correspond to three terminal station apparatuses that perform spatial multiplexing at the same time. The reception weight processing section 170 generates and manages reception weight vectors for multi-user MIMO used for reception on a virtual transmission path corresponding to the first singular value for the three terminal stations that perform spatial multiplexing simultaneously. Specifically, in the reception weight processing section 170, according to the instruction from the communication control circuit 120, the channel information estimation circuit 172 separates the known signal for channel estimation into each subcarrier based on the information input from the FFT circuit 857 in the channel information estimation circuit 172. From the (preamble signal or the like added to the head of the wireless packet), the antenna elements of the terminal station apparatuses 60 corresponding to the terminal station apparatuses 302, 308, and 309 and the antenna elements of the base station apparatus 70 corresponding to the base station apparatus 303 Channel information with element 851 is estimated for each subcarrier, and the estimation result is output to multi-user MIMO reception weight calculation circuit 173. The multi-user MIMO reception weight calculation circuit 173 calculates a reception weight vector to be multiplied for each subcarrier based on the input channel information. Here, equation (78) describes processing using the transmission weight vector v in the case of single-user MIMO. Similarly, if this is replaced with the reception weight vector u in the case of single-user MIMO, equation (78) can be used as it is. Can be applied. At this time, the reception weight vector for combining the signals received by the antenna elements 851-1 to 851-N _MT-Ant differs for each terminal station apparatus 60, and the first reception vector corresponding to the terminal station apparatus 60 to be extracted. The signal processing circuits 114-1 to 114-3 are individually calculated. In the second reception signal processing unit 75 at the subsequent stage of the signal processing, the crosstalk component between the respective signal sequences from the same radio station device is removed. Since it is expected that the mutual interference component between the antennas can be sufficiently reduced, it is not necessary to be aware of multi-user MIMO when calculating the reception weight matrix here, and the MIMO signal detection processing must be performed individually for each wireless station apparatus. Just do it. However, since separate MIMO signal detection processing is required for different wireless station devices, the second received signal processing unit 75 needs to be provided in parallel for the wireless station devices that perform spatial multiplexing at the same time. .

  Thus, information between the first transmission signal processing unit 181 and the second transmission signal processing unit 71 and information between the first reception signal processing unit 185 and the second reception signal processing unit 75 Although there is a difference that is extended to a plurality of systems, spatial multiplex transmission with a single terminal station apparatus using a plurality of first transmission signal processing units 181 and first reception signal processing units 185 is performed. Thus, the configuration is different from that of the related art.

  As described above, the wireless communication system 90 of the tenth embodiment includes a plurality of terminal station apparatuses (second wireless station apparatuses) such as the base station apparatus 303 (first wireless station apparatus) and the terminal station apparatus 308. And The base station device 303 according to the tenth embodiment performs a plurality of first signal processing units 304 having a first antenna element group and signal processing of wireless communication associated with the plurality of first signal processing units. A second signal processing unit 305 (strictly including other base station apparatus functions such as an interface circuit, a MAC layer processing circuit, and a communication control circuit). The terminal station device 308 according to the tenth embodiment includes a second antenna element group and a transmission / reception unit that executes wireless communication with the first signal processing unit via the second antenna element group. The same applies to the terminal station device 302 and the terminal station device 309.

  The first signal processing unit 304 of the tenth embodiment converts the transmission weight vector of the MIMO channel matrix used for wireless communication between the first antenna element group and the second antenna element group into the first weight of the MIMO channel matrix. It is calculated based on at least one of a first right singular vector and a first left singular vector corresponding to one singular value, or at least one of an approximate solution of the first right singular vector and an approximate solution of the first left singular vector. . First signal processing section 304 generates a transmission signal for each terminal station device, and generates a signal vector by multiplying the generated transmission signal by a transmission weight vector. First signal processing section 304 adds and synthesizes signal vectors for terminal station apparatuses spatially multiplexed at the same time. First signal processing section 304 transmits a signal based on the signal vector obtained by addition and synthesis from the first antenna element group to a plurality of terminal station apparatuses at the same time and using the same frequency channel.

  The first signal processing unit 304 of the tenth embodiment converts the reception weight vector of the MIMO channel matrix used for wireless communication between the first antenna element group and the second antenna element group into the first weight of the MIMO channel matrix. A second right singular vector and / or a first left singular vector corresponding to one singular value, or a second based on at least one of an approximate solution of the first right singular vector and an approximate solution of the first left singular vector Is calculated for each wireless station device. First signal processing section 304 multiplies a signal vector based on a signal received via the first antenna element group by a reception weight vector for each terminal station apparatus, thereby forming one system of signal for each terminal station apparatus. Generate a series. The second signal processing unit 305 removes a residual interference component from a signal based on the generated signal sequence.

  That is, the first signal processing unit calculates the transmission weight vector for the MIMO channel matrix used for the wireless communication between the first antenna element group and the second antenna element group for each of the second wireless station devices as the second At least one of the first right singular vector and the first left singular vector corresponding to the first singular value of the MIMO channel matrix for each wireless station device, or an approximate solution of the first right singular vector and the first left singular vector Based on at least one of the approximate solutions, at least one of a first right singular vector and a first left singular vector corresponding to a first singular value of another second wireless station apparatus spatially multiplexed at the same time, or The calculation is performed so as to be orthogonal to at least one of the approximate solution of the first right singular vector and the approximate solution of the first left singular vector. The first signal processing unit generates a transmission signal for each second wireless station device, and generates a transmission signal vector by multiplying the generated transmission signal by a transmission weight vector. The first signal processing unit adds and combines transmission signal vectors for the second wireless station apparatus spatially multiplexed at the same time, and outputs a plurality of signals based on each component of the added and combined signal vector from the corresponding first antenna element. To the second wireless station device.

  Further, the first signal processing unit calculates a reception weight vector for a MIMO channel matrix used for wireless communication between the first antenna element group and the second antenna element group for each of the second wireless station devices as a second weight vector. At least one of the first right singular vector and the first left singular vector corresponding to the first singular value of the MIMO channel matrix for each wireless station device, or an approximate solution of the first right singular vector and the first left singular vector Based on at least one of the approximate solutions, at least one of a first right singular vector and a first left singular vector corresponding to a first singular value of another second wireless station apparatus spatially multiplexed at the same time, or The calculation is performed so as to be orthogonal to at least one of the approximate solution of the first right singular vector and the approximate solution of the first left singular vector. The first signal processing unit multiplies a signal vector based on a signal received via the first antenna element group by a reception weight vector for each second wireless station apparatus, thereby obtaining a second signal for each second wireless station apparatus. , A single signal sequence is generated. The second signal processing unit removes a residual interference component from the generated signal based on the plurality of signal sequences for each first signal processing unit.

  This allows the base station device 70, the terminal station device 60, the wireless communication system 90, and the wireless communication method of the tenth embodiment to increase the transmission capacity by MIMO in an environment where the line-of-sight environment is dominant. The base station device 70, the terminal station device 60, the wireless communication system 90, and the wireless communication method according to the tenth embodiment can increase the transmission capacity in an environment where the line-of-sight environment is dominant even when performing multi-user MIMO. It becomes possible.

[Eleventh embodiment]
[Mobile Fronthaul Construction Method Using Virtual Transmission Path Corresponding to First Singular Value]
(Overview of Basic Principle According to Eleventh Embodiment)
In the fifth embodiment, one-system digital baseband sampling data is multiplied by transmission weights for each antenna system and for each sampling value using time-axis beamforming, whereby a virtual system corresponding to the first singular value is obtained. Directivity was formed for transmission on a transmission line. Since the multiplication of the transmission weight here can be directly performed on the sampling data, the BBU side generates digital baseband sampling data in the same manner as in the mobile fronthaul configuration shown in FIG. All of the above signal processing is implemented, and one system of digital baseband sampling data is transmitted on the optical fiber, and the remaining processing can be performed on the RRH side. On the RRH side, transmission weight multiplication processing for time axis beamforming is required. This transmission weight information is transferred on an optical fiber, and the transferred transmission weight is converted into digital baseband sampling data on the RRH side. Since only multiplication and transmission are performed, all processing depending on the wireless communication system is basically implemented on the BBU side. The only element in the RRH that depends on the radio communication method is that the transmission directivity control has no frequency dependence. In practice, a very narrow directional beam is formed by a multi-element antenna to perform communication. Means to do. This directional beam is ideally a beam destined for the communication partner station in the line-of-sight environment, but if a very strong reflected wave arrives even without line-of-sight, This can be dealt with by forming a directional beam directed at this point. In a situation in which a very large number of reflected waves are collected, although very weak, resulting in a moderate reception level, the line gain of the virtual transmission line corresponding to the first singular value deteriorates compared to the line-of-sight environment. However, originally, there is a tendency to use antenna elements with high directivity gain because of the high frequency band, which has a large distance attenuation and is severe in line design.As a practical problem, it is rare to use a very large number of multiple reflected waves. Is expected. For this reason, in particular, as a mobile fronthaul in a wireless system using a high frequency band, by performing time-axis beamforming using a virtual transmission path corresponding to the first singular value, light is transmitted while utilizing a large-scale antenna. It is possible to reduce the information transmission capacity on the fiber.

  In addition, the information capacity at the time of transmitting the transmission weight information on the optical fiber is estimated. For example, consider the case of using OFDM, and assume that the number of FFT points at that time is, for example, 1024. In this case, assuming a guard interval of 256 samples, for example, the number of samples of one OFDM is 1280. On the other hand, when using, for example, 256 antenna elements, 256 pieces of transmission weight information are required (because the transmission weight vector has 256 dimensions). Assuming that the number of bits of one piece of transmission weight information is the same as the number of bits of one sample, the transmission weight information exactly corresponds to information for 20%. In other words, even if the transmission weight information is transferred every time every OFDM symbol, the transmission weight information can be increased by 20% compared to the configuration of the mobile fronthaul shown in FIG. This is overwhelmingly less than the information volume of 256 times in the configuration of the mobile fronthaul shown in FIG. 13A, and is substantially the same as the configuration shown in FIG. 12A. Furthermore, if the transmission weight information is transferred every few OFDM symbols instead of transferring the transmission weight information every every OFDM symbol, it is possible to further reduce the information amount of the transmission weight information. . Alternatively, the transmission weight information may be transferred only when the transmission weight information is changed. In this way, the signal processing functions depending on the radio transmission system are integrated on the BBU side, and the time axis beam for configuring the virtual transmission path corresponding to the minimum necessary first singular value is arranged on the RRH side. A configuration in which a forming function is mounted can be realized.

  FIG. 76 is a diagram showing an outline of the function allocation of the mobile fronthaul using the virtual transmission path corresponding to the first singular value in the eleventh embodiment. Here, only the function related to signal transmission in the direction from the network side to the user (from the BBU to the RRH direction) is extracted. In the figure, reference numeral 401 denotes a MAC layer processing circuit, reference numeral 402 denotes a transmission signal processing circuit, reference numeral 403 denotes a time axis signal generation circuit, reference numeral 454 denotes an optical interface circuit, reference numeral 455 denotes an optical fiber, reference numeral 456 denotes an optical interface circuit, and reference numeral 457. Is a time axis transmission weight multiplying circuit, 427 is a D / A converter, 428 is an RF processing circuit, 429 is an antenna element, 432 is a transmission weight processing circuit, 441-4 is BBU, and 442-4 is Indicates RRH. The MAC layer processing circuit 401, the transmission signal processing circuit 412, the time axis signal generation circuit 403, and the time axis transmission weight multiplication circuit 457 constitute an area 446-1 and an area 446-2 for performing baseband signal processing related to wireless as a whole.

In comparison with FIGS. 17 and 47 and FIG. 76, the MAC layer processing circuit 78 of FIG. 17 corresponds to the MAC layer processing circuit 401 of FIG. The second transmission signal processing unit 71 in FIG. 17 corresponds to the transmission signal processing circuit 412 in FIG. The IFFT & GI adding circuit 813 in FIG. 47 corresponds to the time axis signal generating circuit 403. The first transmission signal processing circuit 311 in FIG. 47 corresponds to the time axis transmission weight multiplication circuit 457 in FIG. D / A converters 814-1 to 814 -N ′ _BS-Ant in FIG. 47 correspond to D / A converter 427 in FIG. Mixer _{816-1~816-N 'BS-Ant,} filter 817-1~817-N' in FIG. 47 _BS-Ant, high-power amplifier _{818-1~818-N 'BS-Ant} corresponding to the RF processing circuit 428 I do. The antenna elements 819-1 to 819 -N ′ _BS-Ant correspond to the antenna element 429. The first transmission weight processing section 330 in FIG. 47 corresponds to the transmission weight processing circuit 432 in FIG.

  In FIG. 76, when a signal to be transmitted to the BBU 441-1 is input from the network side, the MAC layer processing circuit 401 performs signal processing of the MAC layer, and transmits a frame format used for transmission and reception in a wireless section and flows on the network side. The data frame format is converted and terminated, and a signal in the format of a wireless packet is input to the transmission signal processing circuit 412. The transmission signal processing circuit 412 performs transmission signal processing of a wireless signal. Here, the transmission method in the wireless section is not particularly limited. For example, if OFDM is used, error correction coding, interleaving, modulation processing for each subcarrier, and the like are performed as necessary. The signal generated in this manner is converted to a time-axis signal by the time-axis signal generation circuit 403. For example, taking the case of OFDM as an example, IFFT is performed to convert a signal on the frequency axis into a signal on the time axis, insert a guard interval, and perform waveform shaping processing between symbols. As a result, each sampling value of the digital baseband signal is converted into a signal that is continuous in time series. On the other hand, for example, the reception weight information collected by the reception weight processing circuit on the receiving side of the BBU not shown here is transmitted (or directly) through the communication control circuit or the like not shown here (or directly). Circuit 432. The transmission weight processing circuit 432 performs a calibration process on the reception weight information and the like to calculate a transmission weight. Each sampling value of the digital baseband signal generated by the time axis signal generation circuit 403 and the transmission weight information calculated by the transmission weight processing circuit 432 are converted into a predetermined frame format by the optical interface circuit 454, and the optical signal is converted from the electrical signal to the optical signal. The signal is converted to a signal and output to the optical fiber 455.

  The signal output to the optical fiber 455 is transmitted to the RRH 442-4 side, where the optical interface circuit 456 converts the optical signal into an electric signal, terminates a signal of a predetermined format, and outputs a digital baseband signal. To reproduce the information sequence of the sampling value and the transmission weight information. Both the sampling value and the transmission weight information are input to the time axis transmission weight multiplication circuit 457 as this digital baseband signal. The time axis transmission weight multiplying circuit 457 multiplies the sampling value as a digital baseband signal by the transmission weight of each antenna system in sampling value units. Each sampled value multiplied by the transmission weight is input to a D / A converter 427, which converts a signal for each antenna system into an analog baseband signal at a predetermined clock rate, and an RF processing circuit 458. In each antenna system, the signal is converted into a radio frequency signal by an up-converter, the out-of-band radiation signal is removed by a filter for each antenna system, and then amplified by a high-power amplifier for each antenna system. Send.

  As described above, the function corresponding to the base station apparatus for wireless communication as a whole is accommodated separately in the BBU 441-4 and the RRH 442-4, which are mediated by optical fibers. The feature here is that the radio digital baseband signal processing is centralized in the BBU 441-4 provided in the station on the network side, so that even if there is any change in the radio communication system, all of the BBU 441-4 There is a merit that the change on the side is enough.

  The above description corresponds to the case where only one first transmission signal processing unit 181 in FIG. 17 is provided. It is possible. FIG. 77 is a diagram illustrating an outline (downlink) of the function sharing of the mobile fronthaul using virtual transmission paths corresponding to a plurality of first singular values according to the eleventh embodiment. Here, only the function related to signal transmission (direction from BBU 441-5 to RRH 442-4a, 442-4b) regarding the direction (downlink) from the network side to the user is extracted. In the figure, reference numeral 461 denotes a MAC layer processing circuit, reference numeral 462 denotes a transmission signal processing circuit, reference numerals 403a to 403b denote time axis signal generation circuits, reference numerals 454a to 454b denote optical interface circuits, reference numerals 455a to 455b denote optical fibers, and reference numeral 456a. 456a to 456b are optical interface circuits, 457a to 457b are time axis transmission weight multiplication circuits, 427a to 427b are D / A converters, 428a to 428b are RF processing circuits, 429a to 429b are antenna elements, and 470 is The transmission weight processing circuit, reference numeral 441-5 indicates BBU, and reference numerals 442-4a to 442-4b indicate RRH.

  77, the same components as those in FIG. 76 are denoted by the same reference numerals. Since RRHs 442-4a to 442-4b in FIG. 77 have the same system as the RRH 442-4 in FIG. 76, a suffix a or b is added for identification. Similarly, the time axis signal generation circuits 403a to 403b, the optical interface circuits 454a to 454b, and the optical fibers 455a to 455b are also used for the same number of RRHs 442-4a to 442-4b as those in FIG. The signal processing is the same as that of FIG. 76, but the MAC layer processing circuit 461 and the transmission signal processing circuit 462 process signal sequences for the number of RRHs 442-4a to 442-4b. For example, if spatial multiplexing is performed on one terminal station using MIMO, the MAC layer processing circuit 461 processes the MAC layer as a signal of one terminal station, while the transmission signal processing circuit 462 performs spatial processing on the signal. In consideration of multiplexing, serial-to-parallel conversion is performed on signals of a plurality of systems, and transmission signal processing is performed on each. The transmission weight processing circuit 470 receives the reception weight information collected by the reception weight processing circuit on the receiving side of the BBU not shown here, for example, via a communication control circuit also not shown here (or directly 2) Get information. Processing such as calibration is performed based on the reception weight information for the plurality of systems, transmission weights are calculated, and the transmission weight information is transferred to the RRHs 442-4a to 442-4b corresponding to the transmission weights. Note that the transmission weight processing circuit 470 may acquire the transmission weight similarly to the first transmission weight processing units 330 and 340 in the fifth embodiment.

  Next, FIG. 78 is a diagram illustrating an outline (uplink) of the function allocation of the mobile fronthaul using virtual transmission paths corresponding to a plurality of first singular values according to the eleventh embodiment. Here, only the function related to signal transmission (direction from RRH 444-1a, 444-1b to BBU 443-1) related to the direction (uplink) from the user side to the network is extracted. In the figure, reference numeral 471 denotes a MAC layer processing circuit, reference numeral 472 denotes a reception signal processing circuit, reference numerals 474a to 474b denote optical interface circuits, reference numerals 475a to 475b denote optical fibers, reference numerals 476a to 476b denote optical interface circuits, and reference numerals 473a to 473b. Is a time axis reception weight multiplication circuit, reference numerals 477a to 477b are A / D converters, reference numerals 478a to 478b are RF processing circuits, reference numerals 479a to 479b are antenna elements, reference numeral 480 is a reception weight processing circuit, and reference numerals 481a to 481b are correlations. The calculation circuit 443-1 represents BBU, 444-1a to 444-1b represents RRH. FIG. 78 shows a plurality of systems of RRHs 444-1a to 444-1b as corresponding to FIG. 77, but only one system may be used as in FIG.

  In FIG. 78, when the antenna elements 479a to 479b receive a signal, the RF processing circuits 478a to 478b amplify the signal with a low-noise amplifier for each antenna system, and multiply the signal by a common local oscillator signal as a whole with a mixer. Is down-converted, and an out-of-band signal is removed by a filter to generate an analog baseband signal for each antenna system. This signal is sampled by A / D converters 477a to 477b to generate a data sequence of sample values of a digital baseband signal for each antenna system. On the other hand, the time axis reception weight multiplying circuits 473a to 473b multiply the reception weights and add and synthesize (multiply the reception signal vector by the reception weight vector) to convert it into one system of digital baseband signal. On the other hand, the A / D converters 477a to 477b also output the data sequence of the sample values of the digital baseband signal to the correlation calculation circuits 481a to 481b.

The correlation calculation circuit 481A～481b, the sampled values of the received signals of the second to the _{N Ant} antenna elements in the antenna elements 479A～479b, a complex conjugate value of the sampling values of the received signal of the first antenna element Multiply and add the operation results for a predetermined period. Similarly, the sampling values of the received signals of the first to Nth _Ant elements of the antenna elements 479a to 479b are multiplied by the complex conjugate of the sampled values, and the calculation results for a predetermined period are added. Based on the above results, a correlation operation is performed by equation (52), and the complex conjugate is calculated to calculate the reception weight of each antenna element. The correlation calculation circuits 481a to 481b receive timing indication signals from the reception weight processing circuit 480 of the BBU 443-1 via the optical interface circuits 474a to 474b, the optical fibers 475a to 475b, and the optical interface circuits 476a to 476b. Is done.

  Each time the timing signal is received, the correlation calculation circuits 481a and 481b perform the correlation operation using the equation (52) and reset the memory value of each addition. The correlation calculation circuits 481a to 481b collect the correlation information obtained here for all antenna sequences and transfer them to the reception weight processing circuit 480 via the optical interface circuits 476a to 476b, the optical fibers 475a to 475b, and the optical interface circuits 474a to 474b. I do. The reception weight processing circuit 480 calculates reception weight information based on the correlation information received here, records and manages the calculated reception weight information, and transmits the correlation information or the reception weight information to the transmission weight processing circuit 470 as necessary. A notification is made and a calibration process or the like is performed here and used for calculating the transmission weight.

  Further, the reception weight processing circuit 480 grasps the scheduling information of the reception processing of the BBU 443-1 and the RRHs 442-4a to 442-4b (for example, the communication control circuit or the MAC layer processing circuit 471 not shown here). It grasps by referring to the scheduling information to be managed, etc.), determines at what timing the reception weight should be switched, and issues a timing instruction indicating the switching timing to the optical interface circuits 474a to 474b, the optical fibers 475a to 475b, and the optical interface. The signals are output to the correlation calculation circuits 481a to 481b via the circuits 476a to 476b. This timing instruction takes into account the time lag caused by passing through the optical interface circuits 474a to 474b, the optical fibers 475a to 475b, and the optical interface circuits 476a to 476b. Upon receiving the switching timing instruction, the correlation calculation circuits 481a to 481b output the reception weight information calculated based on the correlation information calculated immediately before to the time axis reception weight multiplication circuits 473a to 473b. The time axis reception weight multiplying circuits 473a to 473b continue to multiply the reception signal by the reception weight information specified last until the switching instruction is given.

FIG. 79 is a diagram showing an outline of the correlation detection circuit 481 (481a, 481b) in the eleventh embodiment. In the drawing, reference numerals 491-2 to 491-N _Ant and reference numerals 495-1 to 495-N _Ant are multipliers, reference numerals 492-2 to 492-N _Ant and reference numerals 496-1 to 496-N _Ant are adders, Reference numerals 493-2 to 493 -N _Ant and reference numerals 497-1 to 497 -N _Ant denote memories, reference numerals 498-1 to 498 -N _Ant denote square root acquisition circuits, and reference numeral 494 denotes a correlation operation control unit. The correlation calculation circuit 481 is connected to the A / D converter 477, the time axis reception weight multiplication circuit 473, and the optical interface circuit 476. Further, it is also connected to a reception weight processing circuit 480 on the BBU 443-1 side via an optical interface circuit 476, an optical fiber 475, and an optical interface circuit 474.

When the sampling value of each antenna system is input from the A / D converter 477, the correlation calculation circuit 481 multiplies the sampling values of the reception signals of the first to Nth _Ant elements in the antenna element 479 by multiplication. vessel in _{491-2~491-N Ant,} multiplied by the sampled value of the second to _{N Ant} antenna elements in the first complex conjugate value of the sampling values of the antenna and the antenna element 479 of the antenna element 479, a multiplication result Is output to the adders 492-2 to 492-N _Ant . Adder _{492-2~492-N Ant} adds the value of the input value and the memory _{493-2~493-N Ant,} stored by entering the addition result to the memory _{493-2~493-N Ant} Let it. By rotating the calculated values between the adders 492-2 to 492-N _Ant and the memories 492-2 to 493-N _Ant sequentially, the correlation results are sequentially added, and the accumulated value is obtained. When an instruction to reset the correlation value is input from the reception weight processing circuit 480 of the BBU 443-1 via the optical interface circuit 474, the optical fiber 475, and the optical interface circuit 476 to the memories 493-2 to 493-N _Ant , Each of the 493-2 to 493 -N _Ant outputs the accumulated value to the correlation operation control unit 494, and resets its own value to zero.

In parallel with the above processing, when the sampling values of the respective antenna systems are input from the A / D converter 477, the sampling values of the reception signals of the first to Nth _Ant elements in the antenna element 479 are multiplied by the multiplier. 495-1 to 495-N _Ant , and the sampling values of the first to N _Ant antennas of the antenna element 479 and their complex conjugate values are multiplied by multipliers 495-1 to 495-N _{Ant to} perform sampling. The square value of the absolute value of the value is output to adders 496-1 to 496-N _Ant . Adder _{496-1~496-N Ant} adds the value of the input value and the memory _{497-1~497-N Ant,} stored by entering the addition result in the memory _{497-1~497-N Ant} Let it. The calculated values are sequentially turned between the adders 496-1 to 496-N _Ant and the memories 497-1 to 497-N _Ant , whereby the square values of the absolute values of the sampled values are sequentially added, and the accumulated value is calculated. It will be required.

When an instruction to reset the correlation value is input from the reception weight processing circuit 480 of the BBU 443-1 via the optical interface circuit 474, the optical fiber 475, and the optical interface circuit 476 to the memories 497-1 to 497-N _Ant , _{497-1~497-N Ant} outputs the accumulated value to the square root acquisition circuit _{498-1~498-N Ant,} reset their values to zero. The square root obtaining circuits 498-1 to 498-N _Ant obtain the square root of the accumulated value and output the value to the correlation operation control unit 494. The correlation operation control unit 494 calculates the reception weight by performing the correlation operation by the equation (52) and further taking the complex conjugate. The reception weight information aggregates information on all antenna systems and transfers the information to the reception weight processing circuit 480 via the optical interface circuit 476, the optical fiber 475, and the optical interface circuit 474. Similarly, this reception weight is also input to the time axis reception weight multiplication circuit 473.

  FIG. 80 is a diagram illustrating an outline (uplink) of another mobile fronthaul function sharing using virtual transmission paths corresponding to a plurality of first singular values according to the eleventh embodiment. Here, only functions related to signal transmission (direction from RRH 444-2a to 444-2b to BBU 443-2) from the user side to the network (uplink) are extracted. In the figure, reference numerals 484a to 484b denote optical interface circuits, reference numerals 485a to 485b denote optical fibers, reference numerals 486a to 486b denote optical interface circuits, reference numeral 482 denotes a reception weight processing circuit, reference numerals 481a to 481b denote correlation calculation circuits, and reference numeral 443-43. 2 represents BBU, and reference numerals 444-2a to 444-2b represent RRH.

  Also, in this figure, the same components as those in FIG. 78 are denoted by the same reference numerals. The difference between FIG. 80 and FIG. 78 is that the reception weight input to the time axis reception weight multiplication circuit 473 is converted from the reception weight processing circuit 482 on the BBU 443-2 side to the optical interface circuit 484, the optical fiber 485, and the optical interface circuit 486. The point is to be notified via. Therefore, it is possible to instruct to use the previously obtained reception weight managed by the reception weight processing circuit 482 without necessarily using the reception weight calculated with reference to the training signal received immediately before. I have.

  As another variation, in FIG. 78, the correlation calculation circuits 481a and 481b are provided with a reception weight storage / management function, and instead of notifying the reception weight information by the optical fiber 475 as shown in FIG. The BBU 443-1 notifies the RRH 444-1a, 444-1b of the identification number of the reception weight managed by the circuit 481, and the reception weight information stored in the correlation calculation circuits 481a, 481b based on the identification number. May be read out and output to the time axis reception weight multiplying circuit 473.

  Next, an embodiment in which the mobile fronthaul configuration in the present embodiment is applied to multi-user MIMO will be described. FIG. 81 is a diagram illustrating an outline (downlink) of function sharing when multiuser MIMO is applied to a mobile fronthaul using virtual transmission paths corresponding to a plurality of first singular values according to the eleventh embodiment. . Here, only the function related to signal transmission (direction from BBU 441-6 to RRH 442-6) related to the direction (downlink) from the network side to the user is extracted. In the figure, reference numeral 463 denotes a transmission signal processing circuit, reference numeral 464 denotes a transmission weight processing circuit, reference numeral 465 denotes an addition / synthesis circuit, reference numeral 441-6 denotes BBU, and reference numeral 442-6 denotes RRH.

  In the figure, the same components as those in FIG. 77 are denoted by the same reference numerals. In the description so far, basically, a plurality of RRHs may perform spatial multiplexing communication with one terminal station apparatus as a single-user MIMO, or perform communication with different terminal station apparatuses on the assumption that mutual interference is limited. I have explained that I can go. On the other hand, in FIG. 81, the one RRH 442-6 is extended to a multi-user MIMO that performs signal transmission to different terminal station devices, and FIG. 74 used for describing the tenth embodiment related to the multi-user MIMO is referred to. While explaining. However, although FIG. 74 is not necessarily premised on time-axis beamforming, the description will be made based on the example of FIG. 74 in the case of performing time-axis beamforming.

In order to support multi-user MIMO, in the configuration of first transmission signal processing section 181 of base station apparatus 70 corresponding to base station apparatus 303 when multi-user MIMO is applied as shown in FIG. The processing corresponding to the transmission signal processing circuits 113-1 to 113-3 is performed by the transmission signal processing circuit 463 in FIG. Also, processing corresponding to the addition / synthesis circuits 812-1 to 812-N'BS- _Ant in FIG. 74 is performed by the addition / synthesis circuit 465 in FIG. Further, the processing corresponding to the transmission weight processing unit 150 in FIG. 74 is configured to be performed by the transmission weight processing circuit 464 in FIG. Further, the outputs of the time axis transmission weight multiplication circuits 457a to 457b of the plurality of RRHs 442-4a to 442-b in FIG. 77 are input to the addition and synthesis circuit 465. The addition / combination circuit 465 adds and combines signals transmitted by the same antenna element and outputs the addition result to the D / A converter 427a. The addition and combining circuit 465 combines the two transmission signals into one signal.

  In the description of FIG. 77, the two RRHs 442-5a to 442-b may transmit signals to one terminal station device or may transmit signals to different terminal station devices. Because of MIMO, different signal sequences are transmitted to different terminal station devices. Except for the difference described above, the same signal processing is performed with the basically same configuration as in FIG. Here, the optical interface circuits 454a to 454b, the optical fibers 455a to 455b, and the optical interface circuits 456a to 456b connect between the BBU 441-6 and the RRH 442-6 via physically different optical fibers and optical interface circuits. Although an example of the case has been described, if an optical fiber capable of realizing a large-capacity transmission is used, it is also possible to physically exchange information by integrating the same optical interface circuit and the same optical fiber. In this sense, what is shown in FIG. 81 may be understood as a “logical transmission path”.

  FIG. 82 is a diagram illustrating an overview (uplink) of function sharing when multiuser MIMO is applied to a mobile fronthaul using virtual transmission paths corresponding to a plurality of first singular values according to the eleventh embodiment. . Here, only the function related to signal transmission (direction from RRH 444-3 to BBU 443-3) in the direction (uplink) from the user side to the network side is extracted. In the figure, reference numeral 466 denotes a reception signal processing circuit, reference numeral 467 denotes a reception weight processing circuit, reference numeral 443-3 denotes BBU, and reference numeral 444-3 denotes RRH. 80, the same components as those in FIG. 80 are denoted by the same reference numerals. Also in this description, extension to multi-user MIMO in which signals from different terminal stations are received by one RRH will be described with reference to FIG. 75 which describes multi-user MIMO. Note that FIG. 75 is not necessarily premised on time-axis beam forming. However, here, description will be made based on the example of FIG. 75 in the case of performing time-axis beam forming.

  In order to support multi-user MIMO, in the configuration of first reception signal processing section 186 of base station apparatus 70 corresponding to base station apparatus 303 when applying multi-user MIMO shown in FIG. The processing corresponding to the received signal processing circuits 114-1 to 114-3 is performed by the time axis reception weight multiplication circuit 483 of FIG. In the fifth embodiment, the first received signal processing unit 385 of the base station device performs signal processing for extracting a virtual transmission path corresponding to the first singular value of the MIMO channel matrix. Also, when multi-user MIMO is applied as in the tenth embodiment, the first received signal processing unit 186 performs signal processing for extracting a virtual transmission path corresponding to the first singular value of the MIMO channel matrix. In the first embodiment, an interference component that cannot be completely removed by this signal processing is removed by a second received signal processing unit 75 at the subsequent stage. On the other hand, in the eleventh embodiment, the second-stage signal processing for removing the interference component that the reception signal processing circuit 466 could not completely remove is performed. By performing the two-stage signal processing in this manner, it is possible to efficiently perform signal separation.

  Further, the processing corresponding to the reception weight processing section 170 of FIG. 75 is configured to be performed by the reception weight processing circuit 467 of FIG. Further, in FIG. 75, a signal is output only to one time-axis reception weight multiplication circuit 473 in one RRH 444-1, whereas in FIG. 82, a plurality of outputs from one A / D converter 477a are output. To the time axis reception weight multiplying circuits 483a to 483b.

  The plurality of time axis reception weight multiplication circuits 483a to 483b multiply the same reception signal vector by the time axis reception weight vectors corresponding to different terminal station devices. The multiplication result is input to the reception signal processing circuit 466 via the optical interface circuits 486a to 486b, the optical fibers 485a to 485b, and the optical interface circuits 484a to 484b. Except for the above difference, the same signal processing is performed with the basically same configuration as in FIG. Here, the optical interface circuits 486a to 486b, the optical fibers 485a to 485b, and the optical interface circuits 484a to 484b connect the RRH 444-3 and the BBU 443-3 via physically different optical fibers and optical interface circuits. Although an example of the case has been described, if an optical fiber capable of realizing a large-capacity transmission is used, it is also possible to physically exchange information by integrating the same optical interface circuit and the same optical fiber. In this sense, what is shown in FIG. 82 may be understood as "logical transmission path".

  In the wireless communication system according to the eleventh embodiment, the BBU includes an optical interface circuit 454 as a first transmission unit and a third transmission unit. The optical interface circuit 454 transmits the sampling data sequence generated by the time axis signal generation circuit 403 and transmission weight information (transmission weight vector) to the RRH. The BBU includes reception weight processing circuits 480, 482, and 467 as reception weight acquisition means. The reception weight processing circuit acquires reception weight information (reception weight vector) used by the RRH for reception from the terminal station device. The reception weight information is calculated based on the correlation value calculated in the RRH. Further, the reception weight processing circuit instructs the RRH to calculate a correlation value and reception weight information.

  In the wireless communication system according to the eleventh embodiment, the RRH includes a time-axis reception weight multiplication circuit 473 as a second transmission unit. The time-base reception weight multiplying circuit 473 sequentially multiplies the reception sampling signal vector sequence, which is the data sequence of the sample values of the digital baseband signal output from the A / D converter 477, by the reception weight for each sampling data, The multiplication results are added and combined to generate one system of sampling data sequence, and one system of sampling data sequence is transmitted to the BBU via the optical interface circuit 456. The RRH includes an optical interface circuit 456 as a fourth transmission unit. The optical interface circuit 456 combines the correlation information calculated by the correlation calculation circuit 481 for all antenna sequences, and transmits the collected information to the BBU via the optical interface circuit 476.

  In the wireless communication system according to the eleventh embodiment, the RRH includes optical interface circuits 456 and 476 as first receiving means and second receiving means. The optical interface circuit 456 receives transmission weight information (transmission weight vector) and a sampling data sequence from the BBU. The optical interface circuit 476 receives reception weight information (reception weight vector) from the BBU. The RRH includes a correlation calculation circuit 481 as reception weight calculation means. The correlation calculating circuit 481 sets one of the plurality of antenna elements 479 as a reference antenna element based on a signal when a known signal transmitted from the terminal station apparatus is received by each of the antenna elements 479, And a known signal received by another antenna element 479 over a predetermined number of samples, and a reception weight vector is calculated based on the calculated correlation value.

  According to the above correspondence, according to the wireless communication system in the eleventh embodiment, even in the downlink from the BBU to the RRH direction, in the uplink from the RRH to the BBU direction, and further including the extension to the multi-user MIMO By providing a multi-element antenna on the RRH, the basic aim of the mobile fronthaul can be achieved without increasing the information capacity flowing on the optical fiber, and with the impact that the amount of information slightly increases compared to the past. The function allocation can be optimized while maintaining it, and the band can be expanded in a situation where the RRH and the terminal station are in a line-of-sight environment and direct waves are dominant.

  In the wireless communication system according to the eleventh embodiment, the configuration has been described in which an optical fiber is used as a communication medium between the BBU and the RRH. A line may be used.

[Supplementary information regarding the relationship between various embodiments]
First, the embodiments of the present invention basically use the virtual transmission paths corresponding to the plurality of first singular values shown in the first embodiment to spatially multiplex a plurality of signal sequences. Features. The relationship with each specific embodiment is summarized below.

(Combination with the second embodiment of the present invention)
The second embodiment of the present invention relates to a specific embodiment relating to optimization of the arrangement interval of the first signal processing unit on the base station apparatus side in order to efficiently realize the first embodiment of the present invention. It is shown. In the case of the fourth embodiment, since the distance between the base station apparatus and the train changes with time, the optimization conditions shown in the second embodiment cannot be used directly, but in other embodiments, In this case, it is possible to utilize the second embodiment. Specifically, in the third embodiment, since the satellite base station device and the control base station device are fixedly installed, the satellite base station device is regarded as a first signal processing unit, and the control base station device is controlled. Is regarded as a terminal station device, and if spatial multiplexing transmission is performed using a virtual transmission path corresponding to a plurality of first singular values between the satellite base station device and the controlling base station device, the satellite base station device is The third embodiment and the second embodiment can be arranged in such a manner that they are arranged in a straight line and the arrangement conditions satisfy predetermined conditions with respect to the average distance between the control base station apparatus and the satellite base station apparatus. Operation combining the mode and the first embodiment is possible. Further, in contrast to an example in which the satellite base station device according to the third embodiment as shown in FIG. 32 is used, the present technology is applied to an access system using the purely first embodiment instead of the satellite base station device. In the application form shown in FIG. 83 in the case of applying the present invention to the first signal processing shown in the second embodiment, the average distance between the first signal processing unit and the terrestrial terminal station apparatus is increased. By optimizing the arrangement intervals of the units, it is possible to stabilize the communication between each terminal station device and the base station device because it is possible to maintain a good orthogonal relationship, although it deviates from the perfect orthogonal condition. become. For this reason, even when the present technology is applied to an access system, operation combining the second embodiment and the first embodiment is possible. In the sense of application to this access system, the eighth embodiment and the tenth embodiment are also applicable, and operation in which the second embodiment and the first embodiment are further combined with these embodiments is also possible. It is.

  In the description of the second embodiment, for example, as illustrated in FIG. 29, the antenna element on the base station apparatus side is replaced with a parabolic antenna, and it is assumed that the terminal station apparatus side has a plurality of antenna elements. However, the base station apparatus side does not necessarily need to include the multi-element antenna group as shown in the first signal processing unit of the first embodiment, and it is substantially an ultra-high gain single antenna element. The terminal station device side can configure a virtual transmission path corresponding to the first singular value by combining with a plurality of antenna elements.

(Combination with Third Embodiment of the Present Invention)
As described above, in the case of the fourth embodiment, since it is specialized in direct communication between the terminal device on the train side and the base station device, the third embodiment of the present invention that performs 2-hop relay transmission Although it is difficult to imagine a combination with the embodiment, similarly to the case where the combination of the third embodiment and the second embodiment of the present invention is possible, other operation conditions of the fourth embodiment are different from those of the other embodiments. It is also applicable for combinations. For example, the third embodiment can be extended to multi-user MIMO transmission as shown in the tenth embodiment in addition to performing the first embodiment by single-user MIMO transmission. And the first embodiment), it is also possible to realize multi-user MIMO while utilizing the satellite base station apparatus of the third embodiment.

(Combination with Fourth Embodiment of the Present Invention)
As described above, the fourth embodiment of the present invention is a kind of unique use form. However, here, the signal processing for forming directivity on the time axis shown in the fifth embodiment may be performed in combination. In the process of channel estimation, it can be used in combination with the sixth and seventh embodiments of the present invention. Here, one-to-one communication between one terminal station device and a base station device on the train side is described as an example. In this sense, single-user MIMO transmission is fundamental. In a case where a plurality of trains are simultaneously present in the same section, besides changing the frequency channel for each of those trains, if the multi-user MIMO shown in the tenth embodiment is used in combination, Communication with a plurality of trains can be realized simultaneously and in parallel on the same channel, thereby making it possible to effectively use frequency resources.

(Combination with the fifth embodiment of the present invention)
The technology intended in the fifth embodiment of the present invention can be applied to a system under any conditions, such as a single carrier transmission system or a multicarrier system such as OFDM as a wireless communication system. In the case of a system based on signal processing on the frequency axis such as OFDM, the direct merit of applying the fifth embodiment of the present invention is small (however, it is not possible to reduce the number of FFT circuits and IFFT circuits mounted). However, for example, when a large-scale antenna is mounted on the RRH side in the mobile fronthaul as in the eleventh embodiment, transmission is performed on an optical fiber even in a multicarrier system such as OFDM. In order to expand the information capacity to a large-scale antenna while compressing the information capacity, a signal on the time axis is transmitted on an optical fiber, and the RRH side uses time-axis beamforming described in the fifth embodiment. In this sense, the eleventh embodiment is based on the premise that the fifth embodiment (and the first embodiment) is operated in combination.

(Combination with the sixth and seventh embodiments of the present invention)
The sixth and seventh embodiments of the present invention are embodiments showing means for feeding back channel information used for forming directivity, and can be used in combination in all embodiments of the present invention. This can be used in a typical fixed installation type terminal station device used in the description of the first embodiment, and can also be applied in an environment where a channel varies with time in an access system. In the eighth embodiment of the present invention, a specific configuration method for simultaneously performing channel estimation between a plurality of adjacent cells or a plurality of terminal station devices in the same cell is assumed, particularly for use in an access system. This is an embodiment assuming a combination of the sixth embodiment and the seventh embodiment of the present invention. Further, the line gain in the channel estimation is normally insufficient for channel estimation between the train-side terminal station device and the base station device shown in the fourth embodiment, which can be used for improving the line gain, It is also possible to operate the fourth embodiment in combination with the sixth and seventh embodiments (and the first embodiment) of the present invention.

  In particular, as a specific embodiment that directly combines the sixth embodiment and the seventh embodiment, training is performed on discrete subcarriers between antenna elements of a terminal station device or a base station device while avoiding overlapping of subcarriers. When assigning to signals, it is assumed that each antenna element of the terminal station apparatus or the base station apparatus is transmitted from the same virtual antenna element, and relative channel information obtained for a plurality of subcarriers from a plurality of different antenna elements. In response to this, a relative weight including subcarriers to which no training signal is assigned is acquired using a least squares method or the like, and a reception weight (and a weight that cancels the complex phase difference of the relative channel information based on the relative channel information) The transmission weight can also be obtained by the calibration process).

(Combination with the eleventh embodiment of the present invention)
The eleventh embodiment of the present invention is similar to the first embodiment of the present invention in which the first signal processing unit and the second signal processing unit are separated, and the space between them is operated using an optical fiber. However, in the first embodiment, various signal processes for forming directivity of transmission and reception are integrated in the first signal processing unit, whereas in the eleventh embodiment, the signal processing corresponds to the second signal processing unit. It is integrated on the BBU side. However, when spatial multiplexing transmission is performed by positively utilizing virtual transmission paths corresponding to a plurality of first singular values, the basic principle is shared. And operation corresponding to the combination of the fifth embodiment and the first embodiment (although the arrangement of the first signal processing unit is strictly different, in the sense of utilizing the virtual transmission path corresponding to the first singular value). It is considered to correspond to the first embodiment). Further, as shown in the eleventh embodiment, the extension to the multi-user MIMO transmission shown in the tenth embodiment is also possible. In this case, the eleventh embodiment, the tenth embodiment, and the The operation corresponds to the combination of the fifth embodiment and the first embodiment. As described above, also regarding the train moving cell shown in the fourth embodiment, the plurality of first signal processing units are regarded as RRHs using the eleventh embodiment, and the second signal processing unit on the BBU side is connected via an optical fiber. With a configuration in which information is exchanged between the second signal processing unit and the second signal processing unit, an operation corresponding to a combination of the eleventh embodiment, the fourth embodiment, and the first embodiment is possible. Naturally, if this is combined with a multi-user MIMO including terminal stations of a plurality of trains, the tenth embodiment will be combined at the same time.

  As described above, various embodiments of the present invention can be operated in combination with each other.

[Other supplementary items of the embodiment according to the present invention]
Hereinafter, some supplementary items related to the embodiment according to the present invention will be described.

In the description of the embodiments of the present invention, the receiving side of the antenna element 851-1~851- (N _'BS-Ant) and the transmission side of the antenna element 819-1~819- (N' _BS-Ant) is Although different numbers have been assigned to the individual elements, the antenna elements (851-1 to 851- (N'BS _-Ant )) or the antenna elements shared by the TDD-SW and the like for the transmission system and the reception system are naturally understood. 819-1 to 819- (N'BS _-Ant )). In particular, when performing implicit feedback, it is essential to share antennas for the transmission system and the reception system, and TDD control that switches the uplink and the downlink in time is fundamental. The switching of the TDD-SW here is managed and controlled by the communication control circuit 120 and the like.

In addition, receiving-side antenna elements 851-1 to 851- (N ′ _BS-Ant ) and transmitting-side antenna elements 819-1 to 819- (N ′ _BS-Ant ) related to the base station apparatus and reception related to the terminal station apparatus. Common numbers are used for the antenna elements 851-1 to 851- (N _MT-Ant ) on the transmission side and the antenna elements 819-1 to 819- (N _MT-Ant ) on the transmission side, respectively, to simplify the description. It is explained. However, requirements such as the size and directional gain of the antenna element of the base station apparatus and the requirements such as the size and directional gain of the antenna element of the terminal station apparatus do not generally match, and the functions are the same. Are given the same numbers, but in actual operation, antenna elements having different conditions in terms of size, directivity gain, etc. may be used. The same applies to the other circuits, although the description using a common number in order to simplify the description, respectively, for example, high-power amplifier 818-1~818- (N _'BS-Ant and _{N MT- Ant} ), low-noise amplifiers 852-1 to 852- (N'BS _-Ant and NMT _-Ant ), etc., also have similar requirements such as amplification factor, power consumption, size and allowable heat generation. Are not the same, and the functions are the same, so the same numbers are given as numbers. However, in actual operation, circuits with different conditions may be used.

  In addition, the transmission weight vector in the embodiment of the present invention means each column vector (or a part of column vectors) of the transmission weight matrix, and is a transmission weight expressed as a vector focusing on one of the spatially multiplexed signal sequences. is there. Specifically, each column vector of the transmission weight matrix when spatially multiplexing a plurality of signal sequences is a vector representation of a weight (a component of the transmission weight vector) focusing on a certain signal sequence among the plurality of signal sequences. The transmission weight matrix is sequentially acquired based on the channel vector or the channel matrix of the terminal station apparatus to be spatially multiplexed. Therefore, generation of a transmission weight vector (and the same also when words such as “calculation”, “determination”, “multiplication”, and “component” follow) are equivalent to generation of a transmission weight matrix as a whole, and in particular, When considering the procedure of sequentially generating the row vector or the column vector of the matrix, it may be described as "generation of transmission weight vector" in a meaning equivalent to "generation of transmission weight matrix".

  Similarly, the reception weight vector in the embodiment of the present invention means each row vector (or a part of the row vector) of the reception weight matrix, and is represented by a vector noting one of the spatially multiplexed signal sequences at the same time. This is the reception weight. Specifically, each row vector of the reception weight matrix when a plurality of signal sequences are spatially multiplexed is a vector representation of a weight (a component of the reception weight vector) focusing on a certain signal sequence among the plurality of signal sequences. In the process of generating the entire reception weight matrix based on the spatially multiplexed channel vectors or channel matrices of the terminal station device, they are sequentially acquired. Therefore, the generation of the reception weight vector (and the same when words such as “calculation”, “determination”, “multiplication”, and “component” follow) are equivalent to the generation of the reception weight matrix as a whole, and in particular, When considering the procedure of sequentially generating the row vector or the column vector of the matrix, it may be described as "generation of reception weight vector" in a meaning equivalent to "generation of reception weight matrix".

  Furthermore, in the embodiments of the present invention, “vector” is used in various forms such as “transmission weight vector”, “reception weight vector”, “channel (information) vector”, “transmission signal vector”, and “reception signal vector”. These are all vectors corresponding to the respective antenna elements, and each of the vectors has "transmission weight", "reception weight", "channel (information)", "transmission signal", and "reception signal" as components. Even if there is no explicit description of “vector” in each embodiment, it is clear that those components are those “vectors” in the embodiment, and if necessary, Should be understood in addition to these contents. Further, the "transmission signal" and the "transmission signal" and "reception signal" in the "reception signal vector" are transmission or reception signals corresponding to each antenna element or based on each antenna element, and are actually transmitted and received. Instead of analog radio frequency signals, digitalized baseband (or intermediate frequency) signals may be used. This signal includes both a signal on the frequency axis and a sampling signal on the time axis. Therefore, in the description of the above-described embodiments, some parts have not been explicitly described, but these signals are not limited to the signals themselves transmitted and received at the radio frequency, but also include the signals and the signals transmitted and received at the radio frequency. Between them, some signal processing may be performed.

  Further, in the description related to the channel estimation in the embodiment of the present invention, a case where the channel information is averaged by a plurality of times, in addition to a case where the channel information is acquired by a single channel estimation, is also described. For example, considering the line gain by utilizing a large-scale antenna element, there is no problem even if the line gain is insufficient in the line design between one antenna and one antenna, while the directivity by the large-scale antenna is not problematic. An appropriate transmission / reception weight is required in order to obtain a transmission gain, and channel information necessary for calculating the transmission / reception weight is basically based on a channel estimation result between one antenna and one antenna. If the line gain is insufficient, the subsequent directivity gain cannot be obtained. As a countermeasure for this, in Non-Patent Document 2 and the like, a training signal is received with a plurality of symbols, and the received signal is averaged to compensate for the shortage of the line gain. As the averaging of the estimation result of the plurality of times, when a periodic training signal is continuous over several symbols, a method of performing short-time averaging over several symbols using the periodicity, and a discrete There is a method of performing long-term averaging in which channel estimation results performed at a time are collected and averaged for a plurality of times. Here, in the case of short-time averaging, since a plurality of symbols to be averaged are continuous, it is considered that all symbol timings are preserved because of their periodicity. It was not necessary to treat it as a relative channel with reference to the complex phase. On the other hand, regarding channel estimation results performed at discrete times, it is not generally necessary that the symbol timings be the same. It was necessary to adopt a configuration for acquiring and averaging. In this case, the complex phases of the channel information of the reference antenna element are all assumed to be zero (that is, take a real value). In calculating the relative channel information, the channel information of all antenna elements is multiplied by Exp (−jθ) for the complex phase θ of the reference antenna, and the channel information of all antenna elements is used as a reference. It may be obtained by dividing by the channel information of the antenna element. Such utilization of relative channel information is generally indispensable for averaging channel estimation results having different symbol timings. However, when symbol timings are common, the relative channel information is generally used for calculating transmission / reception weights. There is no need to use information. It was sufficient to simply calculate the transmission / reception weight for the acquired channel information by using Expressions (1) and (7). However, even if the symbol timing is common, even if the relative channel information is utilized, no problem occurs at all. Therefore, the above description is sometimes described as relative channel information, or simple channel information is used. In some cases, the description has been made by using it as it is. However, the difference is as described above, and it is not essential to use the relative channel information.

  The relative channel information can be treated as channel information obtained by correcting the complex phase with reference to the complex phase of the reference antenna, or the channel information of each antenna element can be used as the reference channel information including the amplitude. It may be obtained by dividing information. For example, the amplitude of each relative channel information has a meaning in the case of the reception weight of the maximum ratio combining expressed by the equation (7), but has the meaning in the case of the transmission and reception weight of the equal gain combining expressed by the equation (1). do not have. Further, actually, in a line-of-sight environment, it is expected that the deviation of the amplitude for each antenna is extremely limited, and in that case, there is no large difference even if all are approximated to the same amplitude. In the first place, the absolute value of the transmission / reception weight has no significant meaning and needs to be separately optimized so as to be an efficient value within a finite number of quantization bits. The discussion is completely different from the embodiment of the present invention, and optimization may be performed in existing technology. In this sense, the magnitude (absolute value) of the transmission / reception weight vector as a vector has no special meaning here, and the transmission / reception weight vector multiplied by an arbitrary coefficient is also assumed here as its statistical property is preserved. Explains.

  By using the above relative channel information, in the embodiment of the present invention, various kinds of signal processing can be simplified. On the other hand, in the conventional MIMO transmission technology, for example, the multiplication process of the reception weight matrix is often described as including the process of directly converting to a state suitable for signal detection. In other words, even for an SISO signal, for signal detection processing, the received signal is divided by the channel estimation result, and the signal points on the constellation in which the I and Q axes are correctly set from the received signal subjected to channel distortion. Had to be converted to In the embodiment of the present invention, the main purpose of the multiplication processing of transmission / reception weights performed by the first signal processing unit is mainly to secure directivity gain and perform signal processing for rough signal separation. By simply allowing a large number of antenna elements to be treated as one virtual antenna element by weight multiplication, the received signal is divided by the channel estimation result, and the I and Q axes are divided from the received signal subjected to channel distortion. Does not include the process of converting into a signal point on a constellation in which is set correctly. However, on the receiving side, it is also possible to perform processing such as signal detection in the subsequent stage, and it is possible to apply the same signal processing as that performed in the conventional MIMO transmission or SISO transmission to these processings. is there. In particular, in the receiving system of the base station device, these processes are performed in the second signal processing circuit.

  Also, when transmitting a training signal composed of limited subcarrier components from different antenna elements on the transmitting station side while avoiding duplication of subcarriers assigned by each antenna element, there is no overlap on the frequency axis The nuclear training signals are combined in space and are received at the receiving station as a training signal containing more subcarrier components. In this sense, the receiving station differs from the training signal transmitted by each antenna element on the transmitting side, but considers a training signal synthesized in space (referred to as a “synthesized training signal”) as if it were a normal training signal. Signal processing can be performed. When using such a combined training signal, it is effective to convert the channel information into relative channel information, which is a relative value with respect to a reference antenna element, and use it.

  In the above description, k (for example, the k-th frequency component, etc.) representing a subcarrier is omitted for simplicity, and further description of individual subcarriers is omitted. Is a wideband system, and all signal processing on channel information, transmission / reception weights, transmission signals and reception signals, etc., is basically the same as the signal processing on the time axis in the fifth embodiment. Specifically, it should be individually defined and processed for each subcarrier on the frequency axis. Therefore, in order to simplify the description, the subscripts that explicitly indicate the subcarriers have been omitted in many descriptions. However, these descriptions are actually performed individually for each subcarrier, and in such a case, the description can be interpreted strictly if understood by adding a subscript indicating the subcarrier. Inside each signal processing circuit, for example, signal processing up to a stage prior to the IFFT processing on the transmission side (for example, assuming an OFDM modulation method, bit string interleaving processing, signal point mapping, signal modulation processing, transmission weight vector The multiplication and the like are all performed for each subcarrier. Similarly, the signal processing after the FFT processing on the receiving side (similarly, if the OFDM modulation method is assumed, the multiplication of the reception weight, the signal detection processing, and the signal Mapping, deinterleaving, etc.) are all performed for each subcarrier.

  In terms of circuit configuration, an individual circuit may be provided for each subcarrier, or the same processing is performed, so that processing is performed serially for each subcarrier and the circuit is shared with the subcarriers It is also possible to convert. Further, a plurality of circuits may be prepared in the middle, subcarriers may be appropriately divided, and a plurality of circuits may perform processing in parallel in a serial manner. These are common to all embodiments.

  In addition, since all subcarriers are used for communication with the same terminal station device in the OFDM modulation scheme, transmission / reception weights (average transmission / reception weight vectors and real-time transmission / reception weight matrices) at that time are common to all subcarriers. The transmission / reception weights for the combined terminal station devices will be used. However, in OFDMA, since assignments to terminal station apparatuses of different combinations are arranged in a patchwork manner on the time axis and the frequency axis, they are assigned for each time (OFDM symbol) and each frequency (subcarrier). It is necessary to use the transmission / reception weight for the terminal station device. However, except for the difference, OFDM and OFDMA can be processed in exactly the same way. In this specification, the description has been made with a focus on OFDM. can do.

  Also, there are various operational variations regarding SC-FDE. However, multiplying the transmission weight vector on the transmission side, and processing the received signal after signals transmitted from each antenna element are combined in space, and In each of the above-described configuration examples, the processing performed by the conventional SC-FDE can be directly applied to any of the reception signal processing after the reception weight is multiplied by the reception weight and the signals of the respective antenna elements are added and combined. Since it has a configuration, it can be applied to all variations of SC-FDE. In this case, after performing signal processing on a single carrier instead of signal processing of the OFDM modulation method, if a downlink is used, FFT processing is performed on a signal on the time axis of the single carrier so that each subcarrier is processed. May be generated, and these signal components may be regarded as signals of each subcarrier generated by the OFDM modulation method and multiplied by the transmission weight vector generated according to the embodiment of the present invention. Similarly, in the case of the uplink, the signal obtained by performing the FFT processing on the received signal is treated in the same manner as in the case of the OFDM modulation method, and the signal is separated by multiplying the reception weight vector generated according to the present invention. By performing IFFT processing on the subcarrier signal, the signal may be converted into a single carrier signal on the time axis. Thus, although some signal processing is different between the OFDM modulation method and the SC-FDE, the generation of transmission / reception weights and the multiplication processing are common, and the embodiment of the present invention is applied to both of these signal methods. Applicable.

  Further, in spatial multiplexing transmission, signals of a plurality of sequences are transmitted in parallel. In the above-described embodiment, processing such as error correction performed on these signal sequences is performed independently for each signal sequence. As described above, on the transmission side, the signal after error correction coding is serially / parallel-converted and separated into spatially multiplexed signal sequences on the transmission side, and the signal on the reception side before error correction processing is performed on the reception side. Parallel / serial conversion, and then error correction processing such as one-system Viterbi decoding may be performed. Furthermore, any error correction processing may be performed regardless of the essence of the embodiment of the present invention, including other variations. In this case, the exchange of signals between the MAC layer processing circuit and the second signal processing unit and the like is not performed by a spatial multiplexing system, for example, information is exchanged as one system signal and serial / multiplexing is performed. A function corresponding to parallel conversion, parallel / serial conversion, and error correction is included in the second signal processing unit and the like.

  In the above description, the right singular vector of the singular value decomposition is used.However, the left singular vector obtained by performing the singular value decomposition of the matrix obtained by transposing the matrix to be subjected to the singular value decomposition, It is equivalent to the right singular vector. Similarly, a vector equivalent to the right singular vector or the left singular vector can be obtained by performing eigenvalue decomposition by multiplying the original channel vector by the Hermitian conjugate vector. In this sense, the use of a vector mathematically equivalent to the “right singular vector” is also within the range of the “right singular vector” intended by the embodiment of the present invention. Similarly, when a mathematically equivalent vector is used as the left singular vector, the range of the “left singular vector” intended by the embodiment of the present invention is used.

  Further, in the above description, only the virtual transmission path corresponding to the first singular value is used between each first signal processing unit on the base station side and the terminal station device, and the singular value after the second singular value is used. Has been described as not using the virtual transmission path corresponding to the above, but when using a shared antenna for a plurality of polarizations such as V polarization and H polarization, for example, one first signal processing unit Since the virtual transmission path corresponding to the first singular value is utilized in the polarization antenna, the virtual transmission path corresponding to the two singular values is utilized in a single first signal processing unit in terms of form. Is equivalent to having In particular, when there is a slight leakage between the polarizations, the first signal processing unit collectively accommodates and processes the two types of polarization antennas, thereby suppressing the crosstalk component between the polarizations. That is, it is possible to use the virtual transmission path corresponding to the first singular value and the second singular value to improve transmission efficiency. In this case, the mathematical expression corresponds to utilizing the first singular value and the second singular value, but in effect, a virtual transmission path corresponding to the first singular value is provided for each polarization antenna group. It is equivalent to using. The intention of the present invention is to use a plurality of antenna groups that are expected to have a very low correlation with each other, such as when utilizing such a polarized antenna. Even when utilizing the same first signal processing unit, including the case where the virtual transmission path corresponding to the first singular value effectively corresponding to each antenna group is utilized in parallel, The point is that spatial multiplexing transmission utilizing a virtual transmission path corresponding to a first singular value in an effective sense is transmitted using a plurality of first signal processing units. In addition, the plurality of first signal processing units included in the base station apparatus basically perform up-conversion and down-conversion between baseband and radio frequencies using individual local oscillator signals. However, it is preferable that the signals of the respective local oscillators behave quasi-synchronously as much as possible. For this purpose, a reference signal such as a common reference clock may be supplied from the third signal processing unit. Alternatively, a reference signal of an intermediate frequency is supplied from the second signal processing unit to the first signal processing unit, and a radio signal for performing up-conversion and down-conversion between the baseband and the radio frequency by, for example, multiplying the reference signal. It is also possible to generate a signal of a frequency. Depending on the value of the intermediate frequency used here and the stability of the circuit for performing the multiplication processing, the uncertainty of the complex phase of the radio signal between the plurality of first signal processing units is limited even in the transmission signal processing. In this case, it is also possible to perform a precoding process between the plurality of first signal processing units in the second signal processing unit. However, basically, at least the reference signal supplied from the second signal processing unit to the plurality of first signal processing units is transmitted on a cable between the plurality of first signal processing units in the second signal processing unit. In order to reduce the loss of the signal, it is fundamental that the frequency of the local signal used in the up-conversion and down-conversion processing between the baseband and the radio frequency should be equal to or less than 1/2 of the frequency of the local signal. The signal processing units become independent. However, when the base station apparatus and the terminal station apparatus are arranged at a short distance, L in Expression (29) becomes small, and as a result, the interval between the first signal processing units can be reduced. Depending on the value of the cable loss, it is also possible to directly supply the signal of the local oscillator from the second signal processing section to the plurality of first signal processing sections.

  In implementing the third embodiment, the control base station not only manages communication between the satellite base station device and the control base station, but also manages communication between the satellite base station device and the terminal station device. Will be done. As described above, when selecting a satellite base station apparatus capable of performing communication with low correlation for a terminal station apparatus of interest from among a plurality of satellite base station apparatuses and performing communication, the controlling base station apparatus The satellite base station to be used is selected as the scheduling process, and the result of the selection is instructed to the satellite base station to perform communication. Further, particularly when communication is performed over a plurality of small cells as shown in FIG. 33, especially when there is a variation in the distance from the satellite base station device to be used and when there is a gain difference in the line design, In some cases, management such as transmission power control is required. These management functions are implemented on the control base station apparatus side, and various wireless communication controls are stably realized.

  In the description of the tenth embodiment, the transmission weight processing circuits 432, 464, and 470 in the BBU are configured to notify the RRH side of the transmission weight via the optical fibers 455, 455a, and 455b, From the reception weight processing circuits 467, 480, 482 to the RRH side via the optical fibers 475a, 475b, 485a, 485b, and the correlation calculation circuits 481a, 481b, 483a, 483b on the RRH side from the optical fibers 475a, 475b. , 485a, 485b, the reception weight information or the channel information of the reception side is notified to the BBU side. Here, the information exchanged between the BBU and the RRH is not necessarily the transmission / reception weight information itself digitally described. It doesn't have to be. For example, if a beam pattern used by the RRH for transmission / reception is prepared in a menu, and coefficients for forming a large number of transmission / reception weights corresponding to each beam pattern are listed in the RRH, the beam pattern is used. May be managed as a codebook, and the codebook value may be exchanged between the BBU and the RRH. In this case, the correlation calculation circuits 481a, 481b, 483a, and 483b on the RRH side compare the calculated correlation value for each antenna element with the coefficient on the codebook, and determine the codebook value with the smallest error from the coefficient. And a function of notifying the selected codebook value.

  Further, particularly when the present invention is used in an access system, when the antenna installation position of the base station apparatus is installed at a low place, the antenna element of the terminal station apparatus and the antenna element of the base station apparatus are placed in a line-of-sight environment. In some cases, it is required that communication can be performed stably even under conditions not described above. In this case, the superiority of the first singular value over the second singular value is reduced, and as a result, sufficient gain is obtained due to the division loss caused by dividing the entire antenna element into a plurality of groups and performing signal processing. In some cases, it cannot be secured. In such a case, for example, it is better to collect all antenna elements in one place and perform MIMO transmission as usual, as compared to a case where the antenna elements are divided into a plurality of first signal processing units. Is expected. When designing a system including such a case, as an example, a number (for example, one) of a plurality of first signal processing units is mounted with a larger number of antenna elements than other first signal processing units. Depending on the situation, the MIMO transmission utilizing the second and subsequent singular values in the special first signal processing unit is performed depending on the situation. It is also possible to operate while switching the mode between the case where MIMO transmission is performed using a dynamic transmission path. For example, when the number of antenna elements of this special first signal processing unit is 256 and the number of antenna elements of the other first signal processing units is 64, it is determined that the prospect of the terminal device can be generally secured. Performs four multiplexing using four virtual transmission paths corresponding to the first singular value, and when the line of sight cannot be secured, normal MIMO communication using a 256-element antenna element of a special first signal processing unit May be performed. At this time, there is no necessity that the special first signal processing unit is installed near other plural first signal processing units. For example, the special first signal processing unit is located at a low place, and The plurality of first signal processing units may be provided at a high place, and may have a physically distinct difference. In this case, the special first signal processing unit installed in a low place uses a virtual transmission path corresponding to the first singular value if there is a low possibility that the view to the terminal station device is secured. May be only the first signal processing unit installed at a high place. By appropriately switching the modes, it is possible to always perform communication under optimum conditions corresponding to the environment. The device corresponding to the special first signal processing unit does not necessarily need to be accommodated in the same second signal processing unit. For example, apart from the base station apparatus that performs MIMO transmission using the virtual transmission path corresponding to the first singular value shown in the embodiment of the present invention, a conventional base station apparatus that also uses the second singular value and subsequent ones is close to the base station apparatus. It is also possible to adopt a configuration in which both terminals cooperate and the terminal station device selects an optimal base station device and performs communication in accordance with the situation.

  Further, in the second embodiment, the provision has been made regarding the case where the first signal processing unit or the directional antenna element on the base station apparatus side is arranged on a straight line. The arrangement of the processing units does not necessarily have to be arranged on a straight line, and may be an arrangement having a two-dimensional spread. Particularly, when the present invention is used in an access system, the correlation between the first signal processing units is reduced by arranging the first signal processing units on the wall surface of the building two-dimensionally. Is also possible.

  In each of the embodiments described above, the configuration has been described on the premise of the implicit feedback of Massive MIMO. However, strictly ignoring the MAC efficiency, there is no problem in a method using explicit feedback using control information.

  In the above description, the multiplication processing of the transmission / reception signal and the transmission / reception weight has been described as being performed on the baseband. However, this shows typical signal processing, and equivalent signal processing is performed on the baseband signal. Of course, it is also possible to implement the present invention on an intermediate IF (Intermediate Frequency) signal between the RF signal and the radio frequency RF signal. In terms of signal processing, it is easier to perform processing in a lower frequency band. However, in the fifth generation mobile communication, communication is performed in a high frequency band such as a millimeter wave band with a bandwidth of several hundred to 1 GHz. Even if the processing is performed in the IF band of several GHz, the difficulty of the processing does not change much. In the specification of the present invention, a baseband signal or a baseband is used in a broad sense of a frequency band in which digital signal processing can be performed. In this sense, an IF different from the baseband in a narrow sense is used. The present invention is essentially applicable to signal processing in a band, and the scope of the present invention covers such a broad baseband signal and baseband.

  Further, in the present invention, since the number of antenna elements provided in the wireless station device is enormous, for example, even if a small number of the antenna elements are to be subjected to exceptional signal processing, for example, It is possible to obtain generally expected characteristics by the effect. However, even if some of such antenna elements are exceptionally processed, the result of signal processing using the remaining antenna elements largely determines the majority of the characteristics. Even if the exception processing is applied, if the effect is rather limited without obtaining the expanded effect by the exception application, the present invention should be covered by the claims.

  The base station device and the terminal station device in the above-described embodiment may be realized by a computer. In that case, a program for realizing this function may be recorded on a computer-readable recording medium, and the program recorded on this recording medium may be read and executed by a computer system. Here, the “computer system” includes an OS and hardware such as peripheral devices. The “computer-readable recording medium” refers to a portable medium such as a flexible disk, a magneto-optical disk, a ROM, and a CD-ROM, and a storage device such as a hard disk built in a computer system. Further, a “computer-readable recording medium” refers to a communication line for transmitting a program via a network such as the Internet or a communication line such as a telephone line. Such a program may include a program that holds a program for a certain period of time, such as a volatile memory in a computer system serving as a server or a client in that case. The program may be for realizing a part of the functions described above, or may be a program that can realize the functions described above in combination with a program already recorded in a computer system, It may be realized using hardware such as a PLD (Programmable Logic Device) or an FPGA (Field Programmable Gate Array).

  As described above, the embodiments of the present invention have been described with reference to the drawings. However, it is apparent that the above embodiments are merely examples of the present invention, and the present invention is not limited to the above embodiments. is there. Therefore, additions, omissions, replacements, and other changes may be made to the components without departing from the technical spirit and scope of the present invention.

1. Base station device,
2. Radio station device,
3 ... line of sight,
4… Stable reflected waves from the structure,
5: multiple reflected waves near the ground
6: multiple reflected waves near the ground,
7 ... Structure,
11 ... transmitting station,
12 ... receiving station,
21 High power amplifier (HPA),
22 Low noise amplifier (LNA),
23: time division switch (TDD-SW),
24 ... antenna element 25 ... wireless module,
26 ... antenna terminal,
27 ... Coaxial cable,
31 ... Controlling base station device,
32 satellite base station device,
33 ... terminal station device,
34 ... Controlling base station device,
35 ... Satellite base station device,
36 ... terminal station device,
37 ... structure,
38 ... structure,
39 ... cell,
40 ... wireless communication system,
41 ... base station device,
50 ... wireless communication system,
51 ... terminal station device,
52 ... wireless communication system,
53 ... Wireless communication system,
60 ... terminal station device,
61 ... transmitter,
65 ... receiving unit,
67 ... Interface circuit,
68 MAC layer processing circuit,
70 ... base station device,
71 second transmission signal processing unit,
75 ... second reception signal processing unit,
77 ... Interface circuit,
78 MAC layer processing circuit,
80 ... base station device,
81: transmitting unit,
85 receiving part,
87 ... Interface circuit,
88 ... MAC layer processing circuit
90 ... wireless communication system,
111: first transmission signal processing circuit,
113 ... Transmission signal processing circuit,
114 ... received signal processing circuit,
120 ... communication control circuit,
121 ... communication control circuit,
130 ... first transmission weight processing unit
131: first channel information acquisition circuit
132 a first channel information storage circuit;
133: first transmission weight calculation circuit
140 ... first transmission weight processing unit,
141 ... channel information acquisition circuit
142 ... channel information storage circuit
143 first transmission weight calculation circuit,
144: first reception weight processing unit;
145 ... Reception signal processing circuit,
146 ... first channel information estimation circuit,
147 first reception weight calculation circuit,
148 ... second transmission signal processing circuit,
149 ... transmission weight calculation circuit,
150: transmission weight processing unit
151 ... channel information acquisition circuit
152 channel information storage circuit,
153: Multi-user MIMO transmission weight calculation circuit
154... A first reception weight processing unit;
155 ... first reception signal processing circuit,
156 first channel information estimation circuit,
157: first reception weight calculation circuit
158 ... first reception signal processing circuit,
159: second reception signal processing circuit,
160 ... first reception weight processing section
161, a first channel information estimation circuit;
162: first reception weight calculation circuit,
170: reception weight processing unit,
172... Channel information estimation circuit
173 ... Multi-user MIMO reception weight calculation circuit
181 ... first transmission signal processing unit,
182 ... first transmission signal processing unit,
185: first reception signal processing unit,
186 ... first received signal processing unit,
190 ... second received signal processing circuit,
191, a channel matrix acquisition circuit,
192: reception weight matrix calculation circuit
193: reception weight matrix multiplication circuit
194 ... signal detection circuit,
201: virtual antenna element,
202: virtual antenna element,
203: virtual antenna element,
204: base station device,
205 ... antenna element group,
206 ... terminal station device,
221-1 to 5 ... antenna elements,
231, 232, 233, 234 ... base station devices,
235, 236, 237, 238, 239 ... terminal station device,
241, 242, 243, 244... Subcarriers,
251-1-3, 253-1-3, 254-1-3, 255-1-3, 256-1-3 ... slots,
252-1 to 3 ... slots for data communication,
257-1-3 ... training signal,
258-1 ... data payload,
291-295 ... plot points,
301 ... base station apparatus,
302 ... terminal station device,
303: base station apparatus,
304: first signal processing unit,
305 ... second signal processing unit,
306 ... base station device,
307 ... parabolic antenna,
308: terminal station device,
309: terminal station device,
310 ... base station apparatus antenna,
311 ... first transmission signal processing circuit,
312 ... terminal station device antenna element group
320: terminal station device antenna element group
330 ... first transmission weight processing unit
331: first transmission signal processing circuit,
332: first channel information acquisition circuit
333: first channel information storage circuit,
334 ... first transmission weight calculation circuit
340... First transmission weight processing unit;
341, a first channel information acquisition circuit;
342 a first channel information storage circuit;
343: a first transmission weight calculation circuit;
348 a second transmission signal processing circuit;
354 first reception weight processing unit,
355 ... first reception signal processing circuit,
356... A first channel information estimation circuit;
357: first reception weight calculation circuit,
358 ... first reception signal processing circuit,
359 ... second reception signal processing circuit,
360: first reception weight processing unit;
361 ... transmitting unit,
362... A first channel information estimation circuit;
363: first reception weight calculation circuit,
365 receiving unit,
381: a first transmission signal processing unit;
385: first reception signal processing unit,
401 ... MAC layer processing circuit,
403, 403a, 403b ... time axis signal generation circuit,
412 ... transmission signal processing circuit,
427, 427a, 427b ... D / A converter,
428, 428a, 428b ... RF processing circuit,
429, 429a, 429b ... antenna elements,
432 ... transmission weight processing circuit,
441-4, 441-5, 441-6 ... BBU,
442-4, 442-4a, 442-4b, 442-6 ... RRH,
443-1, 443-2, 443-3 ... BBU,
444-1a, 444-1b, 444-2a, 444-2b, 444-3 ... RRH,
446-1, 446-2 ... area,
454, 454a, 454b ... optical interface circuit,
455, 455a, 455b ... optical fiber,
456, 456a, 456b ... optical interface circuits 457, 457a, 457b ... time axis transmission weight multiplication circuits
461: MAC layer processing circuit,
462: transmission signal processing circuit,
463 ... transmission signal processing circuit,
464: transmission weight processing circuit,
465 ... addition synthesis circuit,
466 ... Reception signal processing circuit
467: reception weight processing circuit,
470 ... transmission weight processing circuit,
471 MAC layer processing circuit,
472: reception signal processing circuit,
473a, 473b ... time axis reception weight multiplication circuit,
474a, 474b ... optical interface circuit,
475a, 475b ... optical fiber,
476a, 476b ... optical interface circuit,
477a, 477b ... A / D converter,
478a, 478b ... RF processing circuit,
479a, 479b ... antenna elements,
480... Reception weight processing circuit,
481, 481a, 481b... Correlation calculation circuit,
482: reception weight processing circuit
483a, 483b: Time axis reception weight multiplication circuit,
484a, 484b ... optical interface circuit,
485a, 485b ... optical fiber,
486a, 486b ... optical interface circuit,
491-2, 491-3, 491-N _Ant ... Multiplier,
492-2, 492-3, 492-N _Ant ... adder,
493-2, 493-3, 493-N _Ant ... memory,
494: correlation operation control unit,
495-1, 495-2, 495-3, 495-N _Ant ... multiplier,
496-1, 496-2, 496-3, 496-N _Ant ... adder,
497-1, 497-2, 497-3, 497-N _Ant ... memory,
498-1,498-2,498-3,498-N _Ant ... Square root acquisition circuit,
600 ... base station device,
601, 601-1, 601-2, 601-3, 601-4, 601-5, 601-6, 601-7, 601-8 ... first signal processing unit,
602-1, 602-2, 602-3, 602-4, 602-5, 602-6, 602-7, 602-8 ... geometric coordinate information,
603 ... train,
604: terminal station device,
605 ... camera,
606: weight matrix / coordinate database,
607: weight vector / coordinate database,
612 ... Coordinate information acquisition circuit
613: time information acquisition circuit
614 ... communication control circuit,
622 ... Coordinate information acquisition circuit
623: time information acquisition circuit
624: communication control circuit,
700 ... antenna element,
701: TDD-SW,
702: Low noise amplifier (LNA),
703 ... mixer,
704 ... filter,
705 A / D converter,
706 ... FFT circuit,
707: reception weight multiplication circuit,
708: transmission weight multiplication circuit,
709: IFFT & GI addition circuit,
710 ... D / A converter,
711 ... mixer,
712 ... filter,
713: High power amplifier,
714... TDD-SW,
721: antenna element,
722: Low noise amplifier (LNA),
723 ... mixer,
724 ... filter,
725 ... A / D converter,
726 ... FFT circuit,
727: reception weight multiplication circuit,
728: transmission weight multiplication circuit,
729: IFFT & GI addition circuit,
730 ... D / A converter,
731 ... mixer,
732 ... filter,
733 ... High power amplifier,
740: transmission weight processing unit
741 local oscillator 742 local oscillator 743 communication control circuit
781... Scheduling processing circuit
811 ... transmission signal processing circuit,
812 ... addition synthesis circuit,
813: IFFT & GI provision circuit,
814 ... D / A converter,
815: Local oscillator,
816: mixer,
817 ... filter,
818 ... High power amplifier,
819: antenna element,
820: communication control circuit,
821: a first transmission signal processing circuit,
830: transmission weight processing unit
831 ... channel information acquisition circuit,
832: channel information storage circuit,
833: Multi-user MIMO transmission weight calculation circuit
851 ... antenna element,
852: Low noise amplifier (LNA),
853: Local oscillator,
854 ... mixer,
855 ... filter,
856: A / D converter,
857: FFT circuit,
858: reception signal processing circuit,
860: reception weight processing unit
861... Channel information estimation circuit
862 ... Multi-user MIMO reception weight calculation circuit,
881 ... Scheduling processing circuit

Claims (3)

A wireless communication system including a wireless device including a plurality of antenna elements, a baseband device that controls the wireless device, and a terminal station device that wirelessly communicates with the wireless device,
The wireless device,
Radio reception that receives a signal transmitted from the terminal station device through each of the antenna elements and generates a received sampling signal vector sequence that is a sampling data sequence of a baseband signal corresponding to each of the antenna elements from the received signal. Means,
Based on a signal when a known signal transmitted from the terminal station device is received by each of the antenna elements, any one of the plurality of antenna elements is used as a reference antenna element, and the signal received by the reference antenna element is used as the reference antenna element. Correlation calculation means for calculating a correlation value over a predetermined number of samples of a known signal and the known signal received by another antenna element for each antenna element in a time domain,
A reception weight calculation unit that calculates a reception weight vector using the correlation value calculated for each antenna element by the correlation calculation unit,
Multiplication means for sequentially multiplying the reception sampling signal vector sequence by the reception weight vector calculated by the reception weight calculation means for each sampling data,
A second transmission unit that transmits the one-system sampling data sequence generated by the multiplication unit to the baseband device;
With
The baseband device,
Wherein instructs calculation of the correlation value for correlation calculating means instructs the calculation of the pre-Symbol receiving weight vector with respect to the reception weight calculation unit, wherein the sampling data string of the one system received from the wireless device Reception signal processing means for reproducing data transmitted by the terminal station device,
Comprising,
Wireless communication system. There are a plurality of the terminal station devices as a data transmission source,
The multiplying means multiplies each of the reception weight vectors for each of the transmission source terminal station devices and the reception sampling signal vector sequence for each sampling data,
The second transmitting means,
Transmitting the one-system sampling data sequence of the number of the terminal station devices of the transmission source to the baseband device,
The reception signal processing means,
Reproducing the data transmitted by each of the terminal station devices from the one-system sampling data sequence for each of the transmission source terminal station devices,
The wireless communication system according to claim 1. A wireless communication method in a wireless communication system including a wireless device including a plurality of antenna elements, a baseband device that controls the wireless device, and a terminal station device that wirelessly communicates with the wireless device,
An instruction step in which the baseband device instructs the wireless device to calculate a correlation value and a reception weight vector,
When the wireless device receives an instruction from the baseband device, any one of the plurality of antenna elements based on a signal when a known signal transmitted from the terminal station device is received by each of the antenna elements. One as a reference antenna element, and the correlation value over a predetermined number of samples between the known signal received by the reference antenna element and the known signal received by another antenna element is calculated for each of the antenna elements in a time domain. a correlation calculation step of calculating, the
A reception weight calculation step of calculating the reception weight vector using the correlation value calculated for each antenna element by the correlation calculation step,
The wireless device receives a signal transmitted from the terminal station device via each of the antenna elements, and a received sampling signal vector sequence which is a sampling data sequence of a baseband signal corresponding to each of the antenna elements from the received signal. Wireless receiving step of generating
A multiplication step in which the wireless device sequentially multiplies the reception weight vector calculated in the reception weight calculation step with respect to the reception sampling signal vector sequence for each sampling data;
A second transmission step of transmitting, to the baseband device, the one-system sampling data sequence generated in the multiplication step,
A reception signal processing step in which the baseband device reproduces data transmitted by the terminal station device from the one-system sampling data sequence received from the wireless device,
A wireless communication method comprising:

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