JP6669681B2 - Filter circuits, multiplexers and modules - Google Patents

Description

  The present invention relates to a filter circuit, a multiplexer and a module, for example, a filter circuit having an acoustic wave resonator, a multiplexer and a module.

  As the low-pass filter (LPF) and the high-pass filter (HPF), a filter combining an inductor and a capacitor (that is, an LC filter) is used. The LC filter is configured by stacking, for example, ceramic layers. It is known to connect a capacitor and an inductor to an elastic wave filter (for example, Patent Document 1).

JP 2010-41141 A

  In the LC filter, there is a trade-off between the steepness of the cutoff characteristic between the pass band and the stop band and the insertion loss in the pass band. Therefore, if the insertion loss of the desired pass band is secured, the sharpness of the cutoff characteristic is deteriorated.

  The present invention has been made in view of the above problems, and has as its object to increase the steepness of a cutoff characteristic.

The present invention is a first element that is an inductor connected in series between an input terminal and an output terminal, and a capacitor that is connected in parallel with the first element between the input terminal and the output terminal. A second element, a third element which is a capacitor connected in parallel with the first element and in series with the second element between the input terminal and the output terminal, and one end of which is connected to the second element; An elastic wave resonator connected to a first node between the third element and the other end and connected to a ground terminal, wherein the first attenuation pole formed by the resonance frequency of the elastic wave resonator is minimal. Wherein the second attenuation pole closest to the pass band among the attenuation poles formed by the first element, the second element and the third element has a minimum, and the minimum of the first attenuation pole is The filter circuit is located between a pass band and a minimum of the second attenuation pole.

In the above configuration, the second attenuation pole may have a higher frequency than the pass band, and the minimum of the second attenuation pole may be a low-pass filter having a higher frequency than the minimum of the first attenuation pole.

In the above configuration, the second attenuation pole is lower in frequency than the pass band, the minimum of the second attenuation pole, the lower frequency than the minimum of the first attenuation pole may be configured to be a high pass filter.

  In the above configuration, between the input terminal and a second node to which the first element and the second element are commonly connected, and a third node to which the first element and the third element are commonly connected. A configuration may be employed in which a matching circuit is provided on at least one of the output terminal and the output terminal.

The present invention further includes a fourth element and a fifth element, which are one of a capacitor and an inductor, connected in series between an input terminal and an output terminal, and the fourth element and the fifth element are connected between the input terminal and the output terminal. A fourth element connected in parallel with the element and the fifth element, the second element being the other of the capacitor and the inductor, which is different from the fourth element and the fifth element, and the fourth terminal between the input terminal and the output terminal. A third element which is connected in parallel with the element and the fifth element and in series with the second element, is a third element which is different from the fourth element and the fifth element among capacitors and inductors, and one end of which is the second element An acoustic wave resonator connected to a first node between the first element and the third element, and the other end connected to a ground terminal; and one end connected to a fourth node between the fourth element and the fifth element. Connected, the other end Is connected to the ground terminal, a filter circuit comprising a sixth element is the same one as the fourth element and the fifth element of the capacitor and the inductor.

  In the above configuration, the elastic wave resonator may include a plurality of elastic wave resonators connected in series or in parallel between the first node and the ground terminal.

  In the above configuration, the elastic wave resonator may include an IDT.

  In the above configuration, the elastic wave resonator may include a piezoelectric thin-film resonator.

  The present invention is a multiplexer having the above filter circuit.

  The present invention is a module having the above filter circuit.

  ADVANTAGE OF THE INVENTION According to this invention, the steepness of a cutoff characteristic can be improved.

FIG. 1A is a circuit diagram of an LPF according to Comparative Example 1, FIG. 1B is a diagram showing a pass characteristic, and FIG. 1C is a Smith chart showing a reflection characteristic. FIG. 2A is a circuit diagram of an HPF according to Comparative Example 1, FIG. 2B is a diagram showing a pass characteristic, and FIG. 2C is a Smith chart showing a reflection characteristic. FIG. 3A is a plan view of the surface acoustic wave resonator, and FIG. 3B is a cross-sectional view of the piezoelectric thin film resonator. FIG. 4A is a circuit diagram of the LPF according to the first embodiment, FIG. 4B is a diagram illustrating transmission characteristics, and FIG. 4C is a Smith chart illustrating reflection characteristics. FIG. 5 is a circuit diagram illustrating an equivalent circuit of the filter circuit according to the first embodiment. FIG. 6A is a circuit diagram of an LPF according to a first modification of the first embodiment, FIG. 6B is a diagram illustrating a pass characteristic, and FIG. 6C is a Smith chart illustrating a reflection characteristic. FIG. 7A is a circuit diagram of an LPF according to a second modification of the first embodiment, FIG. 7B is a diagram illustrating transmission characteristics, and FIG. 7C is a Smith chart illustrating reflection characteristics. FIG. 8A is a circuit diagram of an LPF according to a third modification of the first embodiment, FIG. 8B is a diagram illustrating a pass characteristic, and FIG. 8C is a Smith chart illustrating a reflection characteristic. FIGS. 9A and 9B are circuit diagrams of a filter circuit according to a fourth modification of the first embodiment. FIG. 10A is a circuit diagram of the HPF according to the second embodiment, FIG. 10B is a diagram illustrating transmission characteristics, and FIG. 10C is a Smith chart illustrating reflection characteristics. FIG. 11A is a circuit diagram of an HPF according to a first modification of the second embodiment, FIG. 11B is a diagram illustrating a pass characteristic, and FIG. 11C is a Smith chart illustrating a reflection characteristic. FIG. 12A is a circuit diagram of an HPF according to a second modification of the second embodiment, FIG. 12B is a diagram illustrating a pass characteristic, and FIG. 12C is a Smith chart illustrating a reflection characteristic. FIG. 13A is a circuit diagram of an HPF according to a third modification of the second embodiment, FIG. 13B is a diagram illustrating transmission characteristics, and FIG. 13C is a Smith chart illustrating reflection characteristics. FIG. 14A is a circuit diagram of an HPF according to a fourth modification of the second embodiment, FIG. 14B is a diagram illustrating a transmission characteristic, and FIG. 14C is a Smith chart illustrating a reflection characteristic. FIG. 15 is a circuit diagram of the diplexer according to the third embodiment. FIG. 16 is a circuit diagram of a front-end circuit including a module according to the fourth embodiment. FIGS. 17A to 17C are cross-sectional views of the module according to the fourth embodiment.

[Comparative Example 1]
Comparative Example 1 is an example of an LC filter. FIG. 1A is a circuit diagram of an LPF according to Comparative Example 1, FIG. 1B is a diagram showing a pass characteristic, and FIG. 1C is a Smith chart showing a reflection characteristic. As shown in FIG. 1A, capacitors C01 and C02 are connected in series between terminals T1 and T2. Inductors L01 and L01 are connected in series between terminals T1 and T2. Capacitors C01 and C02 and inductors L01 and L02 are connected in parallel. The capacitor C03 is connected between the node N1 between the inductors L01 and L02 and the ground terminal.

Simulation was performed on the transmission characteristic S21 of the terminals T1 and T2 and the reflection characteristic S11 of the terminal T1. The simulation conditions of FIGS. 1B and 1C are as follows.
Terminals T1, T2: 50Ω termination C01, C02: 1.8 pF
C03: 1.6 pF
L01, L02: 2nH

In FIGS. 1B and 1C, the frequency at which S21 is the largest is indicated by m1, and the bottom frequency of the attenuation pole higher than the frequency m1 and closest to m1 is indicated by m2. The same applies to the following LPF. The magnitude of S21 and the magnitude / angle of S11 at the frequencies m1 and m2 are as follows.
Frequency m1 = 1.980 GHz, S21 = −0.00 dB, S11 = 0.02 / −6.4 °
Frequency m2 = 2.650 GHz, S21 = -97.8 dB, S11 = 1.0 / 0 °

  At the frequency m1, S21 is large (that is, the loss is small), and S11 is located substantially at the center of the Smith chart. That is, the high-frequency signal input from the terminal T1 is hardly reflected and attenuated by the LPF and is output from the terminal T2. At the frequency m2, S21 is small (that is, the attenuation is large), and the magnitude of S11 is almost 1. That is, the high-frequency signal input from the terminal T1 is almost reflected or attenuated by the LPF, and is hardly output from the terminal T2. The difference between the frequencies m1 and m2 is 670 MHz.

  FIG. 2A is a circuit diagram of an HPF according to Comparative Example 1, FIG. 2B is a diagram showing a pass characteristic, and FIG. 2C is a Smith chart showing a reflection characteristic. As shown in FIG. 2A, an inductor L03 is connected between the node N1 and a ground terminal. Other circuits are the same as those of the LPF, and the description is omitted.

The simulation conditions of FIGS. 2A and 2B are as follows.
Terminals T1, T2: 50Ω termination C01, C02: 1.8 pF
L01, L02: 2nH
L03: 2.2 nF

In FIG. 2B and FIG. 2C, the frequency at which S21 is the largest is indicated by m1, and the bottom frequency of the attenuation pole lower than the frequency m1 and closest to m1 is indicated by m2. The same applies to the following HPF. The magnitude of S21 and the magnitude / angle of S11 at the frequencies m1 and m2 are as follows.
Frequency m1 = 3.620 GHz, S21 = −0.00 dB, S11 = 0.00 / 3.0
Frequency m2 = 2.650 GHz, S21 = -98.1 dB, S11 = 1.0 / 0 °
The difference between the frequencies m1 and m2 is 970 MHz.

  As in Comparative Example 1, in the LC filter, the frequency between the frequencies m1 and m2 is several hundred MHz, and the cutoff characteristics between the passband and the stopband are not steep. When trying to make the frequencies m1 and m2 close to each other, S21 at the frequency m1 decreases (that is, the loss increases). As described above, if an attempt is made to secure the insertion loss in the pass band, the steepness of the cutoff characteristic between the pass band and the stop band cannot be improved.

  Embodiment 1 is an example of a filter circuit having an LPF function. An example of an elastic wave resonator used in the embodiment will be described. FIG. 3A is a plan view of the surface acoustic wave resonator, and FIG. 3B is a cross-sectional view of the piezoelectric thin film resonator. As shown in FIG. 3A, an IDT (Interdigital Transducer) 51 and a reflector 52 are formed on a piezoelectric substrate 50. The IDT 51 has a pair of comb-shaped electrodes 51a facing each other. The comb-shaped electrode 51a has a plurality of electrode fingers 51b and a bus bar 51c connecting the plurality of electrode fingers 51b. The reflectors 52 are provided on both sides of the IDT 51. The IDT 51 excites a surface acoustic wave on the piezoelectric substrate 50. The piezoelectric substrate 50 is, for example, a lithium tantalate substrate or a lithium niobate substrate. The IDT 51 and the reflector 52 are formed of, for example, an aluminum film or a copper film. The piezoelectric substrate 50 may be joined to a lower surface of a supporting substrate such as a sapphire substrate, an alumina substrate, a spinel substrate, or a silicon substrate. A protective film or a temperature compensation film that covers the IDT 50 and the reflector 52 may be provided.

  As shown in FIG. 3B, a piezoelectric film 57 is provided on a substrate 55. A lower electrode 56 and an upper electrode 58 are provided so as to sandwich the piezoelectric film 57. A gap 59 is formed between the lower electrode 56 and the substrate 55. The lower electrode 56 and the upper electrode 58 excite an elastic wave in the thickness longitudinal vibration mode in the piezoelectric film 57. The lower electrode 56 and the upper electrode 58 are, for example, metal films such as a ruthenium film. The piezoelectric film 57 is, for example, an aluminum nitride film. The substrate 55 is, for example, a silicon substrate, a sapphire substrate, an alumina substrate, a spinel substrate, or a glass substrate.

  FIG. 4A is a circuit diagram of the LPF according to the first embodiment, FIG. 4B is a diagram illustrating transmission characteristics, and FIG. 4C is a Smith chart illustrating reflection characteristics. As shown in FIG. 4A, a capacitor C1 is connected in series between terminals T1 and T2. An inductor L2 is connected between the terminals T1 and T2 in parallel with the capacitor C1. An inductor L3 is connected between the terminals T1 and T2 in parallel with the capacitor C1 and in series with the inductor L2. Node N1 is a node between inductors L2 and L3. The node N2 is a node to which the capacitor C1 and the inductor L2 are commonly connected. The node N3 is a node to which the capacitor C1 and the inductor L3 are commonly connected. One end of the elastic wave resonator R1 is connected to the node N1, and the other end is connected to a ground terminal. The elastic wave resonator R1 is a one-port resonator. An LC parallel resonance circuit 10 in which inductors L2 and L3 and capacitor C1 are connected in parallel between nodes N2 and N3 is formed.

The simulation conditions of FIGS. 4B and 4C are as follows.
Terminals T1, T2: 50Ω termination C1: 1.5 pF
L2, L3: 1.5 nH
R1: Simulation was performed using a piezoelectric thin-film resonator having a resonance frequency of 2.26 GHz and an anti-resonance frequency of 2.33 GHz.

  As shown in FIG. 4B, an attenuation pole A1 mainly formed by the resonance frequency of the elastic wave resonator R1 is formed at a frequency m2. Attenuation poles A2 and A3 mainly formed by capacitor C1, inductors L2 and L3 (LC parallel resonance circuit 10) are formed on the higher frequency side than attenuation pole A1.

  For example, when the capacitance of the capacitor C1 is fixed and the inductance of the inductor L1 is increased, the attenuation poles A2 and A3 shift to a lower frequency side, and the attenuation pole A1 shifts to a higher frequency side. When the inductance of the inductor L1 is fixed and the capacitance of the capacitor C1 is increased, the attenuation pole A2 shifts to a lower frequency side, and the attenuation poles A1 and A3 shift to a higher frequency side. As described above, by appropriately setting the capacitor C1 and the inductor L1, an attenuation pole can be formed at a desired frequency.

  Since the attenuation pole A1 is mainly formed by the resonance frequency of the elastic wave resonator R1, the cutoff characteristics can be made steep. If the attenuation poles A1 and A2 approach and the attenuation poles A1 and A2 form one attenuation pole, the steepness of the attenuation pole A1 cannot be obtained. For this reason, it is preferable that each of the attenuation poles A1 and A2 has a minimum.

4B and 4C, the magnitude of S21 and the magnitude / angle of S11 at frequencies m1 and m2 are as follows.
Frequency m1 = 2.223 GHz, S21 = −0.55 dB, S11 = 0.06 / 18 °
Frequency m2 = 2.314 GHz, S21 = -37.1 dB, S11 = 0.98 / 13 °

  In the first embodiment, the loss at the frequency m1 is as good as −0.55 dB, and the difference between the frequencies m1 and m2 is 91 MHz, which can be reduced by almost one digit compared to the first comparative example. As described above, the insertion loss of the pass band can be reduced, and the sharpness of the cutoff characteristic between the pass band and the stop band can be improved.

  FIG. 5 is a circuit diagram illustrating an equivalent circuit of the filter circuit according to the first embodiment. As shown in FIG. 5, the elastic wave resonator R1 is equivalently represented by capacitors C10 and C11 and an inductor L11. The attenuation poles A1 to A3 are determined by the combined impedance of these capacitors C1, C10 and C11 and inductors L2, L3 and L11. Therefore, the bottom frequency of the attenuation pole A1 is formed by the resonance frequency of the elastic wave resonator R1, but is different from the single resonance frequency of the elastic wave resonator R1. The frequencies at the bottoms of the attenuation poles A2 and A3 are formed by the capacitor C1 and the inductors L2 and L3, but are affected by the elastic wave resonator R1.

[Modification 1 of Embodiment 1]
The first modification of the first embodiment is an example in which the inductor and the capacitor of the LC parallel resonance circuit 10 are replaced. FIG. 6A is a circuit diagram of an LPF according to a first modification of the first embodiment, FIG. 6B is a diagram illustrating a pass characteristic, and FIG. 6C is a Smith chart illustrating a reflection characteristic. As shown in FIG. 6A, the capacitor C1 and the inductors L2 and L3 are replaced by the inductor L1 and the capacitors C2 and C3, respectively, as compared with the first embodiment. Other configurations are the same as those of the first embodiment, and a description thereof will be omitted.

The simulation conditions in FIGS. 6B and 6C are as follows.
Terminal T1, T2: 50Ω termination L1: 1.5nH
C2, C3: 5.5 pF
R1: Simulation was performed using a piezoelectric thin-film resonator having a resonance frequency of 2.26 GHz and an anti-resonance frequency of 2.33 GHz.

As shown in FIG. 6B, similarly to the first embodiment, attenuation poles A1 and A2 are formed. No attenuation pole A3 is formed. 6B and 6C, the magnitude of S21 and the magnitude / angle of S11 at the frequencies m1 and m2 are as follows.
Frequency m1 = 2.210 GHz, S21 = −0.31 dB, S11 = 0.8 / 23 °
Frequency m2 = 2.289 GHz, S21 = -30.7 dB, S11 = 0.97 / 69 °
The difference between the frequencies m1 and m2 is 79 MHz.

  In the first modification of the first embodiment, even if the inductor and the capacitor of the LC parallel resonance circuit 10 are exchanged, the insertion loss of the pass band can be reduced and the sharpness of the cutoff characteristic between the pass band and the stop band can be improved. .

[Modification 2 of Embodiment 1]
The second modification of the first embodiment is an example in which a matching circuit is provided. FIG. 7A is a circuit diagram of an LPF according to a second modification of the first embodiment, FIG. 7B is a diagram illustrating transmission characteristics, and FIG. 7C is a Smith chart illustrating reflection characteristics. As shown in FIG. 7A, a matching circuit 12 is provided between a terminal T1 and an inductor L2 as compared with the first embodiment. The matching circuit 12 has an inductor L7 connected in series between the terminal T1 and the node N2, and a capacitor C7 shunt-connected to the node N2. A matching circuit 14 is provided between the inductor L3 and the terminal T2. The matching circuit 14 has an inductor L8 connected between the node N3 and the terminal T2, and a capacitor C8 shunt-connected to the node N3. Other configurations are the same as those of the first embodiment, and a description thereof will be omitted.

The simulation conditions of FIGS. 7B and 7C are as follows.
Terminals T1, T2: 50Ω termination C1: 1.5 pF
C7, C8: 0.8 pF
L2, L3: 1.5 nH
L7, L8: 2.2 nH
R1: Simulation was performed using a piezoelectric thin-film resonator having a resonance frequency of 2.26 GHz and an anti-resonance frequency of 2.33 GHz.

As shown in FIG. 7B, attenuation poles A1 to A3 are formed as in the first embodiment. 7B and 7C, the magnitude of S21 and the magnitude / angle of S11 at the frequencies m1 and m2 are as follows.
Frequency m1 = 2.215 GHz, S21 = −0.40 dB, S11 = 0.04 / −57 °
Frequency m2 = 2.314 GHz, S21 = −34.1 dB, S11 = 0.97 / −68 °
The difference between the frequencies m1 and m2 is 99 MHz.

  In the second modification of the first embodiment, by providing the matching circuits 12 and 14, the insertion loss in the pass band can be made smaller than that of the first embodiment. Further, the pass band can be widened. In addition, the attenuation of the stop band can be increased as compared with the first embodiment.

[Modification 3 of Embodiment 1]
The third modification of the first embodiment is an example in which the capacitor C1 is divided. FIG. 8A is a circuit diagram of an LPF according to a third modification of the first embodiment, FIG. 8B is a diagram illustrating a pass characteristic, and FIG. 8C is a Smith chart illustrating a reflection characteristic. As shown in FIG. 8A, the capacitor C1 is divided into capacitors C4 and C5 in series as compared with the second modification of the first embodiment. Node N4 is a node between capacitors C4 and C5. The capacitor C6 has one end connected to the node N4 and the other end connected to a ground terminal. The other configuration is the same as that of the second modification of the first embodiment, and the description is omitted.

The simulation conditions of FIGS. 8B and 8C are as follows.
Terminals T1, T2: 50Ω termination C4, C5: 3.3 pF
C6: 0.5 pF
C7, C8: 0.6 pF
L2, L3: 1.6 nH
L7, L8: 2.7 nH
R1: Simulation was performed using a piezoelectric thin-film resonator having a resonance frequency of 2.26 GHz and an anti-resonance frequency of 2.33 GHz.

As shown in FIG. 8B, attenuation poles A1 to A3 are formed as in the first embodiment. 8B and 8C, the magnitude of S21 and the magnitude / angle of S11 at frequencies m1 and m2 are as follows.
Frequency m1 = 2.232 GHz, S21 = −0.39 dB, S11 = 0.04 / −90 °
Frequency m2 = 2.350 GHz, S21 = -31.8 dB, S11 = 0.97 / -114 °
The difference between frequencies m1 and m2 is 118 MHz.

  In the third modification of the first embodiment, the attenuation of the stop band can be made larger than that of the second modification of the first embodiment.

[Modification 4 of Embodiment 1]
FIGS. 9A and 9B are circuit diagrams of a filter circuit according to a fourth modification of the first embodiment. As shown in FIG. 9A, the elastic wave resonator R1 may be divided into a plurality of elastic wave resonators R1a and R1b in series. Further, as shown in FIG. 9B, the elastic wave resonator R1 may be divided into a plurality of elastic wave resonators R1a and R1b in parallel. Other configurations are the same as those of the first embodiment, and a description thereof will be omitted.

  Second Embodiment A second embodiment is an example of a filter circuit having an HPF function. FIG. 10A is a circuit diagram of the HPF according to the second embodiment, FIG. 10B is a diagram illustrating transmission characteristics, and FIG. 10C is a Smith chart illustrating reflection characteristics. As shown in FIG. 10A, the circuit is the same as in the first embodiment.

The simulation conditions of FIGS. 10B and 10C are as follows.
Terminal T1, T2: 50Ω termination C1: 3.9 pF
L2, L3: 0.9 nH
R1: Simulation was performed using a piezoelectric thin-film resonator having a resonance frequency of 2.26 GHz and an anti-resonance frequency of 2.33 GHz.

  As shown in FIG. 10B, an attenuation pole A1 mainly formed from the resonance frequency of the elastic wave resonator R1 is formed at the frequency m2. An attenuation pole A2 mainly formed of the capacitor C1, the inductors L2 and L3 (LC parallel resonance circuit 10) is formed on the lower frequency side of the attenuation pole A1.

10 (b) and 10 (c), the magnitude of S21 and the magnitude / angle of S11 at frequencies m1 and m2 are as follows.
Frequency m1 = 2.287 GHz, S21 = −0.19 dB, S11 = 0.02 / −98 °
Frequency m2 = 2.202 GHz, S21 = -27.0 dB, S11 = 0.95 / -108 °

  In the second embodiment, the loss at the frequency m1 is as small as -0.19 dB, and the difference between the frequencies m1 and m2 is 85 MHz, which can be made one digit smaller than that of the comparative example 1. As described above, the insertion loss of the pass band can be reduced, and the sharpness of the cutoff characteristic between the pass band and the stop band can be improved.

[Modification 1 of Embodiment 2]
The first modification of the second embodiment is an example in which the inductor and the capacitor of the LC parallel resonance circuit 10 are replaced. FIG. 11A is a circuit diagram of an HPF according to a first modification of the second embodiment, FIG. 11B is a diagram illustrating a pass characteristic, and FIG. 11C is a Smith chart illustrating a reflection characteristic. As shown in FIG. 11A, the capacitor C1 and the inductors L2 and L3 are replaced by the inductor L1 and the capacitors C2 and C3, respectively, as compared with the first embodiment. The other configuration is the same as that of the second embodiment, and the description is omitted.

The simulation conditions in FIGS. 11B and 11C are as follows.
Terminal T1, T2: 50Ω termination C2, C3: 7.1 pF
L1: 2nH
R1: Simulation was performed using a piezoelectric thin-film resonator having a resonance frequency of 2.26 GHz and an anti-resonance frequency of 2.33 GHz.

As shown in FIG. 11B, attenuation poles A1 and A2 are formed as in the second embodiment. 11B and 11C, the magnitude of S21 and the magnitude / angle of S11 at frequencies m1 and m2 are as follows.
Frequency m1 = 2.291 GHz, S21 = −0.26 dB, S11 = 0.03 / −83 °
Frequency m2 = 2.237 GHz, S21 = -27.7 dB, S11 = 0.96 / -110 °
The difference between the frequencies m1 and m2 is 54 MHz.

  In the first modification of the second embodiment, even if the inductor and the capacitor of the LC parallel resonance circuit 10 are exchanged, the insertion loss in the pass band can be reduced and the sharpness of the cutoff characteristic between the pass band and the stop band can be improved. .

[Modification 2 of Embodiment 2]
The second modification of the second embodiment is an example in which a matching circuit is provided in the second embodiment. FIG. 12A is a circuit diagram of an HPF according to a second modification of the second embodiment, FIG. 12B is a diagram illustrating a pass characteristic, and FIG. 12C is a Smith chart illustrating a reflection characteristic. As shown in FIG. 12A, a matching circuit 12 is provided between a terminal T1 and an inductor L2 as compared with the second embodiment. Matching circuit 12 has inductor L7 shunt-connected to node N2. A matching circuit 14 is provided between the inductor L3 and the terminal T2. Matching circuit 14 has inductor L8 shunt-connected to node N3. The other configuration is the same as that of the second embodiment, and the description is omitted.

The simulation conditions in FIGS. 12B and 12C are as follows.
Terminal T1, T2: 50Ω termination C1: 3.9 pF
L2, L3: 0.9 nH
L7, L8: 2.5 nH
R1: Simulation was performed using a piezoelectric thin-film resonator having a resonance frequency of 2.26 GHz and an anti-resonance frequency of 2.33 GHz.

As shown in FIG. 12B, attenuation poles A1 and A2 are formed as in the second embodiment. 12 (b) and 12 (c), the magnitude of S21 and the magnitude / angle of S11 at frequencies m1 and m2 are as follows.
Frequency m1 = 2.260 GHz, S21 = −0.19 dB, S11 = 0.01 / −8 °
Frequency m2 = 2.202 GHz, S21 = 18.6 dB, S11 = 0.88 / 9 °
The difference between the frequencies m1 and m2 is 58 MHz.

  In the second modification of the second embodiment, by providing the matching circuits 12 and 14, the insertion loss in the pass band can be made smaller than that of the second embodiment. Further, the pass band can be widened. In addition, the attenuation of the stop band can be increased as compared with the first embodiment.

[Modification 3 of Embodiment 2]
The third modification of the second embodiment is an example in which a matching circuit is provided in the first modification of the second embodiment. FIG. 13A is a circuit diagram of an HPF according to a third modification of the second embodiment, FIG. 13B is a diagram illustrating transmission characteristics, and FIG. 13C is a Smith chart illustrating reflection characteristics. As shown in FIG. 13A, a matching circuit 12 is provided between a terminal T1 and a node N2 as compared with the first modification of the second embodiment. A matching circuit 14 is provided between the terminal T2 and the node N3 . Matching circuits 12 and 14 are the same as those of the second modification of the second embodiment. The other configuration is the same as that of the second modification of the second embodiment, and the description is omitted.

The simulation conditions of FIGS. 13B and 13C are as follows.
Terminals T1, T2: 50Ω termination L1: 2.0 nH
C2, C3: 7.1 pF
L7, L8: 2.6 nH
R1: Simulation was performed using a piezoelectric thin-film resonator having a resonance frequency of 2.26 GHz and an anti-resonance frequency of 2.33 GHz.

As shown in FIG. 13B, attenuation poles A1 and A2 are formed as in the second embodiment. 13B and 13C, the magnitude of S21 and the magnitude / angle of S11 at the frequencies m1 and m2 are as follows.
Frequency m1 = 2.276 GHz, S21 = −0.20 dB, S11 = 0.07 / 50 °
Frequency m2 = 2.237 GHz, S21 = -18.9 dB, S11 = 0.88 / -6 °
The difference between the frequencies m1 and m2 is 39 MHz.

  In the third modification of the second embodiment, by providing the matching circuits 12 and 14, the insertion loss in the pass band can be made smaller than that of the first modification of the second embodiment. Further, the pass band can be widened. In addition, the attenuation of the stop band can be increased as compared with the first embodiment.

[Modification 4 of Embodiment 2]
The fourth modification of the second embodiment is an example in which the inductor L1 is divided. FIG. 14A is a circuit diagram of an HPF according to a fourth modification of the second embodiment, FIG. 14B is a diagram illustrating a transmission characteristic, and FIG. 14C is a Smith chart illustrating a reflection characteristic. As shown in FIG. 14A, the inductor L1 is divided into inductors L4 and L5 in series as compared with the third modification of the second embodiment. One end of the inductor L6 is connected to the node N4 and the other end is connected to the ground terminal. The matching circuit 12 has a capacitor C7 connected between the terminal T1 and the node N2. The matching circuit 14 has a capacitor C8 connected between the terminal T2 and the node N3. The other configuration is the same as that of the third modification of the second embodiment, and the description is omitted.

The simulation conditions in FIGS. 14B and 14C are as follows.
Terminals T1, T2: 50Ω termination L4, L5: 1.3 nH
L6: 3.8 nH
L7, L8: 2.9 nH
C2, C3: 4.7 pF
C7, C8: 1.4 pF
R1: Simulation was performed using a piezoelectric thin-film resonator having a resonance frequency of 2.26 GHz and an anti-resonance frequency of 2.33 GHz.

As shown in FIG. 14B, attenuation poles A1 and A2 are formed as in the second embodiment. 14B and 14C, the magnitude of S21 and the magnitude / angle of S11 at the frequencies m1 and m2 are as follows.
Frequency m1 = 2.400 GHz, S21 = -0.12 dB, S11 = 0.01 / 78 °
Frequency m2 = 2.220 GHz, S21 = -21.1 dB, S11 = 0.90 / -139 °
The difference between the frequencies m1 and m2 is 180 MHz.

  In the fourth modification of the second embodiment, the amount of attenuation of the stop band can be made larger than that of the third modification of the second embodiment.

[Modification 5 of Embodiment 2]
Like the modified example 4 of the first embodiment, also in the HPF, as shown in FIG. 9A, the elastic wave resonator R1 may be divided into a plurality of elastic wave resonators R1a and R1b in series. Further, as shown in FIG. 9B, the elastic wave resonator R1 may be divided into a plurality of elastic wave resonators R1a and R1b in parallel.

  According to the first and second embodiments and the modifications thereof, the LC parallel resonance circuit 10 is connected in series between the terminal T1 (input terminal) and the terminal T2 (output terminal). One end of the elastic wave resonator R1 is connected to a node N1 (first node), and the other end is connected to a ground terminal. Here, as an element forming the LC parallel resonance circuit 10, a capacitor C1 or an inductor L1 having one end connected to a node N2 (second node) and the other end connected to a node N3 (third node) is a first element. The capacitor C2 or the inductor L2 having one end connected to the node N2 and the other end connected to the node N1 is defined as a second element. The inductor L3 or the capacitor C3 having one end connected to the node N1 and the other end connected to the node N3 is defined as a third element. In order for the first element to the third element to form the LC parallel resonance circuit 10, as in the first embodiment and its modified examples 2 to 4 and the second embodiment and its modified example 2, each of the first elements has a capacitor C1. At this time, the second and third elements become inductors L2 and L3, respectively. As in the first modification of the first embodiment and the first, third, and fourth modifications of the second embodiment, when the first element is the inductor L1, the second element and the third element are the capacitors C2 and C3, respectively.

  As a result, as in the first and second embodiments and the modifications thereof, the insertion loss at the frequency m1 (pass band) is made similar to those of the first and second comparative examples, and the cutoff characteristic between the pass band and the stop band is sharp. Performance can be improved.

  As in the first embodiment and its modifications, the attenuation poles A2 and A3 formed by the first, second and third elements have a higher frequency than the attenuation pole A1 formed by the elastic wave resonator. Thereby, a filter circuit functioning as an LPF can be realized.

  As in the second embodiment, the attenuation pole A2 formed by the first element, the second element, and the third element has a lower frequency than the attenuation pole A1 formed by the elastic wave resonator. Thereby, a filter circuit functioning as an HPF can be realized.

  The LPF and the HPF may be combined to form a bandpass filter.

  The attenuation pole A1 is mainly formed by the resonance frequency of the elastic wave resonator R1. However, it is affected by the LC parallel resonance circuit 10 and the like. Therefore, the frequency at the bottom of the attenuation pole A1 is different from the resonance frequency when the elastic wave resonator R1 is used alone. The attenuation poles A2 and A3 are mainly formed from the LC parallel resonance circuit 10. However, it is affected by the elastic wave resonator R1. For this reason, the bottom frequencies of the attenuation poles A2 and A3 are different from the resonance frequency when the LC parallel resonance circuit 10 is used alone.

  Since the attenuation pole A1 is mainly formed by the resonance frequency of the elastic wave resonator R1, the steepness of the cutoff characteristic between the pass band and the stop band can be improved. Further, since the attenuation poles A1 and A2 have different minimums, the sharpness of the cutoff characteristics can be further improved. Moreover, the attenuation amount of the stop band can be increased by the attenuation pole A2.

  As in the first and second embodiments and their modifications, the capacitances of the capacitors C2 and C3 may be substantially the same. The inductances of the inductors L2 and L3 may be substantially the same. That is, the reactance of the first element and the reactance of the second element may be substantially equal.

  The capacitances of the capacitors C2 and C3 may be different from each other. The inductances of the inductors L2 and L3 may be different from each other. That is, the reactance of the first element and the reactance of the second element may be different from each other.

  In addition to the LC parallel resonance circuit 10, another LC parallel resonance circuit having a resonance frequency different from that of the LC parallel resonance circuit 10 can be provided between the terminals T1 and T2. Thereby, the stop band can be widened.

  As in the first embodiment and its modified examples 1 to 3 and the second embodiment and its modified examples 1 to 4, only one elastic wave resonator R1 may be connected in series between the node N1 and the ground terminal. As in Modification 4 of Embodiment 1 and Modification 5 of Embodiment 2, a plurality of elastic wave resonators R1a and R1b may be connected in series between the node N1 and the ground terminal. A plurality of elastic wave resonators R1a and R1b may be connected in parallel between the node N1 and the ground terminal.

  When a plurality of elastic wave resonators R1a and R1b are provided, the resonance frequencies of the plurality of elastic wave resonators R1a and R1b may be substantially equal. At least one resonance frequency of the plurality of elastic wave resonators R1a and R1b may be different from the resonance frequency of another elastic wave resonator. By making the resonance frequencies of the elastic wave resonators R1a and R1b different, the stop band can be widened.

  As in Modifications 2 and 3 of Embodiment 1 and Modifications 2 to 4 of Embodiment 2, a matching circuit is provided in at least one of between the terminal T1 and the node N1, and between the node N2 and the terminal T2. You may have. Thereby, the insertion loss of the pass band can be suppressed and the pass band can be widened.

  As in the third modification of the first embodiment, when the first element is a capacitor, the first element includes capacitors C4 (fourth element) and C5 (fifth element) connected in series between the terminals T1 and T2. Element). The capacitor C6 has one end connected to the node N4 (fourth node) and the other end connected to a ground terminal. As in the fourth modification of the second embodiment, when the first element is an inductor, the first element includes inductors L4 (fourth element) and L5 (fifth element) connected in series between terminals T1 and T2. Element). The inductor L6 has one end connected to the node N4 (fourth node) and the other end connected to a ground terminal. Thereby, the attenuation characteristic of the stop band can be improved.

  The elastic wave resonators R1, R1a, and R1b may include an IDT as shown in FIG. 3A, or may include a piezoelectric thin film resonator as shown in FIG. 3B.

  When the elastic wave resonator is a piezoelectric thin film resonator, two elastic wave resonators R1a and R1b are provided in series or in parallel between the node N1 and the ground terminal. The capacitance, the resonance frequency, and the anti-resonance frequency of the elastic wave resonators R1a and R1b are made substantially equal. Further, the polarization directions of the piezoelectric films of the elastic wave resonators R1a and R1b are opposite to each other when viewed from the node N1 or the ground terminal. Thereby, the second harmonic generated by the elastic wave resonators R1a and R1b can be suppressed.

  FIG. 15 is a circuit diagram of the diplexer according to the third embodiment. As shown in FIG. 15, the LPF 60 is connected between the common terminal Ant and the terminal TL. The HPF 62 is connected between the common terminal Ant and the terminal TH. The LPF 60 is a filter circuit according to the first embodiment and its modification, and the HPF 62 is a filter circuit according to the second embodiment and its modification. Thereby, the sharpness of the cutoff characteristics of the LPF 60 and the HPF 62 can be improved. Therefore, demultiplexing with a narrow band interval becomes possible. Either the LPF 60 or the HPF 62 may be the filter circuit of the first or second embodiment. By combining the filter circuits of the first and second embodiments and the modified examples thereof, it is possible to realize a multiplexer such as a duplexer, a triplexer, or a quadplex other than the diplexer. This enables demultiplexing with a narrow band interval as in a system using carrier aggregation or MIMO (Multiple-Input and Multiple-Output).

  FIG. 16 is a circuit diagram of a front-end circuit including a module according to the fourth embodiment. As shown in FIG. 16, the common terminal Ant of the diplexer 30 is connected to the antenna 42. The terminals TL and TH of the diplexer 30 are connected to a duplexer 34 via a switch 32, respectively. The RX terminal of the duplexer 34 is connected to an RFIC (Radio Frequency Integrated Circuit) 40 (broken line). The TX terminal of the duplexer 34 is connected to an RFIC (Radio Frequency Integrated Circuit) 40 via a switch 36 and a power amplifier 38 (solid line).

  The power amplifier 38 amplifies the transmission signal output from the RFIC 40. The switch 36 outputs the amplified transmission signal to one transmission terminal of one or a plurality of duplexers 34. The duplexer 34 outputs a signal in the transmission band among the high-frequency signals input to the transmission terminal to the switch 32 and suppresses signals of other frequencies. The duplexer 34 outputs a signal in a reception band among the high-frequency signals input from the switch 32 to the RFIC 40 and suppresses signals of other frequencies. The RFIC 40 includes a low noise amplifier, and amplifies a signal in a reception band.

  The switch 32 connects one of the one or a plurality of duplexers 34 to the terminal TL or TH. The diplexer 30 outputs or inputs a low-band signal input or output to the terminal TL to the common terminal Ant, and does not input or output a high-band signal to the terminal TL.

  The diplexer 30 outputs or inputs a signal to the common terminal Ant to a low band input or output to or from the terminal TL, and does not input or output a high band signal to the terminal TL. The diplexer 30 causes a high-band signal input or output to the terminal TH to output or input to the common terminal Ant, and does not allow a low-band signal to be input or output to the terminal TH.

  In FIG. 16, the diplexer 30 can be the diplexer of the third embodiment. Further, the filters included in the duplexer 34 can be the filter circuits of the first and second embodiments and their modifications.

  FIGS. 17A to 17C are cross-sectional views of the module according to the fourth embodiment. As shown in FIG. 17A, an acoustic wave resonator 22, a multilayer filter 24, and a chip component 26 are mounted on a printed circuit board 20. A resin sealing portion 28 is provided on the printed circuit board 20 so as to cover the elastic wave resonator 22, the multilayer filter 24, and the chip component 26. The printed board 20 is a board in which wiring is formed on an insulating board such as a glass epoxy resin. The resin sealing portion 28 is a molding resin such as an epoxy resin.

  The elastic wave resonator 22 is provided with at least a part of the filters of the elastic wave resonators R1, R1a and R1b, and / or the duplexer 34 in the first and second embodiments and the modifications thereof. The multilayer filter 24 is provided with at least a part of the LC parallel resonance circuit 10 of the first and second embodiments and the modifications thereof, and / or at least a part of the diplexer 30 and the duplexer 34. The chip component 26 is at least a part of the capacitor and the inductor of the LC parallel resonance circuit 10 and / or the diplexer 30 of the first and second embodiments and the modifications.

  As shown in FIG. 17B, a ceramic substrate 20a such as LTCC (Low Temperature Co-fired Ceramic) or HTCC (High Temperature Co-fired Ceramic) may be used instead of the printed circuit board 20. The other configuration is the same as that of FIG.

  As shown in FIG. 17C, the multilayer filter 24 may not be provided. The other configuration is the same as that of FIG.

  The module may include at least one of the diplexer 30 and the duplexer 34 in FIG.

  As described above, the embodiments of the present invention have been described in detail. However, the present invention is not limited to the specific embodiments, and various modifications and changes may be made within the scope of the present invention described in the appended claims. Changes are possible.

10 LC parallel resonance circuit 12, 14 Matching circuit

Claims (11)

A first element which is an inductor connected in series between the input terminal and the output terminal;
A second element that is a capacitor connected in parallel with the first element between the input terminal and the output terminal;
A third element, which is a capacitor connected between the input terminal and the output terminal in parallel with the first element and in series with the second element;
An elastic wave resonator having one end connected to a first node between the second element and the third element, and the other end connected to a ground terminal;
With
The first attenuation pole formed by the resonance frequency of the elastic wave resonator has a minimum, and the second attenuation pole closest to the pass band among the attenuation poles formed by the first element, the second element, and the third element. The pole has a minimum,
A filter circuit in which the minimum of the first attenuation pole is located between the pass band and the minimum of the second attenuation pole.   2. The filter circuit according to claim 1, wherein the second attenuation pole has a higher frequency than the passband, and the minimum of the second attenuation pole is a low-pass filter whose frequency is higher than the minimum of the first attenuation pole. 3.   The filter circuit according to claim 1, wherein the second attenuation pole has a lower frequency than the pass band, and the minimum of the second attenuation pole is a high-pass filter having a lower frequency than the minimum of the first attenuation pole.   A second node to which the first element and the second element are commonly connected to the input terminal, a third node to which the first element and the third element are commonly connected, and the output terminal; The filter circuit according to any one of claims 1 to 3, further comprising a matching circuit in at least one of: A fourth element and a fifth element each of which is one of a capacitor and an inductor connected in series between the input terminal and the output terminal;
A second element connected in parallel with the fourth element and the fifth element between the input terminal and the output terminal, and a second element that is the other of the capacitor and the inductor that is different from the fourth element and the fifth element;
The fourth element and the fifth element are connected between the input terminal and the output terminal in parallel with the fourth element and the fifth element, and are different from the fourth element and the fifth element among capacitors and inductors. A third element that is the other,
An elastic wave resonator having one end connected to a first node between the second element and the third element, and the other end connected to a ground terminal;
One end is connected to a fourth node between the fourth element and the fifth element, the other end is connected to a ground terminal, and one of the capacitor and the inductor is the same as the fourth element and the fifth element. A filter circuit comprising a sixth element.   The filter circuit according to claim 1, wherein the elastic wave resonator includes a plurality of elastic wave resonators connected in series or in parallel between the first node and the ground terminal.   The filter circuit according to claim 1, wherein the elastic wave resonator includes an IDT.   The filter circuit according to claim 1, wherein the elastic wave resonator includes a piezoelectric thin-film resonator.   The filter circuit according to any one of claims 1 to 8, wherein no capacitor is connected between the first node and the ground terminal in parallel with the acoustic wave resonator. Multiplexer having a filter circuit according to any one of claims 1 9. Module having a filter circuit according to any one of claims 1 9.

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