JP5927411B2 - MOTOR DRIVE DEVICE AND ELECTRIC DEVICE USING THE SAME - Google Patents

Description

  The present invention relates to a motor driving device that drives a brushless DC motor and an electric device using the same.

  Conventionally, this type of motor drive device, as disclosed in, for example, Patent Document 1, at a command speed while correcting speed feedback drive or maintaining a terminal voltage in a predetermined state according to a current value or a drive speed. The motor is driven by switching to either driving (see, for example, Patent Document 1).

  FIG. 15 shows a conventional motor driving apparatus described in the publication. As shown in FIG. A, the terminal voltage detection means 201, the first commutation means 202, the second commutation means 203, the switching means 204, the inverter 205, and the brushless DC motor 206 are configured.

JP 2010-166655 A

  However, in the above-described conventional configuration, the load is such that the current phase is greatly increased with respect to the terminal voltage phase and the motor current zero cross is generated before the switching element having the same phase as the motor current of the inverter 205 is turned off. However, current disturbance occurs under extremely unstable use conditions, and even when driving at high speed, a load that generates a motor current zero cross after the switching element having the same phase as the motor current of the inverter 205 is turned on is very high. Under large conditions, there was a problem that current disturbance occurred because the terminal voltage could not be corrected.

  SUMMARY OF THE INVENTION The present invention solves the above-described conventional problems, and an object of the present invention is to provide a motor drive device that can be stably driven even in a very light and unstable system state or a very heavy load state. And

In order to solve the above-described conventional problems, a motor driving device of the present invention is a motor driving device for driving a brushless DC motor including a rotor and a stator having a three-phase winding, and the three-phase winding An inverter that supplies power to the wire, a terminal voltage acquisition unit that acquires a terminal voltage of the brushless DC motor, a first waveform generation unit that outputs a first waveform signal having a waveform with a conduction angle of 120 degrees or more, A frequency setting unit that is set with a constant duty and changing only the frequency, and an energization angle determination that determines an energization angle between 120 degrees and less than 180 degrees according to the generation timing of the terminal voltage acquired by the terminal voltage acquisition unit. Unit, a second waveform generation unit that outputs a second waveform signal at the conduction angle determined by the previous period conduction angle determination unit and the frequency set by the frequency setting unit, and the terminal voltage acquisition unit acquired A waveform correction unit that outputs a corrected waveform signal obtained by correcting the second waveform signal so that a child voltage approaches a predetermined state, and the first waveform signal and the corrected waveform signal according to the operating state of the brushless DC motor A switching determination unit for switching and outputting,
A drive unit that outputs to the inverter a drive signal for instructing a supply timing of electric power that the inverter supplies to the three-phase winding based on the first waveform signal or the correction waveform signal output from the switching determination unit; It is what you have.

  As a result, the terminal voltage can be kept in an arbitrary state during driving with the second waveform signal, and the phase current 0 cross of the brushless DC motor is generated while the drive signal in phase is off. Therefore, the terminal voltage and the second waveform signal are always controlled so as to approach a predetermined state.

  The motor drive device of the present invention can be stably driven in a wide load range such as a very light load in an unstable system state or a very heavy load in which a current phase and a terminal voltage phase approach each other.

1 is a block diagram of a motor drive device according to Embodiment 1 of the present invention. Timing chart of motor drive device in same embodiment The figure explaining the optimal conduction angle of the motor drive device in the embodiment Another timing chart of the motor drive device in the same embodiment The figure which shows the relationship between the torque at the time of the synchronous drive of the brushless DC motor in the embodiment, and a phase The figure explaining the phase relationship of the phase current and terminal voltage of the brushless DC motor in the embodiment (A) The figure explaining the phase relationship of the brushless DC motor in the embodiment (b) The figure explaining the other phase relationship of the brushless DC motor in the embodiment The characteristic view which shows the relationship between the torque and the phase of the brushless DC motor 4 when the conduction angle of the second waveform signal of the brushless DC motor in the embodiment is 175 degrees The characteristic view which shows the relationship between the torque and the phase of the brushless DC motor 4 when the conduction angle of the second waveform signal of the brushless DC motor in the embodiment is 160 degrees The flowchart which shows operation | movement of the time difference measurement part 11 of the motor drive device in the embodiment. The flowchart which shows operation | movement of the conduction angle determination part 10 of the motor drive device in the embodiment. The flowchart which shows operation | movement of the waveform correction part of the motor drive device in the embodiment The figure which shows the relationship between the rotation speed of the brushless DC motor in the same embodiment, and a duty Sectional drawing of the principal part of the brushless DC motor in the same embodiment Block diagram of a conventional motor drive device

A first invention is a motor drive device for driving a brushless DC motor comprising a rotor and a stator having a three-phase winding, an inverter for supplying power to the three-phase winding, and the brushless DC motor A terminal voltage acquisition unit that acquires the terminal voltage of the terminal, a timing of the terminal voltage change that appears in the terminal voltage when the return current flowing through the brushless DC motor becomes 0 based on the value of the terminal voltage acquisition unit, and the drive signal is turned on or off A time difference measuring unit that calculates a time difference from the timing to be performed, a first waveform generating unit that outputs a first waveform signal having a waveform with a conduction angle of 120 degrees or more, a duty is constant, and only a frequency is changed and set determine a frequency setting unit, the conduction angle so that the time difference measuring unit spike voltage as a reference time difference which occurs measured between less than 180 degrees 120 degrees to A conduction angle determination unit, a second waveform generator for outputting a second waveform signal conduction angle determined by the frequency and the previous SL conduction angle determiner set by the frequency setting unit, the terminal voltage acquisition unit acquires the A waveform correction unit that outputs a corrected waveform signal obtained by correcting the second waveform signal so that the terminal voltage approaches a predetermined state, and the first waveform signal and the correction according to an operating state of the brushless DC motor. Based on the first waveform signal or the corrected waveform signal output from the switching determination unit, the switching determination unit for switching and outputting the waveform signal, and instructing the supply timing of the power supplied to the three-phase winding by the inverter It has a drive part which outputs a drive signal to the inverter.

  As a result, the terminal voltage can be kept in an arbitrary state during driving with the second waveform signal, and the phase current 0 cross of the brushless DC motor is generated while the drive signal in phase is off. The terminal voltage and the second waveform signal are always controlled so as to approach a predetermined state, so that a very light load in an unstable system state and a very close load where the current phase and the terminal voltage phase approach each other. Stable driving can be performed over a wide load range such as in a heavy state.

Further, it is only necessary to detect the induced voltage 0 cross as employed in the first waveform generator and output the high and low values, and to observe the timing of the change. It can be realized by an algorithm, and can further improve the software quality and construct a system at a lower cost.

In the second invention, in particular, the time difference measured by the time difference measuring unit of the first invention is a spike voltage generation period, and the energization angle determining unit is based on a first threshold time in which the time difference is determined. by was narrowing the conduction angle when it becomes rather small, even abrupt load change at low loads by a margin in time, the drive signal of the current zero crossings is in phase of the brushless DC motor Thus, the terminal voltage can be more reliably corrected so as to approach a predetermined state at a low load, and further stable driving is possible.

According to a third aspect of the invention, in particular, the time difference measured by the time difference measurement unit of the first invention is a spike voltage off period, and the energization angle determination unit is a second threshold time in which the time difference is determined. Since the energization angle is narrowed when it becomes smaller, the drive signal having the current 0 cross of the brushless DC motor having the same phase can be obtained by giving a margin to the time even in the case of a sudden load fluctuation at a high load. Thus, the terminal voltage can be corrected so as to approach the predetermined state more reliably at a high load, and further stable driving is possible.

In the fourth aspect of the invention, in particular, the first waveform generation unit of the first to third aspects of the invention acquires a zero cross point of the terminal voltage acquired by the terminal voltage acquisition unit as position information of the rotor, and the rotation By outputting the first waveform signal based on the position information of the child, it is not necessary to provide a special circuit for acquiring the position signal for driving by the first waveform generator, and the configuration is very inexpensive. can do.

According to a fifth aspect of the invention, in particular, the brushless DC motor according to the first to fourth aspects of the present invention is constructed by embedding a permanent magnet in an iron core and further having a saliency. Since the reluctance torque due to the saliency can be effectively utilized together with the magnet torque due to the permanent magnet, the high-efficiency high-speed driving performance can be further extended along with the high-efficiency driving at low speed.

In the sixth aspect of the invention, in particular, the brushless DC motor of the first to fifth aspects of the invention drives the compressor, and the compressor is a load having a relatively large inertia. Because accuracy is not required and the phase difference fluctuation cycle is slower than the motor drive cycle, it is possible to operate with a reduced number of corrections, and the motor can be operated with a control device that is slower and less expensive. A drive device can be provided. Further, even when the same compressor as that of the conventional motor drive device is used, the refrigeration capacity can be increased, so that the high capacity refrigeration cycle can be reduced in size and price. Furthermore, if the motor driving device of the present invention is replaced with a refrigeration cycle using a conventional motor driving device, a more efficient motor can be used.

In the seventh aspect of the invention, in particular, since the compressor of the sixth aspect of the invention is a reciprocating compressor, the rotor is structurally connected with a metallic and heavy crankshaft and piston, so that the inertia is reduced. It is very large and the speed fluctuation in a short time at high speed drives a very small load, so the change in phase difference between current and terminal voltage is small, and at low speed it is highly efficient according to torque pulsation. High refrigeration capacity can be output with stable driving at high speed.

In the eighth aspect of the invention, the refrigerant used in the compressor of the sixth aspect or the seventh aspect is R600a, so that the cylinder volume is increased in order to obtain the refrigerating capacity, the inertia is increased, the speed and Stable driving that does not easily fluctuate depending on the load is possible.

The ninth aspect of the invention is an electric device provided with a brushless DC motor driven by the motor driving device of the first to eighth aspects of the invention. Therefore, more stable driving is possible. In addition, high-efficiency operation can be performed when stable, and refrigeration capacity can be improved by high-speed driving during rapid cooling. In addition, when used in an air conditioner as an electrical device, it can handle a wide driving range from the lowest load during cooling to the maximum load during heating, and can reduce power consumption particularly at low loads below the rating. .

  Hereinafter, embodiments of the present invention will be described with reference to the drawings. Note that the present invention is not limited to the present embodiment.

(Embodiment 1)
FIG. 1 is a block diagram of a motor drive device according to Embodiment 1 of the present invention. In FIG. 1, an AC power source 1 is a general commercial power source, and in Japan, a power source of 50 or 60 Hz with an effective value of 100V. The motor driving device 22 is connected to the AC power source 1 and drives the brushless DC motor 4. Hereinafter, the motor drive device 22 will be described.

  The rectifying and smoothing circuit 2 rectifies and smoothes AC power into DC power with the AC power supply 1 as an input, and includes four rectifying diodes 2a to 2d connected in a bridge and smoothing capacitors 2e and 2f. In the present embodiment, the rectifying / smoothing circuit 2 is configured by a voltage doubler rectifying circuit, but the rectifying / smoothing circuit 2 may be configured by a full-wave rectifying circuit. Further, in the present embodiment, the AC power source 1 is a single-phase AC power source, but when the AC power source 1 is a three-phase AC power source, the rectifying and smoothing circuit 2 is configured by a three-phase rectifying and smoothing circuit.

  The inverter 3 converts the DC power from the rectifying / smoothing circuit 2 into AC power. The inverter 3 is configured by connecting six switching elements 3a to 3f in a three-phase bridge. The six return current diodes 3g to 3l are connected to the switching elements 3a to 3f in the reverse direction.

The brushless DC motor 4 includes a rotor 4a having a permanent magnet and a stator 4b having a three-phase winding. The brushless DC motor 4 rotates the rotor 4a when the three-phase alternating current generated by the inverter 3 flows in the three-phase winding of the stator 4b.

  The terminal voltage acquisition unit 5 acquires the terminal voltage of the brushless DC motor 4. The terminal voltage acquisition unit 5 detects the current zero cross of the phase current of the brushless DC motor 4 by acquiring the terminal voltage of the inverter 3, for example. Specifically, the terminal voltage acquisition unit 5 determines whether or not there is a current flowing through the return current diode (for example, the diode 3h) of the inverter 3, that is, the point at which the current flow switches from positive to negative, or from negative to positive. Detect zero cross point. The terminal voltage acquisition unit 5 detects this zero cross point as the zero cross point of the phase current of the brushless DC motor 4.

  The terminal voltage acquisition unit 5 detects the relative magnetic pole position of the rotor 4 a of the brushless DC motor 4. Specifically, the terminal voltage acquisition unit 5 detects the relative rotational position of the rotor 4a based on the induced voltage generated in the three-phase winding of the stator 4b. As another position detection method, there is a method of estimating the magnetic pole position by performing vector calculation on the detection result of the motor current (phase current or bus current).

  As described above, the terminal voltage acquisition unit 5 detects the magnetic pole position of the brushless DC motor 4 and the zero cross point of the phase current.

  The speed detector 6 detects the speed (that is, the rotational speed) of the brushless DC motor 4 based on the position information detected by the terminal voltage acquisition unit 5. For example, it can be easily detected by measuring a signal from the terminal voltage acquisition unit 5 generated at a constant period.

  The frequency command unit 7 receives a speed for performing a required output from the system state and the like, and calculates a motor driving frequency from the speed and the number of stations of the brushless DC motor 4 known in advance. The calculation result is output as a command frequency to the first waveform generation unit 8, the frequency setting unit 9, and the conduction angle determination unit 10.

  The first waveform generation unit 8 generates a first waveform signal for driving the switching elements 3 a to 3 f of the inverter 3. The first waveform signal is a rectangular wave signal with an energization angle of 120 degrees to 150 degrees. In order to smoothly drive the brushless DC motor 4 having the three-phase winding, the energization angle needs to be 120 degrees or more. On the other hand, in order for the terminal voltage acquisition unit 5 to detect the position based on the induced voltage, an interval of 30 degrees or more is required as the ON / OFF interval of the switching element. For this reason, the upper limit of the conduction angle is 150 degrees obtained by subtracting 30 degrees from 180 degrees. The first waveform signal may be a waveform conforming to a rectangular wave, even if it is other than a rectangular wave. For example, it is a trapezoidal wave having a slope at the rise / fall of the waveform.

  The first waveform generation unit 8 generates a first waveform signal based on the position information of the rotor 4 a detected by the terminal voltage acquisition unit 5. The first waveform generation unit 8 further performs pulse width modulation (PWM) duty control in order to keep the speed commanded by the frequency command unit 7. As a result, the brushless DC motor 4 is efficiently driven with an optimum duty based on the rotational position.

  The frequency setting unit 9 has a constant duty and sets the frequency by changing only the frequency according to the command frequency from the frequency command unit 7.

The time difference measuring unit 11 calculates the difference between the time when the current zero cross point is acquired from the terminal voltage acquisition unit 5 and the time when the drive signal output from the drive unit 15 to the inverter 3 first turns on after the current zero crossing. Calculate and output as spike voltage off period. In addition, the difference between the time when the current zero cross point is acquired from the terminal voltage acquisition unit 5 and the time when the drive signal generated last before the current zero crossing is turned off is calculated and used as the spike voltage generation period. Output. Here, the spike voltage is a voltage generated while the energy stored in the brushless DC motor is flowing in the return current diode (for example, 3h) of the inverter 3 while the switching element is turned off.

  The conduction angle determination unit 10 acquires a spike voltage off period and a spike voltage generation period that are outputs from the time difference measurement unit 11.

  When the spike voltage off period is smaller than the first threshold time and when the spike voltage generation period is smaller than the second threshold time, the energization angle is narrowed. In this embodiment, two kinds of time differences are used, but it is also possible to control only by the spike voltage generation period. For example, a period in which the drive signal of a specific phase (for example, the U phase) is turned off is calculated from the conduction angle and the frequency, and a value obtained by subtracting the first threshold time of the present embodiment from the off period is calculated. This is provided as a third threshold time, and if the spike voltage generation time is not between the second threshold time and the third threshold time, it can be realized by narrowing the energization angle. When the threshold time difference is one type, the types measured by the time difference measuring unit 11 are reduced, so that even a microcomputer with a low processing speed can be easily realized. The same control can be performed only during the spike voltage off period.

  The first threshold time and the second threshold time indicate how much the spike voltage off period and the spike voltage generation period change when the load driven by the brushless DC motor 4 or the voltage of the AC power supply 1 changes, respectively. And set the maximum change time.

  When extending the energization angle, the spike voltage off period is greater than the result of adding a predetermined value to the first threshold time, and the spike voltage generation period is greater than the result of adding the predetermined value to the second threshold time. Suppose. By providing hysteresis in the threshold time difference in this way, the frequency of changing the energization angle is reduced, and more stable driving is possible. The predetermined value that becomes the hysteresis may be 1.5 times the maximum time fluctuation, which is the same as the value of each threshold time. As a result, the energization angle does not immediately narrow even for the maximum fluctuation. The range of the conduction angle width is 120 or more and less than 180 degrees because it is not necessary to provide an off period of 30 degrees unlike the first waveform signal. In the present embodiment, since a threshold time difference is provided, it is about 175 degrees.

  The second waveform generation unit 12 generates a second waveform signal for driving the switching elements 3 a to 3 f of the inverter 3 based on the frequency from the frequency setting unit 9 and the conduction angle determined by the conduction angle determination unit 10. Generate. The second waveform signal may be a waveform that conforms to a rectangular wave. For example, it may be a sine wave or a distorted wave. In the present embodiment, the duty is at a maximum or close to the maximum (a constant duty of 90 to 100%).

  The waveform correction unit 13 receives the second waveform signal output from the second waveform generation unit 12 and the spike voltage generation period output from the time difference measurement unit 11 as inputs. After correcting the waveform, the waveform correction unit 13 calculates the average of the received spike voltage generation period. In the present embodiment, the average of the past 10 times is used. Although the average number of times is 10, it is determined from the response speed of the system and the load fluctuation cycle. The difference between the average of the previous spike voltage generation period and the spike voltage generation period input from the time difference measurement unit 11 is calculated, the waveform proportional to the calculation result is corrected for the second waveform signal, and the switching determination unit 14 Output to. Since the average of the spike voltage generation period is calculated after correction, the average may be 0 for the first time and the difference may increase, resulting in an excessive correction amount. However, the spike voltage being driven by the first waveform signal may be increased. This can be prevented by measuring the occurrence period in advance and inputting it to calculate the average, or by setting the correction amount to 0 for the first time. In this embodiment, the initial correction amount is assumed to be zero.

  The switching determination unit 14 determines whether the rotation speed of the rotor 4a is low or high, and switches the waveform signal input to the drive unit 15 between the first waveform signal and the second waveform signal. Specifically, the first waveform signal is selected when the speed is low, and the second waveform signal is selected and output when the speed is high. Here, the determination of whether the rotation speed is low or high can be made based on the actual speed detected by the speed detector 6. In addition, the determination of whether the speed is low or high can also be made based on the set rotational speed and the duty. For example, when the duty is maximum (generally 100%), the speed is maximum, so the switching determination unit 14 switches the waveform signal to the second waveform signal.

  Based on the waveform signal output from the switching determination unit 14, the drive unit 15 outputs a drive signal that instructs the supply timing of the power that the inverter 3 supplies to the three-phase winding of the brushless DC motor 4. Specifically, the drive signal turns on or off (hereinafter referred to as on / off) the switching elements 3a to 3f of the inverter 3. As a result, optimum AC power is applied to the stator 4b, the rotor 4a rotates, and the brushless DC motor 4 is driven.

  Next, an electric device using the motor driving device 22 in the present embodiment will be described. A refrigerator 21 will be described as an example of the electric device.

  Although the compressor 21 is mounted on the refrigerator 21, the rotational motion of the rotor 4a of the brushless DC motor 4 is converted into reciprocating motion by a crankshaft (not shown). A piston (not shown) connected to the crankshaft reciprocates in a cylinder (not shown) to compress the refrigerant in the cylinder. That is, the compressor 17 is configured by the brushless DC motor 4 and the crankshaft, piston, and cylinder.

  As a compression method (mechanism method) of the compressor 17, an arbitrary method such as a rotary type or a scroll type is used. In this embodiment, the case of the reciprocating type will be described. The reciprocating compressor 17 has a large inertia. For this reason, when the brushless DC motor 4 of the compressor 17 is driven synchronously, the drive of the compressor 17 is stabilized.

  The refrigerant used for the compressor 17 is generally R134a or the like, but in the present embodiment, R600a is used as the refrigerant. R600a has a lower global warming potential than R134a, but has a low refrigeration capacity. In the present embodiment, the compressor 17 is constituted by a reciprocating compressor, and the cylinder volume is increased in order to ensure the refrigerating capacity. Since the compressor 17 having a large cylinder volume has a large inertia, the brushless DC motor 4 is rotated by the inertia even when the power supply voltage is lowered. As a result, fluctuations in the rotational speed are reduced, and more stable synchronous driving is possible. However, since the compressor 17 with a large cylinder volume has a large load, it is difficult to drive with the conventional motor drive device. The motor drive device 22 according to the present embodiment is optimal for driving the compressor 17 using R600a because the drive range particularly at high loads is expanded.

  The refrigerant compressed by the compressor 17 constitutes a refrigeration cycle in which the refrigerant passes through the condenser 18, the decompressor 19, and the evaporator 20 in this order and returns to the compressor 17 again. At this time, since the condenser 18 radiates heat and the evaporator 20 absorbs heat, cooling and heating can be performed. A refrigerator 21 is configured with this refrigeration cycle. Here, as another example of the electric device, an air conditioner is provided with a blower in the condenser 18 or the evaporator 20.

The operation of the motor drive device 22 configured as described above will be described. First, the operation when the speed of the brushless DC motor 4 is low (at the time of low speed) will be described. FIG. 2 is a timing chart of the motor driving device 22 in the present embodiment. In particular, FIG. 2 is a timing diagram of signals for driving the inverter 3 at a low speed. The signal for driving the inverter 3 is a drive signal output from the drive unit 15 in order to turn on / off the switching elements 3 a to 3 f of the inverter 3. In this case, the drive signal is obtained based on the first waveform signal. The first waveform signal is output from the first waveform generation unit 8 based on the output of the terminal voltage acquisition unit 5.

  In FIG. 2, signals U, V, W, X, Y, and Z are drive signals for turning on / off switching elements 3a, 3c, 3e, 3b, 3d, and 3f, respectively. Waveforms Iu, Iv, and Iw are respectively U-phase, V-phase, and W-phase current waveforms of the winding of the stator 4b. Here, in driving at low speed, commutation is sequentially performed in intervals of 120 degrees based on the signal from the terminal voltage acquisition unit 5. The signals U, V, and W perform duty control by PWM control. Waveforms Iu, Iv, and Iw, which are U-phase, V-phase, and W-phase current waveforms, are sawtooth waveforms as shown in FIG. In this case, commutation is performed at an optimal timing based on the output of the terminal voltage acquisition unit 5. For this reason, the brushless DC motor 4 is driven most efficiently.

  Next, the optimum energization angle will be described with reference to FIG. FIG. 3 is a diagram for explaining an optimum energization angle of the motor drive device 22 in the present embodiment. In particular, FIG. 3 shows the relationship between the energization angle at low speed and the efficiency. In FIG. 3, line A shows circuit efficiency, line B shows motor efficiency, and line C shows total efficiency (product of circuit efficiency A and motor efficiency B). As shown in FIG. 3, when the energization angle is larger than 120 degrees, the motor efficiency B is improved. This is because the effective value of the phase current of the motor decreases (that is, the power factor increases) and the motor efficiency B increases as the copper loss of the motor decreases as the conduction angle increases. However, when the energization angle is larger than 120 degrees, the number of times of switching increases, and the switching loss may increase. In such a case, the circuit efficiency A decreases. From the relationship between the circuit efficiency A and the motor efficiency B, there is a conduction angle at which the overall efficiency C is the best. In the present embodiment, 130 degrees is the conduction angle at which the overall efficiency C is the best.

  Next, the operation when the speed of the brushless DC motor 4 is high (at high speed) will be described. FIG. 4 is a timing chart of the motor drive device 22 in the present embodiment. In particular, FIG. 4 is a timing diagram of drive signals for driving the inverter 3 at high speed. In this case, the drive signal is obtained based on the second waveform signal. The second waveform signal is output from the second waveform generation unit 12 based on the output of the frequency setting unit 9.

  Signals U, V, W, X, Y, Z and waveforms Iu, Iv, Iw in FIG. 4 are the same as those in FIG. Each signal U, V, W, X, Y, Z is commutated by outputting a predetermined frequency based on the output of the frequency setting unit 9. In this case, the conduction angle is 120 degrees or more and less than 180 degrees. FIG. 4 shows a case where the conduction angle is 150 degrees. By increasing the conduction angle, the current waveforms Iu, Iv, and Iw of each phase approximate to a sine wave.

  By increasing the frequency while keeping the duty constant, the rotational speed is significantly increased compared to the conventional case. When the rotational speed is increased, the motor is driven as a synchronous motor, and the current increases as the drive frequency increases. In this case, the peak current is suppressed by expanding the conduction angle to less than the maximum of 180 degrees. Therefore, even if the brushless DC motor 4 is driven at a higher current, it is operated without overcurrent protection.

Here, the second waveform signal generated by the second waveform generator 12 will be described. FIG. 5 is a diagram showing the relationship between torque and phase when the brushless DC motor 4 is driven synchronously. In FIG. 5, the horizontal axis represents the motor torque, and the vertical axis represents the phase difference based on the phase of the induced voltage. When the phase is positive, the phase is positive with respect to the phase of the induced voltage. 5 showing the stable state in the synchronous drive, the line D1 indicates the phase of the phase current of the brushless DC motor 4, and the line E1 indicates the phase of the terminal voltage of the brushless DC motor 4. Here, since the phase of the phase current is ahead of the phase of the terminal voltage, it can be seen that the brushless DC motor 4 is driven at high speed by synchronous driving. As is clear from the relationship between the phase of the phase current and the phase of the terminal voltage shown in FIG. 5, there is little change in the phase of the phase current with respect to the load torque. On the other hand, since the phase of the terminal voltage changes linearly, the phase difference between the phase current and the terminal voltage changes almost linearly according to the load torque.

  Thus, in synchronous driving, the driving of the brushless DC motor 4 is stabilized in relation to the phase of the appropriate phase current and the phase of the terminal voltage in accordance with the driving speed and the load. FIG. 6 shows the relationship between the terminal voltage phase and the phase current phase in this case. In particular, FIG. 6 is a vector diagram showing the relationship between the phase of the phase current due to the load and the phase of the terminal voltage on the dq plane.

  In the synchronous drive, when the load increases, the phase of the terminal voltage vector Vt changes in the advance direction while keeping the magnitude almost constant. Referring to FIG. 6, the terminal voltage vector Vt rotates in the direction of arrow F. On the other hand, when the load increases, the current vector I changes in magnitude as the load increases (for example, the current increases as the load increases) while maintaining a substantially constant phase. Referring to FIG. 6, the current vector I extends in the direction of arrow G. In this way, the phase relationship between the vectors is determined in an appropriate state in which the voltage vector and the current vector are in accordance with the driving environment (input voltage, load torque, driving speed, etc.).

  Here, a temporal change in phase at a certain load or speed when the brushless DC motor 4 is synchronously driven in an open loop will be described with reference to the drawings. FIG. 7 is a diagram for explaining the phase relationship of the brushless DC motor 4. In particular, FIG. 7 shows the relationship between the phase of the phase current of the brushless DC motor 4 and the phase of the terminal voltage. 7A and 7B, the horizontal axis represents time, and the vertical axis represents the phase based on the phase of the induced voltage (that is, the phase difference from the induced voltage). In both figures, line D2 indicates the phase of the phase current, line E2 indicates the phase of the terminal voltage, and line H2 indicates the phase difference between the phase of the phase current and the phase of the terminal voltage. FIG. 7A shows a driving state with a low load, and FIG. 7B shows a driving state with a high load. Further, since the phase of the phase current is advanced from the phase of the terminal voltage in both FIG. 7A and FIG. 7B due to the difference from the phase of the induced voltage, the brushless DC motor 4 is very It can be seen that it is driven at high speed.

  As shown in FIG. 7A, in the synchronous drive when the load is small with respect to the drive speed, the rotor 4a is delayed by an angle corresponding to the load with respect to the commutation. That is, when viewed from the rotor 4a, commutation advances and becomes a phase, and a predetermined relationship is maintained. That is, when viewed from the induced voltage, the phase of the terminal voltage and the phase current becomes a leading phase, and a predetermined relationship is maintained. Since this is the same state as the magnetic flux weakening control, high-speed driving is possible.

  On the other hand, as shown in FIG. 7B, when the load is large with respect to the driving speed, the rotor 4a is delayed with respect to the commutation so that the magnetic flux is weakened, and the rotor 4a is synchronized with the commutation cycle. Accelerate as you do. Thereafter, due to the acceleration of the rotor 4a, the phase current decreases due to the decrease of the lead phase of the terminal voltage, and the rotor 4a decelerates. This state is repeated, and the rotor 4a repeats this acceleration and deceleration. As a result, the drive state (drive speed) is not stabilized eventually. That is, as shown in FIG. 7B, the rotation of the brushless DC motor 4 fluctuates with respect to the commutation performed at a constant cycle. For this reason, when the phase of the induced voltage is used as a reference, the phase of the terminal voltage varies. In such a driving state, the rotation of the brushless DC motor 4 fluctuates, and a swell sound is generated accordingly. Further, since the current pulsates, it is determined that the current is an overcurrent, and the brushless DC motor 4 may be stopped.

Therefore, when the brushless DC motor 4 is synchronously driven in an open loop, the brushless DC motor 4 is stably driven when the load is small. However, the above disadvantage occurs when the load is large. That is, when the brushless DC motor 4 is synchronously driven in an open loop, it cannot be driven at high speed / high load, and the driving range is not expanded.

  Therefore, the motor drive device 22 in the present embodiment drives the brushless DC motor 4 in a state in which the phase of the phase current and the phase of the terminal voltage are kept in a phase relationship corresponding to the load as shown in FIG. . A method for maintaining the phase relationship between the phase of the phase current and the phase of the terminal voltage will be described below.

  The motor driving device 22 detects the reference phase of the terminal voltage (that is, the commutation reference position of the drive signal) and the reference point of the phase of the phase current, and based on this, the commutation timing in the open-loop synchronous drive (with a constant cycle). (Commutation) is corrected to determine the commutation timing while maintaining the phase relationship between the phase of the phase current and the phase of the terminal voltage. Specifically, the waveform correction unit 13 determines the commutation timing that maintains the above phase relationship.

  As the reference phase of the terminal voltage, timing at which the drive signal is turned on or off can be used. On the other hand, the reference point of the current phase is the most representative point at the current zero cross, and has a correlation with the position of the rotor 4a. Therefore, the second waveform signal corrected based on the current zero cross has a waveform having a predetermined relationship with the position of the rotor detected by the terminal voltage acquisition unit. In order to grasp this predetermined relationship, the time difference measuring unit 11 sets a spike voltage generation period which is a time from the turn-off of the drive signal which is the reference phase of the terminal voltage to the occurrence of the current zero cross which is the reference of the phase current phase. measure. Then, the waveform correction unit 13 outputs the second waveform signal corrected based on the spike voltage generation period to the drive unit 15. However, the spike voltage generation period cannot be measured depending on the load state depending on the conduction angle. Therefore, the energization angle determination unit 10 appropriately controls the energization angle based on the information of the time difference measurement unit 11 so that the correction based on the phase difference can be performed.

  The generation of spike voltage will be described with reference to FIGS.

  FIG. 8 is a characteristic diagram showing the relationship between the torque when the conduction angle of the second waveform signal is 175 degrees and the phase of the brushless DC motor 4.

  FIG. 9 is a characteristic diagram showing the relationship between the torque when the conduction angle of the second waveform signal is 160 degrees and the phase of the brushless DC motor 4.

  8 and 9, the horizontal axis indicates the motor torque, and the vertical axis indicates the phase difference based on the phase of the induced voltage. When the phase is positive, the phase is positive with respect to the phase of the induced voltage. 8 indicates the phase of the phase current of the brushless DC motor 4, and the line E3 indicates the phase of the terminal voltage of the brushless DC motor 4. 9 indicates the phase of the phase current of the brushless DC motor 4, and the line E4 indicates the phase of the terminal voltage of the brushless DC motor 4. However, the terminal voltage phase difference is assumed to be 0 degree of the timing when the drive signal on the U phase is turned on.

  In FIG. 8, the torque indicated by L1 to the torque indicated by L4 is the torque required in the system. At L2, the phase difference between the line E3 and the line D3 is +5 degrees, which is the same as the period during which the drive signal is off, so the current 0 cross point coincides with the timing at which the drive signal is turned off. That is, no spike voltage is generated with torque between L1 and L2, and correction cannot be performed. In the torque indicated by L3, since the line D3 and the line E3 are equal, the current 0 cross point coincides with the drive signal being turned on. That is, with the torque between L3 and L4, the spike and the terminal voltage generated by the drive signal are connected, and the spike voltage cannot be detected off. Thus, when the energization angle is 175 degrees, correction based on the spike voltage cannot be performed in L1 to L2 and L3 to L4, and the current waveform may be disturbed and vibration may occur.

  On the other hand, in FIG. 9, the torque indicated by L6 is equal to L1 in FIG. 8, and the torque indicated by L7 is equal to L4 in FIG. That is, the torque range is from L6 to L7. In L5, the phase difference between the line D4 and the line E4 is +20 degrees, which is the same as the period in which the drive signal at the conduction angle of 160 degrees is off, so the current 0 cross point coincides with the timing at which the drive signal is turned off. . Compared with the energization angle of 175 degrees, the spike does not disappear until the low torque, and the correction can be made below the necessary minimum torque indicated by L6. In the torque indicated by L8, since the phase difference between the line D4 and the line E4 is equal, the current 0 cross point and the drive signal OFF coincide. L8 becomes larger than the required maximum torque. This is because, by narrowing the energization angle, the phase difference of the phase current with respect to the induced voltage is stabilized in a large state. By narrowing the energization angle in this way, a section in which the spike voltage is turned off can be created and correction can be made.

  Next, specific operations of the time difference measurement unit 11, the conduction angle determination unit 10, and the waveform correction unit 13 will be described with reference to the flowcharts of FIGS. 10, 11, and 12.

  FIG. 10 is a flowchart showing the operation of the time difference measuring unit 11. FIG. 11 is a flowchart showing the operation of the conduction angle determination unit 10. FIG. 12 is a flowchart showing the operation of the waveform correction unit.

  First, in step 101 of FIG. 10, the spike generation flag indicating that the spike voltage has been generated and the spike OFF flag indicating that the spike voltage has been turned off are cleared, and the process proceeds to step 102.

  In step 102, whether or not a certain switching element is turned off, that is, the off timing of the switching element is waited for. In the present embodiment, the switching element on the lower side of the U phase, that is, the switching element 3b of the inverter 3 is waited for the off timing. If the switching element 3b does not change from on to off (No in step 102), the process returns to step 102 again and continues to wait in step 102 until the switching element 3b is turned on. On the other hand, when the switching element 3b is turned off (Yes in step 102), the process proceeds to step 103. In step 103, the time difference measuring unit 11 starts a timer for measuring time, and proceeds to step 104.

  In step 104, it is determined whether or not the spike voltage of the specific phase is turned off by the input from the terminal voltage acquisition unit 5, and the spike OFF flag that is a flag indicating that the spike voltage is turned off is not set. If the spike OFF flag is cleared, but the U-phase spike voltage is being generated in which the specific phase is the specific phase in the present embodiment (No in step 104), the process proceeds to step 105.

  In step 105, it is determined whether the upper switching element of the specific phase is turned on. Since the specific phase is the U phase, if the switching element 3a on the upper side of the U phase remains off (No in step 105), the process returns to step 104 again.

  In step 104, it is determined again whether the spike voltage of the specific phase is turned off and the spike OFF flag is not set. The spike off flag is not set as before. However, when the spike voltage of the U phase that is the specific phase is turned off (Yes in step 104), the process proceeds to step 106.

In step 106, since the spike is turned off, the time when the spike is turned off is recorded as the spike generation period. When the current phase advances greatly and no spike has occurred, a value of almost zero is recorded. Although it is actually 0, it is a very short time that exists as a steady value, and is a process that is not different from 0 in the process. Then, the spike OFF flag is set, and the process proceeds to step 105 again.

  In step 105, it is determined again whether the switching element 3a is turned on. When the switching element 3a is kept off (No in Step 105), the process proceeds to Step 104.

  In step 104, the process proceeds to step 106 in the previous step 104, and since the spike OFF flag is set in step 106, the process proceeds to step 105. That is, the condition is satisfied only for the first time when it is determined that the spike is off, and the process proceeds to step 106 and then proceeds to step 105 in step 104 after the next time.

  Step 105 determines whether or not the switching element 3a is turned on. If the switching element 3a is turned on (Yes in Step 105), the process proceeds to Step 107.

  In step 107, the timer value is recorded and recorded as the SW element off period, which is the time from when the switching element 3b is turned off to when the switching element 3a is turned on. Then, the process proceeds to Step 108.

  In step 108, it is determined whether the spike OFF flag is set. In the terminal voltage state, the spike OFF flag is set in step 106, and the spike OFF flag is set (Yes in step 108), so the routine proceeds to step 109.

  In step 109, the spike voltage off period is calculated. Since the spike voltage off period is a period from when the spike is turned off to when the drive signal is turned on, the result of subtracting the spike voltage generation period from the SW element off period calculated in step 107 is obtained. This is stored and the process proceeds to Step 110. In step 110, a spike voltage generation period and a spike voltage off period are output, and the process ends.

  On the other hand, if the spike voltage is not turned off before the switching element 3a is turned on, the process does not proceed to step 106 in step 104, but reaches step 108. In step 108, since the flag is not set in step 106, the routine proceeds to step 111.

  In step 111, the spike voltage off period is set to 0, the SW element off period is set as the spike voltage generation period, and the process proceeds to step 110. In step 110, a spike voltage generation period and a spike voltage off period are output, and the process ends.

  The conduction angle is basically controlled so that the spike voltage can be detected to be off. However, even if the spike voltage cannot be detected to be off due to an unexpected voltage fluctuation or the like, the minimum correction can be performed as a protection operation.

  After the time difference measuring unit 11 completes the measurement of the spike voltage generation period and the spike voltage off period, the flow of the conduction angle determination unit 10 shown in FIG. 11 is performed.

First, in step 201, it is determined whether or not the spike voltage generation period input from the time difference measuring unit 11 is smaller than a predetermined first threshold time. The first threshold time is set by preliminarily grasping the maximum difference between the current phase and the terminal voltage phase that fluctuate during one current cycle, and converting it into time. Further, the first threshold time is assumed to be at least longer than the spike voltage generation period recorded when the spike voltage is not generated and the step 106 is reached in the flow of FIG. As a result of the determination, if the spike voltage generation period is smaller than the first threshold time (Yes in step 201), the process proceeds to step 202.

  In step 202, when the fluctuation of the difference between the current phase and the terminal voltage phase becomes the maximum, the spike voltage may not be generated, so the energization angle is narrowed. And the energization angle after a change is output and a process is complete | finished.

  On the other hand, when the spike voltage generation period is longer than the first threshold time in Step 201 (No in Step 201), the process proceeds to Step 203.

  In step 203, it is determined whether or not the spike voltage off period input from the time difference measuring unit 11 is smaller than a predetermined second threshold time. Similar to the first threshold time, the maximum difference between the current phase and the terminal voltage phase, which fluctuates during one current cycle, is grasped in advance and set by converting into time, and the first threshold time and the second threshold time are set. The threshold time is set to the same value. As a result of the determination, if the spike voltage off period is smaller than the second threshold time (Yes in Step 203), the process proceeds to Step 204.

  In step 204, when the fluctuation of the difference between the current phase and the terminal voltage phase becomes the maximum, the spike voltage may not be able to detect the off state, so the conduction angle is narrowed. And the energization angle after a change is output and a process is complete | finished.

  On the other hand, when the spike voltage off period is longer than the second threshold time in Step 203 (No in Step 203), the process proceeds to Step 205.

  In step 205, it is determined whether or not the spike voltage off period is smaller than a value obtained by multiplying the second threshold time by 2.5. When the value to be compared is the second threshold time, there is a high possibility that the energization angle is widened immediately after the energization angle is narrowed, and an oscillation state is entered. A value of 2.5 times is set as a hysteresis for preventing this. By setting the width for enlarging the energization angle to 1 degree so that the spike voltage off period is less than 0.5 times the second threshold time and setting the value to be compared to 2.5 times the second threshold time, Even when the fluctuation range is the maximum after increasing the spike voltage, the spike voltage off period is reliably longer than the second threshold time at the next determination, and the energization angle does not narrow immediately. If the spike voltage off period is less than 2.5 times the second threshold time (No in Step 205), the energization angle is not changed and the process ends. On the other hand, when the spike voltage off period is longer than 2.5 times the second threshold time (Yes in Step 205), the process proceeds to Step 206.

  In step 206, it is determined whether or not the spike voltage generation period is shorter than a value obtained by multiplying the first threshold time by 2.5. For the same reason that 2.5 times the second threshold time is set in step 205, the first threshold time is also multiplied by 2.5 and compared with the spike voltage generation period. If the spike voltage generation period is long as a result of the comparison (No in step 206), the energization angle is not changed and the process is terminated. On the other hand, when the spike voltage generation period is longer than 2.5 times the first threshold time (Yes in Step 206), the process proceeds to Step 207.

  In step 207, the spike voltage is generated even when the energization angle is widened, and the off timing can be detected, so the energization angle is widened.

When the terminal voltage acquisition unit 5 detects the rising of the terminal voltage before the U-phase upper switching element (3a) is turned on, the induced voltage phase is advanced from the terminal voltage phase. The energization angle may be narrowed because it is determined that it is excessively applied. This determination condition is determined before step 205, and if the condition is satisfied, the determination after step 205 is not passed. As a result, application of an excessive voltage can be suppressed, and thus efficient operation can be performed.

  After the energization angle is determined by the energization changing unit B, the flow of the waveform correction unit 13 shown in FIG. 12 is performed.

  In step 301, the difference between the spike voltage generation time measured by the time difference measuring unit 11 and the average time of the past 10 times is calculated, and the process proceeds to step 302. In step 302, the commutation timing correction amount is calculated based on the difference calculated in step 301, and the process proceeds to step 303.

  Here, the correction of the commutation timing is to correct the commutation timing with respect to the basic commutation cycle based on the frequency set by the frequency setting unit 9, that is, the command speed. Therefore, when a large correction amount is added, overcurrent or step-out occurs. Therefore, when calculating the correction amount, the calculation is performed after adding a low-pass filter or the like to suppress rapid fluctuations in the commutation timing. As a result, even when the zero cross of the current is erroneously detected due to the influence of noise or the like, the influence on the correction amount is reduced and the driving stability is further improved. Furthermore, since a sudden change is suppressed in the calculation of the correction amount, the change in the commutation timing for accelerating / decelerating the brushless DC motor 4 also becomes gentle. For this reason, even when the command speed is greatly changed and the frequency (commutation cycle) by the frequency setting unit 9 is significantly changed, the change of the commutation timing becomes gradual and the acceleration / deceleration becomes smooth.

  More specifically, the commutation timing correction is to always bring the spike voltage generation period closer to the average time of the spike voltage generation period. For example, when the rotational speed of the rotor 4a decreases due to an increase in the load, the phase of the phase current moves in the delay direction with reference to the phase of the terminal voltage. For this reason, the spike voltage generation period measured this time is longer than the average time of the spike voltage generation period. In this case, the second waveform generation unit 12 corrects the commutation timing so that the commutation timing is delayed from the timing of the commutation cycle based on the rotation speed (the number of rotations). That is, since the measurement time becomes longer due to the phase of the phase current being delayed, the second waveform generator 12 delays the commutation timing to delay the phase of the terminal voltage, and the spike that is the phase difference from the phase of the phase current. Move the voltage generation period closer to the average time.

  On the contrary, when the rotational speed of the rotor 4a is increased due to a decrease in the load, the phase of the phase current moves in the advance direction with reference to the phase of the terminal voltage. For this reason, the spike voltage generation period measured this time is shorter than the average time of the spike voltage generation period. In this case, the second waveform generation unit 12 once corrects the commutation timing so that the commutation timing is earlier than the timing of the commutation cycle based on the rotation speed. That is, since the measurement time is shortened because the phase of the phase current has become earlier, the second waveform generator 12 advances the phase of the terminal voltage by advancing the commutation timing, and sets the spike voltage generation period to the average time. Move closer.

  Further, the waveform correction unit 13 corrects the commutation timing as an arbitrary timing of the specific phase (for example, only the switching element on the upper side of the U phase) (for example, once per rotation of the rotor 4a). The commutation is performed temporally with a commutation cycle based on the target rotational speed. Thereby, the phase relationship between the phase of the phase current and the phase of the terminal voltage is optimally maintained according to the load, and the driving speed of the brushless DC motor 4 is maintained.

  Next, in step 303, the average time is updated in consideration of the spike voltage generation period input this time, and the process proceeds to step 304. In step 304, the commutation timing is determined by adding a correction amount to the second waveform signal set by the second waveform generator 12.

  That is, the commutation timing is obtained by adding a correction amount to the frequency set by the second waveform generation unit 12 so that the phase of the phase current and the phase of the terminal voltage always have an average phase difference. Determined with reference to phase. Therefore, when the load increases, the phase difference, which is the difference between the phase of the phase current and the commutation timing, is narrowed (the spike voltage generation period is increased). On the other hand, the brushless DC motor 4 is based on a state in which the average time used as a correction reference is small (the average of the spike voltage generation period is large) and the phase difference is narrower than before the load is increased. Driven. As a result, the brushless DC motor 4 is driven with a larger advance angle, and the output torque is increased and the required output torque is ensured by improving the flux-weakening effect.

  Conversely, when the load is reduced, the phase difference that is the difference between the phase of the phase current and the commutation timing is widened (spike voltage generation period is reduced). On the other hand, the average time used as a reference for correction is large (the average of the spike voltage generation period is small), and the brushless DC motor 4 is based on a state in which the phase difference is widened compared to before the load is reduced. Driven. As a result, the brushless DC motor 4 is driven at a smaller advance angle, and the output torque is reduced due to the reduction of the flux-weakening effect, so that an excessive torque is not output. As described above, a drive that ensures a necessary output and does not generate an extra output is performed.

  As described above, the operation shown in FIGS. 10, 11, and 12 is performed once in one cycle of the electrical angle, thereby enabling stable driving at high speed.

  In the present embodiment, since the commutation cycle is corrected only at the off timing of the switching element 3a on the upper side of the U phase, a case where correction is performed once during one electrical angle cycle is described. However, the correction timing may be set in consideration of the application of the motor driving device 22 and the inertia of the brushless DC motor 4. For example, the correction may be performed once per rotation of the rotor 4a, corrected twice during one electrical angle cycle, or corrected every time each switching element is turned on.

  Next, the switching operation by the switching determination unit 14 will be described. FIG. 13 is a diagram showing the relationship between the rotational speed and the duty of the brushless DC motor 4 in the present embodiment.

  In FIG. 13, when the rotational speed of the brushless DC motor 4, that is, the rotational speed of the rotor 4 a is 50 r / s or less, the brushless DC motor 4 is driven based on the first waveform signal from the first waveform generator 8. Is done. The duty is adjusted to the most efficient value according to the rotational speed by feedback control.

  The rotation speed is 50 r / s, the duty is 100%, and the drive based on the first waveform generator 8 cannot be rotated further. That is, the limit is reached. Therefore, above the rotational speed of 50 r / s, the duty is constant, and only the frequency (that is, the commutation cycle) is increased, and the brushless DC motor 4 is driven.

  On the other hand, in the driving based on the corrected waveform signal obtained by correcting the second waveform signal, when the conduction voltage is 120 degrees and the induced voltage 0 cross is detected by the terminal voltage acquisition unit 5, the first waveform generation unit 8 Since the torque can be sufficiently driven by the first waveform signal, the drive is switched to the drive based on the first waveform generator 8.

  As described above, the first waveform signal generated by the first waveform generation unit 8 and the corrected waveform signal obtained by correcting the second waveform signal generated by the second waveform generation unit 12 by the waveform correction unit 13 are appropriately switched. It enables driving from low speed and low load to high speed and high negative.

  Next, the structure of the brushless DC motor 4 of the present embodiment will be described. FIG. 14 is a cross-sectional view showing a cross section perpendicular to the rotation axis of the rotor of the brushless DC motor 4 in the present embodiment.

  The rotor 4a includes an iron core 4g and four magnets 4c to 4f. The iron core 4g is configured by stacking punched thin steel sheets of about 0.35 to 0.5 mm. As the magnets 4c to 4f, arc-shaped ferrite permanent magnets are often used, and as illustrated, the magnets 4c to 4f are arranged symmetrically with the arc-shaped concave portions facing outward. On the other hand, when rare earth permanent magnets such as neodymium are used as the magnets 4c to 4f, they may be flat.

  In the rotor 4a having such a structure, an axis extending from the center of the rotor 4a toward the center of one magnet (for example, 4f) is defined as a d-axis, and one magnet (for example, 4f) is connected to the center of the rotor 4a. The axis that goes to the magnet adjacent to (for example, 4c) is the q axis. The inductance Ld in the d-axis direction and the inductance Lq in the q-axis direction have opposite saliency and are different. That is, this can effectively use torque (reluctance torque) using reverse saliency as well as torque (magnet torque) due to magnetic flux of the magnet. Therefore, torque can be used more effectively as a motor. As a result, a highly efficient motor is obtained as the present embodiment.

  In the control of the present embodiment, when driving is performed by the frequency setting unit 9 and the second waveform generation unit 12, the phase current becomes a leading phase. For this reason, since this reluctance torque is largely used, it can be driven at a higher rotation than a motor without reverse saliency.

  Further, the brushless DC motor 4 of the present embodiment has a rotor 4a in which permanent magnets 4c to 4f are embedded in an iron core 4g and has saliency. In addition to the magnet torque of the permanent magnet, reluctance torque due to saliency is used. This not only improves efficiency at low speeds, but also improves high-speed drive performance. Also, if a rare earth magnet such as neodymium is used as the permanent magnet to increase the ratio of magnet torque, or the difference between inductances Ld and Lq is increased to increase the ratio of reluctance torque, the optimum conduction angle can be changed. Can increase efficiency.

  Next, the case where the compressor 17 is driven using the motor drive device 22 of the present embodiment for the refrigerator 21 and the air conditioner will be described. In the case of a conventional motor driving device, it is necessary to use a brushless DC motor that secures a necessary torque by reducing the number of windings in order to support high-speed / high-load driving. Such a brushless DC motor has a large motor noise. If the motor drive device 22 of the present embodiment is used, even if the brushless DC motor 4 in which the winding amount of the winding is increased and the torque is reduced is used, it can be driven at high speed / high load. As a result, the duty when the rotational speed is low can be made larger than when the conventional motor driving device is used. Therefore, motor noise, particularly carrier sound (corresponding to a frequency in PWM control, for example, 3 kHz) can be reduced.

  In addition, since the compressor 17 is a reciprocating compressor, the inertia is larger and the torque pulsation at a high speed is small. Therefore, the compressor 17 can be stably operated at a high speed. Further, when the compressor 17 is mounted on the refrigerator 21, the load of the refrigerator 21 is not suddenly changed, so that the change in the phase difference between the phase of the phase current and the phase of the terminal voltage is small, and more stable driving is possible. .

In the case of driving the compressor 17 of the air conditioner using the motor driving device 22 of the present embodiment, it is possible to cope with a wide driving range from the lowest load during cooling to the maximum load during heating. It is possible to reduce power consumption at a low load below the rating.

  As described above, the present invention is a motor drive device for driving the brushless DC motor 4 including the rotor 4a and the stator 4b having the three-phase winding, and the power is supplied to the three-phase winding of the stator 4b. An inverter 3 that supplies a voltage, a terminal voltage acquisition unit 5 that acquires a terminal voltage of the brushless DC motor 4, a first waveform generation unit 8 that outputs a first waveform signal having a waveform with a conduction angle of 120 degrees or more, The frequency setting unit 9 that is set with a constant duty and changing only the frequency, and the energization angle determination that determines the energization angle between 120 degrees and less than 180 degrees according to the generation timing of the terminal voltage acquired by the terminal voltage acquisition unit 5. Unit 10, a second waveform generation unit 12 that outputs a second waveform signal at the frequency set by frequency setting unit 9 and the conduction angle determined by conduction angle determination unit 10, and the terminal voltage acquired by terminal voltage acquisition unit 5 Where A waveform correction unit 13 that outputs a corrected waveform signal obtained by correcting the second waveform signal so as to approach the state of the above, and a switch that switches and outputs the first waveform signal and the corrected waveform signal according to the operating state of the brushless DC motor 4 Based on the determination unit 14 and the first waveform signal or the correction waveform signal output from the switching determination unit 14, a drive signal that indicates the supply timing of the power that the inverter 3 supplies to the three-phase winding of the stator 4 b is invertered 3 has a drive unit 15 for outputting to the output.

  As a result, the terminal voltage can be maintained at an arbitrary state during driving with the correction waveform signal based on the second waveform signal, and the drive signal having the same phase current 0 cross of the brushless DC motor 4 is being turned off. The terminal voltage and the second waveform signal are always controlled so as to approach the predetermined state, and the very light load in the unstable system state and the current phase and the terminal voltage phase are very close. It can be driven stably over a wide load range such as when the load is heavy.

  In addition, the present embodiment includes a time difference measuring unit 11 that calculates a time difference between the terminal voltage change timing that appears in the terminal voltage when the return current flowing through the brushless DC motor 4 becomes 0 and the drive signal on or off. By changing the timing of generating the terminal voltage to a change in time difference, a simple circuit that detects the induced voltage zero crossing as employed in the first waveform generator 8 and outputs a binary value of high and low, and its change It is only necessary to observe it, and it can be realized by a simpler algorithm, and it is possible to further improve software quality and to construct a system at a lower cost.

  Further, the present embodiment realizes the predetermined state of the terminal voltage by simple calculation by assuming that the average of the time difference and the difference of the time difference are 0, and the software algorithm is simplified. An inexpensive microcomputer with improved software quality, such as improved maintainability, and low computing performance can be used.

  Further, in the present embodiment, since the energization angle determination unit 10 narrows the energization angle when the time difference becomes larger than the determined first threshold time, the load fluctuation is abrupt with a low load. In addition, by providing a margin in time, the current zero cross of the brushless DC motor 4 is prevented from appearing before the on-phase drive signal is turned on, and the terminal voltage approaches the predetermined state more reliably at low load. Thus, more stable driving is possible.

  Further, in the present embodiment, the energization angle determination unit 10 narrows the energization angle when the time difference measured by the time difference measurement unit 11 becomes smaller than the second threshold time, so that the energization angle is suddenly increased with a high load. Even when the load fluctuates, a margin is provided in the time to prevent the current zero cross of the brushless DC motor 4 from appearing before the in-phase drive signal is turned off. Correction can be made so as to approach a predetermined state, and further stable driving is possible.

  Further, in the present embodiment, the first waveform generation unit 8 acquires the zero cross point of the terminal voltage acquired by the terminal voltage acquisition unit 5 as the position information of the rotor 4a, and the first waveform generator 8 based on the position information of the rotor 4a By outputting this waveform signal, it is not necessary to provide a special circuit for acquiring a position signal for driving by the first waveform generator 8, and it can be constructed at a very low cost.

  Further, in the present embodiment, the rotor 4a of the brushless DC motor 4 is configured by embedding the permanent magnets 4c to 4f in the iron core 4g and further having saliency. Since the reluctance torque due to the saliency as well as the magnet torque by the permanent magnets 4c to 4f can be used effectively, the high-efficiency high-speed drive performance can be further extended along with the high-efficiency drive at low speed.

  Further, in the present embodiment, since the brushless DC motor 4 drives the compressor 17, the compressor 17 is a load having a relatively large inertia, so that high control accuracy like a servo motor is required. Since the phase difference fluctuation cycle is slower than the drive cycle of the brushless DC motor 4, it is possible to operate with the number of corrections reduced, and the motor drive device can be operated with a slower and cheaper control device. Can provide. Further, even when the same compressor as that of the conventional motor drive device is used, the refrigeration capacity can be increased, so that the high capacity refrigeration cycle can be reduced in size and price. Furthermore, if the motor driving device of the present invention is replaced with a refrigeration cycle using a conventional motor driving device, a more efficient motor can be used.

  Further, in the present embodiment, since the compressor 17 is a reciprocating compressor, the rotor is structurally connected with a metal and a heavy crankshaft and piston, so that inertia is very large. At high speed, speed fluctuation in a short time will drive a very small load, so the change in phase difference between current and terminal voltage will be small, and at low speed, highly efficient operation according to torque pulsation will be performed, High speed can output high refrigeration capacity with stable driving.

  Further, in the present embodiment, since the refrigerant used in the compressor 17 is R600a, the cylinder volume is increased in order to obtain the refrigerating capacity, the inertia is increased, and the stable drive that is not easily fluctuated depending on the speed and load. Is possible.

  In addition, since the refrigerator 21 (electric device) including the brushless DC motor 4 driven by the motor driving device of the present embodiment is used, the load fluctuation is not steep, so that more stable driving is possible. In addition, high-efficiency operation can be performed when stable, and refrigeration capacity can be improved by high-speed driving during rapid cooling. In addition, when used in an air conditioner as an electrical device, it can handle a wide driving range from the lowest load during cooling to the maximum load during heating, and can reduce power consumption particularly at low loads below the rating. .

  The motor drive device of the present invention is intended to improve the drive stability of a brushless DC motor at high speed / high load and to extend the drive range. Thereby, it can be applied not only to refrigerators and air conditioners but also to higher efficiency of compressors in vending machines, showcases, and heat pump water heaters. In addition, the present invention can be applied to energy saving of electric devices using brushless DC motors such as washing machines, vacuum cleaners, and pumps.

3 Inverter 4 Brushless DC motor 4a Rotor 4b Stator 4c, 4d, 4e, 4f Magnet (permanent magnet)
4 g Iron core 5 Terminal voltage acquisition unit 6 Speed detection unit 7 Frequency command unit 8 First waveform generation unit 9 Frequency setting unit 10 Energization angle determination unit 11 Time difference measurement unit 12 Second waveform generation unit 13 Waveform correction unit 14 Switch determination unit 15 Drive Part 17 Compressor 21 Refrigerator (electric equipment)
22 Motor drive device

Claims (9)

A motor driving device for driving a brushless DC motor comprising a rotor and a stator having a three-phase winding, wherein an inverter for supplying power to the three-phase winding and a terminal voltage of the brushless DC motor are obtained. The time difference between the terminal voltage acquisition unit that performs and the timing of the terminal voltage change that appears in the terminal voltage when the return current flowing through the brushless DC motor becomes 0 from the value of the terminal voltage acquisition unit and the timing that the drive signal is turned on or off A time difference measuring unit that calculates a first waveform generating unit that outputs a first waveform signal having a waveform with an energization angle of 120 degrees or more, a frequency setting unit that sets only a frequency with a constant duty , the conduction angle to determine the conduction angle for less than 180 degrees 120 degrees so that the spike voltage as a reference for the time difference the time difference measuring unit measures occurs Tough and, a second waveform generator for outputting a second waveform signal conduction angle determined by the frequency and the previous SL conduction angle determiner set by the frequency setting unit, the terminal to which the terminal voltage acquisition unit acquires A waveform correction unit that outputs a correction waveform signal obtained by correcting the second waveform signal so that the voltage approaches a predetermined state, and the first waveform signal and the correction waveform signal according to the operating state of the brushless DC motor. A switching determination unit for switching and outputting, and a drive signal for instructing a supply timing of power supplied to the three-phase winding by the inverter based on the first waveform signal or the correction waveform signal output from the switching determination unit And a drive unit that outputs to the inverter. Wherein said time difference the time difference measuring unit is measured as a spike voltage generating period, the conduction angle determination unit that the narrowing the conduction angle when the time difference becomes rather smaller than the first threshold time determined Item 2. The motor drive device according to Item 1 . The time difference measured by the time difference measurement unit is a spike voltage off period, and the energization angle determination unit narrows the energization angle when the time difference becomes smaller than a determined second threshold time. Item 2. The motor drive device according to Item 1 . The first waveform generation unit acquires a zero cross point of the terminal voltage acquired by the terminal voltage acquisition unit as position information of the rotor, and outputs the first waveform signal based on the position information of the rotor. The motor drive device of any one of Claims 1-3 . The brushless DC motor of the rotor is constructed by embedding a permanent magnet in the iron core, further the motor driving apparatus according to any one of claims 1 to 4 having saliency. The brushless DC motor is a motor driving apparatus according to any one of claims 1 to 5 for driving the compressor. The motor driving device according to claim 6 , wherein the compressor is a reciprocating compressor. The motor driving device according to claim 6 or 7 , wherein the refrigerant used in the compressor is R600a. Electrical device with a brushless DC motor driven by a motor driving device according to any one of claims 1-8.

Images