JP5625008B2 - Power converter, electric motor drive system, conveyor, lifting device - Google Patents

Description

  The present invention relates to a power converter, an electric motor drive system including the power converter, a transporter, and a lifting device.

  In a drive system for an AC electric motor (hereinafter simply referred to as “motor”) such as a synchronous motor or an induction motor, a voltage-type inverter is represented as a motor driving device that converts DC power into AC power to drive the motor. A power converter is frequently used. In order to improve the performance of such a power conversion device for driving a motor, it is necessary to accurately detect the magnetic pole position of the rotor, the rotation speed, and the like as control information for the rotor of the motor. Recent power converters do not actually measure the rotational state of the rotor by attaching a position sensor, speed detector, etc. to the motor, but estimate the rotational state of the rotor from back electromotive voltage information generated in the motor. Therefore, a control method that performs highly accurate control amount estimation is used.

  However, the motor control method for estimating the rotation state of the rotor from the back electromotive voltage information is difficult to apply because the back electromotive voltage becomes absolutely small near the extremely low speed of the motor. It is. Therefore, as a control amount estimation method at a low speed, there is a method using the saliency of the electric motor.

  Patent Document 1 describes a magnetic pole position detection device that estimates a magnetic pole position that represents the rotation state of a rotor, particularly using the saliency of a permanent magnet synchronous motor. This magnetic pole position detection device generates an alternating magnetic field in a predetermined phase of the motor, detects a high-frequency current (or voltage) of a component orthogonal to this phase, and estimates the magnetic pole position of the motor rotor based on this Calculate. This technique estimates the magnetic pole position of the motor rotor by using the characteristic (electric saliency) that the electric inductance of the motor rotor changes with respect to the superimposed phase. That is, the magnetic pole position is estimated based on the saliency by measuring the inductance from the correlation between the high frequency voltage and the pulsating current.

  With this method, it is possible to accurately estimate the operation information of the electric motor without using a sensor for detecting the rotation state of the rotor. Thereby, the cost of a sensor and a cable for outputting a detection signal of the sensor, and the trouble of installing them can be reduced. Furthermore, inappropriate behavior of the motor drive due to sensor assembly error, noise due to the surrounding environment, sensor failure, and the like can be suppressed.

Japanese Patent No. 331472

  As in the technique described in Patent Document 1, a motor control method that generates a high-frequency current by adding a high-frequency voltage to a voltage command uses a local inductance near the current operating point. However, in general, in a motor, when the load increases and the amount of current increases, a magnetic flux saturation phenomenon occurs, and the saliency of local inductance tends to decrease nonlinearly. Therefore, the sensitivity of magnetic pole position detection using saliency deteriorates in a high load region, and as a result, there is a problem that current pulsation and torque vibration increase. In some cases, there is a risk of step-out.

  The present invention has been made with respect to the above-described problems, and its main purpose is to prevent out-of-step in a high load region and to keep the drive of the motor safe. The purpose is to estimate the detection sensitivity of the magnetic pole position by polarity.

A power conversion device according to an aspect of the present invention includes a voltage conversion unit that converts a DC voltage into an AC voltage and outputs the voltage to an AC motor having saliency, a current detection unit that detects a current flowing through the AC motor, and a current detection unit. Current extraction means for extracting a high-frequency current from the current detected by the method, and estimating the magnetic pole position of the rotor of the AC motor based on the high-frequency current extracted by the current extraction means, and a period different from the rotation period of the rotor The magnetic pole position is determined based on the superposed voltage phase adjusting means for outputting the high frequency voltage phase command value for adjusting the phase for superimposing the alternating voltage changing in step, and the high frequency voltage phase command value output by the superposed voltage phase adjusting means. estimated, a current control means for outputting a fundamental wave voltage instruction for controlling the current flowing to the AC motor, the stator coordinates relative to the stator of the AC motor Rights As the trajectory of the high-frequency current on the draw phase of two or more different segments with each other, the high-frequency voltage phase and superimposing the voltage phase shifting means for adjusting the command value, a high frequency voltage phase command value adjusted by the superimposed voltage phase shifting means Based on the above, the voltage superimposing means for superimposing the alternating voltage on the fundamental voltage command and outputting it to the voltage converting means, the high frequency current extracted by the current extracting means, and the high frequency voltage phase command adjusted by the superposed voltage phase shifting means Sensitivity calculating means for calculating the sensitivity to the estimation of the magnetic pole position based on the value.
A power conversion device according to another aspect of the present invention includes a voltage conversion unit that converts a DC voltage into an AC voltage and outputs the voltage to an AC motor having saliency, a current detection unit that detects a current flowing through the AC motor, a current Current extraction means for extracting a high-frequency current from the current detected by the detection means, and estimating the magnetic pole position of the rotor of the AC motor based on the high-frequency current extracted by the current extraction means and the rotation period of the rotor Based on the superposed voltage phase adjusting means for outputting a high frequency voltage phase command value for adjusting the phase on which the alternating voltage changing at different periods is superposed, and the high frequency voltage phase command value outputted by the superposed voltage phase adjusting means, the magnetic pole Current control means for estimating a position and outputting a fundamental voltage command for controlling a current flowing through the AC motor, and at least a predetermined phase shift amount of 2 or more By switching periodically and adding to the high-frequency voltage phase command value, the superimposed voltage phase shift means for adjusting the high-frequency voltage phase command value, and the alternating voltage based on the high-frequency voltage phase command value adjusted by the superimposed voltage phase shift means Based on the voltage superimposing means that superimposes the voltage on the fundamental voltage command and outputs it to the voltage converting means, the high-frequency current extracted by the current extracting means, and the high-frequency voltage phase command value adjusted by the superimposed voltage phase shift means And sensitivity calculating means for calculating the sensitivity to the estimation of the magnetic pole position.
An electric motor starting system according to the present invention includes the above power conversion device and an AC electric motor.
The conveyance machine by this invention is provided with the said power converter device, an alternating current motor, and the conveyance part which operate | moves using the driving force generated by the alternating current motor.
A lifting device according to the present invention includes the power conversion device, an AC motor, a lifting unit, and a winding mechanism that moves the lifting unit up and down using a driving force generated by the AC motor.

  According to the present invention, it is possible to estimate the magnetic pole position detection sensitivity based on the saliency of the electric motor.

It is a lineblock diagram of electric motor drive system 110a by a 1st embodiment of the present invention. It is explanatory drawing of a fundamental wave current and a high frequency current. It is explanatory drawing of the relationship between the fundamental wave current vector I1 and the fundamental wave voltage vector V1, and the relationship between the high frequency current vector Ih and the high frequency voltage vector Vh. It is a schematic diagram of the graph showing the relationship between phase difference (theta) ivh and phase difference (theta) vhd. It is a block diagram of the electric motor drive system 110b by a comparative example. It is a figure which shows an example of the high frequency voltage vector Vh and the high frequency current vector Ih in a comparative example. It is explanatory drawing of the superimposition voltage phase shift means. It is a figure which shows an example of the high frequency voltage vector Vh and the high frequency current vector Ih in this invention. It is principle explanatory drawing of the sensitivity calculation in this invention. It is a block diagram of the electric motor drive system 110c by 2nd Embodiment of this invention. It is a block diagram of the electric motor drive system 110d by 3rd Embodiment of this invention. It is a block diagram of the electric motor drive system 110e by 4th Embodiment of this invention. It is a block diagram of the conveying machine 130 by 5th Embodiment of this invention. It is a block diagram of the raising / lowering apparatus 140 by 6th Embodiment of this invention. It is a principle explanatory view of sensitivity detection error by a transient phenomenon.

  Hereinafter, first to sixth embodiments of the present invention will be described in detail with reference to the drawings. In the following description, the same reference numerals are given to the components common to the drawings, and the description of the overlapping components is omitted.

(First embodiment)
FIG. 1 is a configuration diagram of an electric motor drive system 110a according to a first embodiment of the present invention.

  In FIG. 1, the electric motor drive system 110 a includes a power conversion device 101 a and an electric motor 1. The power conversion device 101a includes an electric motor control device 100a, a voltage conversion unit 3, and a current detection unit 2.

  First, the definition of symbols used to describe the operation of the electric motor drive system 110a will be described. First, the fundamental wave current and the high frequency current that flow from the power converter 101a to the motor 1 when the motor 1 is driven in the motor drive system 110a will be described with reference to the explanatory diagram of FIG.

  FIG. 2A schematically shows an example of the waveform of each phase of the three-phase alternating current flowing from the power converter 101a to the electric motor 1. FIG. As shown in this figure, in the u-phase AC current Iu, the v-phase AC current Iv, and the w-phase AC current Iw, high-frequency currents are superimposed on sine waves that are 120 ° apart from each other. As will be described later, this high-frequency current is generated by a superimposed voltage that is superimposed in the motor control device 100a in order to estimate the magnetic pole position of the rotor of the motor 1.

  FIG. 2B is a current locus obtained by converting one cycle of the three-phase alternating current waveform shown in FIG. 2A into a stator coordinate system by a general three-phase two-phase conversion. The current trajectory in FIG. 2B can be considered separately into a circular orbit component corresponding to the sine wave component in FIG. 2A and a vibration component that vibrates at a high frequency with respect to the circular orbit component. . Hereinafter, this circular orbit component is referred to as a fundamental wave component, and the vibration component is referred to as a high frequency component.

The current at each moment in the current locus in FIG. 2B can be represented by a vector I from the origin. This current vector I is expressed by the following equation (1). In Expression (1), I1 is a fundamental wave current vector representing the fundamental wave component of the current, and Ih is a high frequency current vector representing the high frequency component of the current. That is, the three-phase alternating current flowing from the power converter 101a to the electric motor 1 can be expressed as the rotation of the fundamental current vector I1 plus the vibration of the high-frequency current vector Ih.
I = Il + Ih (1)

The three-phase AC voltage applied from the power conversion device 101a to the electric motor 1 is also expressed by the following equation (2) by converting it into the stator coordinate system, as in the three-phase AC current described above. It can be represented by a voltage vector V. In Equation (2), V1 is a fundamental voltage vector that represents a fundamental component of voltage, and Vh is a high-frequency voltage vector that represents a high-frequency component of voltage.
V = Vl + Vh (2)

  Next, the relationship between the fundamental wave current vector I1 and the fundamental wave voltage vector V1 and the relationship between the high frequency current vector Ih and the high frequency voltage vector Vh will be described with reference to the explanatory diagram of FIG.

  FIG. 3A schematically shows a fundamental wave current vector I1 at a certain moment in the current locus of FIG. 2B and a fundamental wave voltage vector V1 corresponding thereto. As shown in FIG. 3A, the phases of the fundamental wave current vector I1 and the fundamental wave voltage vector V1 with respect to the α phase (u phase) are defined as a fundamental wave current phase θi1 and a fundamental wave voltage phase θv1, respectively.

  Here, it is assumed that the electric motor 1 is a synchronous motor using a permanent magnet as a rotor. In this case, the angular velocity of the fundamental wave current phase θi1, that is, the rotational speed of the fundamental wave current vector I1, and the angular velocity of the fundamental wave voltage vector V1, that is, the rotational velocity of the fundamental wave voltage vector V1, are the rotational speed of the rotor of the motor 1. It almost coincides with ωr. In FIG. 3A, the respective norms of the fundamental wave current vector I1 and the fundamental wave voltage vector V1 are normalized so that different physical quantities of voltage and current are schematically represented on the same stator coordinate plane. doing.

  FIG. 3B schematically shows a high-frequency current vector Ih at a certain moment in the current locus of FIG. 2B and a corresponding high-frequency voltage vector Vh. As shown in FIG. 3B, the phases of the high-frequency current vector Ih and the high-frequency voltage vector Vh with respect to the α phase (u phase) are defined as a high-frequency current phase θih and a high-frequency voltage phase θvh, respectively.

  Here, it is assumed that the motor control device 100a is a voltage-type inverter that modulates and outputs a voltage command using a modulation signal such as PWM. In this case, the high-frequency current vector Ih and the high-frequency voltage vector Vh include a component due to the modulation signal in addition to the above-described component due to the superimposed voltage. However, it is assumed here that the component due to the superimposed voltage is dominant, and the component due to the modulation signal can be ignored. Further, it is assumed that the vibration frequency of the high-frequency current vector Ih and the high-frequency voltage vector Vh is sufficiently larger than the fundamental frequency that is a value near the rotation speed ωr. Under these assumptions, the high-frequency current vector Ih and the high-frequency voltage vector Vh observed on the stator coordinate plane are represented by the fundamental current vector I1 and the fundamental voltage vector V1, as shown in FIG. A linear trajectory is drawn around each end point (operation point). That is, the end point of the fundamental wave current vector I1 is defined as the start point of the high-frequency current vector Ih. Further, the end point of the fundamental wave voltage vector V1 is defined as the start point of the high-frequency voltage vector Vh.

  Next, the detection principle of the magnetic pole position of the electric motor 1 based on the saliency in the present invention will be described. The relationship between the voltage vector V and the current vector I in the electric motor 1 can be expressed by the following equation (3). In Expression (3), Φ represents the main magnetic flux vector of the electric motor 1, and r represents the resistance value of the electric motor 1.

... (3)

  If attention is paid only to the high frequency component in the expression (3), the voltage drop due to the resistance value r can be sufficiently ignored. Therefore, the fluctuation of the magnetic flux due to the high frequency voltage vector Vh can be approximated by the product of the local inductance matrix L and the high frequency current vector Ih. Therefore, Formula (3) can be transformed into the following Formula (4).

... (4)

  Here, when the structure of the rotor of the electric motor 1 is not symmetric with respect to the position direction, the inductance matrix L of Expression (4) has anisotropy corresponding to the position of the rotor. Therefore, when the electrical angle phase corresponding to the magnetic pole position of the rotor with respect to the α phase (u phase) is defined as the magnetic pole position θd, the inductance matrix L can be expressed as a function of the magnetic pole position θd. In many of the permanent magnet synchronous motors having the asymmetric rotor shape as described above, the inductance in the direction parallel to the magnetic flux by the magnet embedded in the rotor is minimized, and the inductance in the direction orthogonal to the direction is maximized. It has special characteristics. Such a characteristic is called saliency.

When the electric motor 1 has the saliency as described above, the high-frequency voltage phase θvh and the high-frequency current phase θih do not necessarily match as shown in FIG. Therefore, the difference between the high frequency current phase θih and the high frequency voltage phase θvh in this case is defined as a phase difference θivh. Further, the difference between the high frequency voltage phase θvh and the magnetic pole position θd is defined as a phase difference θvhd. That is, the phase differences θivh and θvhd are expressed by the following equations (5) and (6), respectively.
θivh = θih−θvh (5)
θvhd = θvh−θd (6)

  FIG. 4 is a schematic diagram of a graph showing the relationship between the phase difference θivh and the phase difference θvhd. It is known that when a graph showing these relationships is created with the phase difference θvhd as the horizontal axis and the phase difference θivh as the vertical axis, a graph close to a sine wave as shown in FIG. From the graph of FIG. 4A, it can be seen that θivh = 0 when θvhd = 0.

In the graph of FIG. 4A, the vicinity of the zero point can be approximated as a straight line. The slope of this straight line is defined as the magnetic pole position estimation sensitivity KΔθ (<0) based on the saliency. In the vicinity of the zero point, the following relationship (7) is established between the phase difference θivh and the phase difference θvhd using this sensitivity KΔθ.
θivh = KΔθ × θvhd (7)

  From equation (7), θvhd = 0 can be achieved by adjusting the phase of the superimposed voltage so that θivh = 0. Thereby, it can be seen from the equation (6) that the high-frequency voltage phase θvh and the magnetic pole position θd can always be matched. That is, the magnetic pole position θd can be known from the value of the high-frequency voltage phase θvh.

  Here, the relationship between the phase difference θivh and the phase difference θvhd as described above varies depending on the magnitude of the saliency of the electric motor 1. This is shown in FIG. FIG. 4B shows that the absolute value of the sensitivity KΔθ increases as the saliency increases, and the absolute value of the sensitivity KΔθ decreases as the saliency decreases.

  As described above, the saliency of the electric motor 1 is the anisotropy of inductance in the electric motor 1. When the output torque increases in the electric motor 1 and the amount of current increases, the current magnetic flux linked to the rotor gradually increases accordingly. At this time, the inductance of the rotor gradually decreases due to the magnetic saturation phenomenon. In particular, the phase in which the current is efficiently converted into torque (effective current direction) and the phase in which the inductance is maximized are substantially the same in the motor 1, and the inductance in the phase direction increases as the current increases. descend. As described above, when the inductance of the electric motor 1 is reduced, the saliency, that is, the anisotropy of the rotor inductance is also relatively lost. Therefore, when the torque is increased during the operation of the electric motor 1, the saliency deteriorates from a certain operating point, and the absolute value of the sensitivity KΔθ decreases. As a result, when the sensitivity KΔθ falls below a predetermined detection limit, the electric motor 1 cannot be controlled, and there is a risk that step-out occurs.

  Therefore, the motor drive system 110a according to the present invention estimates the sensitivity KΔθ in order to solve such a problem, and outputs a warning indicating insufficient sensitivity when the estimation result falls below a predetermined reference value. Thereby, the step-out in the high load region is prevented in advance, and the drive of the electric motor 1 is maintained safely. This point will be described in detail later.

  The above is the magnetic pole position detection principle based on the saliency in the present invention.

  Here, returning to FIG. 1, the operation of each component of the motor drive system 110a will be described.

  The current detection means 2 detects the instantaneous value of the three-phase alternating current flowing from the voltage conversion means 3 to the motor 1, that is, the current vector I described above, and outputs the detection result to the current extraction means 6. The current detection means 2 is realized by a current sensor using a Hall element, for example.

The voltage conversion means 3 converts a DC voltage from a DC power supply (not shown) into a three-phase AC voltage based on the voltage vector command V ^* generated by the motor control device 100a, and outputs it to the motor 1. At this time, the three-phase AC voltage output from the voltage conversion means 3 to the electric motor 1 is represented by the voltage vector V described above. The voltage conversion means 3 is realized by an inverter using, for example, a MOSFET (Metal-Oxide-Semiconductor Field-Effect Transistor) or an IGBT (Insulated Gate Bipolar Transistor) as a switching element.

  The electric motor 1 is a three-phase synchronous motor and operates with a three-phase AC voltage from the voltage conversion means 3. The electric motor 1 is a permanent magnet synchronous motor having the saliency as described above, and is configured such that a rotor in which a plurality of permanent magnets are incorporated rotates inside a stator. The details of the configuration of the electric motor 1 are not shown.

  The motor control device 100a includes a torque command generation unit 4, a current control unit 5, a current extraction unit 6, a superimposed voltage phase adjusting unit 7, a superimposed voltage phase shift unit 8, a voltage superimposing unit 9, a sensitivity calculating unit 10, and an alarm issuing unit 11. It has. The motor control device 100a includes a ROM (Read Only Memory), a RAM (Random Access Memory), a CPU (Central Processing Unit), a program, and the like. In other words, each of the above-described units included in the motor control device 100a is realized as a process executed by the CPU according to a program.

  Here, in order to explain the characteristics of the motor control device 100a in an easy-to-understand manner, the control method of the motor 1 when the estimation of the sensitivity KΔθ according to the present invention is not applied will be described first as a comparative example. FIG. 5 is a configuration diagram of an electric motor drive system 110b according to a comparative example. The electric motor drive system 110b includes the electric motor 1 common to FIG. 1 and a power conversion device 101b. The power conversion device 101b includes a voltage conversion unit 3 and a current detection unit 2 that are common to those in FIG. 1, and an electric motor control device 100b.

  The motor control device 100b includes torque command generation means 4, current control means 5, current extraction means 6, superposed voltage phase adjustment means 7 and voltage superposition means 9 as components common to the motor control apparatus 100a of FIG. Yes. On the other hand, the superimposed voltage phase shift means 8, the sensitivity calculation means 10 and the alarm issuing means 11 of FIG. 1 are not provided. That is, the motor control device 100a in FIG. 1 is characterized in that it includes a superimposed voltage phase shift means 8, a sensitivity calculation means 10, and an alarm issuing means 11 as compared with the motor control device 100b in FIG.

  Next, the operation of each component of the motor control device 100b in FIG. 5 will be described.

The torque command generation means 4 outputs a torque command τ ^* for the electric motor 1 to the current control means 5 based on a request from the host system.

  The current extraction unit 6 extracts the fundamental wave current vector I1 and the high frequency current vector Ih from the current vector I detected by the current detection unit 2. Then, the fundamental current vector I 1 is output to the current control unit 5, and the high-frequency current vector Ih is output to the superimposed voltage phase adjustment unit 7. A specific method for extracting the fundamental wave current vector I1 and the high-frequency current vector Ih from the current vector I in the current extraction means 6 will be described in detail later.

The superposed voltage phase adjusting unit 7 uses the high frequency current vector Ih from the current extracting unit 6 to detect the high frequency voltage phase command value θvh as a target value for the high frequency voltage phase θvh according to the magnetic pole position detection principle based on the saliency as described above. ^{* Is} determined. Specifically, the high-frequency current phase θih is obtained from the high-frequency current vector Ih, and based on this, the high-frequency voltage phase command value θvh ^* is output so that the high-frequency current phase θih matches the high-frequency voltage phase θvh. That is, the phase difference θivh between the high-frequency current phase θih and the high-frequency voltage phase θvh becomes 0, whereby θvhd = 0, so that the magnetic pole position θd matches the high-frequency voltage phase θvh ^*. To decide. Then, the determined high-frequency voltage phase command value θvh ^* is output to the current control means 5 and the voltage superimposing means 9. As a result, the high-frequency voltage phase command value θvh ^* is output to the current control means 5 as information for estimating the magnetic pole position θd. For the voltage superimposing means 9, as information for adjusting the phase for superimposing an alternating voltage as described later on the fundamental wave voltage vector command V 1 ^* from the current control means 5, a high-frequency voltage phase command value is used. θvh ^* is output.

The current control unit 5 calculates a fundamental wave current vector command I1 ^* as a target value for the fundamental wave current vector I1 based on the torque command τ ^* input from the torque command generation unit 4. Then, the fundamental wave voltage vector command V1 ^* as a target value for the fundamental wave voltage vector V1 is set so that the fundamental wave current vector I1 included in the current vector I flowing through the motor 1 coincides with the fundamental wave current vector command I1 ^*. Determine and output.

The current control means 5 estimates the magnetic pole position θd based on the high-frequency voltage phase command value θvh ^* from the superimposed voltage phase adjustment means 7 when calculating the fundamental wave current vector command I1 ^* . That is, the torque output from the electric motor 1 that is a synchronous motor is expressed as a function of the fundamental wave current vector I1 and the magnetic pole position θd. Therefore, in order to keep the torque constant, it is necessary to synchronize the fundamental wave current phase θi1 and the magnetic pole position θd to make the difference constant and to control the amplitude of the fundamental wave current vector I1. In order to do this, it is necessary to estimate the magnetic pole position θd. Here, since the high-frequency voltage phase command value θvh ^* is output from the superimposed voltage phase adjusting means 7 as described above, it can be regarded as substantially coincident with the magnetic pole position θd. Therefore, the magnetic pole position θd can be estimated from the high-frequency voltage phase command value θvh ^* , and the fundamental current vector command I1 ^* can be calculated.

The voltage superimposing means 9 adjusts and outputs the high frequency voltage vector command Vh ^* as a target value for the high frequency voltage vector Vh based on the high frequency voltage phase command value θvh ^* from the superimposed voltage phase adjusting means 7. Here, the norm waveform of the high-frequency voltage vector Vh is a rectangular wave that changes periodically, as will be described later. The frequency of this rectangular wave, that is, the vibration frequency of the high-frequency voltage vector Vh is sufficiently larger than the rotational speed ωr of the rotor as described above. The voltage superimposing means 9 determines and outputs a high-frequency voltage vector command Vh ^* so that an alternating voltage corresponding to such a rectangular wave is output with respect to the high-frequency voltage phase command value θvh ^* .

The fundamental voltage vector command V1 ^* from the current control unit 5 and the high frequency voltage vector command Vh ^* from the voltage superimposing unit 9 are added and output to the voltage conversion unit 3 as the voltage vector command V ^* . That is, in the voltage vector command V ^* output from the motor control device 100a to the voltage conversion means 3, the voltage superimposing means 9 performs an alternating voltage corresponding to the high-frequency voltage vector command Vh ^* with respect to the fundamental voltage vector command V1 ^* . Are superimposed.

  Next, the operations of the voltage superimposing means 9 and the current extracting means 6 that play a central role in detecting the magnetic pole position based on the saliency will be described in detail. As described below, the magnetic pole positions of the electric motor 1 can be detected by operating in cooperation with each other.

  As described above, the saliency of the electric motor 1 is a property of inductance in the electric motor 1. In order to obtain inductance information from equation (4), it is necessary to obtain a voltage vector Vh and a current vector Ih generated thereby. In order to realize this, in the motor control device 100b, the high-frequency voltage vector Vh is a rectangular wave, and the phase of the rectangular wave and the detection timing of the current vector I are synchronized.

  FIG. 6 is a diagram illustrating an example of the high-frequency voltage vector Vh and the high-frequency current vector Ih in the motor control device 100b of the comparative example. FIG. 6A shows a norm waveform example of the high-frequency voltage vector Vh and the high-frequency current vector Ih. In these norm waveforms, the displacement in the phase direction that is inverted by 180 ° from the high-frequency voltage phase θvh and the high-frequency current phase θih is shown as a negative value. FIGS. 6B and 6C show examples of the high-frequency voltage vector Vh and the high-frequency current vector Ih on the stator coordinate plane with reference to the stator of the electric motor 1 corresponding to FIG. Each is shown.

  In FIG. 6A, when the first period is divided into four periods (1) to (4), the norm of the high-frequency voltage vector Vh shows a positive value in the periods (1) and (2). ing. During these periods, the norm of the high-frequency current vector Ih increases at a rate of change corresponding to the local inductance. On the other hand, in the periods (3) and (4), the norm of the high-frequency voltage vector Vh shows a negative value. In these periods, the norm of the high-frequency current vector Ih decreases at a reduction rate corresponding to the local inductance.

Here, in the voltage superimposing means 9 and the current extracting means 6, the timing at which the high-frequency voltage vector command Vh ^* is changed in accordance with the periods (1) to (4) and the current detected by the current detecting means 2. The timing for sampling the vector I is synchronized. This is repeated at each cycle. That is, the current extraction unit 6 acquires the current vector I in accordance with the timing of each current vector detection point shown on the norm waveform of the high-frequency current vector Ih in FIG. In this way, the current extraction means 6 can alternately acquire the upper and lower peak values and the center value of the high-frequency current vector Ih as the current vector I.

  The current extraction unit 6 can extract the high-frequency current vector Ih and the fundamental current vector I1 from the current vector I by acquiring the current vector I as described above. That is, the high-frequency current vector Ih can be obtained by subtracting the current vector I detected last time from the current vector I detected this time. Further, by calculating the average of the current vector I within a predetermined period, the fundamental wave current vector I1 can be obtained by canceling the high frequency component from the current vector I. Thus, by extracting the high-frequency current vector Ih and the fundamental wave current vector I1 from the current vector I, they can be obtained separately from each other.

  As described above, the high-frequency voltage vector Vh is output as a rectangular wave, and the voltage superimposing means 9 and the current extracting means 6 are operated in cooperation, whereby the high-frequency current vector Ih and the fundamental current are detected from the detected current vector I. Vector I1 can be determined. As a result, the magnetic pole position of the electric motor 1 can be detected by the magnetic pole position detection principle based on the saliency as described above.

  The electric motor drive system 110b according to the comparative example has the configuration as described above.

  Next, the electric motor drive system 110a according to the first embodiment of the present invention shown in FIG. 1 will be described focusing on the difference from the electric motor drive system 110b of FIG. 5 according to the above comparative example. In the motor drive system 110a of FIG. 1, the motor control device 100a further includes a superimposed voltage phase shift means 8, a sensitivity calculation means 10, and an alarm issuing means 11 in addition to the components included in the motor control device 100b of FIG. Yes.

The superposed voltage phase shift means 8 receives the high frequency voltage phase command value θvh ^* from the superposed voltage phase adjustment means 7. The superimposed voltage phase shift means 8 adjusts the high frequency voltage phase command value θvh ^* by adding a predetermined phase shift amount. Then, the adjusted value is output to the voltage superimposing means 9 and the sensitivity calculating means 10 as the shifted high-frequency voltage phase command value θvh ^** .

  FIG. 7 is an explanatory diagram of the superimposed voltage phase shift means 8. FIG. 7A is a diagram showing an example of the internal configuration of the superimposed voltage phase shift means 8.

  As shown in FIG. 7A, the superimposed voltage phase shift unit 8 includes a superimposed pattern generation unit 71 and a signal selector 72. The signal selector 72 selects any one of the three shift amount candidate values θvh1, θvh2, and θvh3 and outputs it as a phase shift amount. The shift amount candidate value selected by the signal selector 72 is sequentially switched according to the superposition pattern signal input from the superposition pattern generation means 71.

  The superposition pattern generation means 71 outputs a superposition pattern signal for determining a shift amount candidate value to be selected to the signal selector 72. At this time, the superimposition pattern generation means 71 changes the superposition pattern signal at the sampling period when the current extraction means 6 samples the current vector I as described above, or at a positive multiple. Thereby, the phase shift amount output from the signal selector 72 is sequentially switched between θvh1, θvh2, and θvh3.

The phase shift amount output from the signal selector 72 is added to the high-frequency voltage phase command value θvh ^* input to the superimposed voltage phase shift means 8. Then, it is output from the superimposed voltage phase shift means 8 as the high frequency voltage phase command value θvh ^** after the shift.

FIG. 7B is a diagram showing an example of input / output signals of the superimposed voltage phase shift means 8 of FIG. Here, an example is shown in which θvh1 = 20 °, θvh2 = 0 °, and θvh3 = −20 °. As shown in FIG. 7B, the high-frequency voltage phase command value θvh ^* that is an input signal to the superimposed voltage phase shift means 8 increases in synchronization with the drive frequency of the electric motor 1. In contrast, the high-frequency voltage phase command value Shitavh ^** is the shifted is the output signal from the superimposed voltage phase shifting means 8, phase by respectively predetermined amounts in positive and negative directions is shifted.

  In the above example, the example in which any one of the three shift amount candidate values θvh1, θvh2, and θvh3 is selected as the phase shift amount has been described. Further, as a specific example of the phase shift amount, the example in which θvh1 = 20 °, θvh2 = 0 °, and θvh3 = −20 ° has been described. However, the phase shift amount applicable in the superimposed voltage phase shift means 8 is not limited to these examples. If it is a real number within a range of −45 ° to 45 °, the superimposed voltage phase shift means 8 sets at least two or more arbitrary values as shift amount candidate values, and selects one of them as a phase shift amount. Is possible.

Here, in the motor drive system 110b (motor control device 100b) of FIG. 5 according to the comparative example, the voltage superimposing means 9 has a high frequency based on the high frequency voltage phase command value θvh ^* from the superimposed voltage phase adjusting means 7 as described above. The voltage vector command Vh ^* was adjusted. On the other hand, in the electric motor drive system 110a (electric motor control device 100a) of FIG. 1 according to the present invention, the voltage superimposing means 9 is the high-frequency voltage phase after the shift output from the superposed voltage phase shifting means 8 as described above. Based on the command value θvh ^** , the high-frequency voltage vector command Vh ^* is adjusted.

FIG. 8 is a diagram illustrating an example of the high-frequency voltage vector Vh and the high-frequency current vector Ih in the motor control device 100a of the present invention. FIG. 8A shows a norm waveform example of the high-frequency voltage vector Vh and the high-frequency current vector Ih similar to FIG. 6A and an example of the shifted high-frequency voltage phase command value θvh ^** . In FIG. 8A, the high-frequency voltage phase command value θvh ^** after the shift corresponds to the example of FIG. 7B. FIGS. 8B and 8C show examples of the high-frequency voltage vector Vh and the high-frequency current vector Ih on the stator coordinate plane corresponding to FIG. 8A, respectively.

In FIG. 8 (a), when divided into eight periods of each waveform as shown in FIG. (1) to (8), (1), be θvh ^** = θvh ^* + θvh2 the period (2) , (3), it is a θvh ^** = θvh ^* + θvh1 in the period of (4). Also, (5), in the period (6) is ^{θvh ** = θvh * + θvh2,} (7), a θvh ^** = θvh ^* + θvh3 in the period (8). Here, as described above, θvh1 = 20 °, θvh2 = 0 °, and θvh3 = −20 °. That is, θvh ^** = θvh ^* in the periods (1), (2), (5), and (6).

As shown in FIGS. 8B and 8C, the periods (1), (2), (5), (6), the periods (3), (4), (7), In the period (8), the slopes of the high-frequency voltage vector Vh and the high-frequency current vector Ih are different from each other. That is, the gradients of these vectors change according to the change in the amount of phase shift superimposed on the high-frequency voltage phase command value θvh ^** after the shift. Note that the inclination of these vectors represents the phase of each vector. That is, the slope of the high frequency voltage vector Vh represents the high frequency voltage phase θvh, and the slope of the high frequency current vector Ih represents the high frequency current phase θih.

The superimposed voltage phase shift means 8 adjusts the high frequency voltage phase command value θvh ^* from the superimposed voltage phase adjustment means 7 by changing the phase shift amount as described above, and the shifted high frequency voltage phase command value θvh ^*. Output as ^* . Thereby, as shown in FIGS. 8B and 8C, two or more trajectories of the high-frequency voltage vector Vh and the high-frequency current vector Ih have different inclinations (that is, phases) on the stator coordinate plane. Draw a line segment. The superimposed voltage phase shift means 8 can adjust the high-frequency voltage phase command value θvh ^* in this way.

Here, the high-frequency voltage vector Vh and the high-frequency current vector Ih corresponding to the periods (3) and (4) are represented as a high-frequency voltage vector Vh1 and a high-frequency current vector Ih1, respectively. The high-frequency voltage vector Vh and the high-frequency current vector Ih corresponding to the periods (1), (2), (5), and (6) are represented as a high-frequency voltage vector Vh2 and a high-frequency current vector Ih2, respectively (7), The high-frequency voltage vector Vh and the high-frequency current vector Ih corresponding to the period (8) are represented as a high-frequency voltage vector Vh3 and a high-frequency current vector Ih3, respectively. In the motor control device 100a of the present invention, these vectors appear alternately as the amount of phase shift superimposed on the shifted high-frequency voltage phase command value θvh ^** periodically changes.

The sensitivity calculation means 10 obtains high-frequency current phases θih1, θih2, and θih3 for the high-frequency current vectors Ih1, Ih2, and Ih3 extracted by the current extraction means 6, respectively. Then, using each of these high-frequency current phases θih1, θih2, θih3 and the shifted high-frequency voltage phase command value θvh ^** outputted from the superimposed voltage phase shift means 8 at that time, a phase difference between them is obtained. θivh1, θivh2, and θivh3 are respectively calculated. Here, the shifted high-frequency voltage phase command value θvh ^** represents the phase of each of the high-frequency voltage vectors Vh1, Vh2, and Vh3. Therefore, the phase differences θivh1, θivh2, and θivh3 calculated as described above represent the phase differences between the high-frequency current vectors Ih1, Ih2, and Ih3 and the corresponding high-frequency voltage vectors Vh1, Vh2, and Vh3, respectively. Yes.

  After calculating the phase differences θivh1, θivh2, and θivh3 as described above, the sensitivity calculation means 10 subsequently uses the phase differences θivh1 and θivh3 to respond to the saliency of the electric motor 1 by the method described below. The estimated sensitivity KΔθ is estimated.

When the superimposed voltage phase adjusting means 7 is operating normally according to the magnetic pole position detection principle based on the saliency, the phase difference θvhd between the high-frequency voltage phase θvh and the magnetic pole position θd is 0 as described above. That is, the high-frequency voltage phase command value θvh ^* output from the superimposed voltage phase adjusting means 7 matches the magnetic pole position θd. At this time, the phase difference θivh1 represents the phase difference θivh between the high-frequency current phase θivh and the high-frequency voltage phase θvh when the high-frequency voltage phase θvh is shifted from the magnetic pole position θd by the phase shift amount θvh1. Similarly, the phase difference θivh3 represents the phase difference θivh between the high-frequency current phase θivh and the high-frequency voltage phase θvh when the high-frequency voltage phase θvh is shifted from the magnetic pole position θd by the phase shift amount θvh3.

  9 illustrates the relationship between the high-frequency voltage phase θvh and the phase differences θivh1 and θivh3 as described above with respect to the phase difference θivh and the phase difference θvhd shown in FIG. This is shown on the graph. In FIG. 9, the phase difference θivh1 corresponds to a point 91 on the graph. The coordinates of this point 91 can be represented by (θvh1, θivh1). The phase difference θivh3 corresponds to a point 92 on the graph. The coordinates of the point 92 can be expressed by (θvh3, θivh3).

Here, as described above, the sensitivity KΔθ is defined by the slope of a straight line near the zero point in the graph of FIG. That is, the sensitivity KΔθ can be calculated by the following equation (8) as the slope of the linear function connecting the points 91 and 92.
KΔθ = (θivh3-θivh1) / (θvh3-θvh1) (8)

  The sensitivity calculation means 10 estimates the sensitivity KΔθ according to the saliency of the electric motor 1 by calculating the sensitivity KΔθ as described above. Then, the calculated sensitivity KΔθ is output to the alarm issuing means 11.

In the above example, θvh2 = 0 °. Therefore, the superimposed voltage phase adjusting means 7 can adjust the high-frequency voltage phase command value θvh ^* according to the magnetic pole position detection principle based on the saliency as described above by using the phase difference θivh2 at this time. Thereby, the electric motor 1 can be driven by the same method as that described in the electric motor drive system 110b of FIG. 5 according to the comparative example.

  The alarm issuing means 11 compares the absolute value of the sensitivity KΔθ from the sensitivity calculation means 10 with a predetermined reference value. As a result, when the absolute value of the sensitivity KΔθ is lower than the reference value, it is determined that the sensitivity is insufficient, and an alarm indicating the fact is output. For example, by outputting an alarm sound or turning on an alarm lamp, the system user is warned and urged to take measures according to the situation. In addition, by outputting an alarm signal to the host system, the host system that receives the alarm signal may perform necessary control such as torque limitation.

  As described above, in the electric motor drive system 110a according to the present embodiment, the sensitivity calculation means 10 continuously calculates the sensitivity KΔθ while the electric motor 1 is being driven, and if the absolute value falls below a predetermined reference value, an alarm is issued. An alarm is output by the issuing means 11. As a result, it is possible to detect a lack of sensitivity associated with the nonlinear characteristics of the electric motor 1 due to magnetic saturation or the like, and to notify the system user or the host system. Therefore, it is possible for the system user and the host system to prevent step-out in advance by determining that it is difficult to continue the operation of the electric motor 1 due to insufficient sensitivity and taking appropriate measures. .

  In the above description, the sensitivity KΔθ is calculated from the phase differences θivh1 and θivh3. However, as is apparent from FIG. 9, other combinations of phase differences, for example, combinations of θivh1 and θivh2, or θivh2 and θivh3 Sensitivity KΔθ can also be obtained from a set or the like. That is, the sensitivity KΔθ can be calculated from any two phase differences. Therefore, the phase shift amount switched by the superimposed voltage phase shift means 8 can be an arbitrary value of two or more. However, it is preferable to select one or more values as the phase shift amount from positive and negative values within a range of −45 ° to 45 °. In any case, the same effect as described in the above embodiment can be obtained.

  Also, the phase shift amount switching pattern is not limited to the example shown in FIG. That is, any switching pattern can be used as long as two or more phase shift amounts are switched without deviation in the sampling period of the current vector I by the current extraction means 6 or a positive multiple of the sampling period. In any case, the same effect as described in the above embodiment can be obtained.

  According to 1st Embodiment of this invention demonstrated above, there can exist the following effect.

(1) The power conversion device 101a includes the voltage conversion means 3 and the motor control device 100a. In the motor control device 100a, the current extraction unit 6 extracts a high-frequency current vector Ih from the current vector I detected by the current detection unit 2. Based on the high-frequency current vector Ih, the superimposed voltage phase adjusting means 7 estimates the magnetic pole position θd of the rotor of the electric motor 1 and superimposes a phase for superimposing an alternating voltage that changes at a period different from the rotation period of the rotor. A high-frequency voltage phase command value θvh ^* for adjustment is output. Based on the high frequency voltage phase command value θvh ^* , the current control means 5 estimates the magnetic pole position θd and outputs a fundamental voltage vector command V1 ^* for controlling the current vector I flowing through the electric motor 1. Also, superimposed voltage phase shifting means 8 adjusts the high-frequency voltage phase command value Shitavh ^*, and outputs a high-frequency voltage phase command value after the shift θvh ^**. Based on the shifted high-frequency voltage phase command value θvh ^** , the voltage superimposing means 9 superimposes the alternating voltage on the fundamental voltage vector command V1 ^* and outputs it to the voltage converting means 3. The sensitivity calculation means 10 determines the magnetic pole position θd based on the high-frequency current vector Ih extracted by the current extraction means 6 and the shifted high-frequency voltage phase command value θvh ^** adjusted by the superimposed voltage phase shift means 8. A sensitivity KΔθ for estimation is calculated. Since it did in this way, the detection sensitivity of magnetic pole position (theta) d by the saliency of the electric motor 1 can be estimated.

(2) As shown in FIG. 8C, the superposed voltage phase shift means 8 has two or more loci whose high-frequency current vectors Ih have different phases on the stator coordinate plane with the stator of the electric motor 1 as a reference. The high frequency voltage phase command value θvh ^* is adjusted so as to draw a line segment. Specifically, the phase shift amount Shitavh1, by adding the θvh2 and high-frequency voltage phase command value periodically switched at least two or more of θvh3 θvh ^*, and to adjust the high-frequency voltage phase command value Shitavh ^* . As a result, the sensitivity calculation means 10 can calculate the high-frequency current phases θih1, θih2, and θih3 for the high-frequency current vectors Ih1, Ih2, and Ih3, respectively, and can calculate the sensitivity KΔθ based on these.

(3) The superimposed voltage phase shift means 8 periodically switches the phase shift amounts θvh1 and θvh3 and adds them to the high frequency voltage phase command value θvh ^* . At this time, the voltage superimposing means 9 superimposes the high-frequency voltage vector Vh1 on the fundamental voltage vector command V1 ^* according to the shifted high-frequency voltage phase command value θvh ^** added with the phase shift amount θvh1, and also performs phase shift. The high-frequency voltage vector Vh3 is superimposed on the fundamental voltage vector command V1 ^* in accordance with the shifted high-frequency voltage phase command value θvh ^** added with the amount θvh3. The sensitivity calculation means 10 calculates the phase difference θivh1 between the high-frequency current vector Ih1 and the high-frequency voltage vector Vh1, and the phase difference θivh3 between the high-frequency current vector Ih3 and the high-frequency voltage vector Vh3, and these calculated positions. Based on the phase difference, the sensitivity KΔθ is calculated using the above equation (8). That is, as shown in FIG. 9, by calculating the inclination between the coordinate point 91 corresponding to the set of the phase shift amount θvh1 and the phase difference θivh1, and the coordinate point 92 corresponding to the set of the phase shift amount θvh3 and the phase difference θivh3. Then, the sensitivity KΔθ is calculated. Since it did in this way, sensitivity Kdeltatheta can be correctly calculated by simple calculation.

(4) The superimposed voltage phase shift means 8 can select one or more values as a phase shift amount from positive and negative values within a range of −45 ° to 45 °. In this way, the sensitivity calculation means 10 can reliably calculate the sensitivity KΔθ.

(5) In the motor control device 100a, the alarm issuing means 11 outputs an alarm based on the sensitivity KΔθ calculated by the sensitivity calculating means 10. Since it did in this way, it warns that it is difficult to maintain the driving | operation of the electric motor 1 for lack of sensitivity to a system user and a high-order system, and prevents step-out beforehand by an appropriate measure. Can be urged.

(Second Embodiment)
FIG. 10 is an overall configuration diagram of an electric motor drive system 110c according to the second embodiment of the present invention. The electric motor drive system 110 c includes a power conversion device 101 c and the electric motor 1. The power conversion device 101 c includes an electric motor control device 100 c, a voltage conversion unit 3, and a current detection unit 2. The electric motor 1, the voltage conversion means 3, and the current detection means 2 are the same as the electric motor drive system 110a shown in FIG. Hereinafter, the difference between the motor control device 100c of the present embodiment and the motor control device 100a described in the first embodiment will be described.

  In FIG. 10, the motor control device 100c is different from the motor control device 100a shown in FIG. 1 in that a superimposed voltage amplitude adjusting means 12 is added.

The superposed voltage amplitude adjusting means 12 determines the amplitude of the high frequency voltage vector command Vh ^* output from the voltage superposing means 9 based on the high frequency current vector Ih from the current extracting means 6 and the sensitivity KΔθ from the sensitivity calculating means 10. A high frequency voltage amplitude command VhAmp ^* for adjustment is determined. Then, by outputting the determined high-frequency voltage amplitude command VhAmp ^* to the voltage superimposing means 9, the alternating voltage by the rectangular wave as described above is superimposed on the fundamental voltage vector command V1 ^* from the current control means 5. Adjust the amplitude. Thereby, the amplitude of the alternating voltage, which was a fixed value in the first embodiment, is adjusted according to the magnitude of the sensitivity KΔθ in the present embodiment.

Here, the principle of determining the high frequency voltage amplitude command VhAmp ^* by the superimposed voltage amplitude adjusting means 12 will be described. In order for the superimposed voltage phase adjusting means 7 to correctly calculate the high frequency current phase θih, the norm of the high frequency current vector Ih needs to be sufficiently larger than the detection error of the current vector I by the current detecting means 2. However, when the norm of the high-frequency current vector Ih is large, there are disadvantages such as torque pulsation increases in the electric motor 1 and electromagnetic noise increases. Therefore, it is desirable to limit the norm of the high-frequency current vector Ih to a minimum size within a range necessary for calculating the high-frequency current phase θih.

  By the way, as the absolute value of the sensitivity KΔθ becomes smaller, the high-frequency current phase θih and the high-frequency voltage phase θvh corresponding to a certain axis error (error of the high-frequency voltage phase θvh with respect to the magnetic pole position θd) are correspondingly obtained as shown in FIG. The absolute value of the phase difference θivh is small. Therefore, the influence of the norm of the high-frequency current vector Ih on the detection accuracy of the phase difference θivh is relatively large. Therefore, in such a case, the norm of the high-frequency current vector Ih needs to be relatively large in order to calculate the high-frequency current phase θih. On the other hand, when the absolute value of the sensitivity KΔθ is large, the absolute value of the phase difference θivh with respect to the axis error is correspondingly increased. Therefore, the magnitude of the norm of the high-frequency current vector Ih is detected by the phase difference θivh. The effect on accuracy is relatively small. Therefore, in this case, the norm of the high-frequency current vector Ih may be small. Thus, it can be seen that the minimum norm of the high-frequency current vector Ih necessary for calculating the high-frequency current phase θih is determined according to the sensitivity KΔθ.

Therefore, the superimposed voltage amplitude adjusting means 12 calculates a high frequency current amplitude command IhAmp ^* as a command value for the norm of the high frequency current vector Ih using the following equation (9) based on the sensitivity KΔθ.

... (9)

  In the above formula (9), K is an arbitrary constant. The value of the constant K may be changed according to the situation. For example, when it is desired to accurately detect the magnetic pole position θd by increasing the detection accuracy of the phase difference θivh, it is important to increase K or to suppress the torque pulsation and electromagnetic noise of the motor 1 at a higher frequency than the accuracy of the magnetic pole position θd. If desired, K can be reduced.

Further, the superimposed voltage amplitude adjusting unit 12 calculates a high frequency current amplitude IhAmp representing the norm based on the high frequency current vector Ih from the current extracting unit 6. Then, the high-frequency voltage amplitude command VhAmp ^* is determined so that the high-frequency current amplitude IhAmp matches the high-frequency current amplitude command IhAmp ^*, and is output to the voltage superimposing means 9.

According to the second embodiment of the present invention described above, in the motor control device 100c, the superimposed voltage amplitude adjusting means 12 is controlled by the voltage superimposing means 9 based on the sensitivity KΔθ calculated by the sensitivity calculating means 10. The amplitude of the alternating voltage superimposed on the vector command V1 ^* is adjusted. Since it did in this way, the alternating voltage superimposed on fundamental wave voltage vector instruction | command V1 ^* can be optimized dynamically according to sensitivity K (DELTA) (theta). Thus, for example, under operating conditions where the absolute value of the sensitivity KΔθ is high, the amount of alternating voltage superimposed is reduced to suppress noise, and conversely, under the operating conditions where the absolute value of the sensitivity KΔθ is small, the alternating voltage superimposed amount is increased to increase the necessary position. Accuracy can be ensured.

(Third embodiment)
FIG. 11 is an overall configuration diagram of an electric motor drive system 110d according to the third embodiment of the present invention. The electric motor drive system 110d includes a power conversion device 101d and the electric motor 1. The power conversion device 101d includes an electric motor control device 100d, a voltage conversion unit 3, and a current detection unit 2. The electric motor 1, the voltage conversion means 3, and the current detection means 2 are the same as the electric motor drive system 110a shown in FIG. Hereinafter, the difference between the motor control device 100d of this embodiment and the motor control device 100a described in the first embodiment will be described.

  In FIG. 11, the motor control device 100 d is different from the motor control device 100 a shown in FIG. 1 in that the alarm control means 11 is not provided and the sensitivity KΔθ from the sensitivity calculation means 10 is input to the torque command generation means 4. It is a point that has been.

In the present embodiment, the torque command generation means 4 adjusts the torque command τ ^* for the electric motor 1 based on the sensitivity KΔθ from the sensitivity calculation means 10. Thus, when the current vector I flowing through the electric motor 1 becomes too large and the sensitivity KΔθ decreases, the fundamental wave voltage vector command V1 ^* output from the current control unit 5 to the voltage conversion unit 3 is adjusted to adjust the current vector I. To prevent a decrease in sensitivity KΔθ. The principle will be described below.

  In general, the decrease in the absolute value of the sensitivity KΔθ occurs when a magnetic saturation phenomenon occurs due to an increase in the amount of current in the electric motor 1 as described above, and thereby the saliency deteriorates. Therefore, when the sensitivity KΔθ is deteriorated, the absolute value of the fundamental wave current vector I1 is not increased any more, so that the sensitivity KΔθ can be prevented from being lost and the subsequent step-out can be prevented from occurring. .

Therefore, in the present embodiment, when the absolute value of the sensitivity KΔθ falls below a predetermined value, the torque command generating means 4 restricts the absolute value of the torque command τ ^{* from} increasing further. Alternatively, the absolute value of the torque command τ ^* is decreased. The current control means 5 that has received the torque command τ ^* suppressed in this way outputs a fundamental voltage vector command V1 ^* corresponding to the torque command τ ^* after the suppression. As a result, the decrease in the sensitivity KΔθ is suppressed and the operation of the electric motor 1 can be maintained.

According to the third embodiment of the present invention described above, in the motor control device 100d, the current control unit 5 is controlled by the torque command generation unit 4 based on the sensitivity KΔθ calculated by the sensitivity calculation unit 10. In response to τ ^* , the fundamental voltage vector command V1 ^* is adjusted and output so as to limit the current vector I flowing through the electric motor 1. Since it did in this way, it can prevent that sensitivity K (DELTA) (theta) reduces more than needed, and can prevent the out-of-step of the electric motor 1 beforehand.

In the third embodiment described above, the fundamental voltage vector output from the current controller 5 is suppressed by the torque command generator 4 suppressing the torque command τ ^* based on the sensitivity KΔθ from the sensitivity calculator 10. The command V1 ^* is adjusted to suppress the decrease in sensitivity KΔθ. However, such an operation may be performed only by the current control means 5. For example, in the current control means 5, the absolute value of the fundamental wave vector I1 when sensitivity is lowered is set in advance. Based on the sensitivity KΔθ from the sensitivity calculation means 10, it is determined whether or not the absolute value of the sensitivity KΔθ is below a predetermined value. If the absolute value is below the predetermined value, the torque command τ ^* from the torque command generation means 4 is determined ^. Regardless, the fundamental voltage vector command V1 ^* is determined based on the preset absolute value of the fundamental current vector I1 when the sensitivity is lowered. Even if it does in this way, the effect similar to the above can be acquired.

(Fourth embodiment)
FIG. 12 is an overall configuration diagram of an electric motor drive system 110e according to the fourth embodiment of the present invention. The electric motor drive system 110e includes a power conversion device 101e and the electric motor 1. The power conversion device 101e includes an electric motor control device 100e, a voltage conversion unit 3, and a current detection unit 2. The electric motor 1, the voltage conversion means 3, and the current detection means 2 are the same as the electric motor drive system 110a shown in FIG. Hereinafter, the difference between the motor control device 100e of the present embodiment and the motor control device 100a described in the first embodiment will be described.

  In FIG. 12, the motor control device 100e is different from the motor control device 100a shown in FIG. 1 in that a reactive current command generating means 13 is added.

The reactive current command generation means 13 compares the absolute value of the sensitivity KΔθ from the sensitivity calculation means 10 with a predetermined reference value. As a result, when the absolute value of the sensitivity KΔθ is lower than the reference value, a predetermined signal is output to the current control unit 5. Upon receiving this signal, the current control means 5 adjusts and outputs the value of the fundamental voltage vector command V1 ^* so that the reactive current in the current vector I increases. Thereby, when the absolute value of the sensitivity KΔθ is lowered, the reactive current is increased in the current vector I. The reactive current is a current that does not contribute to the generation of torque in the electric motor 1 and is a loss during operation.

  Here, the reason why the reactive current is increased in the current vector I when the absolute value of the sensitivity KΔθ decreases will be described. The decrease in the absolute value of the sensitivity KΔθ occurs when the magnetic flux in the phase direction of the effective current is saturated in the electric motor 1. Therefore, in such a case, the reactive current that hardly occurs during normal operation is intentionally increased, thereby further increasing the magnetic flux saturation in the phase direction of the reactive current. As a result, anisotropy of inductance is relatively generated, and the sensitivity KΔθ can be recovered.

  Therefore, in the present embodiment, the reactive current command generation means 13 compares the absolute value of the sensitivity KΔθ with a predetermined reference value, and outputs a predetermined signal when the absolute value of the sensitivity KΔθ falls below the reference value. In response to this signal, the current control means 5 is operated so as to increase the reactive current command until the sensitivity KΔθ maintains the reference value. As a result, it is possible to output a higher torque from the electric motor 1 than in the third embodiment while preventing step-out. However, in this case, the operating efficiency of the electric motor 1 deteriorates as the reactive current increases.

According to the fourth embodiment of the present invention described above, in the motor control device 100e, the current control means 5 is a signal output from the reactive current command generation means 13 based on the sensitivity KΔθ calculated by the sensitivity calculation means 10. Accordingly, the fundamental voltage vector command V1 ^* is adjusted and output so as to increase the reactive current flowing through the electric motor 1. Since it did in this way, it can prevent that sensitivity K (DELTA) (theta) reduces more than needed, and can prevent the out-of-step of the electric motor 1 beforehand. Furthermore, while performing this, a higher torque can be output from the electric motor 1 than the electric motor control device 100d described in the third embodiment.

(Fifth embodiment)
FIG. 13 is an overall configuration diagram of a conveyor 130 according to the fifth embodiment of the present invention. The transport machine 130 includes the electric motor 1, a power conversion device 131, a power voltage mechanism 132, and a transport unit 133. As the power conversion device 131, any of the power conversion devices 101a, 101c, 101d, and 101e described in the first to fourth embodiments can be used. The electric motor 1 is the same as that described in the first to fourth embodiments.

  The power conversion device 131 controls the operation of the electric motor 1 by the method described in the first to fourth embodiments. The driving force generated by the electric motor 1 is transmitted to the transport unit 133 via the power transmission mechanism 132. The transport unit 133 operates using this driving force to transport an installed baggage or the like from a predetermined position to another predetermined position.

  As described above, the present invention can also be applied to a conveyor.

(Sixth embodiment)
FIG. 14 is an overall configuration diagram of an elevating device 140 according to a sixth embodiment of the present invention. The elevating device 140 includes the electric motor 1, the power conversion device 141, a winding mechanism 142, a wire 143, and an elevating unit 144. Note that any of the power conversion devices 101a, 101c, 101d, and 101e described in the first to fourth embodiments can be used as the power conversion device 141. The electric motor 1 is the same as that described in the first to fourth embodiments.

  The power conversion device 141 controls the operation of the electric motor 1 by the method described in the first to fourth embodiments. The hoisting mechanism 142 moves the elevating unit 144 up and down by hoisting and lowering the wire 143 connected to the elevating unit 144 using the driving force generated by the electric motor 1.

  As described above, the present invention can also be applied to a lifting device.

(Modification)
In each of the embodiments described above, if the axis error, that is, the error of the high-frequency voltage phase θvh with respect to the magnetic pole position θd is large due to a transient phenomenon, it is difficult to calculate the sensitivity KΔθ correctly. Therefore, in such a case, the calculation of the sensitivity KΔθ by the sensitivity calculation means 10 may be stopped. This point will be described in detail below with reference to FIG.

  FIG. 15 is a diagram for explaining the principle of sensitivity detection error due to a transient phenomenon. This figure shows the relationship between the high-frequency voltage phase θvh and the phase difference θivh when the axis error is caused by a transient phenomenon on the graph of the phase difference θivh and the phase difference θvhd as shown in FIGS. Is.

In the same manner as described in the first embodiment, when the superimposed voltage signal is changed by sequentially switching the phase shift amounts θvh1, θvh2, and θvh3 in the superimposed voltage phase shift means 8, θvh2 = 0 ° as described above. The superimposed voltage phase adjusting means 7 uses the phase difference θivh2 between the high-frequency current vector Ih2 and the high-frequency voltage vector Vh2 at this time to adjust the high-frequency voltage phase command value θvh ^* according to the magnetic pole position detection principle based on the saliency ^. To do.

  Here, as shown in FIG. 15, it is assumed that the phase difference θvhd between the high-frequency voltage phase θvh and the magnetic pole position θd is not 0 due to an axis error caused by a transient phenomenon. In this case, the phase difference θivh1 with respect to the phase shift amount θvh1 corresponds to a point 151 on the graph, and the phase difference θivh3 with respect to the phase shift amount θvh3 corresponds to a point 152 on the graph. Therefore, as described in the first embodiment, when the sensitivity KΔθ is obtained from the slope of the linear function connecting these points, a value different from the actual value is calculated. That is, when a large axis error due to a transient phenomenon occurs, the phase differences θivh1 and θivh3 detected with respect to the phase shift amounts θvh1 and θivh3 deviate from the linear approximation region near the origin. Therefore, it is theoretically difficult to calculate the correct sensitivity KΔθ.

  Therefore, the sensitivity calculation means 10 obtains the phase difference θivh2 between the high-frequency current vector Ih2 and the high-frequency voltage vector Vh2 when the phase shift amount θvh2 is 0, and compares the absolute value with a predetermined reference value. Here, if no axis error due to a transient phenomenon occurs, the phase difference θivh2 should be almost zero. Therefore, when the absolute value of the phase difference θivh2 exceeds the reference value, an axis error due to a transient phenomenon has occurred, and the adjustment of the superimposed phase of the alternating voltage by the superimposed voltage phase adjusting means 7 has not converged. It can be judged. Therefore, in this case, since the sensitivity KΔθ cannot be calculated correctly, it is preferable to stop the output by stopping the calculation of the sensitivity KΔθ by the sensitivity calculation means 10.

  According to the modification described above, it is possible to suppress the calculation error of the sensitivity KΔθ due to the transient phenomenon.

  Each embodiment and modification described above are merely examples, and the present invention is not limited to these contents as long as the features of the invention are not impaired.

DESCRIPTION OF SYMBOLS 1 Electric motor 2 Current detection means 3 Voltage conversion means 4 Torque command generation means 5 Current control means 6 Current extraction means 7 Superposed voltage phase adjustment means 8 Superposed voltage phase shift means 9 Voltage superposition means 10 Sensitivity calculation means 11 Alarm issuing means 12 Superposed voltage Amplitude adjusting means 13 Reactive current command generating means 71 Superposed pattern generating means 72 Signal selectors 100a, 100b, 100c, 100d, 100e Electric motor control devices 101a, 101b, 101c, 101d, 101e Power converters 110a, 110b, 110c, 110d, 110e Electric motor drive system 130 Conveyor 140 Lifting device

Claims (12)

A voltage conversion means for converting a DC voltage into an AC voltage and outputting it to an AC motor having saliency;
Current detecting means for detecting a current flowing in the AC motor;
Current extraction means for extracting a high-frequency current from the current detected by the current detection means;
In order to estimate the magnetic pole position of the rotor of the AC motor based on the high-frequency current extracted by the current extraction means, and to adjust the phase for superimposing the alternating voltage that changes at a period different from the rotation period of the rotor Superimposed voltage phase adjusting means for outputting a high frequency voltage phase command value of
Current control means for estimating the magnetic pole position based on the high-frequency voltage phase command value output by the superimposed voltage phase adjustment means and outputting a fundamental voltage command for controlling the current flowing through the AC motor;
Superimposed voltage phase shift means for adjusting the high-frequency voltage phase command value so that the locus of the high-frequency current draws two or more line segments having different phases on a stator coordinate plane with respect to the stator of the AC motor. When,
Based on the high-frequency voltage phase command value adjusted by the superimposed voltage phase shift unit, a voltage superimposing unit that superimposes the alternating voltage on the fundamental voltage command and outputs it to the voltage conversion unit;
Sensitivity calculation means for calculating the sensitivity to the estimation of the magnetic pole position based on the high frequency current extracted by the current extraction means and the high frequency voltage phase command value adjusted by the superimposed voltage phase shift means. A power conversion device. A voltage conversion means for converting a DC voltage into an AC voltage and outputting it to an AC motor having saliency;
Current detecting means for detecting a current flowing in the AC motor;
Current extraction means for extracting a high-frequency current from the current detected by the current detection means;
In order to estimate the magnetic pole position of the rotor of the AC motor based on the high-frequency current extracted by the current extraction means, and to adjust the phase for superimposing the alternating voltage that changes at a period different from the rotation period of the rotor Superimposed voltage phase adjusting means for outputting a high frequency voltage phase command value of
Current control means for estimating the magnetic pole position based on the high-frequency voltage phase command value output by the superimposed voltage phase adjustment means and outputting a fundamental voltage command for controlling the current flowing through the AC motor;
By adding to the high-frequency voltage phase command value 2 or more predetermined phase shift amount periodically switched even without low, the superimposed voltage phase shifting means for adjusting the high-frequency voltage phase command value,
Based on the high-frequency voltage phase command value adjusted by the superimposed voltage phase shift unit, a voltage superimposing unit that superimposes the alternating voltage on the fundamental voltage command and outputs it to the voltage conversion unit;
Sensitivity calculation means for calculating the sensitivity to the estimation of the magnetic pole position based on the high frequency current extracted by the current extraction means and the high frequency voltage phase command value adjusted by the superimposed voltage phase shift means. A power conversion device. The power conversion device according to claim 2 ,
The superimposed voltage phase shift means periodically switches the first phase shift amount and the second phase shift amount as the phase shift amount, and adds the phase shift amount to the high frequency voltage phase command value,
The voltage superimposing unit superimposes the first high frequency voltage as the alternating voltage on the fundamental voltage command according to the high frequency voltage phase command value to which the first phase shift amount is added, and the second voltage voltage command. In accordance with the high-frequency voltage phase command value to which the phase shift amount is added, the second high-frequency voltage as the alternating voltage is superimposed on the fundamental voltage command,
The sensitivity calculation means calculates a first phase difference between the high-frequency current and the first high-frequency voltage, and a second phase difference between the high-frequency current and the second high-frequency voltage. A power conversion device that calculates the sensitivity based on the calculated first phase difference and second phase difference. The power conversion device according to claim 3 ,
The sensitivity calculation unit is configured to respond to a first coordinate point corresponding to the set of the first phase shift amount and the first phase difference, and to a set of the second phase shift amount and the second phase difference. A power conversion device, wherein the sensitivity is calculated by calculating an inclination with respect to the second coordinate point. The power conversion device according to claim 2 ,
The superposed voltage phase shift means selects one or more values from positive and negative values within a range of −45 ° to 45 °, respectively, as the phase shift amount. In the power converter according to any one of claims 1 to 5 ,
The power conversion device further comprising alarm issuing means for outputting an alarm based on the sensitivity calculated by the sensitivity calculating means. In the power converter according to any one of claims 1 to 6 ,
The power conversion device further comprising: a superimposed voltage amplitude adjusting unit that adjusts an amplitude of the alternating voltage superimposed on the fundamental voltage command by the voltage superimposing unit based on the sensitivity calculated by the sensitivity calculating unit. . In the power converter device according to any one of claims 1 to 7 ,
The power control device, wherein the current control unit adjusts and outputs the fundamental voltage command so as to limit a current flowing through the AC motor based on the sensitivity calculated by the sensitivity calculation unit. In the power converter according to any one of claims 1 to 8 ,
The current control means adjusts and outputs the fundamental voltage command so as to increase the reactive current flowing through the AC motor based on the sensitivity calculated by the sensitivity calculation means. . The power conversion device according to any one of claims 1 to 9 ,
An electric motor drive system comprising the AC electric motor. The power conversion device according to any one of claims 1 to 9 ,
The AC motor;
A transport unit that operates using a driving force generated by the AC motor. The power conversion device according to any one of claims 1 to 9 ,
The AC motor;
Elevating part;
An elevating apparatus comprising: a hoisting mechanism that moves the elevating part up and down using a driving force generated by the AC motor.