JP5488044B2 - Torque ripple suppression control device and control method for rotating electrical machine - Google Patents

Description

  The present invention relates to a control device and a control method for automatically suppressing torque ripple (torque pulsation) of a rotary electric machine in a torque control device of a rotary electric machine such as a motor, and more particularly to non-interference control of compensation current for torque ripple suppression. About.

  In principle, the motor generates torque ripple (torque pulsation), causing various problems such as vibration / noise, adverse effects on riding comfort, and electrical / mechanical resonance. In particular, an embedded magnet PM motor (IPMSM), which has become popular in recent years, generates a combination of cogging torque ripple and reluctance torque ripple. As a countermeasure, various methods for superimposing a compensation current for canceling the torque ripple on the torque command have been studied.

  However, for example, in the method of performing feedforward compensation using a mathematical analysis model, there is a concern about the influence of analysis errors. Further, in the method of storing the feedback learning control result at the steady operating point and performing feedforward compensation, it takes time to appropriately adjust the control parameter for each operating point, and online compensation becomes difficult. Further, the method for reducing the current ripple is not always optimally suppressed from the viewpoint of torque ripple. In addition, torque ripple observer compensation methods have been studied, but the characteristics during variable speed operation and on-line feedback suppression verification have been insufficient.

  The inventors of the present application have already proposed a feedback suppression control method using an axial torque meter for the purpose of highly accurately suppressing torque ripple, which is a cause of electrical / mechanical resonance, in response to the above problems (for example, Non-Patent Document 1). reference). This control method focuses on the periodicity of torque ripple and constructs a control system that compensates for each pulsation frequency component, and also has a function that automatically adjusts parameters to respond quickly to changes in the operating state using the system identification result. The details of this control method will be described below.

(1) Basic Configuration of Torque Ripple Suppression Control Device FIG. 4 is a basic configuration diagram of a conventional torque ripple suppression control device. The figure is applied to a system in which a load is torque-excited by a motor.

A motor 1 that is a torque ripple generation source and some load device 2 are coupled by a shaft 3, and its shaft torque is measured by a torque meter 4 and input to a torque ripple suppression device 5. Moreover, the rotor position (phase) information of a motor is input using rotational position sensors 6, such as a rotary encoder. The torque ripple suppression device 5 includes torque pulsation suppression means, and gives the inverter 7 a command value obtained by adding a torque pulsation compensation current to the current command value generated based on the torque command value (or speed command value). In the example of FIG. 4, in consideration of the current vector control by the inverter 7, the d-axis and q-axis current command values id ^* and iq ^* on the rotation coordinates (orthogonal dq axes) synchronized with the rotation of the motor are given. Yes.

It is known that torque ripple (torque pulsation) periodically occurs according to the rotor position due to the structure of the motor. Therefore, means for extracting a torque ripple frequency component in synchronism with the motor rotation is used to convert the torque ripple of arbitrary order n into cosine and sine coefficients T _An and T _Bn [Nm]. The exact measurement means for torque ripple frequency components includes Fourier transform, but if importance is placed on the ease of operation, a low-pass filter can be passed through a single-phase harmonic rotation coordinate system based on the rotation phase θ [rad]. A torque ripple frequency component can be extracted.

The torque ripple suppression device 5 performs torque ripple suppression control using the cosine and sine coefficients T _An and T _Bn described above, and the cosine / sine coefficients I _An and I _Bn [ampere] of the compensation current iqc ^* [ampere] of an arbitrary frequency component. Is generated. Conversion to the compensation current iqc ^* is obtained by the following equation (2) using the same rotation phase θ as that at the time of conversion, and this compensation current is superposed on the q-axis current command value and normal vector control is performed.

In FIG. 5, the cosine and sine coefficients T _An and T _Bn of the torque ripple are obtained from the shaft torque T _det and the rotational phase θ, and the compensation current cosine coefficient I _An and the sine coefficient I _Bn are obtained from these and the rotational speed ω. It is a control block diagram of the torque ripple suppression apparatus which calculates | requires the compensation current of torque periodic external suppression. The symbols in the figure have the following meanings.

T ^* : Torque command value, T _det : Torque detection value, T _An : n-th order torque pulsation extraction component (cosine coefficient), T _Bn : n-th order torque pulsation extraction component (sine coefficient), ω: Rotation speed detection value, θ : Rotation phase detection value, iqc ^* : torque ripple compensation current, id: d-axis current detection value, id ^* : d-axis current command value, iq: q-axis current detection value, iq ^* : q-axis current command value, iu. iv. iw: u, v, w-phase current, iqo ^* : q-axis current command value (before compensation current superposition), I _An : n-order compensation current cosine coefficient, l _Bn : n-order compensation current sine coefficient, abz: rotation sensor signal . Note that the subscript n is an nth-order component torque ripple.

In FIG. 5, the command value conversion unit 11 converts the torque command value T ^* into d-axis and q-axis current command values Id ^* and Iqo ^* in the rotation dq coordinate system in vector control. In general, conversion formulas and tables that realize maximum torque / current control are used. Current vector control unit 12 suppresses the torque ripple those superimposed torque pulsation compensation current iqc ^* the q-axis current command value Iqo ^* as q-axis current command value iq ^*. In the example of FIG. 5, the compensation current i _qc is superimposed on the q-axis current command value, but it may be applied to the d-axis current or both the d-axis and the q-axis. Alternatively, in a system where interference of dq axis current does not become a problem, a torque pulsation compensation signal may be directly superimposed on the torque command value.

  The current vector control unit 12 drives the load device 14 by performing a current vector control operation on a general orthogonal rotating coordinate system d-axis q-axis and driving the motor (IPMSM) 13 by vector control. The coordinate conversion unit 15 converts the three-phase alternating currents iu, iv, iw and the motor rotation phase θ detected by the current sensor 16 into currents id, iq in a dq axis orthogonal rotation coordinate system synchronized with the motor rotation coordinates. The rotation phase / speed detection unit 17 converts the rotation sensor signal abz of the rotation position sensor 18 such as an encoder into information on the rotation speed ω and the rotation phase θ.

The torque pulsation frequency component extraction unit 19 extracts torque periodic disturbance for each pulsation frequency component from the shaft torque detection value T _det detected by the shaft torque meter 20 and the rotation phase θ. A typical extraction means is Fourier transform. The method of extracting the pulsation component is arbitrary, but the emphasis is placed on ease of calculation, and the shaft torque detection value T _det [Nm] is multiplied by the nth-order cosine wave and sine wave based on the rotational phase θ, and each is passed through a low-frequency range. By applying the filter, the approximate Fourier transform shown in the equations (3) to (5) is performed. This is called torque ripple synchronous coordinate transformation.

£: Laplace transform, G _F : Pulsation extraction filter, ω _f : Pulsation extraction low-pass filter cut-off frequency [rad / s], s: Laplace operator The torque ripple suppression control unit 21 performs extraction / expression using equations (3) to (5) _Torque ripple suppression control is performed with each order component using the converted T _An and T _Bn, and the cosine coefficient I _An ^* [A] and the sine of the n-th order frequency component i _qcn ^* [A] of the compensation current i _qc ^* [A]. A coefficient I _Bn ^* [A] is generated. The conversion to the n-th order compensation current i _qcn ^* is calculated using the same rotation phase θ as in the torque ripple synchronous coordinate conversion, as in the equation (2).

The torque command T ^* [Nm] is converted into dq-axis current command values i _d ^* and i _q0 ^* that realize the maximum torque / current control, and the combined value i _qc ^* of each compensation current generated by the equation (2) ^. _Is superposed on i _q0 ^* to perform vector control. Basically, the contents of calculation processing performed by the torque ripple suppression control device are extraction of shaft torque pulsation component, torque ripple suppression, and compensation current signal generation, and other processing is performed by a general inverter.

  As a representative form of the torque ripple suppression control unit 21, a periodic disturbance observer compensation method or a compensation current Fourier coefficient learning control method can be used.

The compensation current generator 22 generates a compensation current command value iqc ^* by the above equation (2) and superimposes it on the q-axis current command value.

(2) System Identification The system configuration shown in FIG. 4 is a multi-inertia torsional resonance system due to the moment of inertia of the PM motor 1, the load device 2, the torque meter 4, the coupling 3, and the like. When the detected value of the shaft torque is fed back, since there are a plurality of resonance / anti-resonance frequencies, the suppression control parameter must be appropriately determined according to the operation state. If the learning time of the control parameter is long, there is a risk of increasing the electric / mechanical resonance phenomenon, so a quick automatic adjustment function is necessary.

Therefore, in Non-Patent Document 1, in order to derive a variable nominal control parameter adapted to the rotational speed change, the system transfer characteristic from the output value to the input value of the torque ripple suppressing device 5 in FIG. 4, that is, the compensation current command in FIG. The frequency transfer characteristic from the value iqc ^* to the detection value T _det of the shaft torque detector 4 is identified. The system identification method is arbitrary, but the shaft torque detection value T _det when a Gaussian noise signal is given to iqc ^* in an open loop is measured for 20 seconds at a calculation period of 100 μs, and the frequency is calculated from the ratio of input and output power spectral densities. The results of non-parametric estimation of the transfer function are shown in FIG. 6 (actual machine characteristics including mechanical system, inverter current response, torque meter response, dead time, etc.). In addition, the result of parametric identification in the case of approximating a four-inertia system from the tendency of frequency transfer characteristics is also shown. There are various optimization methods for approximation, but errors in amplitude characteristics up to 1 kHz in the frequency domain were evaluated, and constrained nonlinear minimization (sequential quadratic programming) was performed.

In FIG. 5, in order to construct a control system with coordinates synchronized with the torque ripple frequency, only an arbitrary frequency transfer characteristic is extracted from the system identification result of FIG. In the steady state, the amplitude / phase transfer characteristic of the system synchronized with the torque ripple frequency can be expressed by a one-dimensional complex vector. Therefore, the system characteristic _Psys in the control system of FIG. 5 is defined as the following equation (6).

P _Am : Real part of the system characteristic, P _Bm : Imaginary part of the system characteristic, m: Frequency element number of the system identification table For example, the system characteristic from 1 to 1000 [Hz] is expressed by equation (3) for each IHz. In this case, the system identification table can be constructed from the elements of 1000 complex vectors. Only one complex vector is always used in the control system, and P _Am and P _Bm are instantaneously read from the identification table according to changes in the rotational speed (torque ripple frequency change), and converted into complex vectors by linear interpolation. The identified results are applied to suppression control. Since the axes of the real part and the imaginary part are defined with reference to the rotation phase, the cosine coefficient in equation (1) corresponds to the real part component and the sine coefficient corresponds to the imaginary part component.

(3) Compensation current Fourier coefficient learning control method This is a torque ripple suppression control method described as method 1 in Non-Patent Document 1 described above, which is obtained as a Fourier coefficient of a torque ripple frequency component, and from this, the compensation current is calculated by the above equation (2). Find iqc ^* . In this control method, a system transfer function of a frequency component synchronized with the torque ripple frequency is expressed by a one-dimensional complex vector, and a real part and an imaginary part of the torque ripple of an arbitrary frequency component are extracted by Fourier transform or the like. By applying the cosine and sine Fourier coefficients to the real and imaginary parts of the complex vector, a feedback suppression control system is constructed.

  The compensation current coefficient is obtained by an IP (proportional / integral) learning control method. The proportional / integral gain is determined by the model matching method so that the closed loop characteristic of the IP suppression control system matches the pole arrangement of an arbitrary standard system reference model. In addition, since the parameters are automatically adapted using the system identification result and the rotation speed information, the implementation to the multi-inertia resonance system is facilitated.

  Saves the amplitude and phase of the compensation current signal when suppression is completed at an arbitrary steady operating point (steady torque and steady rotational speed), and implements it at multiple operating points to create a two-dimensional table of torque and rotational speed Implement as At this time, it is possible to input the torque / rotational speed information into the table, generate a compensation current from the read compensation current amplitude / phase data, and suppress feedforward.

FIG. 7 is a calculation block diagram of Fourier coefficient learning control, in which 31 represents an IP control unit, 32 represents a non-interacting unit, 33 represents an actual system, and 34 represents a torque ripple extracting unit. This figure represents only the control system synchronized with the torque ripple frequency component, and uses the identification result of equation (6) as a nominal model. In this method, torque ripple is suppressed by an IP control method in which a proportional term in which cosine (real part) / sine (imaginary part) coefficient command values T _An ^* and T _Bn ^* of the torque ripple component are set to 0 is provided in the preceding stage. When the compensation current is also expressed as a complex vector and superimposed on the system, the torque detection value T _det is expressed by Equation (7), and the compensation current I _An . It can be seen that I _Bn interfere with each other.

  Therefore, as shown in FIG. 7, by providing a speed adaptive type non-interference term using the system identification result, the suppression control system of the real part and the imaginary part is made non-interactive. The closed loop transfer function from the target value to the detected value after decoupling is given by equation (8).

When the target value response of equation (8) is adapted to the pole placement of the binomial coefficient standard system shown in equation (10), equation (11) can be derived from the coefficient comparison of equations (8) and (10). Is given as a nominal proportional gain K _p and an integral gain K _i .

ω _c : Desired learning control response frequency [rad / s], where ω _c <ω _f / 2.

Since the proportional gain K _{p in the} equation (11) includes components of the real part P _Am and the imaginary part P _Bm of the system identification model that changes according to the speed, even if the rotational speed (torque ripple frequency) changes (10) Serves to maintain the pole placement response of the equation. The learning control response is arbitrarily determined within a range that satisfies the condition. Although details of the disturbance response are omitted, the derivation result of the gain equation is equivalent to the equation (11).

Kanno et al., “Examination of torque ripple suppression control method focusing on periodic disturbance of PM motor”, 2009 IEEJ Industrial Application Division Conference, I-615-618, Date: August 31-September 2, 2009 Sun, venue: Mie University

  In general, in a device that detects torque ripple and suppresses torque ripple by feedback control over the entire frequency band, the feedback control response decreases due to the influence of computation dead time in the high frequency band, and disturbance suppression performance in the high frequency band decreases. It becomes difficult to generate a desired compensation current.

  On the other hand, the periodic learning method obtained by the above-described IP (proportional / integral) learning control method generates a sine wave / cosine wave having the same frequency as that of the periodic disturbance, and then generates the sine / cosine coefficient (amplitude / cosine coefficient). (Equivalent to phase). Therefore, this method is easier to deal with high frequency disturbances than a controller that collectively generates compensation currents in all frequency bands, and can be expected to improve periodic disturbance suppression performance. Further, if a suppression controller is configured in parallel for each target order, torque ripple suppression of a plurality of orders can be handled at the same time.

Further, in FIG. 5, in order to construct a control system with coordinates synchronized with the torque ripple frequency, system transfer characteristics from the torque ripple compensation current command value i _qc ^* to the shaft torque detection value T _det are identified. From this system identification result, only an arbitrary frequency transfer characteristic is extracted and feedback controlled. In the steady state, the amplitude / phase transfer characteristic of the system synchronized with the torque ripple frequency can be expressed by a one-dimensional complex vector. Therefore, the frequency transfer function of the real system is expressed by the complex vector P (jω) related to the rotational speed ω as shown in the equation (12). Can be defined.

P _A (ω): Real part of real system, P _B (ω): Imaginary part of real system In Non-Patent Document 1, in order to construct a control system with torque ripple synchronous coordinates, an arbitrary n-th order component is obtained from equation (12). Only the frequency transfer characteristic is extracted, and an actual system of torque ripple synchronous coordinates is expressed by equation (13). That is, it is noted that the amplitude / phase characteristics of an arbitrary n-order component system can be expressed by a simple one-dimensional complex vector P _n . As in the equations (2) to (5), the real part / imaginary part axes are defined with reference to the rotational phase θ, the cosine coefficient corresponds to the real part component, and the sine coefficient corresponds to the imaginary part component.

P _An : Real system n-order component real part, P _Bn : Real system n-order component imaginary part The system identification result in the control system of FIG. 5 can be similarly defined as a formula (14) with a one-dimensional complex vector. it can. In addition, in the estimated value in the equation (14) and the following equations, the “^” symbol is attached to the top of P, but in the specification, it is written as “P ^”.

P ^ _An : n-order component real part estimated value of identification result, P ^ _Bn : n-order component imaginary part estimated value of identification result For example, when the system identification result from 1 to 1000 Hz is expressed by a complex vector for each IHz, A table composed of 1000 one-dimensional complex vector elements can be constructed. It is also possible to express the identification result by an approximate expression.

  From the above, even in a complex system, the system model is always a simple one-dimensional complex vector, and equation (14) can be changed instantaneously according to changes in motor rotation speed ω (torque ripple frequency change). It can easily handle variable speed operation.

Here, in the torque ripple suppression control block shown in FIG. 7, the division P _Bn / P _An is used in the non-interacting unit. However, when the system real part P _An becomes an operation state that becomes zero, the division P _Bn / P _An has a zero denominator P _An, which results in an overflow in the divider or data processing step that performs this calculation, which could cause the operation to stop or stop.

Also, the determination of the integral gain K _i, the denominator (2 [omega _c - [omega] _f) is 0 in (8), becomes zero split, the decision not.

An object of the present invention is to provide a non-interacting control and the integral gain K _i dynamoelectric machine torque ripple suppression control apparatus and a control method capable torque ripple suppression control that avoids the inconveniences caused by division by zero operation in the determination of.

In the present invention, in order to solve the above-described problem, the non-interacting unit includes the n-order component real part P of the identification result of the rotating electrical machine system included in the n-order compensation current cosine coefficient I _An and the sine coefficient I _Bn of the torque ripple. to avoid by P _an = 0 non-interference calculation eliminates division by zero (P _Bn / P _an) by by components of _an and the imaginary part P _Bn, I-P control unit division by zero in the determination of the integral gain K _i And is characterized by the following apparatus and method.

(1) The torque command value or speed command value of the rotating electrical machine system is converted into the d and q axis current command values of the rotating coordinate system in vector control, and the frequency transfer function of the rotating electrical machine control system is arbitrarily determined by system identification. Modeling represented by a complex vector of frequency components is performed, and the real and imaginary parts of the torque ripple of an arbitrary frequency component are extracted using this model, and the above d, q so as to suppress the real and imaginary parts of the torque ripple. A torque ripple suppression control device for a rotating electrical machine that superimposes a feedback compensation current on a shaft current command value,
Torque ripple seeking shaft torque detection value T _det pulsating extraction filter G _F and the rotational phase cosine coefficients of n order frequency components different torque pulsation torque ripple of the rotary electric machine system has a theta T _An sine coefficient T _Bn rotary electric machine An extractor;
I is obtained from the cosine coefficient T _An , the sine coefficient T _Bn, and the rotational speed ω of the rotating electrical machine to obtain the nth-order compensation current cosine coefficient I _An ′ and sine coefficient I _Bn ′ of the torque ripple by IP (proportional / integral) learning control -P control unit;
N-order component real part estimated value P ^ _An and n-order component imaginary part estimated value P ^ _Bn of the identification result of the rotating electrical machine system included in the cosine coefficient I _An ′ and sine coefficient I _Bn ′,

A non-interacting unit that obtains a cosine coefficient I _An ^* and a sine coefficient I _Bn ^* that are made non-interfering with each other in the calculation of
And a compensation current generator that obtains a feedback compensation current for suppressing periodic disturbance of the rotating electrical machine system from the cosine coefficient I _An ^* , the sine coefficient I _Bn ^*, and the rotational phase θ of the rotating electrical machine.

  (2) The IP control unit

The proportional gain K _p and the integral gain K _i are determined from the above.

  (3) The IP control unit

The proportional gain K _p and the integral gain K _i are determined from the above.

(4) The torque command value or speed command value of the rotating electrical machine system is converted into the d and q axis current command values of the rotating coordinate system in vector control, and the frequency transfer function of the rotating electrical machine control system is arbitrarily determined by system identification. Modeling represented by a complex vector of frequency components is performed, and the real and imaginary parts of the torque ripple of an arbitrary frequency component are extracted using this model, and the above d, q so as to suppress the real and imaginary parts of the torque ripple. A torque ripple suppression control method for a rotating electrical machine that superimposes a feedback compensation current on a shaft current command value,
Torque ripple extraction unit, cosine coefficient T _An sine coefficients of the shaft torque detection value T _det pulsating extraction filter G _F and the rotary electric machine system n order frequency components different torque pulsations torque ripple having from rotational phase θ of the rotary electric machine _Find T _Bn ,
The IP control unit calculates the n-th compensation current cosine coefficient I _An 'and the sine coefficient I _Bn ' of the torque ripple from the cosine coefficient T _An and the sine coefficient T _Bn and the rotational speed ω of the rotating electrical machine by IP (proportional / Integral) learning control,
The non-interacting unit includes a real part estimated value P ^ _An and an nth order component imaginary part estimated value P ^ of the identification result of the rotating electrical machine system included in the cosine coefficient I _An 'and the sine coefficient I _Bn '. _Bn

The cosine coefficient I _An ^* and the sine coefficient I _Bn ^* which are made non-interfering with each other in the calculation of are obtained, and these are used as the torque ripple compensation current command value.
The compensation current generator obtains a feedback compensation current for suppressing periodic disturbance of the rotating electrical machine system from the cosine coefficient I _An ^* , the sine coefficient I _Bn ^*, and the rotational phase θ of the rotating electrical machine.

As described above, according to the present invention, the non-interacting unit performs the real part P _An of the n-order component of the identification result of the rotating electrical machine system included in the n-order compensation current cosine coefficient I _An and the sine coefficient I _Bn of the torque ripple. since the imaginary part and by P _an = 0 by P _Bn to avoid a non-interference calculation eliminates division by zero _{(P Bn / P an),} I-P controller with eliminates division by zero in the determination of the integral gain K _i It can torque ripple suppression control that avoids the inconveniences caused by division by zero operation in the determination of non-interacting control and the integral gain K _i.

FIG. 4 is a calculation block diagram of Fourier coefficient learning control (first embodiment). The torque ripple suppression test result by Embodiment 1. FIG. FIG. 4 is a calculation block diagram of Fourier coefficient learning control (second embodiment). The basic block diagram of the conventional torque ripple suppression control apparatus. The control block diagram of the conventional torque ripple suppression apparatus. System identification result by nonparametric estimation of frequency transfer function. The operation block diagram of Fourier coefficient learning control.

(Embodiment 1)
FIG. 1 is a calculation block diagram of Fourier coefficient learning control in the present embodiment. In the figure, the torque ripple frequency component extraction unit (19 in FIG. 5) is configured as a calculation block according to equations (3) to (5). The torque ripple suppression control unit (21) includes a proportional leading IP control unit and a non-interacting unit in torque ripple synchronous coordinates, and a cosine coefficient (real part) T _An ^* and a sine coefficient (imaginary part) of the torque ripple command value. ) Input 0 to T _Bn ^* to suppress it.

  Note that the reason why the proportional precedence type IP control is adopted is to facilitate the transient response improvement at the time of the torque ripple suppression start command and the calculation when matching the closed loop response after decoupling with the reference model ( Later).

By the way, when the n-order compensation current i _qcn ^* shown in the above equation (2) is expressed by a complex vector, it is converted as follows from Euler's formula.

Similarly, the torque ripple detection value T _detn of the nth order component is converted as follows.

  Here, if the real system of Formula (13) is used, the following relationship is established.

di _n : n-th order periodic disturbance current Considering a state in which the periodic disturbance current is removed by the compensation current, using the system identification result (14), the relationship of the expression (17) is expressed as follows.

  Here, when Expressions (14) and (15) are substituted into Expression (18) and the extraction filter characteristics shown in Expression (5) are taken into consideration, the following is obtained.

When the complex Fourier coefficient n-order component T _Cn (n> 0) of the equation (16) is extracted, it is as follows.

Similarly, when the complex Fourier coefficient n-order component T _Cn (n> 0) of the equation (19) is extracted, it is as follows.

  By comparing the coefficients of the real part and the imaginary part of the equations (20) and (21), the following relational expression is derived.

As can be seen from the equation (22), it can be seen that the real part and the imaginary part of the torque detection value T _det include the real part I _An ^* and the imaginary part I _Bn ^* of the compensation current.

  In order to cancel the interference between the real part and the imaginary part, the non-interacting part in FIG. 1 performs a non-interacting calculation shown in the equation (23).

  When the identification result is positive and the equation (23) is substituted into the equation (21) to make it non-interfering, the equation (24) is replaced, and the compensation currents for the real part and the imaginary part can be controlled independently and identically. . The fact that the real part and the imaginary part can be designed independently can avoid the problem of 0%.

  Next, when the closed-loop transfer function of the target value response after decoupling is calculated in the same manner as in the equation (8), the equation (25) is obtained.

  This target value response is adapted to the pole arrangement of the binomial coefficient standard system shown in the equation (26) by the model matching method. In the present embodiment, the equation (26) is adapted, but it is also possible to adapt it to another reference model such as a Butterworth standard system.

ω _c : Desired learning control response frequency [rad / s]
(25) from the coefficient comparison equation (26) becomes the determining the proportional gain K _p and the integral gain K _i (27) below.

In addition, the periodic disturbance response at this time is expressed by the equation (28), and the pole arrangement coincides with the target value response of the equation (26). Than the final value theorem, the torque ripple is seen to converge to zero even if the input cycle disturbance real part dI _An and the imaginary part dI _Bn any order.

The proportional gain K _{p of the} equation (28) applied to this control system includes the real part estimated value P ^ _An and the imaginary part estimated value P ^ _Bn of the system identification result of the expression (14). Similarly, the identification result is used for the non-interference term in the equation (23). That is, these are automatically adjusted according to the motor rotational speed ω, and function to always maintain the standard system pole arrangement shown in the equation (26).

  From the above, in this embodiment, torque ripple suppression control that avoids the problem of zero percent in non-interacting control can be performed, and periodic disturbance suppression that can be adapted to operating point fluctuations due to variable speed operation is achieved. Labor-saving by adjustment can also be expected.

  FIG. 2 shows a torque ripple suppression test result by the torque ripple suppression device configured by the calculation block shown in FIG. 1 and obtains a torque ripple suppression result of the sixth-order component.

(Embodiment 2)
In the configuration of the first embodiment, paying attention to the denominator of K _{i in} the equation (27), there is a possibility that it becomes zero when ω _c = ω _f / 2. Since ω _c and ω _f are fixed values determined arbitrarily by the user, there is no problem if they are set so as to avoid the singular point, but in this embodiment, a control configuration in which no singular point is generated is proposed, and the response frequency Increase the degree of freedom.

  FIG. 3 is a calculation block diagram of Fourier coefficient learning control in the present embodiment. 1 is different from FIG. 1 in the calculation block configuration of the IP controller.

  When the closed-loop transfer function of the target value response according to the configuration of the IP control unit is calculated, equation (29) is obtained.

When the proportional gain K _p and the integral gain K _i are obtained from the coefficient comparison of the equations (29) and (26) so that the standard system pole arrangement design is obtained, the equation (30) is obtained.

As can be seen from the integral gain K _{i in} equation (30), zero division by a singular point like K _i in equation (27) of the first embodiment does not occur.

Therefore, the controller can be operated without any problem even _if ω _c = ω _f / 2. Note that, when ω _c = ω _f / 2, since the proportional gain K _p = 0, only the integral control functions.

DESCRIPTION OF SYMBOLS 11 Command value conversion part 12 Current vector control part 13 Motor 14 Load apparatus 15 Coordinate conversion part 16 Current sensor 17 Rotation phase / speed detection part 18 Rotation position sensor 19 Torque pulsation frequency component extraction part 20 Shaft torque detector 21 Torque ripple suppression control part 22 Compensation current generation unit 31 IP control unit 32 Decoupling unit 33 Real system 34 Torque ripple extraction unit

Claims (4)

The torque command value or the speed command value of the rotating electrical machine system is converted into the d and q axis current command values of the rotating coordinate system in vector control, and the frequency transfer function of the rotating electrical machine control system is changed to an arbitrary frequency component by system identification. Modeling is performed with a complex vector, and the real and imaginary parts of the torque ripple of any frequency component are extracted using this model, and the d and q-axis current commands are used to suppress the real and imaginary parts of the torque ripple. A torque ripple suppression control device for a rotating electrical machine that superimposes a feedback compensation current on a value,
Torque ripple seeking shaft torque detection value T _det pulsating extraction filter G _F and the rotational phase cosine coefficients of n order frequency components different torque pulsation torque ripple of the rotary electric machine system has a theta T _An sine coefficient T _Bn rotary electric machine An extractor;
I is obtained from the cosine coefficient T _An , the sine coefficient T _Bn, and the rotational speed ω of the rotating electrical machine to obtain the nth-order compensation current cosine coefficient I _An ′ and sine coefficient I _Bn ′ of the torque ripple by IP (proportional / integral) learning control -P control unit;
N-order component real part estimated value P ^ _An and n-order component imaginary part estimated value P ^ _Bn of the identification result of the rotating electrical machine system included in the cosine coefficient I _An ′ and sine coefficient I _Bn ′,

A non-interacting unit that obtains a cosine coefficient I _An ^* and a sine coefficient I _Bn ^* that are made non-interfering with each other in the calculation of
And a compensation current generator for obtaining a feedback compensation current for suppressing periodic disturbance of the rotating electrical machine system from the cosine coefficient I _An ^* , the sine coefficient I _Bn ^* and the rotational phase θ of the rotating electrical machine. Torque ripple suppression control device for electric machines. The IP controller is

The torque ripple suppression control device for a rotary electric machine according to claim 1, wherein the proportional gain K _p and the integral gain K _i are determined from The IP controller is

The torque ripple suppression control device for a rotary electric machine according to claim 1, wherein the proportional gain K _p and the integral gain K _i are determined from The torque command value or the speed command value of the rotating electrical machine system is converted into the d and q axis current command values of the rotating coordinate system in vector control, and the frequency transfer function of the rotating electrical machine control system is changed to an arbitrary frequency component by system identification. Modeling is performed with a complex vector, and the real and imaginary parts of the torque ripple of any frequency component are extracted using this model, and the d and q-axis current commands are used to suppress the real and imaginary parts of the torque ripple. A torque ripple suppression control method for a rotating electrical machine that superimposes a feedback compensation current on a value,
Torque ripple extraction unit, cosine coefficient T _An sine coefficients of the shaft torque detection value T _det pulsating extraction filter G _F and the rotary electric machine system n order frequency components different torque pulsations torque ripple having from rotational phase θ of the rotary electric machine _Find T _Bn ,
The IP control unit calculates the n-th compensation current cosine coefficient I _An 'and the sine coefficient I _Bn ' of the torque ripple from the cosine coefficient T _An and the sine coefficient T _Bn and the rotational speed ω of the rotating electrical machine by IP (proportional / Integral) learning control,
The non-interacting unit includes a real part estimated value P ^ _An and an nth order component imaginary part estimated value P ^ of the identification result of the rotating electrical machine system included in the cosine coefficient I _An 'and the sine coefficient I _Bn '. _Bn

The cosine coefficient I _An ^* and the sine coefficient I _Bn ^* which are made non-interfering with each other in the calculation of are obtained, and these are used as the torque ripple compensation current command value.
A compensation current generator obtains a feedback compensation current for suppressing periodic disturbance of a rotating electrical machine system from the cosine coefficient I _An ^* , the sine coefficient I _Bn ^* and the rotational phase θ of the rotating electrical machine. Torque ripple suppression control method.

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