JP5363129B2 - Inverter control device - Google Patents

Description

  The present invention relates to an inverter control device for efficiently controlling the rotation of a motor without using a position sensor or a speed sensor.

  AC motors such as synchronous motors and induction motors rotate the rotor by generating a rotating magnetic field in the stator. A permanent magnet synchronous motor having a permanent magnet in the rotor generates the largest torque when the angle formed between the magnetic flux generated by the permanent magnet of the rotor and the rotating magnetic field generated by the stator is 90 degrees. Therefore, in order to rotate the rotor efficiently, it is necessary to detect the magnetic pole position of the rotor.

  On the other hand, the induction motor does not use a permanent magnet for the rotor, and can be rotated without detecting the magnetic pole position of the rotor in principle. However, recently, in order to improve the responsiveness to torque control, a vector control technique using a speed sensor that detects the rotational speed of the rotor has been developed.

  A position sensor and a speed sensor such as a rotary encoder, a hall element, and a resolver are used to detect the magnetic pole position and the rotational speed of the rotor. However, these sensors are generally expensive, and the entire device including the motor and the inverter becomes expensive. In addition, the sensor may not be installed depending on the use environment of the motor, such as operation under high temperature. Therefore, various sensorless motor driving methods for operating a motor without using a position sensor or a speed sensor have been proposed.

  Patent Document 1 discloses an inverter control device that controls an inverter without detecting the position of a rotor based on a current flowing from the inverter to the motor. FIG. 1 is a block diagram showing a conventional sensorless inverter control device described in Patent Document 1. In FIG. Currents Iu, Iv, and Iw supplied from the inverter 1 to the motor M are detected by the current detector 2, and the detected three-phase current amounts Iu, Iv, and Iw are sent to the three-phase to two-phase coordinate conversion unit 5. It is done.

  In the three-phase to two-phase coordinate conversion unit 5, the current amounts Iu, Iv, and Iw are expressed as current vectors on the three-phase fixed coordinate system, and these current vectors Iu, Iv, and Iw are expressed in the α-β coordinate system ( It is converted into current vectors Iα and Iβ on the two-phase fixed coordinate system (see FIG. 2). The α axis of the α-β coordinate system coincides with the direction of the u phase, and the β axis is orthogonal to the α axis. This three-phase to two-phase conversion is performed using a known formula.

  Further, the three-phase to two-phase coordinate conversion unit 5 converts the current vectors Iα and Iβ into current vectors Iγ and Iδ on the γ-δ coordinate system (two-phase rotation coordinate system) using the estimated rotation angle θ ( (See FIG. 3). The γ-axis of the γ-δ coordinate system is placed at a phase advanced by a phase difference angle (described later) from the rotor position of the motor M, and the δ-axis is placed perpendicular to the γ-axis. The current vector Iδ is an effective current having the same phase as the output voltage of the inverter 1, and the current vector Iγ is a reactive component current having a phase delayed by 90 degrees from the phase of the output voltage of the inverter 1. The estimated rotation angle θ is a phase difference from the u phase to the γ axis. The conversion from the fixed coordinate system to the rotating coordinate system shown in FIG. 3 is performed using a known formula. By converting the fixed coordinate system to the rotating coordinate system in this way, the alternating current when viewed from the rotor can be handled as a direct current.

The three-phase to two-phase coordinate conversion unit 5 outputs values Iγ and Iδ as current amounts to the first current control unit 7 and the second current control unit 8, respectively. The first current control unit 7 generates the voltage command value Vγ to the inverter 1 so that the deviation between the current amount Iγ and the target reactive component current Iγ * becomes zero. The second current control unit 8 generates the current amount Iδ. Then, the voltage command value Vδ to the inverter 1 is generated so that the deviation from the target current value Iδ ^* from the speed control unit becomes zero.

The gain generator 11 controls the proportional gain Kg so as to lead the voltage command value Vγ to zero. In the multiplier 12, the proportional gain Kg obtained by the gain generator 11 is multiplied by the voltage command value Vδ from the second current controller 8, thereby obtaining the estimated angular frequency (estimated rotational speed) ωe of the rotor. . This estimated angular frequency ωe is sent to the speed control unit 14. The speed control unit 14 generates a target torque current value Iδ ^* such that the deviation between the estimated angular frequency ωe and a predetermined target angular frequency (target speed) ω ^* [rad / sec] is zero, This is sent to the current control unit 8.

  The estimated angular frequency ωe is also sent to the integrator 16, where an estimated rotational angle (estimated phase difference) θ is calculated by integrating the estimated angular frequency ωe with a predetermined period T [sec]. This estimated rotation angle is sent to the above-described three-phase / two-phase coordinate conversion unit 5 and two-phase / three-phase coordinate conversion unit 6. The two-phase to three-phase coordinate conversion unit 6 uses the estimated rotation angle θ to output the voltage command values Vγ and Vδ output from the second current control unit 8 and the first current control unit 7 on the α-β coordinate system. The voltage vectors Vα and Vβ are converted into three-phase voltage vectors Vu, Vv and Vw on the three-phase fixed coordinate system. These transformations are performed using known formulas.

  The two-phase / three-phase coordinate conversion unit 6 outputs voltage command values Vu, Vv, Vw to the inverter 1. The inverter 1 is a PWM (Pulse Width Modulation) type voltage source inverter 1. The inverter 1 has a power switching element such as a power transistor, a power MOS, and an IGBT. These power switching elements constitute, for example, a 6-arm type bridge circuit, and DC power from a power source (not shown) has an arbitrary frequency and voltage by switching the power switching element by pulse width modulation control. Convert to AC power. The inverter 1 thus generates a three-phase voltage corresponding to the voltage command values Vu, Vv, Vw. The generated three-phase voltage is applied to the stator winding of the motor M.

  Next, sensorless vector control for controlling the permanent magnet synchronous motor using the inverter control device shown in FIG. 1 will be described. FIG. 4 is a diagram showing the relationship among the current vectors Iγ and Iδ, the voltage command values Vγ and Vδ, and the induced voltage. In FIG. 4, the γ-δ coordinate system and the dq coordinate system coexist. The dq coordinate system is an orthogonal coordinate system set according to the direction of the rotor of the motor M. More specifically, the d axis indicates the magnetic flux direction of the rotor, and the q axis is orthogonal to the d axis. Therefore, the dq coordinate system can be referred to as a motor shaft, and the γ-δ coordinate system can be referred to as an inverter shaft. Symbol Ea is an induced voltage generated in the stator winding by the rotation of the rotor. This induced voltage Ea is generated in the q-axis direction. The phase difference angle (load angle) β is a phase difference between the output voltage of the inverter 1 (voltage applied to the motor M) and the induced voltage Ea.

The voltage command values Vγ and Vδ described above are expressed by the following equations.
Vγ = RIγ + L (dIγ / dt) −ωLIδ + Easinβ (1)
Vδ = RIδ + L (dIδ / dt) + ωLIγ + Ecosβ (2)
Here, R is an internal resistance specific to the motor, and L is an inductance specific to the motor.

In the steady state, the differential terms of the above formulas (1) and (2) are zero, so that the following formula is obtained.
Vγ = RIγ−ωLIδ + Easinβ (3)
Vδ = RIδ + ωLIγ + Ecosβ (4)

When the phase difference angle β approximates to zero, the above equations (3) and (4) are expressed as follows.
Vγ = RIγ−ωLIδ (5)
Vδ = RIδ + ωLIγ + Ea (6)

As described above, since Vγ and Iγ are controlled to be zero, the following equations are derived from the equations (5) and (6).
Vδ = RIδ + Ea (7)

Further, assuming that the voltage drop due to the internal resistance R of the motor M is zero, Expression (7) is expressed as follows.
Vδ = Ea (8)
Here, since the induced voltage Ea is the product of the angular frequency and the magnetic flux of the rotor, that is, Ea = ωψ, the following equation is derived from the equation (8).
ω = Vδ / ψ (9)
Here, if 1 / ψ is set as Kg,
ω = VδKg (10)

  As described above, when β and Iγ are set to zero, the γ-δ coordinate system and the dq coordinate system coincide with each other, and the voltage command value (output voltage of the inverter 1) Vδ and current (output current of the inverter 1). The phase of Iδ matches. As a result, the vector diagram shown in FIG. 5 is finally obtained. Thus, the rotor rotates synchronously with the output voltage and the output current in phase.

  However, the actual phase of the rotor may deviate from the δ axis due to fluctuations in the load applied to the rotor. 6A shows a state where the phase of the rotor is delayed with respect to the δ axis, FIG. 6B shows a state where the phase of the rotor matches the δ axis, and FIG. This shows a state where the phase of is advanced with respect to the δ axis. In the state shown in FIG. 6A, Vγ (> 0) and Iγ (> 0) are generated, and in the state shown in FIG. 6C, Vγ (<0) and Iγ (<0) are generated. On the other hand, in the state shown in FIG. 6B, Vγ and Iγ are not generated. Therefore, the phase and rotational speed of the rotor can be estimated by controlling the rotational speed so that the voltage command value Vγ is led to zero by the gain generator 11.

  However, this sensorless vector control has the following problems. FIG. 7 shows a state where the permanent magnet synchronous motor is operating in an ideal state. In this ideal state, the direction of the current I is a direction perpendicular to the magnetic flux direction of the rotor, and in this state, the torque of the rotor becomes maximum. As shown in FIG. 7, the phase of the current I is delayed by the phase difference angle β from the phase of the voltage Vδ. This is a phase lag caused by the inductance component of the motor M. In the sensorless vector control described above, control is performed on the assumption that the phase difference angle β is zero, but in reality, the phase difference angle β always exists. Furthermore, in the sensorless vector control described above, since the current Iγ is controlled to zero, the current I and the voltage Vδ have the same phase. Therefore, the vector diagram of FIG. 5 is actually represented as the vector diagram shown in FIG.

  As can be seen from FIG. 8, the actual current I is decomposed into a current component Id along the d-axis and a current component Iq along the q-axis. Of these, the current component Id is a current that flows parallel to the direction of the magnetic flux of the motor M, and is a negative current. That is, if the current Iγ is controlled to be zero so that the current I and the voltage Vδ are in phase, the current Id is actually caused to flow in a direction that cancels the magnetic flux of the rotor. For this reason, when the magnetic flux of the rotor is weakened, the torque is weakened, and the rotor and the rotating magnetic field are easily out of synchronization. Furthermore, in order to obtain the same current (= torque) as the current I ′ shown in FIG. 8, it is necessary to increase the current I. That is, in order to obtain the same current (= torque), the inverter I requires a larger inverter output current than the current I ′. As described above, if the inverter shaft and the motor shaft are in phase, the efficiency of the motor and the inverter is reduced as a result.

Patent No. 40222630

  The present invention has been made to solve the above-described conventional problems, and an object of the present invention is to provide an inverter control device that enables efficient motor operation.

  One aspect of the present invention is an inverter control device that controls an inverter for driving a motor, the current detector detecting a three-phase current supplied from the inverter to the motor, and a current detector. A three-phase to two-phase coordinate conversion unit for converting the three-phase current into a magnetizing current and a torque current in a rotating coordinate system, and a first voltage command value for setting the deviation between the magnetizing current and the first target current value to zero A first current control unit that calculates a second voltage command value for making a deviation between the torque current and the second target current value zero, and the first and second voltages A two-phase / three-phase coordinate converter for converting a command value into a three-phase voltage command value on a three-phase fixed coordinate system, and gain generation for generating a magnetizing current adjustment gain value for setting the first voltage command value to zero The magnetizing current adjustment gain value and the first A multiplier for calculating an estimated angular frequency of the motor from a voltage command value; a speed control unit for calculating the second target current value for making a deviation between the estimated angular frequency and a predetermined target angular frequency zero; And a gain control unit that calculates the first target current value for setting the deviation between the magnetizing current adjustment gain value and the target gain value to zero.

In a preferred aspect of the present invention, the magnetizing current adjustment gain value is a variable determined from the angular frequency of the motor and the second voltage command value .
In a preferred aspect of the present invention, the target gain value is a value determined from a voltage-frequency characteristic of the motor.
In a preferred aspect of the present invention, the target gain value is calculated from a rated voltage of the motor and a rated frequency of the motor.
In a preferred aspect of the present invention, the target gain value is a value calculated in advance from characteristics of a load connected to the motor.

According to a preferred aspect of the present invention, there is provided an output voltage calculation unit for calculating an output voltage command value for the inverter, and a first target gain value for setting a deviation between the output voltage command value and a predetermined output voltage upper limit value to zero. The target gain calculation unit for calculating the value, the second target gain value calculated in advance from the rated voltage of the motor and the rated frequency of the motor, and the first target gain value are compared, and the larger value is obtained. And a comparator that outputs the target gain value to the gain control unit.
In a preferred aspect of the present invention, the predetermined output voltage upper limit value is one of a rated voltage of the motor and an upper limit output voltage of the inverter.

In a preferred aspect of the present invention, the apparatus further includes a lower limiter that defines a lower limit value of the first target current value calculated by the gain control unit.
A preferred aspect of the present invention further includes a step-out determination unit that determines that step-out has occurred when the magnetization current adjustment gain value generated by the gain generation unit reaches a predetermined threshold value. And
In a preferred aspect of the present invention, the operation of the motor is stopped when it is determined that the step-out has occurred.
A preferred aspect of the present invention further includes a comparator that compares the first target current value calculated by the gain control unit with a predetermined target excitation current value and selects a smaller value. The 1 current control unit calculates the first voltage command value for making a deviation between the magnetization current and the first target current value or the target excitation current value selected by the comparator zero. Features.

According to the present invention, the magnetizing current is not controlled to zero, and the output voltage of the inverter is controlled so as to be an optimum value corresponding to the rotational speed of the motor. Therefore, efficient motor operation becomes possible.
In addition, according to the present invention, since motor-specific parameters such as the winding resistance value of the motor that varies with temperature and the inductance value of the motor that varies with load are not used, the operating characteristics do not change during motor driving. Further, since parameters specific to the motor are not used, it is not necessary to measure these parameters in advance, and it is not necessary to change the parameters depending on the operation state.
In addition, according to the present invention, the magnetizing current can be passed to the motor as a magnetic flux axis current in the case of a synchronous motor and as an exciting current in the case of an induction motor, so that the inverter can be controlled without selecting the type of motor. can do.
In addition, according to the present invention, when the motor is driven at a variable speed, the output voltage is constantly monitored in the entire variable speed range, and the inverter is controlled so that the output voltage matches the frequency of the motor. A stable motor drive can be realized.

It is a block diagram which shows the conventional sensorless type inverter control apparatus. It is a figure for demonstrating a mode that the electric current vector on a three-phase fixed coordinate system is converted into the electric current vector on a two-phase fixed coordinate system. It is a figure for demonstrating a mode that the electric current vector on a two-phase fixed coordinate system is converted into the electric current vector on a two-phase rotational coordinate system. It is a figure which shows the relationship between electric current vector Igamma, Idelta, voltage command value Vgamma, Vdelta, and an induced voltage. It is a figure which shows the vector produced | generated by the conventional vector control method. 6A shows a state where the phase of the rotor is delayed with respect to the δ axis, FIG. 6B shows a state where the phase of the rotor is synchronized with the δ axis, and FIG. FIG. 6 is a diagram showing a state in which the phase of is advanced with respect to the δ axis. It is a figure which shows the state which the permanent magnet synchronous motor is drive | operating in an ideal state. It is a figure showing the actual vector including a motor and an inverter when controlling so that electric current value Iγ may become zero. It is a block diagram which shows the sensorless type inverter control apparatus which concerns on one Embodiment of this invention. FIG. 10A and FIG. 10B are diagrams showing how the magnitude of the output voltage Vδ changes by controlling the magnetizing current value Iγ. It is a graph which shows the change of the motor characteristic when the magnetizing current adjustment gain value Pvf deviates from the ideal value. It is a block diagram which shows the sensorless type inverter control apparatus which concerns on other embodiment of this invention. It is a block diagram which shows the sensorless type inverter control apparatus which concerns on further another embodiment of this invention. It is a block diagram which shows the sensorless type inverter control apparatus which concerns on further another embodiment of this invention. It is a block diagram which shows the sensorless type inverter control apparatus which concerns on further another embodiment of this invention.

Hereinafter, embodiments of the present invention will be described.
FIG. 9 is a block diagram showing a sensorless inverter control apparatus according to an embodiment of the present invention. In the present specification and drawings, the same or corresponding elements are denoted by the same reference numerals, and redundant description thereof is omitted. The vector control of the inverter control device according to the present embodiment is performed according to the vector diagram shown in FIG.

  As shown in FIG. 9, the inverter control apparatus according to the present embodiment is a current detector (for example, DC Current Transformer or shunt resistor) that detects three-phase currents Iu, Iv, and Iw supplied from the inverter 1 to the motor M. 2, a three-phase to two-phase coordinate conversion unit 5 that converts the three-phase currents Iu, Iv, and Iw detected by the current detector 2 into current vectors Iγ and Iδ in the rotating coordinate system, and a current amount Iγ that is a magnetization current The first current control unit 7 that calculates a voltage command value Vγ for making the deviation between the current value Iγ * and the target current value Iγ * zero, and the deviation between the current amount Iδ that is an effective current (torque current) and the target current value Iδ * A second current control unit 8 that calculates a voltage command value Vδ for setting the voltage to zero, and the voltage command values Vγ and Vδ output from the second current control unit 8 and the first current control unit 7 as a three-phase fixed coordinate system Upper three-phase voltage vector (three-phase voltage command value) Vu, V and a two-phase / three-phase coordinate conversion unit 6 for converting into v and Vw. The first current control unit 7 and the second current control unit 8 each have a PI control unit. Here, the magnetizing current means a magnetic flux axis current in the case of a permanent magnet type synchronous motor, and means an exciting current in the case of an induction motor.

  The inverter control device further includes a gain generation unit 11 that generates a magnetizing current adjustment gain value Pvf for setting the voltage command value Vγ to zero. A zero value is input to the gain generator 11 as the target magnetization voltage Vγ *. The magnetizing current adjustment gain value Pvf is a variable derived from the voltage-frequency characteristics of the motor M, and the ratio (Pvf = ω) between the angular frequency of the motor M and the voltage applied to the motor M (that is, the voltage command value Vδ). / Vδ).

  The voltage-frequency characteristic of the motor means, for example, the relationship between the rated voltage of the motor and the rated frequency of the motor. When the rated voltage is applied to the motor while the motor is rotating at the rated frequency, the motor operating efficiency is best.

  In this motor drive, if the current amount Iγ is controlled so that the ratio between the frequency of the motor M and the voltage applied to the motor M always maintains the theoretical value (ideal value) of the magnetizing current adjustment gain value Pvf, the motor M can be operated efficiently. Therefore, the theoretical value of the magnetizing current adjustment gain value Pvf can be said to be a gain value indicating the relationship between the frequency and the voltage when the motor M operates most efficiently.

  The inverter control device according to the present embodiment further multiplies the magnetizing current adjustment gain value Pvf output from the gain generation unit 11 by the voltage command value Vδ from the second current control unit 8 to estimate the estimated rotational speed ω [ rad / sec], a speed controller 14 for calculating a target torque current value Iδ * for making the deviation between the estimated rotational speed ωe and the target speed ω * to be zero, and an estimated rotational speed ωe. The integrator 16 for calculating the estimated rotation angle θ by integrating at a predetermined period T [sec], the deviation between the magnetizing current adjustment gain value Pvf output from the gain generation unit 11 and the target gain value Pvf * is zero. And a gain control unit 20 for calculating a target current value (target magnetization current value) Iγ *.

  When the voltage-frequency characteristic of the motor M in the load characteristic connected to the motor M is known, the target gain value Pvf * input to the gain control unit 20 is determined from the voltage-frequency characteristic. The However, when the load connected to the motor M is unknown, the target gain value Pvf * for the load characteristics cannot be determined. Therefore, when the load characteristic is unknown, the target gain value Pvf * is determined from the voltage-frequency characteristic of the motor M.

This inverter control device does not require motor-specific parameters such as the winding resistance value of the motor M that varies with temperature and the inductance value of the motor M that varies with the load, and the rated voltage and rated frequency displayed on the motor M are not required. The target gain value Pvf * is determined only from the above. Specifically, when the rated voltage of the motor M is Vconst and the rated frequency of the motor M is Fconst, the target gain value Pvf * is obtained as shown in Expression (11).
Pvf * = 2π × Fconst / Vconst (11)

  The gain generation unit 11 generates a magnetization current adjustment gain value Pvf for setting the voltage command value Vγ to zero using PI control or a PLL (Phase-Locked Loop) technique. For example, in the case of the motor M having the above rated voltage and rated frequency, when the phase of the output current I of the inverter 1 coincides with the q axis (that is, when Id = 0), or the ideal value of the voltage and the actual value. When the output voltages match, the magnetizing current adjustment gain value Pvf output from the gain generator 11 is a theoretical value (ideal value). When the phase of the output current I of the inverter 1 is delayed from the q axis (that is, when Id> 0), or when the actual output voltage is larger than the ideal value of the voltage, the magnetization output from the gain generator 11 The current adjustment gain value Pvf is smaller than the theoretical value (ideal value). In this case, the gain control unit 20 calculates the target current value Iγ * so that the deviation between the target gain value Pvf * and the magnetizing current adjustment gain value Pvf becomes zero. As a result, the target current value Iγ * decreases, and the output voltage of the inverter 1 decreases. When the phase of the output current I of the inverter 1 is advanced from the q axis (that is, when Id <0), or when the actual output voltage is smaller than the ideal value of the voltage, the magnetization output from the gain generator 11 The current adjustment gain value Pvf is larger than the theoretical value (ideal value). In this case, the gain control unit 20 calculates the target current value Iγ * so that the deviation between the target gain value Pvf * and the magnetizing current adjustment gain value Pvf becomes zero. As a result, the target current value Iγ * increases and the output voltage of the inverter 1 increases.

  FIGS. 10A and 10B are diagrams showing how the magnitude of the voltage command value Vδ changes by controlling the magnetizing current value Iγ. In FIG. 10A, the magnetizing current value Iγ is appropriately commanded. For this reason, the inverter output current I flows perpendicularly to the rotor. That is, the motor shaft is accurately estimated, and the voltage command value Vδ is output according to the voltage-frequency characteristics of the motor M. On the other hand, as shown in FIG. 10B, when the magnetizing current value Iγ is not properly commanded, the inverter output current I ′ does not flow perpendicularly to the rotor and is divided into the torque current Iq and the magnetizing current Id. Will be. That is, a deviation occurs in the estimation of the motor shaft, a negative current flows on the d-axis of the rotor, and a component -ωLId is generated. As a result, a voltage command value Vδ smaller than the ideal voltage determined from the voltage-frequency characteristics of the motor M is output.

Specifically, when there is a deviation in the estimation of the motor shaft, the third term ωLId in the following equation (13) is in the negative direction (that is, Id <0), so the voltage command value Vδ is small.
Vγ = RId + L (dId / dt) −ωLIq + Easinβ (12)
Vδ = RIq + L (dIq / dt) + ωLId + Ecosβ (13)
That is, by controlling the magnetizing current value Iγ so that the magnetizing current adjustment gain value Pvf becomes the target gain value Pvf * calculated in advance from the voltage-frequency characteristics of the motor M, the d axis and q axis that are motor axes Can be estimated. Therefore, the motor M can be driven efficiently.

  When the target current value Iγ * is controlled so that the deviation between the magnetization current adjustment gain value Pvf output from the gain generator 11 and the target gain value Pvf * is zero, the voltage command value Vγ converges to zero. That is, the magnetizing current value Iγ is controlled so that the output voltage of the inverter 1 when the motor M is rotating at a certain rotational speed matches the ideal value of the voltage determined from the voltage-frequency characteristics of the motor M. . Therefore, the most efficient voltage command value corresponding to the rotation speed of the motor M is input to the inverter 1, and the inverter 1 can drive the motor M efficiently.

  The target gain value Pvf * is a predetermined value transmitted from the target gain input unit 31 to the gain control unit 20. The target speed ω * is a predetermined value transmitted from the target speed input unit 32 to the speed control unit 14. The target magnetization voltage Vγ * is a predetermined value (zero in this embodiment) transmitted from the target magnetization voltage input unit 33 to the gain generation unit 11. The target gain input unit 31, the target speed input unit 32, and the target magnetization voltage input unit 33 may each include a storage unit that stores the predetermined value, and are obtained by calculation according to the operating state of the motor M. May be.

  FIG. 11 is a graph showing changes in motor characteristics when the magnetizing current adjustment gain value Pvf deviates from the ideal value when the command speed of the motor M is Fdemand. When the load characteristic connected to the motor M is unknown, the rated voltage of the motor M is Vconst, and the rated frequency is Fconst, the target gain value Pvf * is calculated from the above equation (11). Here, the ideal target gain value calculated from Equation (11) is shown in FIG. 11 as Pvf * A. Further, Pvf_B in FIG. 11 indicates a case where the magnetizing current adjustment gain value Pvf output from the gain generation unit 11 is higher than the ideal target gain value Pvf * A. That is, in the case of Pvf_B, it means that a voltage command value Vδ lower than the ideal voltage command value in the voltage-frequency characteristic of the motor M is output to the inverter 1. Therefore, the voltage command value Vδ is increased by controlling the gain control unit 20 to increase the magnetizing current value Iγ so that the gain control unit 20 brings the magnetizing current adjustment gain value Pvf closer to the ideal target gain value Pvf * A. On the other hand, Pvf_C in FIG. 11 indicates a case where the magnetizing current adjustment gain value Pvf output from the gain generation unit 11 is lower than the ideal target gain value Pvf * A. That is, in the case of Pvf_C, it means that a voltage command value Vδ higher than the ideal voltage command value in the voltage-frequency characteristic of the motor M is output to the inverter 1. Accordingly, the voltage command value Vδ is lowered by controlling the magnetizing current value Iγ to be small so that the gain controller 20 brings the magnetizing current adjustment gain value Pvf closer to the ideal target gain value Pvf * A. Thus, the voltage command value Vδ corresponding to the target speed can be obtained according to the characteristics of the motor M, so that the motor M can be operated efficiently.

  The voltage command values Vγ and Vδ calculated by the first current control unit 7 and the second current control unit 8 are sent to the two-phase / three-phase coordinate conversion unit 6, where the voltage command values Vγ and Vδ are three-phase. Are converted into voltage command values Vu, Vv, and Vw. The two-phase / three-phase coordinate conversion unit 6 outputs voltage command values Vu, Vv, Vw to the inverter 1. The inverter 1 is a PWM (Pulse Width Modulation) type voltage source inverter 1. The inverter 1 converts a DC voltage from a DC power supply (not shown) into a pulse voltage by an ON / OFF operation of the switching element, and generates a three-phase voltage corresponding to the voltage command values Vu, Vv, and Vw. The generated three-phase voltage is applied to the stator winding of the motor M.

  Next, another embodiment of the present invention will be described with reference to FIG. FIG. 12 is a block diagram showing a sensorless inverter control apparatus according to another embodiment of the present invention. In addition, since the structure of this embodiment which is not demonstrated in particular is the same as that of embodiment mentioned above, the overlapping description is abbreviate | omitted.

  When it is necessary to rotate the motor M at a high speed, the terminal voltage of the motor M (that is, the induced voltage of the motor M) increases, so that the voltage command value output from the inverter control device is the rated voltage of the motor M or the inverter. 1 may exceed the preset upper limit output voltage. Therefore, the inverter control apparatus according to the present embodiment, when the voltage command value that exceeds the rated voltage of the motor M or the upper limit output voltage of the inverter 1 is calculated, the terminal voltage of the motor M is the rated voltage of the motor M or The target current value Iγ * is controlled so as to be equal to or lower than the upper limit output voltage set in the inverter 1.

  As shown in FIG. 12, the inverter control device according to this embodiment includes an output voltage calculation unit 22 that calculates an output voltage command value Vout to the inverter 1 from a voltage command value Vγ and a voltage command value Vδ, and an output voltage command. A target gain calculation unit 24 for calculating a first target gain value Pvf * 1 for setting a deviation between the value Vout and a preset output voltage upper limit value to zero; a command voltage and a command frequency of the motor M (for example, rated The second target gain value (rated gain value) Pvf * 2 calculated in advance from the voltage and the rated frequency) is compared with the first target gain value Pvf * 1 calculated by the target gain calculator 24, and the larger one is calculated. And a comparator 23 that selects the target gain value Pvf * of the gain controller 20.

The output voltage calculation unit 22 calculates the output voltage command value Vout from the following equation.
Vout = (Vγ ² + Vδ ² ) ^1/2
The output voltage calculation unit 22 may calculate the output voltage command value Vout from the voltage command values on the fixed coordinates as well as the voltage command values Vγ and Vδ on the rotation coordinates, and hardware such as a pulse transformer may be used. It may be used to calculate the output voltage command value Vout.

  The predetermined output voltage upper limit value input to the target gain calculation unit 24 is a smaller value of the rated voltage of the motor M and the upper limit output voltage set in the inverter 1. The target gain value Pvf * selected by the comparator 23 is input to the gain control unit 20 described above. The gain control unit 20 sets the target current value Iγ so that the deviation between the target gain value Pvf * input via the comparator 23 and the magnetization current adjustment gain value Pvf output from the gain generation unit 11 becomes zero. * Control.

  When the output voltage command value Vout calculated by the output voltage calculation unit 22 exceeds the rated voltage of the motor M or the upper limit output voltage of the inverter 1, the comparator 23 sets the first target gain value Pvf * 1 to the gain control unit 20. Output to. Therefore, the target gain value Pvf * of the gain control unit 20 is increased. As a result, the output voltage of the inverter 1 is maintained below the output voltage upper limit value while allowing the motor M to rotate at high speed. As described above, according to the present embodiment, even when the motor M is rotated at a high speed, the terminal voltage of the motor M is maintained below a predetermined upper limit value by controlling the target gain value Pvf *. Can do.

  The output voltage upper limit value is a predetermined value transmitted from the output voltage upper limit value input unit 34 to the target gain calculation unit 24. The second target gain value Pvf * 2 is a predetermined value transmitted from the target gain input unit 35 to the comparator 23. The output voltage upper limit value input unit 34 and the target gain input unit 35 may each have a storage unit for storing the predetermined value, or may be obtained by calculation according to the operating state of the motor M.

  Next, another embodiment of the present invention will be described with reference to FIG. FIG. 13 is a block diagram showing a sensorless inverter control apparatus according to still another embodiment of the present invention. The configuration of the present embodiment that is not particularly described is the same as that of the above-described embodiment shown in FIG.

  As shown in FIG. 13, a lower limiter 25 is provided between the gain control unit 20 and the first current control unit 7. The lower limiter 25 defines a lower limit value of the target current value Iγ * from the gain control unit 20. By providing the lower limiter 25, the magnetizing current Iγ equal to or higher than the lower limit value always flows. As a result, since the current vector Id in the d-axis direction always exists, the position of the rotor can be easily estimated even in a region where the current is small such as a low speed region. Therefore, controllability in the low speed region is improved.

  The lower limit value is a predetermined value transmitted from the lower limit value input unit 38 to the lower limit limiter 25. The lower limit input unit 38 may include a storage unit that stores the predetermined value, or may be obtained by calculation in accordance with the operating state of the motor M.

  Next, another embodiment of the present invention will be described with reference to FIG. FIG. 14 is a block diagram showing a sensorless inverter control apparatus according to still another embodiment of the present invention. The configuration of the present embodiment that is not particularly described is the same as that of the above-described embodiment shown in FIG.

  As shown in FIG. 14, the inverter control device according to the present embodiment further includes a step-out determination unit 27 connected to the gain generation unit 11. The step-out determination unit 27 detects that the magnetizing current adjustment gain value Pvf generated by the gain generation unit 11 has reached a predetermined threshold value, thereby rotating the rotor rotational speed and the rotating magnetic field generated by the stator. Is determined to be out of synchronization with the rotation speed. When it is determined that the step-out has occurred, the operation of the motor M is stopped. Therefore, it is possible to prevent the inverter 1 from being destroyed and the motor M from operating abnormally due to the flow of an abnormal current caused by the step-out of the motor M.

  The threshold value is a predetermined value transmitted from the threshold value input unit 39 to the step-out determination unit 27. The threshold value input unit 39 may include a storage unit that stores the predetermined value, or may be obtained by calculation in accordance with the operating state of the motor M.

  Next, another embodiment of the present invention will be described with reference to FIG. FIG. 15 is a block diagram showing a sensorless inverter control apparatus according to still another embodiment of the present invention. The configuration of the present embodiment that is not particularly described is the same as that of the above-described embodiment shown in FIG.

  Although the sensorless inverter control device according to the above-described embodiment can be applied to both driving of a synchronous motor and an induction motor, the sensorless inverter control device according to the present embodiment is particularly preferably applied to driving of an induction motor. be able to. When driving an induction motor, an excitation current must be passed through the motor. Therefore, the induction motor is driven by setting the target current value iγ * to a value larger than zero.

  As shown in FIG. 15, a comparator 29 is provided between the gain control unit 20 and the first current control unit 7. The comparator 29 compares a predetermined target excitation current value iγ * 1 with the target current value iγ * 2 from the gain control unit 20, and the smaller one is the target current value iγ of the first current control unit 7. Select as *. The target excitation current value iγ * 1 is a value larger than zero. The selected target current value iγ * is input to the first current control unit 7.

  The target excitation current value iγ * is a predetermined value transmitted from the target excitation current input unit 40 to the comparator 29. The target excitation current input unit 40 may include a storage unit that stores the predetermined value, or may be obtained by calculation in accordance with the operating state of the motor M.

  The embodiment described above is described for the purpose of enabling the person having ordinary knowledge in the technical field to which the present invention belongs to implement the present invention. Various modifications of the above embodiment can be naturally made by those skilled in the art, and the technical idea of the present invention can be applied to other embodiments. Therefore, the present invention should not be limited to the described embodiments, but should be the widest scope according to the technical idea defined by the claims.

DESCRIPTION OF SYMBOLS 1 Inverter 2 Current detector 5 3 phase-2 phase coordinate transformation part 6 2 phase-3 phase coordinate transformation part 7 1st current control part 8 2nd current control part 11 Gain production | generation part 12 Multiplier 14 Speed control part 16 Integrator 20 gain control unit 22 output voltage calculation unit 23 comparator 24 target gain calculation unit 25 lower limit limiter 27 step-out determination unit 29 comparator 31 target gain input unit 32 target speed input unit 33 target magnetization voltage input unit 34 output voltage upper limit value input Unit 35 target gain input unit 38 lower limit value input unit 39 threshold value input unit 40 target excitation current input unit

Claims (11)

An inverter control device for controlling an inverter for driving a motor,
A current detector for detecting a three-phase current supplied from the inverter to the motor;
A three-phase to two-phase coordinate conversion unit for converting the three-phase current detected by the current detector into a magnetizing current and a torque current in a rotating coordinate system;
A first current control unit that calculates a first voltage command value for making a deviation between the magnetizing current and the first target current value zero;
A second current control unit for calculating a second voltage command value for making the deviation between the torque current and the second target current value zero;
A two-phase to three-phase coordinate converter for converting the first and second voltage command values into a three-phase voltage command value on a three-phase fixed coordinate system;
A gain generator for generating a magnetizing current adjustment gain value for setting the first voltage command value to zero;
A multiplier for calculating an estimated angular frequency of the motor from the magnetizing current adjustment gain value and the second voltage command value;
A speed control unit that calculates the second target current value for setting a deviation between the estimated angular frequency and a predetermined target angular frequency to zero;
An inverter control device comprising: a gain control unit that calculates the first target current value for setting a deviation between the magnetizing current adjustment gain value and the target gain value to zero. The inverter control device according to claim 1, wherein the magnetizing current adjustment gain value is a variable determined from an angular frequency of the motor and the second voltage command value .   The inverter control device according to claim 1, wherein the target gain value is a value determined from a voltage-frequency characteristic of the motor.   The inverter control device according to claim 3, wherein the target gain value is calculated from a rated voltage of the motor and a rated frequency of the motor.   The inverter control device according to claim 1, wherein the target gain value is a value calculated in advance from characteristics of a load connected to the motor. An output voltage calculation unit for calculating an output voltage command value to the inverter;
A target gain calculation unit for calculating a first target gain value for making a deviation between the output voltage command value and a predetermined output voltage upper limit value zero;
The gain control unit compares the second target gain value calculated in advance from the rated voltage of the motor and the rated frequency of the motor with the first target gain value, and sets the larger value as the target gain value. The inverter control device according to claim 1, further comprising a comparator that outputs to the inverter.   The inverter control apparatus according to claim 6, wherein the predetermined output voltage upper limit value is one of a rated voltage of the motor and an upper limit output voltage of the inverter.   The inverter control device according to claim 1, further comprising a lower limiter that defines a lower limit value of the first target current value calculated by the gain control unit.   The step-out determination unit that determines that step-out has occurred when the magnetizing current adjustment gain value generated by the gain generation unit reaches a predetermined threshold value. Inverter control device. The inverter control device according to claim 9, wherein when it is determined that the step-out has occurred, the operation of the motor is stopped. A comparator for comparing the first target current value calculated by the gain control unit with a predetermined target excitation current value and selecting a smaller value;
The first current control unit calculates the first voltage command value for making a deviation between the magnetization current and the first target current value or the target excitation current value selected by the comparator zero. The inverter control device according to claim 1.

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