JP5237210B2 - Position detection device - Google Patents

Description

  The present invention relates to a position detection apparatus that can be applied to both a rotational position and a linear position, such as a rotational position detector such as a resolver or a synchro, or a linear position detector according to a similar position detection principle, and more particularly to an electrical phase difference. The present invention relates to a position detection device that performs absolute position detection based on the above.

  An inductive rotational position detector that produces two-phase output (sine phase and cosine phase output) with one-phase excitation input is known as a “resolver”, and three-phase output (120 degrees with one-phase excitation input) What produces the three shifted phases) is known as “synchro”. The oldest conventional resolver has a secondary winding (sine pole and cosine pole) orthogonal to each other at a mechanical angle of 90 degrees on the stator side, and a primary winding on the rotor side. (The relationship between primary and secondary can be reversed). This type of resolver is disadvantageous because it requires a brush to make electrical contact with the rotor primary winding. On the other hand, the existence of a brushless resolver that does not require a brush is also known. The brushless resolver is provided with a rotary transformer in place of the brush on the rotor side. On the other hand, by changing the excitation method of the resolver so that a one-phase output is generated by a two-phase excitation input, an output signal including an electrical phase shift angle corresponding to the rotation angle is obtained, thereby obtaining digital data of the detected angle. The scheme is also known.

  Further, as a phase difference detection method in a position detection device using these electrical phase detection principles, it is shown in the following Patent Document 1 proposed by the inventors of the present application and Patent Document 2 corresponding to the US Patent. Other techniques are also known.

  In the techniques disclosed in Patent Documents 1 and 2, excitation is performed by a reference AC signal (for example, sin ωt), and amplitude modulation is performed using the first function value (for example, sine function value) corresponding to the detection target position as an amplitude coefficient. A first AC output signal (for example, sin θ sin ωt) and a second AC output signal (for example, cos θ sin ωt) that is amplitude-modulated using the second function value (for example, cosine function value) corresponding to the detection target position as an amplitude coefficient are output. Based on the first and second AC output signals output from a position sensor (for example, a resolver type sensor), positive and negative ones with respect to the reference AC signal by a shift amount (θ) corresponding to the position to be detected. Shift corresponding to the same detection target position as the first electrical AC signal (for example, sin (ωt + θ)) having the electrical phase angle shifted in the direction A second electrical AC signal (eg, sin (ωt−θ)) having an electrical phase angle shifted in other positive and negative directions relative to the reference AC signal by an amount, and In addition, by adding and synthesizing the positive and negative phase shift detection values (+ θ and −θ) included in the second electrical AC signal, highly accurate detection data from which the temperature characteristic error component has been canceled out is obtained. Trying to get.

  Patent Document 3 below discloses a position detection device that employs a phase difference detection method as described in Patent Documents 1 and 2 above. The system configuration of the position detection device shown therein is composed of a position sensor unit and a calculation device (computer or the like). In the position sensor unit, the first and second electrical AC signals (sin (ωt + θ) and sin (ωt−θ)) is synchronized with the zero-cross phase, and a zero-cross detection signal (latch timing signal) is output to an arithmetic device (such as a computer). The phase shift value including the count is calculated. However, in this configuration, two output wirings corresponding to two zero cross detection signals (latch timing signals) are necessary.

  Japanese Patent Application Laid-Open No. 2004-228842 is configured to output one PWM signal having a pulse width corresponding to the electrical phase difference at the detection position, thereby reducing the number of output wires and simplifying the configuration. A position detecting device is shown. That is, the PWM corresponding to the time difference Δt between the zero cross detection signal (latch timing signal) Lp and Lm synchronized with the zero cross phase of the first and second electrical AC signals (sin (ωt + θ) and sin (ωt−θ)). A signal is output. Incidentally, in this case, a PWM signal having no problem can be obtained if the electrical phase difference at the detection position corresponds to a phase difference of 180 degrees or less. The time difference Δt from the zero cross detection signal (latch timing signal) Lp of the static AC signal (sin (ωt + θ)) to the zero cross detection signal (latch timing signal) Lm of the second electrical AC signal (sin (ωt−θ)). The responded PWM signal indicates only a phase difference of less than 180 degrees, and accurate position detection cannot be performed. Therefore, in Patent Document 4, it is determined whether or not the detected electrical phase difference is within 180 degrees, but a specific configuration is not shown. In addition, the configuration of determining whether or not the detected electrical phase difference is within 180 degrees is essentially such a determination for a detection apparatus configured to be able to detect only within 180 degrees in the first place. Unless an appropriate solution was proposed, what to do was difficult to achieve.

JP-A-9-126809 US Pat. No. 5,710,509 JP 2002-131084 A JP 2006-214948 A

  The present invention has been made in view of the above-described points. The electrical AC signal sin (ωt + θ) phase-shifted by the phase shift amount θ corresponding to the detection target position and the electrical AC signal sin delayed-shifted. An object of the present invention is to provide a position detection device capable of accurately measuring a phase time difference Δt of 180 degrees or more in a position detection device of a type that detects a phase time difference Δt with respect to (ωt−θ).

  The position detection device according to the present invention is excited by a reference AC signal, and is modulated by using a first function value corresponding to the detection target position as an amplitude coefficient, and the first AC output signal corresponding to the detection target position. A position sensor that outputs a second AC output signal that is amplitude-modulated with the function value of 2 as an amplitude coefficient, and based on the first and second AC output signals, the reference is shifted by a shift amount corresponding to the detection target position. A first electrical AC signal having an electrical phase angle shifted in one direction positive and negative with respect to the AC signal, and positive and negative with respect to the reference AC signal by a shift amount corresponding to the same detection target position. A circuit for generating a second electrical AC signal having an electrical phase angle shifted in the negative negative direction, and a timing of a reference phase of the first electrical AC signal as a starting point, Reference phase The timing of the reference phase of the second electrical AC signal after the elapse of timing is used as an end point, and the first electrical AC signal and the second electrical AC signal are transmitted at a time interval from the start point to the end point. And a phase output circuit that generates a phase output signal indicating an electrical phase difference.

  The second electrical AC signal (sin (ωt−) after the timing of the reference phase of the reference AC signal (sin ωt) has elapsed, starting from the reference phase timing of the first electrical AC signal (sin (ωt + θ)). A phase output signal indicating an electrical phase difference between the first electrical AC signal and the second electrical AC signal at a time interval Δt from the start point to the end point with the reference phase timing of θ)) as an end point. Since it is generated, a phase time difference Δt of 180 degrees or more can be accurately detected.

The figure which shows one Example of this invention comprised as a rotational position detection apparatus of a leakage magnetic flux detection type.

The front schematic diagram which shows the example which made the shape of the high magnetic permeability magnetic body different from the example of FIG. 1 as a modification of a coil part.

The figure which shows the example of the electric circuit for a detection contained in the position detection apparatus of this invention.

The block diagram which shows an example of the system formed by connecting the output of the position detection apparatus of this invention to a microcomputer.

FIG. 4 is a timing chart for explaining that temperature drift compensation of detection data can be performed by the detection circuit configuration of FIG. 3.

The block diagram which shows another structural example of the detection system formed by connecting the output of the position detection apparatus of this invention to a microcomputer.

The timing chart which shows the operation example which counts the time difference of an electrical phase angle.

The flowchart which shows an example of the process which performs the time difference count operation | movement according to FIG.

FIG. 8 is a logic circuit diagram showing an example of a PWM conversion circuit configured to perform a time difference counting operation according to FIG. 7.

The figure which shows another structural example of the coil part of a leakage position detection type rotational position detection apparatus.

The figure which shows another example of arrangement | positioning of a coil and a high magnetic permeability magnetic body.

The figure which shows another structural example of the position detection apparatus which concerns on this invention.

  FIG. 1 is a diagram showing a configuration example of a position sensor 10 applied in a position detection device according to an embodiment of the present invention configured as a leakage magnetic flux detection type rotational position detection device, and (a) is a schematic front view. (B) is a schematic side view. The rotor (rotor) 2 attached to the rotation shaft 1 to be detected includes a predetermined magnetic response member. As an example of the magnetic response member, 180 N poles and S poles of the permanent magnet MG are provided. The thing which consists of the structure arrange | positioned by the space | interval of a degree is used. In that case, the magnetic field generated by the permanent magnet MG of the rotor 2 rotates for two cycles per rotation of the rotor 2 of the rotating shaft 1. The position sensor 10 detects the leakage magnetic flux of the permanent magnet MG of the rotor 2, and determines the rotational position of the rotating shaft 1 by determining the current angle of the magnetic field generated by the permanent magnet MG of the rotor 2.

  In FIG. 1A, the position sensor 10 includes, as the stator 3, four-pole flat coils 11, 12, 13, and 14 arranged at a predetermined angular interval on a ring-shaped circuit board 15. For example, the flat coils 11, 12, 13, and 14 of each pole are made of spiral printed wiring, and a plurality of spiral printed wiring coils are stacked to increase the inductance. As shown in FIG. 1B, the ring-shaped circuit board 15 of the stator 3 is fitted around the rotor 2. As shown in FIG. 1B, high permeability magnetic bodies 16 are arranged on both surfaces of the substrate 15 corresponding to the arrangement of the flat coils 11, 12, 13, 14. Each high magnetic permeability magnetic body 16 is made of a high magnetic permeability material such as amorphous, and forms a thin plate or thin film. Such a high magnetic permeability material functions as a flux gate. That is, when the magnetic pole of the permanent magnet MG comes close to the high magnetic permeability material, the high magnetic permeability material has a property of causing magnetic saturation under the influence of a large magnetic flux and greatly reducing the magnetic permeability. Accordingly, the high permeability magnetic body 16 in each flat coil 11, 12, 13, 14 changes the permeability according to the proximity of the magnetic poles S, N of the permanent magnet MG of the rotor 2, and the corresponding coil 11, 12. , 13 and 14 cause an impedance change corresponding thereto. By this flux gate principle, the magnetic flux leaking from the permanent magnet of the rotor 2 can be detected by the coils 11, 12, 13, 14. The high magnetic permeability magnetic body 16 may be disposed only on one side of the substrate 15 in correspondence with the arrangement of the flat coils 11, 12, 13, and 14.

  For example, as shown in the figure, when the magnetic field due to the magnetic poles S and N of the permanent magnet of the rotor 2 is rotated for two cycles per rotation of the rotor 2, the rotation of the rotor 2 is assumed to be φ. Each high permeability magnetic body 16 in each flat coil 11, 12, 13, 14 has a periodicity twice as large as φ according to the leakage flux of the permanent magnet corresponding to the angle φ (that is, the rotation angle Magnetic saturation (correlated with φ) is generated, and the impedance of each of the coils 11, 12, 13, and 14 is changed in accordance with the periodic change of the magnetic saturation.

  Here, by appropriately setting the shape of the high-permeability magnetic body 16, the characteristic of the impedance change generated in the first coil 11 (sine pole S) according to the rotation angle φ of the rotor 2 is equivalent to the sine function sin θ. Or it can be made close to it. Note that θ = 2φ. Further, in the second coil 12 (minus sign pole -S) arranged at a mechanical angle of about 90 degrees clockwise from the first coil 11, the characteristic of impedance change according to the rotation angle φ of the rotor 2 is present. It can be made equal to or close to the minus sine function −sin θ. Similarly, in the third coil 13 (cosine pole C) arranged with a mechanical angle of about 225 degrees clockwise from the first coil 11, the characteristic of impedance change according to the rotation angle φ of the rotor 2 is cosine. It can be made to be equivalent to or close to the function cos θ. Further, in the fourth coil 14 (minus cosine pole -C) arranged with a mechanical angle of about 315 degrees clockwise from the first coil 11, the characteristic of impedance change according to the rotation angle φ of the rotor 2 is present. It can be made equal to or close to the minus cosine function −cos θ. In addition, the mechanical arrangement | positioning of each coil 11-14 is not restricted to the said example, A design change is possible suitably. In short, it is sufficient that the characteristics of the impedance change corresponding to the rotation angle φ of the rotor 2 in each of the coils 11 to 14 are sin θ, −sin θ, cos θ, and −cos θ, respectively.

  In addition, the shape of the high magnetic permeability magnetic body 16 is substantially circular according to the circular shape of each flat coil 11, 12, 13, 14 in FIG. Even such a substantially circular shape is a shape that gradually changes with respect to the displacement direction of the rotor 2, so that the change characteristic of magnetic saturation exhibits a sinusoidal characteristic with the displacement of the adjacent rotor 2. May be possible. However, more preferably, as shown in FIG. 2, the shape of the high permeability magnetic body 16 may have a tapered shape like an eye contour. In any case, as described above, the characteristics of the impedance change corresponding to the rotation angle φ of the rotor 2 in each of the coils 11 to 14 are sin θ, −sin θ, cos θ, and −cos θ, respectively, or as close to them as possible. Thus, the shape of the high permeability magnetic body 16 is a shape that gradually changes in the displacement direction of the rotor 2.

  As described above, the first and second coils 11 and 12 are arranged so as to be in opposite phases to the change in leakage magnetic flux generated from the rotor 2, and the third and fourth coils 13 and 14 are also formed. These are arranged so as to be in opposite phases to the change in leakage magnetic flux generated from the rotor 2. Then, the set of the first and second coils 11 and 12 and the set of the third and fourth coils 13 and 14 have a predetermined phase (π / 2) with respect to the change in leakage magnetic flux generated from the rotor 2. It is arranged so as to be displaced.

FIG. 3 shows an example of an electric circuit for detection applied to the position sensor 10. In FIG. 3, the coils 11 to 14 are equivalently shown as variable inductance elements. Each of the coils 11 to 14 is excited in one phase by a predetermined high-frequency AC signal (indicated by Esinωt for convenience) given from the reference AC signal source 40. The voltages Va, Vc, Vb, and Vd generated in the coils 11 to 14 are equivalent to the coils according to the angle variable θ (= 2φ) corresponding to the rotational position φ of the rotor 2 to be detected, as described below. The magnitude | size according to the impedance value for every 11-14 is shown. P ₀ is a constant element of the magnetic circuit.
Va = (P ₀ + sin θ) sin ωt
Vb = (P ₀ + cos θ) sinωt
Vc = (P ₀ -sin θ) sin ωt
Vd = (P ₀ −cos θ) sinωt

The analog computing unit 31 obtains the difference between the output voltage Va of the coil 11 corresponding to the sine phase and the output voltage Vc of the coil 12 corresponding to the minus sine phase that changes differentially with respect to the angle, as described below. An AC output signal having an amplitude coefficient with a sine function characteristic of the variable θ is generated.
Va−Vc = (P ₀ + sin θ) sin ωt− (P ₀ −sin θ) sin ωt
= 2sinθsinωt
The analog computing unit 32 obtains the difference between the output voltage Vb of the coil 13 corresponding to the cosine phase and the output voltage Vd of the coil 14 corresponding to the minus cosine phase that changes differentially with respect to the angle, as described below. An AC output signal having an amplitude coefficient of the cosine function characteristic of the variable θ is generated.
Vb−Vd = (P ₀ + cos θ) sin ωt− (P ₀ −cos θ) sin ωt
= 2 cos θ sin ωt

  In this way, two AC output signals “2 sin θ sin ωt” and “2 cos θ sin ωt” that are respectively amplitude-modulated by two periodic amplitude functions (sin θ and cos θ) including the angular variable θ correlated with the rotation position φ to be detected are obtained (hereinafter referred to as “2 cos θ sin ωt”) The coefficient “2” is omitted.) This is equivalent to the sine phase output signal sin θ sin ωt and the cosine phase output signal cos θ sin ωt of a detector conventionally known as a resolver. The names sine phase and cosine phase, and the sine and cosine representations of the amplitude functions of the two AC output signals are for convenience. If either one is a sine and the other is a cosine, it will You can say. That is, Va−Vc = cos θ sin ωt, and Vb−Vd = sin θ sin ωt may be expressed.

  The reference AC signal source 40 and the analog arithmetic units 31 and 32 excite the coils 11 to 14 with a predetermined reference AC signal, and output signals corresponding to the impedances of the coils 11 to 14 are output to the coils 11 to 14. And the first and second coils 11 and 12 are differentially combined to generate a first combined output AC signal sinθsinωt, and the outputs of the third and fourth coils 13 and 14 This corresponds to a sensor circuit that differentially combines the signals to generate a second combined output AC signal cos θ sin ωt.

  Here, the compensation of the temperature drift characteristic will be described. The impedances of the coils 11 to 14 change according to the temperature, and the output voltages Va to Vd also change. However, in the AC output signals sin θ sin ωt and cos θ sin ωt having the sine and cosine function characteristics obtained by calculating and synthesizing these, the temperature drift error of the coil is compensated by the calculation of “Va−Vc” and “Vb−Vd”. It is not affected by the coil impedance change due to. Therefore, accurate detection is possible. In addition, temperature drift characteristics in other circuit parts such as the reference AC signal source 40 are automatically compensated as will be described later.

  In this embodiment, the rotational position is detected by the phase detection method based on the two AC output signals sin θ sin ωt and cos θ sin ωt output from the calculators 31 and 32. As a phase detection method in this case, for example, a technique disclosed in Japanese Patent Laid-Open No. 9-126809 may be appropriately used. For example, one AC output signal sin θ sin ωt is electrically shifted 90 degrees by the shift circuit 33 to generate an AC signal sin θ cos ωt, and the other AC output signal cos θ sin ωt is added and synthesized by the analog adder 34. An AC signal (a signal obtained by converting the phase component θ into an AC phase shift), which is sin (ωt + θ), is phase-shifted in the plus direction (advanced phase) according to θ.

  On the other hand, by subtracting and synthesizing the AC output signal sin θ cos ωt output from the shift circuit 33 and the other AC output signal cos θ sin ωt by the analog subtractor 36, sin (ωt−θ), which is a negative direction according to θ ( An AC signal (a signal obtained by converting the phase component θ into an AC phase shift) that is phase-shifted to a late phase is generated.

  The shift circuit 33, the analog adder 34, and the analog subtractor 36 are based on the first and second combined output AC signals sin θ cos ωt and cos θ sin ωt, and are based on a shift amount θ corresponding to the rotational position φ of the rotor 2. The first AC signal sin (ωt + θ) having an electrical phase angle shifted in one direction positive and negative with respect to the AC signal sin ωt, and the reference AC by the shift amount θ corresponding to the same rotation position φ to be detected. This corresponds to a signal generation circuit that generates a second electrical AC signal sin (ωt−θ) having an electrical phase angle shifted in the positive and negative directions with respect to the signal sinωt.

  Then, the zero cross of the AC detection signal sin (ωt + θ) shifted in phase is detected by the comparator 35, and a zero cross detection pulse Lp is generated. Further, the zero cross of the AC detection signal sin (ωt−θ) shifted in phase is detected by the comparator 37 to generate a zero cross detection pulse Lm. The zero cross detection pulse Lp indicates the timing of the reference phase (phase angle 0) of the first electrical AC signal sin (ωt + θ). The zero cross detection pulse Lm indicates the timing of the reference phase (phase angle 0) of the second electrical AC signal sin (ωt−θ).

  The phase shift zero-cross detection pulse Lp output from the comparator 35 indicates the phase shift amount θ in the phase-shifted AC detection signal sin (ωt + θ) as the advance time position with respect to the zero phase point of the reference AC signal sin ωt. Corresponds to the signal.

  Further, the late-shift zero-cross detection pulse Lm output from the comparator 37 determines the phase shift amount θ in the late-shifted AC detection signal sin (ωt−θ) from the zero-phase point of the reference AC signal sinωt. It corresponds to a timing signal indicated by a time position.

  Thus, both the phase shift zero cross detection pulse Lp and the phase shift zero cross detection pulse Lm are detection data indicating the phase shift amount θ corresponding to the rotational position φ of the rotor 2 in the time position. Therefore, in principle, either one of the phase shift zero cross detection pulse Lp and the phase shift zero cross detection pulse Lm may be output as a detection signal of the rotational position φ of the rotor 2. However, in order to obtain detection data compensated for temperature drift as described later, it is preferable to use both.

  Each of the circuits 31 to 37 and 40 shown in FIG. 3 is accommodated as a unit on one circuit board 15 and grouped as a circuit unit. The circuit unit and the position sensor 10 are accommodated in the casing and attached to the rotary shaft 1 as a detection target. FIG. 4 shows a system configuration example in which a position detection unit 100 including a position sensor 10 and a circuit unit is connected to a microcomputer 20 for using the detection output. The microcomputer 20 and the circuit shown in FIG. 3 need only be connected by at least a DC power supply line and two output lines 21a and 21b. The phase shift zero cross detection pulse Lp and the phase shift zero cross detection pulse Lm are output to the two output lines 21a and 21b, respectively. The microcomputer 20 has a plurality of input ports for timing signal capture, and the output lines 21a and 21b are connected to the input ports, respectively. The microcomputer 20 counts the time difference Δt between the two timing signals (pulses Lp and Lm) given from the lines 21a and 21b connected to the input port, whereby the phase angle data θ corresponding to the rotational position φ of the rotor 2 is obtained. Is measured digitally. In the case of this example, the phase angle data θ represents the rotational position as an absolute value within the one cycle range with the rotation range of the rotor 2 of 180 degrees as one cycle. A rotation range exceeding 180 degrees can be detected by incrementing the number of cycles.

  The comparators 35 and 37 and the microcomputer 20 count Δt are the reference phase of the reference AC signal sin ωt, the reference phase of the first electrical AC signal sin (ωt + θ), and the second electrical AC signal sin (ωt− A phase output circuit that generates a phase output signal indicating an electrical phase difference between the first electrical AC signal sin (ωt + θ) and the second electrical AC signal sin (ωt−θ) based on the reference phase of θ). It corresponds to. Of course, instead of the microcomputer 20, a digital circuit that counts the time difference Δt between the pulses Lp and Lm may be used.

  In the microcomputer 20, it is only necessary to count the time difference Δt between the two timing signals (pulses Lp and Lm) given from the lines 21 a and 21 b, and the microcomputer 20 generates the reference AC signal source 40 provided in the sensor 10. There is no need to know the zero phase time point of the reference AC signal sinωt. Therefore, the processing / configuration for time measurement on the computer 20 side is simplified. On the other hand, in the circuit in the sensor 10 housed in the casing, the reference AC signal source 40 is based on a DC power source supplied from the outside by an analog oscillation circuit or using a sine wave function generator or the like. It is only necessary to generate the AC signal sin ωt, and it is not necessary to provide this to the microcomputer 20 as a reference signal for synchronization. Therefore, the configuration of the external terminals can be simplified, leading to cost reduction.

  Here, compensation for temperature drift characteristics will be described again. Due to the temperature drift characteristic, for example, when the frequency or amplitude level of the AC signal generated by the reference AC signal source 40 varies, or when the impedance of other circuit elements or signal lines varies, the detected phase advance shift is detected. Each phase shift value θ in the zero-cross detection pulse Lp and the zero-shift detection pulse Lm of the late shift includes an error ε due to temperature drift characteristics. However, since this error ε appears in the same direction and the same direction (same value and same sign) in both detection pulses Lp and Lm, the error ε is automatically canceled at the time difference Δt between the two detection pulses Lp and Lm (timing signal). Will be. Therefore, it is possible to perform highly accurate detection without being affected by impedance change of a circuit or the like due to temperature drift.

  FIG. 5 is a timing chart schematically showing the temperature compensation. (A) shows an example of the generation timing of detection pulses Lp, Lm (timing signal) when there is no error ε due to temperature drift, and the time difference Δt is theoretically 2θ, indicating an accurate rotational position. (B) shows an example of generation timing of detection pulses Lp and Lm (timing signal) when there is an error ε due to temperature drift. In this case, the phase detection pulse Lp is generated ahead of the zero phase time point of the reference AC signal by an advance time corresponding to “+ θ−ε” including the error ε, and the phase detection pulse Lm is The delay time is delayed by a delay time corresponding to “−θ−ε” including the error ε with respect to the zero phase time point of the reference AC signal. However, even if each of the two detection pulses Lp and Lm (timing signal) includes the error ε in this way, the error ε is automatically canceled at the time difference Δt between the two, and the theory indicates an accurate rotational position. This corresponds to the value 2θ. Thus, temperature drift compensation has been achieved.

  FIG. 6 shows another example of a circuit configuration mounted in the position detection unit 100. In the example of FIG. 6, the circuit configuration in the position detection unit 100 has a pulse width corresponding to the time difference Δt between the advanced detection pulse Lp and the delayed detection pulse Lm in addition to the circuit configuration of FIG. A PWM conversion circuit 38 for forming the variable pulse width signal PWM is further provided. A variable pulse width signal PWM having a pulse width corresponding to the time difference Δt formed by the PWM conversion circuit 38 is output via one output line 21 c and input to the microcomputer 20. The microcomputer 20 has an input port for capturing a PWM signal, and the output line 21c is connected to the input port. The microcomputer 20 digitally measures the phase angle data θ corresponding to the rotational position φ of the rotor 2 by counting the pulse time width Δt of the PWM signal from the line 21c connected to the input port. In this circuit configuration example, since only one output line 21c is required, the configuration can be further simplified.

  By the way, as described above, when counting the time difference Δt between the pulses Lp and Lm, as shown in FIG. 7A, the first electrical alternating current with respect to the reference phase (0 degree) of the reference alternating current signal sinωt. The electrical phase angle shift + θ (that is, the timing of Lp) of the signal sin (ωt + θ) and the electrical phase angle shift −θ (that is, the timing of Lm) of the second electrical AC signal sin (ωt−θ) are within 180 degrees. Then, accurate phase angle shift data θ (that is, 2θ) can be obtained by simply counting the time difference Δt between the pulses Lp and Lm.

On the other hand, as shown in FIG. 7B, the electrical phase angle shift + θ (that is, Lp ₁ of the first electrical AC signal sin (ωt + θ) with respect to the reference phase (0 degree) of the reference AC signal sin ωt. Timing) and the electrical phase angle shift −θ (that is, the timing of Lm ₁ ) of the second electrical AC signal sin (ωt−θ) is 180 degrees or more, simply calculate the time difference Δt between the pulses Lp and Lm. If counted, the time difference between the timing of the pulse Lp ₁ of the current cycle and the pulse Lm ₀ of the previous cycle is counted, and accurate phase angle shift data θ (that is, 2θ) cannot be obtained. Therefore, even in such a case, it is a new problem to obtain accurate phase angle shift data θ (that is, 2θ).

In order to solve such a problem, in the embodiment of FIG. 4, a reference zero cross pulse R ₀ indicating the reference phase (0 degree) of the reference AC signal sin ωt is generated on the position detection unit 100 side, and this is generated by a microcomputer. For example, the time difference Δt may be counted by the procedure shown in FIG. 8 on the microcomputer 20 side. In FIG. 8, in essence, the timing of the reference phase of the first AC signal sin (ωt + θ) (that is, the timing of Lp) starts and the timing of the reference phase of the reference AC signal sinωt (R ₀ ) has elapsed. The second electrical AC signal sin (ωt−θ) of the reference phase (that is, the timing of Lm) is set as the end point, and the first electrical AC signal and the second signal at the time interval Δt from the start point to the end point. A phase output signal indicating the electrical phase difference (2θ) of the electrical AC signal is generated.

The process of FIG. 8 starts at the generation timing of the reference zero cross pulse _R0 . In step S1, it is determined whether Δt is being counted (a count flag is set). If YES, go to step S2 to determine whether the generation timing of Lm has arrived. When the generation timing of Lm is reached, the counting operation of Δt is finished in step S3. In this case, the count data of Δt is stored as rotation position detection data, and the count flag is reset. Thereafter, in step S4, it is determined whether the generation timing of Lp has arrived. If the generation timing of Lp has arrived, the count operation of Δt is started in step S5, and the count flag is set. The process proceeds to steps S4 and S5 through YES in step S3, as in the example of FIG. 7A, the first electrical AC signal with respect to the reference phase (0 degree) of the reference AC signal sin ωt. The electrical phase angle shift + θ (that is, the timing of Lp) of sin (ωt + θ) and the electrical phase angle shift −θ (that is, the timing of Lm) of the second electrical AC signal sin (ωt−θ) are within 180 degrees. This is the case.

If NO in step S4, it is determined in step S6 whether the generation timing of the reference zero cross pulse _R0 has arrived. If not, the process returns to step S4. That is, the processing is performed according to the earlier occurrence of Lp or R ₀ by circulating between step S4 and step S6. If Lp occurs before R ₀ occurs, the process proceeds to step S5 as described above.

On the other hand, if _R0 occurs before Lp occurs, YES is determined in step S6, and the routine jumps to the start of the routine of FIG. In this case, since Δt is not being counted (the count flag is reset), step S1 is NO and the process proceeds to step S4. In step S4, it is determined whether the generation timing of Lp has arrived. If the generation timing of Lp has arrived, the count operation of Δt is started in step S5, and the count flag is set. Then, this routine is ended, and the process waits until R ₀ occurs next time. The processing of steps S4 and S5 from NO in step S1 is performed for the first electrical AC signal sin with respect to the reference phase (0 degrees) of the reference AC signal sin ωt, as in the example of FIG. 7B. When (ωt + θ) electrical phase angle shift + θ (that is, Lp timing) and second electrical AC signal sin (ωt−θ) electrical phase angle shift −θ (that is, Lm timing) are 180 degrees or more. It is.

On the other hand, in order to solve the above-described problems, in the embodiment of FIG. 6, the PWM conversion circuit 38 provided on the position detection unit 100 side uses the first method similar to the processing concept of FIG. The second electrical AC signal sin () after the elapse of the reference phase timing (R ₀ ) of the reference AC signal sin ωt, starting from the reference phase timing (ie, Lp timing) of the electrical AC signal sin (ωt + θ). The logic circuit may be configured to generate the PWM signal such that the timing of the reference phase (ωt−θ) (that is, the timing of Lm) is the end point. For this purpose, the PWM conversion circuit 38 may be configured as shown in FIG. 9, for example. In FIG. 9, the zero cross detection pulse Lp of the first electrical AC signal sin (ωt + θ) is input to the set input S of the flip-flop 381, and the set output Q of the flip-flop 381 is input to the AND gate 382. A pulse R ₀ generated at the timing of the reference phase (0 degree) of the AC signal sin ωt is input to the other input of the AND gate 382, and the output of the AND gate 382 is input to the set input S of the flip-flop 383. The set output Q of the flip-flop 383 is input to the AND gate 384, and the zero cross detection pulse Lm of the second electric AC signal sin (ωt−θ) is input to the other input of the AND gate 384, The output is input to the reset input R of the flip-flop 381. The set output Q of the flip-flop 381 is output to the output line 21c as a PWM signal indicating a time difference Δt according to the rotational position. A signal obtained by inverting the set output Q of the flip-flop 381 by the inverter 385 is input to the reset input R of the flip-flop 383. According to this configuration, the flip-flop 381 is set at the generation timing of the zero-crossing detection pulse Lp of the first electrical AC signal sin (ωt + θ), and the reference phase (0 degree) timing is set while the flip-flop 381 is set. When the pulse R ₀ is generated, the AND condition of the AND gate 382 is satisfied, the flip-flop 383 is set, and the zero cross of the second electrical AC signal sin (ωt−θ) is set with the flip-flop 383 set. When the detection pulse Lm is generated, the AND condition of the AND gate 384 is satisfied and the flip-flop 381 is reset. As a result, the second phase after the reference phase timing (R ₀ ) of the reference AC signal sin ωt has elapsed with the reference phase timing (that is, Lp timing) of the first electrical AC signal sin (ωt + θ) as the starting point. The PWM signal can be generated so that the timing of the reference phase of the electrical AC signal sin (ωt−θ) (that is, the timing of Lm) is the end point. When the PWM signal falls to 0, the flip-flop 383 is reset by the output 1 of the inverter 385.

  The configuration of the detection coils 11 to 14 in the position sensor 10 is not limited to the flat coil type formed on the printed circuit board as in the above embodiment, and may be any type. For example, FIG. 10 is of a type in which coils 11 to 14 are wound around a core 26 of a high permeability magnetic material. Or as shown in FIG. 11, the structure which arrange | positions the high permeability magnetic body 16 of predetermined shape adjacent to the single coil 11 may be sufficient. In the fluxgate type sensor, since the detection can be performed in response to the leakage magnetic flux sensitively, the relationship between the direction of the leakage magnetic flux generated from the rotor 2 that is the generation source of the leakage magnetic flux and the direction of the axis of the coil Has a high degree of freedom and a high tolerance of coil arrangement.

  The present invention is not limited to the above-described fluxgate type position sensor, and can be applied to any type of position sensor. For example, as a magnetic response member of a position sensor, not a permanent magnet as described above, but a known magnetic material (high magnetic material such as iron) or diamagnetic material (a highly conductive and non-magnetic material such as copper). Material) or a combination of a magnetic material and a diamagnetic material may be used.

  FIG. 12 is a schematic view showing an embodiment of a position sensor applicable to the present invention, and includes four-pole (S, -S, C, -C) coils 11 to 14 arranged on a stator, and a rotor. An example of a physical arrangement relationship with a magnetic response member 300 (magnetic body or diamagnetic body or a combination of a magnetic body and a diamagnetic body) formed and arranged on the surface of 200 is shown by a schematic front view. . A magnetic response member 300 having a predetermined shape (for example, an eccentric shape) is formed on the surface of the rotor 200, and the magnetic response member 300 rotates in accordance with the rotation of the rotor 2. The stator includes four coils 11 to 14 as detection coils. In this example, the magnetic flux passing through the coils 11 to 14 is directed in the axial direction of the rotary shaft 1. The stator and the rotor 200 are a state in which the end surfaces of the coil cores (for example, magnetic cores such as iron cores) of the coils 11 to 14 of the stator and the magnetic response member 300 formed on the surface of the rotor 200 are spaced apart from each other. In other words, the gaps are formed between the end faces of the coil cores of the coils 11 to 14 and the surface of the magnetic response member 300 of the rotor 200 so as to be opposed to each other, and the rotor 200 is located with respect to the stator. Rotates without contact. When the area of the end surface of the coil core of each of the coils 11 to 14 facing the magnetic response member 300 changes according to the rotational position of the rotor 200, the rotational angle of the rotary shaft 1 can be detected. In FIG. 12, in the stator, the coil 11 of the sine pole S and the coil 12 of the minus sine pole -S are arranged at a mechanical angle of 180 degrees, and the coil 13 of the cosine pole C and the minus cosine pole -C. The coil 14 is also arranged at a mechanical angle of 180 degrees, and the coil 11 of the sine pole S and the coil 13 of the cosine pole C are arranged at a mechanical angle of 90 degrees. As a result, the rotation angle φ of 360 degrees of the rotating shaft 1 is detected in an absolute manner at an electrical phase angle θ of 360 degrees. That is, the relationship θ = φ.

  The present invention is not limited to rotational position detection, and can be applied to various types of position detection devices such as linear position detection, inclination angle detection, and twist angle detection. In the above embodiment, the detection coil is configured to detect the impedance of an AC-excited coil. However, the configuration is not limited to this, and the primary coil and the induction output that are AC-excited for each pole are used. You may employ | adopt the structure containing the secondary coil to produce.

2 Rotor
3 Stator 10 Sensor (position detection device)
11, 12, 13, 14 Coil 15 Circuit board 16 High permeability magnetic body MG Permanent magnet 300 Magnetic response member

Claims (2)

A first AC output signal excited by a reference AC signal and amplitude-modulated with a first function value corresponding to the detection target position as an amplitude coefficient and an amplitude with the second function value corresponding to the detection target position as an amplitude coefficient A position sensor for outputting a modulated second AC output signal;
Based on the first and second AC output signals, a first electric having an electrical phase angle shifted in one of positive and negative directions with respect to the reference AC signal by a shift amount corresponding to the detection target position. And a second electrical AC signal having an electrical phase angle shifted in other positive and negative directions with respect to the reference AC signal by a shift amount corresponding to the same detection target position. Circuit,
The starting point is the timing of the reference phase of the first electrical AC signal, and the starting point is the timing of the reference phase of the second electrical AC signal after the timing of the reference phase of the reference AC signal has elapsed. And a phase output circuit that generates a phase output signal indicating an electrical phase difference between the first electrical AC signal and the second electrical AC signal at a time interval from the first to the end point. Detection device.   The position detection device according to claim 1, wherein the phase output circuit generates a PWM signal indicating a phase difference between the first electrical AC signal and the second electrical AC signal as the phase output signal.

Images