JP5178335B2 - AC motor control device - Google Patents

Description

  The present invention relates to a current estimation method in an AC motor control device.

  Conventionally, as a means for obtaining current information necessary for controlling an electric motor, there is a one-shunt current detection method in which a phase current sensor is omitted and a DC shunt current of an inverter is detected to reproduce the phase current of the electric motor.

  However, with the one-shunt current detection method, it is necessary to detect different two-phase currents with fine timing, and if the pulsed current becomes smaller than the minimum detectable width, the current cannot be detected. There is.

  Therefore, methods for obtaining current information for two phases have been studied by creating conditions that allow current detection, such as by creating a voltage command for correcting the pulse width of the inverter or by switching the inverter switching pattern. It was.

  However, these methods do not change the point of managing fine timing in order to detect current. For this reason, the calculation to correct the voltage command and the calculation to switch the switching pattern increase the amount of calculation, so that sufficient calculation capability and a highly accurate timer to detect at precise timing are required. I had to choose a functional microcomputer.

  However, there is a problem that the cost becomes high in a high-performance microcomputer. In addition, when a fan or pump drive is considered as an application, it also hinders circuit scale reduction.

Against this background, a current detection method that simplifies the current detection timing has been desired.
In Patent Document 1, as a method for simplifying the detection timing, a method of fixing the timing is proposed, and a current value in the vicinity of the intermediate time of each conduction period is sampled with respect to a pulsed current. A method of performing motor control using the current value I0s obtained in this manner is disclosed.

  This current value I0s is the current of the maximum phase of the voltage command, detects the effective component and the invalid component from the current value I0s, and then estimates the d-axis current as the excitation component and the q-axis current as the torque component. Calculation is performed to control the motor.

JP 2007-221999 A

  In the conventional method, the current for the effective portion and the portion for the ineffective portion are converted into the d-axis current as the excitation component and the q-axis current as the torque component based on the voltage phase. However, during the acceleration / deceleration operation of the motor, an axis error that is a phase deviation between the reference axis of control and the magnetic flux axis of the motor is generated. Therefore, the detected current value on the control axis is the d-axis current on the motor shaft and the q-axis. It will be different from the current. As a result, an error is generated in the current components of the control shaft and the motor shaft, so that acceleration / deceleration performance and operation efficiency are deteriorated. It was necessary to solve these problems.

  In order to solve the above-mentioned problem, the motor control device of the present invention newly considers a shaft error in addition to the voltage phase when calculating the current detection value on the control shaft. By considering the axis error, the current value matching the d-axis current and the q-axis current on the motor shaft is estimated even during acceleration / deceleration operation.

  According to the present invention, it is possible to accurately estimate and calculate a d-axis current that is an excitation component of a motor and a q-axis current that is a torque component. As a result, the performance of acceleration / deceleration operation is improved, and highly responsive operation characteristics and high efficiency operation can be realized.

  The present invention detects the effective current and reactive current of the motor from the phase of the output voltage of the inverter from the input current detection value of the inverter that drives the AC motor, and the excitation current component and the torque current component of the current detection value are detected. A vector control device that performs a current estimation calculation, wherein a current detection value of an effective part and an ineffective part of a motor, and a phase value of an inverter output voltage that is a first phase value, a magnetic flux axis of the motor, a control reference axis, The second phase value obtained by adding these phase error values is used to estimate at least one of the d-axis current that is the excitation component of the motor and the q-axis current that is the torque component.

  In addition, the present invention is characterized in that the second phase value is a low-pass filter value obtained by adding the first phase value and the phase error value.

  Further, the phase of the inverter output voltage is obtained by the arc tangent of the ratio between the d-axis voltage command value and the q-axis voltage command value obtained by vector control calculation.

  Further, the phase error value is calculated using a voltage command value calculated by vector control, a frequency calculation value or a frequency command value, and a current estimation value of the d-axis current or the q-axis current obtained by the estimation calculation. It is characterized by that.

  Further, according to the present invention, the calculation of the vector control of the inverter calculates the voltage command value of the d-axis and the q-axis using the current estimated value of the d-axis current or the q-axis current, And calculating the d-axis and q-axis voltage command values based on the deviation between the current command value of the q-axis and the current estimate value of the d-axis current or the q-axis current, or the d-axis and Based on the deviation between the q-axis current command value and the d-axis current or the estimated current value of the q-axis current, the second d-axis and q-axis current command values are calculated, and the second current command value and the motor constant are calculated. And a voltage command value for the d-axis and the q-axis is calculated using the estimated frequency value.

  Hereinafter, the details of the present invention will be described with reference to the drawings.

  In the following embodiments, a permanent magnet type synchronous motor will be described as an AC motor. However, other AC motors such as an induction motor and a reluctance motor can be similarly realized.

[First Embodiment]
A first embodiment of the present invention will be described with reference to the configuration diagram of FIG.

The configuration of the control device of this embodiment is
A frequency command generator 1 for giving a frequency command ωr ^* to the motor;
A controller 2 that calculates an AC applied voltage of the electric motor, converts it into a pulse width modulated (PWM) signal, and outputs it;
An inverter 3 driven by the PWM signal;
A DC power supply 4 for supplying power to the inverter 3;
A permanent magnet type electric motor 5 to be controlled;
It comprises current detecting means 6 for detecting a DC bus current I0 which is an input current supplied from the DC power supply 4 to the inverter.

The controller 2 includes a vector calculation unit 50a for calculating the d-axis and q-axis voltage command values Vdc ^* and Vqc ^* from the frequency command ωr ^* and the current detection values Idc and Iqc given from the host, and the phase θc of the control axis. A coordinate conversion unit 61 is used to calculate a three-phase AC applied voltage from the d-axis and q-axis voltage command values, and a PWM calculation unit 62 is used to convert the AC applied voltage into a PWM signal.

  The current detection values Idc and Iqc of the controller 2 calculate the effective current Ia and the reactive current Ir by the effective and reactive current calculation means 14 from the current I0s obtained by detecting the DC bus current I0 by the sample and hold circuit 13, and the dq axis Calculation is performed by the current calculation means 10a.

  Similarly, for the phase θc of the control axis, first, an axis error Δθ, which is a phase deviation between the control reference axis (dc axis) and the magnetic flux axis (d axis) of the motor, is determined by the axis error calculation unit 15 as a motor constant and a voltage command. The axis error estimated value Δθc is calculated from the value, the electrical angular velocity ω1 and the current detection value, the electrical angular velocity ω1 is controlled by the PLL 16 so that Δθc is zero, and the ω1 is integrated by the integrator 17 to control θc. And the phase.

The dq-axis current calculation means 10a uses the phase φ obtained by the arc tangent of the ratio between the d-axis voltage command value Vdc ^* and the q-axis voltage command value Vqc ^* , and the axis error estimated value Δθc to determine the current coordinate conversion means 11. Estimates at least one of a detection value Idc corresponding to the d-axis current or a detection value Iqc corresponding to the q-axis current from the current Ia and the current Ir which is ineffective.

The vector calculation unit 50a converts the frequency command ωr ^* into the electrical angular frequency ω1 ^* by the conversion gain 51 using the number of poles P of the motor of the electric motor 5, and the current command calculation 52 is the current detection value Idc and Iqc or one of them. Is used to calculate current command values Id ^* and Iq ^* , and the voltage command calculation unit 53 uses the ω1 ^* , Id ^* , Iq ^* and motor constants to calculate the d-axis and q-axis voltage command values Vdc ^* and Vqc. Calculate ^* .

  The inverter 3 includes a main circuit 31 and a gate driver 32 that generates a gate signal to the main circuit. The DC power supply 4 that supplies power to the inverter 3 includes an AC power supply 41, a diode bridge circuit 42 that rectifies the AC, and a smoothing capacitor 43 that suppresses pulsating components included in the DC power supply.

  Next, the “principle of control operation” will be described with reference to FIG.

Current command calculation means 52 creates command value Iq ^* based on detection value Iqc.

In normal operation, the current command is generated as an output of a low-pass filter in which Id ^* is set to zero and Iq ^* is input Iqc.

The voltage command calculation unit 52 uses the current commands Id ^* and Iq ^* , the electrical angular frequency command ω1 ^*, and the motor constant to calculate the voltage command values Vdc ^* and Vqc ^* for the d-axis and the q-axis according to (Equation 1). Calculate.

The motor constants are R ^* : resistance set value Ld ^* : d-axis inductance set value Lq ^* : q-axis inductance set value Ke ^* : induced voltage constant set value.

The coordinate conversion unit 61 outputs AC voltage command values Vu ^* , Vv ^* , Vw ^* corresponding to the AC applied voltage.

  The axis error calculation unit 15 calculates an axis error Δθ, which is a difference between the phase θc of the base axis of the control axis and the phase θ of the magnetic flux axis of the motor, as Δθc by (Equation 2), and estimates Δθ.

  The PLL 16 calculates and outputs the electrical angular velocity ω1 so that the estimated axis error value Δθc is zero. The integrator 17 integrates ω1 and outputs the phase θdc of the control axis.

  Here, a method of calculating the effective current Ia and the reactive current Ir will be described.

  The sample and hold circuit 13 samples the current in the vicinity of an intermediate point in the energization period of I0 and detects I0s.

  The effective and reactive current calculation means 14 obtains the effective current Ia and the reactive current Ir by integrating the detected I0s in an electrical angle period of 60 degrees by the method described in Japanese Patent Application Laid-Open No. 2007-221999.

  Specifically, by using the periodic functions Fc and Fs for cutting out the effective current and the reactive current, the products of the detection current I0s are integrated, and the average values Iam and Irm for the 60-degree period are obtained.

  The effective current Ia and the reactive current Ir are obtained from these average values using (Equation 3) and (Equation 4).

  Next, the current coordinate conversion means 13 that characterizes the present invention will be described.

  In the current coordinate conversion means 11, as shown in (Equation 5), the d-axis and q-axis with respect to the control axis are used from the effective current Ia and the reactive current Ir using the voltage phase φ and the axis error estimated value Δθc. An estimation calculation of the current detection values Idc and Iqc is performed.

Here, the voltage phase φ is a phase difference angle between the primary voltage vector V obtained by synthesizing Vdc ^* and Vqc ^* and the qc axis, and is expressed by (Equation 6).

  Here, the phase relationship between voltage and current will be described.

  FIG. 2 is a vector diagram showing the phase relationship between voltage and current.

  First, the magnetic flux-torque axis is defined as the dq axis for the motor axis and the dc-qc axis for the control axis. In FIG. 2, since the estimated shaft error value Δθc = 0 is considered, the motor shaft and the control shaft coincide with each other.

  In addition, the phase coincident with the primary voltage vector V is defined as an effective a-axis, and the axis delayed by 90 ° from the a-axis is defined as an ineffective r-axis. Ia and Ir obtained by the effective and reactive current calculation means 14 are equivalent to a current vector obtained by resolving the primary current I1 along the ar axis.

  Here, consider the case of Δθc = 0 in (Equation 5). Under this assumption, Idc and Iqc can be obtained from Ia and Ir.

  Thus, although it is impossible to directly obtain the current detection values of Idc and Iqc from I1, it is possible through Ia and Ir.

  Next, consider a case where an axis error Δθ occurs during the acceleration / deceleration operation of the electric motor.

  FIG. 3 is a vector diagram showing the phase relationship between voltage and current when an axis error occurs and Δθc ≠ 0. In (Equation 5), if Idc and Iqc are obtained without considering Δθc, the current vectors of Idc and Iqc shown in FIG. 3 are obtained, and a current component corresponding to the dq axis cannot be obtained.

  Therefore, it is considered that a current component corresponding to the dq axis is obtained by considering Δθc in (Equation 5).

  FIG. 4 is a vector diagram showing the phase relationship between voltage and current when the generated axis error is taken into consideration.

  In FIG. 4, Ia and Ir are the same as those in FIG. 3, but by considering Δθc in addition to φ, the current components corresponding to the dq axes can be obtained as Iqc and Idc.

  The effect of the present invention will be described.

  First, FIG. 5 shows control characteristics during acceleration operation in the prior art.

  (A) Speed, (b) d-axis current component, (c) q-axis current component, (d) active current and reactive current, and (e) axis error are shown in order from the top of FIG.

  Although the q-axis current component increases in relation to the speed, a load is applied to the motor in proportion to the speed.

  It can be seen that there is an error between the motor currents Id and Iq during acceleration and the current detection values Idc and Iqc. This is because the axis error Δθ shown in FIG.

  On the other hand, FIG. 6 shows control characteristics during acceleration operation when the present invention is used.

  Considering the shaft error Δθ that occurs during acceleration, the currents Id and Iq of the motor are approximated to the currents Id and Iq by performing an estimation calculation of the motor currents Id and Iq using the effective component current Ia, the reactive component Ir, and the shaft error estimated value Δθc. The matched current detection values Idc and Iqc can be obtained.

  In the present invention, in (Equation 5), it is possible to detect a current value that coincides with Id and Iq of the motor by taking into account the shaft error that occurs during acceleration / deceleration operation by the estimated value Δθc.

  As a result, the performance of acceleration / deceleration operation is improved, and high-responsive operation characteristics and high-efficiency operation can be realized.

[Second Embodiment]
Another embodiment of the present invention is shown by replacing the dq-axis current calculation means 10a shown in FIG. 1 with a dq-axis current calculation means 10b.

  FIG. 7 shows the dq axis current calculation means 10b.

In the figure, the arc tangent element 12 for obtaining the current coordinate conversion means 11 and the phase φ of the inverter output voltage from Vd ^* and Vq ^* is the same as in FIG.

  In FIG. 7, a low-pass filter element 18 is added to the input of the current coordinate conversion means 11.

  The low-pass filter element 18 is a first-order lag element, and the time constant T is preferably set to a response equal to or shorter than the calculation period of the axis error calculation.

  By adding the low-pass filter element 18, the high-frequency component of the axis error estimated value Δθc can be removed, and response (follow-up) can be made to the low-frequency component of the axis error that occurs during acceleration / deceleration. .

[Third Embodiment]
The present embodiment is an example in which the present invention is applied to a control device provided with a speed controller and a current controller.

  FIG. 8 shows a vector calculation unit 50b in which the vector calculation unit 50a is replaced in the configuration of the controller 2 shown in FIG. 1 and a speed controller 54 and a current controller 55 are added.

The speed controller 54 outputs a torque current command Iq ^* by proportional + integral (or integral) calculation so that the deviation between the electrical angular velocity command ω1 ^* and the electrical angular velocity ω1 is zero.

The current controller 55 performs the second current command Id ^** and Iq ^* by proportional + integral (or integral) calculation so that the deviation between the current commands Id ^* and Iq ^* and the detected current values Idc and Iqc is zero ^{. * Is} output.

The voltage command calculation unit 53 outputs the voltage commands Vdc ^** and Vqc ^{** according} to (Equation 7) using the second current commands Id ^** and Iq ^** , the electrical angular velocity ω1 and the motor constant.

  In FIG. 8, the current controller 55 is connected to the voltage command calculation unit 53 in a column, but may be connected in parallel.

  Further, in the configuration of the controller 2, the primary delay element 18 may be added by replacing the dq-axis current calculation means 10a shown in FIG. 1 with a dq-axis current calculation means 10b shown in FIG.

Further, although the speed control system for giving the frequency command is shown, the frequency command generator 1, the conversion gain 51, and the speed controller 54 are removed, and the q-axis current command Iq ^* and the d-axis current command Id ^* corresponding to the torque command are changed. It is good also as a structure of the electric current control system (what is called torque control system) to give.

  By adding the current controller 55 in the present embodiment, it becomes possible to perform current feedback control that detects the d-axis current or the q-axis current of the motor, and high torque response characteristics and high-efficiency operation are achieved. It becomes possible.

[Fourth Embodiment]
FIG. 9 shows an example in which the first to third embodiments are modularized.

  In this embodiment, the speed command generator 1 and the controller 2 are configured by a microcomputer, and the inverter 3 is formed as an IC module. As a module, the current detection means 6 and the diode bridge circuit 42 for rectifying the alternating current to the inverter 3 and the smoothing capacitor 43 Are produced on the same substrate.

  The output of the inverter 3 is connected to a permanent magnet type electric motor 5 to be controlled and driven.

  The modularization allows the IC module to be handled as a single component, facilitating assembly and reducing the size of the device.

[Fifth Embodiment]
FIG. 10 shows an example in which the fan 301 is driven by the motor drive system 300.

  The motor drive system 300 includes a motor 302 and a motor control device 303.

  The motor control device 303 has a configuration excluding the AC power supply 41 and the electric motor 5 shown in FIG. 1 and has the form described in the first to fourth embodiments.

  By driving the fan 301, air cooling such as a heat radiating fin or a heat exchanger or dust collection using a dust collection filter is performed.

  By driving the fan according to this embodiment, high response characteristics and high-efficiency operation are possible, and excellent cooling performance and dust collection capability can be obtained.

[Sixth Embodiment]
FIG. 11 shows an example in which the pump 404 is driven by the motor drive system 404. The pump 404 is connected to the hydraulic circuit 410 by a pipe 406 and supplies oil to the circuit. In this embodiment, oil will be described as an example, but the medium may be water instead of oil.

  The hydraulic circuit 410 includes a relief valve 407 that keeps the hydraulic pressure below a set value, a cylinder 409 that operates as an actuator, a solenoid valve 406 that switches the hydraulic circuit, and a tank 405 that stores oil. The hydraulic circuit 410 works by pressure oil supplied from the pump 404. I do.

  The pump 404 and the motor control system 400 constitute an electric pump system 401. The motor control system 400 includes a motor 403 that is a drive source of the pump 404 and a motor control device 402 that drives the motor.

  The motor control device 402 has a configuration excluding the AC power supply 41 and the electric motor 5 shown in FIG. 1 and has the form described in the first to fourth embodiments.

  In the hydraulic circuit 410, when the circuit is switched by the solenoid valve 406, the load on the pump 404 is changed, resulting in a load disturbance of the motor 403.

  By driving the pump according to this embodiment, high response characteristics and highly efficient operation are possible, and excellent responsiveness to such load disturbance can be obtained.

[Seventh Embodiment]
FIG. 12 shows an example in which the present invention is applied to an air conditioner outdoor unit. In the air conditioner outdoor unit 500, the configuration related to the driving of the motor is a motor control device 501, a compressor 502, and an outdoor unit fan 503.

  The compressor 502 incorporates a motor as a drive source, compresses and expands the refrigerant, and adjusts the indoor temperature by a thermal cycle.

  The motor control device 501 has a configuration excluding the AC power supply 41 and the electric motor 5 of FIG. 1, and has the form described in the first to fourth embodiments.

  By driving the compressor according to this embodiment, high response characteristics and high-efficiency operation are possible, and excellent temperature control capability can be obtained.

  In addition to the outdoor unit fan 503, the air conditioner includes a fan that blows indoor air to the air conditioner indoor unit, both of which are application examples of the fifth embodiment that drives the fan.

[Eighth Embodiment]
FIG. 13 shows an example in which the present invention is applied to a washing machine 601. In the washing machine 601, a washing tub 606 and a stirring blade (pulsator) 605 are provided in a water receiving tank 607, and the washing tank 606 and the stirring blade 605 are driven by a motor control system 600. Which of the washing tub 606 and the stirring blade 605 is driven is switched by the clutch unit 604 during the washing process. Note that the clutch portion 604 may or may not have a speed reduction mechanism. The motor control system 600 includes a drive motor 603 and a motor control device 602 that drives the motor 603.

  A washing machine is characterized in that the load torque and the moment of inertia vary greatly depending on the amount and type of laundry.

  According to the present embodiment, sufficient responsiveness can be obtained with respect to such changes in load torque and moment of inertia, washing performance is improved, and washing time can be shortened.

  Further, the washing / drying machine having a dryer function includes a fan for blowing air, and driving of the fan is an application example of the fifth embodiment.

The block diagram of a control apparatus. The vector diagram which shows the phase relationship of the voltage and electric current when an axis error does not generate | occur | produce. The vector diagram which shows the phase relationship of the voltage and electric current in case an axis | shaft error generate | occur | produces. The vector diagram which shows the phase relationship of the voltage and electric current when the generated axis | shaft error is considered. Operation waveforms during acceleration in the prior art. The operation waveform at the time of acceleration in the present invention. The block diagram of a dq axis calculating means. The block diagram of a vector calculating part. The schematic diagram of the AC motor control apparatus at the time of modularizing. The schematic diagram of a fan drive system. The schematic diagram of a pump drive system. Schematic diagram of an air conditioner outdoor unit. The schematic diagram of a washing machine.

Explanation of symbols

P Number of motor poles Δθ Axis error Δθc Axis error estimated value which is phase deviation between reference axis (dc axis) of control and magnetic flux axis (d axis) of motor

Claims (6)

From the input current detection value of the inverter that drives the AC motor,
Detect the effective and reactive currents of the motor from the phase of the inverter output voltage,
A vector control device that performs current estimation calculation of a d-axis current as an excitation component and a q-axis current as a torque component from the detected current value,
A low-pass filter of a phase error value between the magnetic flux axis of the motor and the control reference axis is added to the current detection value of the effective and ineffective portions of the motor and the phase value of the inverter output voltage which is the first phase value. A control apparatus for an AC motor, wherein at least one of a d-axis current and a q-axis current of the motor is estimated using a second phase value obtained by adding the values . The control device according to claim 1,
An AC motor control apparatus characterized in that the phase of the inverter output voltage is obtained by an arc tangent of a ratio of a d-axis voltage command value and a q-axis voltage command value obtained by vector control calculation . The control device according to claim 1,
The phase error value is calculated using a voltage command value calculated by vector control, a frequency calculation value or frequency command value, and a d-axis or q-axis current estimation value obtained by estimation calculation. A control device for an AC electric motor that is characterized . The control device according to claim 1,
An AC motor control apparatus characterized in that the vector control calculation of the inverter calculates a d-axis and q-axis voltage command value using the d-axis or q-axis current estimation value . The control device according to claim 1,
Based on the deviation between the d-axis and q-axis current command values given from the host and the estimated current value of the d-axis or q-axis, the inverter vector control calculation is based on the d-axis and q-axis voltage commands. A control apparatus for an AC motor , characterized by calculating a value . The control device according to claim 1,
Based on the deviation between the d-axis and q-axis current command values given from the host and the estimated current value of the d-axis or q-axis, the calculation of the vector control of the inverter is based on the second d-axis and q-axis. An AC motor control apparatus , comprising: calculating a current command value, and calculating a d-axis and q-axis voltage command value using the second current command value, a motor constant, and a frequency estimation value .

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