JP5042879B2 - Switching power supply - Google Patents

Description

  The present invention relates to a switching power supply device that outputs a DC stabilized voltage / current that is insulated from an input power supply side, and in particular, can achieve a reduction in switching element loss, a reduction in size and cost, and stable startability. About.

  Conventionally, a composite resonance type series converter circuit is known as a small-sized and high-efficiency switching power supply as described above. FIG. 12 shows a main circuit configuration according to the prior art, and FIG. 13 shows waveforms of main parts.

  In this switching power supply device, a series circuit of switching elements Q1 and Q2 (hereinafter referred to as power MOSFET) is connected between both terminals of the DC input power supply E, and a capacitor C0 is connected. A series resonant circuit of an inductor L, a primary winding L11 of the output transformer T, and a capacitor C1 is formed between the connection point of the power MOSFETs Q1 and Q2 and one end of the DC input power supply E, and the power MOSFET Q1 Alternatively, a capacitor C2 is connected in parallel with either Q2 (in FIG. 12, one end of the DC input power supply E is connected to the low voltage side, and the capacitor C2 is connected in parallel to the power MOSFET Q2). Also, diodes D1 and D2 are connected in antiparallel to power MOSFETs Q1 and Q2, respectively (in many cases, they are also used as body diodes for power MOSFETs Q1 and Q2).

  Further, an intermediate tap is provided in the output winding of the output transformer T and divided into two (L21, L22), and a full-wave rectifier circuit is formed by diodes D3 and D4 that rectify their outputs, and between the intermediate taps. A smoothing capacitor C3 and a DC load Load are connected. The power MOSFETs Q1 and Q2 are alternately turned ON / OFF at a preset frequency in consideration of the complex resonance condition by the control unit 1 shown as a block. Therefore, the control unit 1 has a high-frequency oscillation function, a function of alternately driving the two power MOSFETs Q1 and Q2, and a function of setting a dead time period for turning off the two power MOSFETs Q1 and Q2. A feedforward, feedback control function and output variable function for controlling voltage, current, and power are provided.

  Referring to FIG. 13, Vg1 and Vg2 indicate drive signals for power MOSFETs Q1 and Q2 set in advance by control unit 1. A dead time period in which both are turned ON / OFF alternately and both are turned OFF is set. VQ1, IQ1 and VQ2, IQ2 indicate the drain-source voltage and drain current of the power MOSFETs Q1, Q2. When the drive signal Vg1 is High, the drain current IQ1 flows through the power MOSFET Q1, and when it is Low, the voltage VQ1 substantially equal to the DC input power source E is applied (the same applies to the power MOSFET Q2). In the dead time period, the drain-source voltages VQ1 and VQ2 have rising and falling waveforms with an arbitrary slope due to the effect of the capacitor C2, the inductor L, and the excitation inductance of the output transformer T. The drain currents IQ1 and IQ2 have a series resonance current waveform set approximately by the inductor L and the capacitor C1, and these combined currents are the series resonance of the inductor L, the primary winding L11 of the output transformer T, and the capacitor C1. It becomes the current of the circuit. VC1 represents a voltage waveform of the capacitor C1, and has a waveform delayed in phase from the current of the series resonance circuit. ID3 and ID4 indicate current waveforms of the output rectifier diodes D3 and D4. The relationship between the drive frequency of the power MOSFETs Q1 and Q2 and the series resonance frequency of the inductor L and the capacitor C1 is expressed as “resonance frequency> drive frequency. By satisfying this condition, it is possible to set the current of one of the diodes D3 and D4 to flow and then the other current starts to flow. During the period when both diode currents do not flow, power is transmitted to the output side. Not. That is, during the period in which no current flows through the diodes D3 and D4, the secondary side of the output transformer T is considered to be unloaded, and the primary side exciting inductance L of the transformer T is inserted in series into the primary side series resonance circuit. As a result of the switching of the series resonance condition, inflection points are also found in the waveforms of the drain currents IQ1 and IQ2.

  In such a complex resonance type series converter, ZVS (zero voltage switching), that is, a condition setting such that a current starts to flow after the applied voltage of the switching elements Q1 and Q2 is reduced is possible, and switching loss is extremely small. Since the recovery loss of the secondary side rectifier diodes D3 and D4 can be avoided, high frequency can be achieved with high efficiency. Further, since the voltage / current waveform at the time of switching is stable and the ringing of the secondary side rectifier diodes D3 and D4 can be suppressed, the noise is also excellent.

  The above-described prior art is a so-called separately-excited switching power supply that drives the two switching elements Q1 and Q2 by setting the frequency in advance with an oscillator while having such various features. A level shifter is required for the potential side switching elements Q1 and Q2. From the viewpoint of frequency followability and loss, there is a technical problem with high frequency, and from a cost viewpoint, for example, Patent Documents 2 to 4 Self-excited studies are also being conducted as shown.

  FIG. 14 is an electric circuit diagram showing a known example of the current feedback type self-excited composite resonance series converter. The main circuit configuration is substantially the same as that shown in FIG. 12, but in this prior art, the switching elements Q1 and Q2 (configured by a power MOSFET) are driven by the secondary winding l2 and the tertiary winding l3 of the current feedback transformer CT. , And an appropriate gate drive signal while shaping the waveform by resistors R1 to R3, R4 to R6 and capacitors C4 and C5. In the current feedback type, a bipolar transistor is often used as the switching element. However, when it is desired to set the operating frequency high, the power MOSFET may be used as described above. In such a configuration, in order to drive a voltage-driven power MOSFET by feeding back a sinusoidal resonance current, the gate circuit needs to be devised, and the loss tends to increase.

On the other hand, FIG. 15 is an electric circuit diagram showing a known example of a voltage feedback type self-excited composite resonance series converter. In this prior art, the auxiliary elements L13 and L12 of the output transformer T1 are used for driving the switching elements Q1 and Q2 (configured by a power MOSFET), respectively, and resistors R1 to R3, R4 to R6 and capacitors C4 and C5 are used. An appropriate gate drive signal is obtained while shaping the waveform by such means. Thus, when a feedback signal is obtained from the auxiliary windings L12 and L13 of the output transformer T1, a rectangular-wave feedback voltage is obtained. Therefore, it can be said that this method is suitable for the power MOSFETs Q1 and Q2 that are voltage-driven elements.
Japanese Patent No. 2734296 Japanese Patent No. 3371595 JP 2002-262568 A JP 2006-129548 A

  However, the above-described prior art simply oscillates with the feedback windings L13 and L12 of the output transformer T1, so that there is a problem that the switching OFF operation is particularly slow and the switching loss is increased. In the case of the self-excited type, the smooth switching OFF operation affects the stability, so various ideas are added, but the circuit configuration becomes larger and the superiority over the other-excited type is scarce as it tries to obtain a stable operation with high accuracy. Become.

  The object of the present invention is to greatly simplify the circuit and reduce the cost of a self-excited composite resonant series converter of voltage feedback type compared to other types of excitation while maintaining the original characteristics of low loss and low noise. It is to provide a switching power supply device that can be realized.

  In the switching power supply device of the present invention, a series circuit composed of first and second switching elements is connected between both terminals of the DC input power supply, and the connection point between the first and second switching elements and the DC input power supply A series circuit composed of an inductor, a capacitor, and a primary winding of a transformer is connected to one terminal, and switching is performed by the start-up circuit applying a start-up pulse to one of the first and second switching elements. First, the secondary side induced current of the transformer obtained by the switching is rectified and smoothed by a diode and a smoothing capacitor, and the secondary side induced current stops flowing due to the completion of charging of the smoothing capacitor. And a second control circuit detects from a drop in the voltage of one auxiliary winding of the transformer and detects the first and second switches. Switching is continued by turning off the ON side of the chucking element and turning on the OFF side of the first and second switching elements by increasing the other auxiliary winding voltage. A switching power supply comprising a voltage feedback type self-excited composite resonant series converter configured as described above, wherein the smoothing capacitors are in parallel with each other, a first capacitor having a relatively small capacity, and a second capacitor having a large capacity. And is provided between the first capacitor and the second capacitor, with the first capacitor as the secondary winding side and the second capacitor as the output end side. Is maintained at a high impedance until the charge amount of the first capacitor reaches a predetermined level or more.

  According to the above configuration, in the switching power supply device including the voltage feedback self-excited complex resonance series converter that uses the induced voltage of the auxiliary winding for ON / OFF driving of the switching element, the secondary-side smoothing capacitors are connected to each other. In parallel, the first capacitor provided on the secondary winding side with a relatively small capacity and the second capacitor provided on the output end side with a large capacity are divided into the first capacitor between them. An impedance circuit that maintains a high impedance until the charge amount of the capacitor reaches a predetermined level or more is provided.

  Therefore, the secondary current generated by the voltage induced on the secondary side at the time of start-up does not flow rapidly to the load side by the impedance circuit, charges the first capacitor, and 1 during the charging through the auxiliary winding. Of the first and second switching elements, the ON side is turned OFF and the OFF side is turned ON by the voltage signal fed back to the next side. In this way, in the voltage feedback self-excited complex resonance series converter, which can realize low-loss and low-noise, and further simplification of the circuit and cost reduction compared to the separately excited type, the startability can be improved. it can.

  Preferably, the impedance circuit is interposed between a series circuit of a Zener diode and a resistor provided in parallel with the first capacitor, and one terminal of the first and second capacitors, and the Zener diode and the resistor And a switch that is turned on when the voltage at the connection point of the switch becomes a certain value or more.

  Preferably, the impedance circuit is interposed between a series circuit of first and second impedance elements provided in parallel with the first capacitor and one terminal of the first and second capacitors, And a switch that is turned on when the divided voltage by the first and second impedance elements exceeds a certain value.

  More preferably, the impedance circuit is interposed between a series time constant circuit of a resistor and a capacitor provided in parallel with the first capacitor, and one terminal of the first and second capacitors, and the capacitor And a switch that is turned on when the charging voltage becomes equal to or higher than a certain value.

  Further preferably, an impedance element is further provided in parallel with the switch, and the impedance element is inserted between the first and second capacitors immediately after startup, and when the charging voltage of the capacitor becomes a predetermined value or more, the switch The impedance element is short-circuited.

  More preferably, the impedance circuit is composed of an impedance element provided in series with the second capacitor.

  Further preferably, the impedance circuit is interposed in series with the second capacitor and a series circuit of a Zener diode and a resistor provided in parallel with the first capacitor, and is connected to a connection point of the Zener diode and the resistor. It is characterized by comprising a switch that turns on when the voltage exceeds a certain value.

  The switching power supply device of the present invention further includes an abnormality detection circuit that turns off the switch when a load abnormality is detected.

  According to said structure, the switch which isolate | separates a large capacity | capacitance 2nd capacitor | condenser at the time of starting as mentioned above can be combined with the abnormality protection switch which isolate | separates this load, when the abnormality of a load is detected.

  Furthermore, in the switching power supply device of the present invention, the impedance circuit includes a switch that disconnects the first capacitor and the second capacitor, and the switch further includes a control circuit, a choke coil, and a diode. The step-down or step-up chopper circuit is configured.

  According to the above configuration, there is no need to feed back the secondary signal to the primary side for constant voltage or constant current control, and the operation waveform of the self-excited composite resonance series converter can be kept optimal. Become.

  In the switching power supply device of the present invention, the load is an LED.

  According to the above configuration, when an LED is used as a load, no current flows until a voltage equal to or higher than the forward voltage of the LED (n times that in the case of a series configuration) is applied. Therefore, the start-up property of the voltage feedback type self-excited composite resonance series converter is improved, and the rush current to the LED generated when the LED load is attached / detached is suppressed.

  As described above, the switching power supply according to the present invention is a switching power supply comprising a voltage feedback type self-excited complex resonance series converter that uses the induced voltage of the auxiliary winding for ON / OFF driving of the switching element. The smoothing capacitor is divided into a first capacitor provided on the secondary winding side with a relatively small capacity in parallel with each other, and a second capacitor provided on the output end side with a large capacity, In the meantime, an impedance circuit is provided that maintains a high impedance until the charge amount of the first capacitor becomes equal to or higher than a predetermined level.

  Therefore, the secondary current generated by the voltage induced on the secondary side at start-up does not flow rapidly to the load side by the impedance circuit, charges the first capacitor, and passes through the auxiliary winding during the charging. By the voltage signal fed back to the primary side, the ON side of the first and second switching elements is driven OFF, and the OFF side is driven ON. In this way, in the voltage feedback self-excited complex resonance series converter, which can realize low-loss and low-noise, and further simplification of the circuit and cost reduction compared to the separately excited type, the startability can be improved. it can.

  In addition, as described above, the switching power supply device of the present invention turns off the switch provided as an impedance circuit between the first and second capacitors when the abnormality detection circuit detects a load abnormality.

  Therefore, the switch that disconnects the large-capacity second capacitor at the time of start-up can also be used as an abnormality protection switch that disconnects the load when an abnormality of the load is detected.

  Furthermore, in the switching power supply device of the present invention, as described above, the impedance circuit is configured by a switch that separates the first capacitor and the second capacitor, and the control circuit, choke coil is included in the switch. And a diode are further provided to constitute a step-down or step-up chopper circuit.

  Therefore, it is not necessary to feed back the secondary side signal to the primary side for constant voltage and constant current control, and the operation waveform of the self-excited composite resonance series converter can be kept optimal.

  Moreover, the switching power supply device of this invention makes load into LED as mentioned above.

  Therefore, current does not flow until a voltage higher than the forward voltage of the LED is applied, and it is in a so-called no-load state at the time of start-up, so that the start-up property of the voltage feedback type self-excited complex resonance series converter is improved. There is an effect of suppressing the rush current to the LED generated when the LED load is attached / detached.

  FIG. 16 is a diagram showing a circuit configuration of an improved voltage feedback type self-excited complex resonance series converter which is a premise of the present invention, and FIG. 17 is a diagram showing main operation waveforms thereof. This switching power supply device is intended to reduce the loss of the switching elements Q1 and Q2, and to reduce the size and cost, and to increase the operation frequency. In FIG. 16, a series circuit of switching elements Q1 and Q2 (hereinafter referred to as power MOSFET) is connected between both terminals of the DC input power supply E, and a capacitor C0 is connected. A series resonant circuit of an inductor L, a primary winding L11 of the output transformer T1, and a capacitor C1 is formed between the connection point of the power MOSFETs Q1 and Q2 and one end of the DC input power supply E, and the power MOSFET Q1 , Q2 and a capacitor C2 are connected in parallel (FIG. 16 shows an example in which one end of the DC input power source E is connected to the low voltage side and the capacitor C2 is connected to the power MOSFET Q2 in parallel). Also, diodes D1 and D2 are connected in antiparallel to power MOSFETs Q1 and Q2, respectively. Capacitor C2 may be substituted by the junction capacitance of power MOSFETs Q1 and Q2, and diodes D1 and D2 may also be used as body diodes of power MOSFETs Q1 and Q2.

  Further, an intermediate tap is provided in the output winding of the output transformer T1 and divided into two (L21, L22), and a full-wave rectifier circuit is formed by diodes D3 and D4 that rectify their outputs, and between the intermediate taps. A smoothing capacitor C3 and a DC load Load are connected. Further, an auxiliary winding L12 is provided in the output transformer T1, and the reverse polarity side of the primary main winding L11 is connected to the gate of the power MOSFET Q2 via the gate resistor R12, and the power MOSFET Q2 is generated by the voltage generated in the auxiliary winding L12. Is configured to be driven. Similarly, for the gate drive circuit of the high-voltage side power MOSFET Q1, a second auxiliary winding L13 is provided in the output transformer T1, and the same polarity side as the primary main winding L11 is connected to the power MOSFET Q1 via the gate resistor R11. The power MOSFET Q1 is configured to be connected to the gate and driven by the voltage generated in the second auxiliary winding L13. Thus, the power MOSFETs Q1 and Q2 are alternately turned on and off by the feedback voltages from the two auxiliary windings L13 and L12, and self-oscillate.

  Furthermore, the power MOSFETs Q1 and Q2 are provided with control circuits Cont1 and Cont2 for setting the OFF timing. These control circuits Cont1 and Cont2 are short-circuited between the gates and sources of the power MOSFETs Q1 and Q2, and the peak hold circuit composed of a switch element SW1 for turning off the power MOSFETs Q1 and Q2, a diode D5, and a capacitor C4. And a comparator Comp for detecting the reverse voltage of the diode D5 and turning on the switch element SW1, and a delay circuit 2 and a switch SW2 for discharging the charge of the capacitor C4 in preparation for the next cycle. Configured.

  The circuit operation will be described with reference to FIG. In the figure, VQ1 and VQ2 are drain-source voltages of the power MOSFETs Q1 and Q2, IQ1 and IQ2 are drain currents of the power MOSFETs Q1 and Q2, VC1 is a voltage of the capacitor C1, ID3 and ID4 are diode currents of the output rectifier diodes D3 and D4, VL11 is a voltage of the primary winding L11 of the output transformer T1, VL13 is a voltage of the auxiliary winding L13 of the output transformer T1, and cont1 and cont2 are signals for driving the switch element SW1 in the control circuits Cont1 and Cont2, respectively. These are approximated to the operation waveforms of the separately excited composite resonant series converter described in FIG.

  That is, the voltage drop point of the auxiliary windings L13 and L12 generated at the timing when the currents ID3 and ID4 of the secondary side rectifier diodes D3 and D4 are interrupted is obtained by a peak hold circuit constituted by the diode D5 and the capacitor C4. The DC voltage is detected by comparing the coil voltage as it is with a comparator Comp. At that time, the switch element SW1 is turned on to turn off the power MOSFETs Q1 and Q2. This sets a switching-off condition, which enables stable self-excited oscillation and makes the gate voltage fall steep to reduce the switching loss of the power MOSFETs Q1 and Q2. After the turn-off, when a predetermined time elapses in the delay circuit 2, the switch SW2 is turned on and the capacitor C4 is discharged.

  Here, in the configuration of FIG. 15 described above, switching is inverted in conjunction with the resonance operation of the main circuit, and there is no particular problem when the feedback windings L13 and L12 include a stable signal accompanying the resonance operation. However, when the resonance is unstable, abnormal oscillation occurs. On the other hand, in the case of the composite resonance of the present invention, the main resonance is set as the current resonance (series resonance), and the switching is not turned off during that time, and the switching is turned off during the subsequent period in which no current flows through the diodes D3 and D4. Thus, it is possible to ensure the stability of the operation against the instability of the resonance. However, during the period in which no current flows through the diodes D3 and D4, the energy stored in the inductor L charges the capacitor C2 in parallel with the switching element Q2. However, the resonance is weak and generally the energy of the inductor L is released. It becomes slow switching at the stage where it is done. On the other hand, in the configuration of FIG. 16, the switching OFF is not performed by the switch element SW1 from the comparator Comp at a timing immediately after the period in which no current flows in the diodes D3 and D4, rather than waiting for the slow switching OFF. I do. As a result, the switching loss is reduced as described above.

FIG. 18 is a circuit diagram of the control circuits Conta and Cont, which is a specific configuration example of the control circuits Cont1 and Cont2 of FIG. In the control circuit Conta shown in FIG. 18A, the voltage feedback signal inputted between the terminals P11-P13; P21-P23 is divided by the resistors r1 and r2, and the divided voltage is minus-terminal of the comparator IC1. To supply. The divided voltage charges the capacitor c2 via the diode d5 and the Zener diode zd, and supplies the charging voltage to the + terminal of the comparator IC1. With such a configuration, when the feedback voltage decreases below a predetermined value, the potential at the voltage dividing point also decreases, the potential at the − terminal becomes equal to or less than the potential at the + terminal, and the output of the comparator IC1 becomes high level. As a result, the switch element q1 is turned ON, and the terminals P12-P13; P22-P23 are short-circuited, that is, the power MOSFETs Q1, Q2 can be turned OFF. In the figure, the diode d2 and the capacitor c1 form a power source for the comparator IC1, and the diodes d3 and d4 and the resistor r3 form a circuit for extracting the charge of the capacitor c2. Such a diode d5 and a Zener diode zd increase the degree of freedom in setting the threshold value of the comparator IC1.

  Further, in the control circuit Contb shown in FIG. 18B, a voltage feedback signal input between the terminals P11-P13; P21-P23 is accumulated in the capacitor c11 via the resistor r11 and the diode d11, and the feedback When the voltage drops below the gate threshold voltage of the MOSFET q11, the MOSFET q11 is turned on to release the charge of the capacitor c11 and the switch element q12 composed of a pnp bipolar transistor is turned on by the discharge current, Terminals P12-P13; P22-P23 are short-circuited, that is, the power MOSFETs Q1, Q2 are turned OFF. In addition, the correspondence of each terminal P11-P13; P21-P23 is shown in FIG.

  FIG. 19 is a diagram showing an actual circuit configuration of FIG. 16 described above. Although stable steady operation is possible with the configuration of FIG. 16, the configuration of FIG. 16 has a problem in the startability when input power is first supplied. Specifically, the DC input power supply E is configured by rectifying and smoothing the commercial power supply Vac with the rectifying bridge DB and the capacitor C0, and the capacitor C1 is charged in advance via the starting resistor R0 in parallel with the power MOSFET. When a starting circuit (relaxation oscillation circuit) IGN comprising a resistor R10, a capacitor C10 and a diac Q10 oscillates and its output is supplied to the gate terminal of the power MOSFET Q2, the power MOSFET Q2 is turned on momentarily, and the charge of the capacitor C1 is output. Since it is discharged through the primary winding L11 of the transformer T1, an electromotive force is generated in the auxiliary winding L13 of the output transformer T1 in the direction of turning on the power MOSFET Q1, and the operation of continuous self-oscillation is started. .

  However, at the time of startup, there is no accumulated charge in the output-side smoothing capacitor C3, and the induced voltages of the auxiliary windings L13 and L12 of the output transformer T1 due to the startup pulse are easily absorbed. That is, after the smoothing capacitor C3 is fully charged, a voltage is induced in the auxiliary windings L13 and L12, but the drive signals (voltage pulses) for the power MOSFETs Q1 and Q2 cannot be obtained until then. In addition, even if the gate circuit includes a smoothing element by the capacitor C4, it becomes a factor to absorb the induced voltage, and it is difficult to obtain a feedback signal that reaches the gate threshold voltage of the power MOSFETs Q1 and Q2.

  For this reason, FIG. 20 shows a case proposed as one attempt (for digital control of the IEICE Technical Report EE2006-53 (2007-01) self-excited drive type resonant converter). This prior art uses digital control by the digital signal processor 3 to change the pulse condition while confirming the start-up state, so that it can be easily solved for analog control requiring complicated control. Yes. However, one of the main purposes of constructing the composite resonance series converter by self-excitation is to reduce the size and cost by a simple configuration. From this viewpoint, a simple method other than digitization is desired.

  FIG. 1 is a block diagram showing an electrical configuration of the switching power supply device according to the embodiment of the present invention. This switching power supply apparatus is similar to the switching power supply apparatus shown in FIGS. 16 and 19 described above, and corresponding portions are denoted by the same reference numerals. Also in this switching power supply device, a series circuit of switching elements Q1 and Q2 (hereinafter referred to as power MOSFET) is connected between both terminals of the DC input power supply E, and a capacitor C0 is connected. A series resonance circuit of an inductor L, a primary winding L11 of the output transformer T1, and a capacitor C1 is connected between a connection point of the power MOSFETs Q1 and Q2 and one end of the DC input power supply E, and the power MOSFET Q1. Alternatively, a capacitor C2 is connected in parallel with either one of Q2 and a starting resistor R0 is connected in parallel with the other. (FIG. 1 shows an example in which one end of the DC input power source E is connected to the low voltage side, the capacitor C2 is connected to the power MOSFET Q2, and the starting resistor R0 is connected in parallel to the power MOSFET Q1). As described above, the capacitor C2 may be substituted by the junction capacitance of the power MOSFETs Q1 and Q2, and the diodes D1 and D2 may also be shared by the body diodes of the power MOSFETs Q1 and Q2.

  Furthermore, an intermediate tap is provided in the output winding of the output transformer T1 and divided into two (L21, L22), and a full-wave rectifier circuit is formed by diodes D3 and D4 that rectify their outputs. In this form, a first capacitor C31 having a small capacity is connected to the intermediate tap, and a second capacitor C32 having a large capacity and a DC load Load are further connected via the impedance circuit Z. That is, in the premise configuration shown in FIGS. 16 and 19, the smoothing capacitor C3 is divided into C31 and C32, and the impedance circuit Z is interposed between them. The output transformer T1 is provided with auxiliary windings L13 and L12, and drives the power MOSFETs Q1 and Q2 via the gate resistors R11 and R12 and the control circuits Cont1 and Cont2, respectively. The control circuits Cont1, Cont2 and the starting circuit IGN constitute a circuit similar to that described with reference to FIG.

  In the switching power supply configured as described above, when a starting pulse is supplied from the starting circuit IGN to the gate of the power MOSFET Q2 in a state where the capacitor C2 is charged from the DC input power source E via the starting resistor R0, the power The MOSFET Q2 is turned ON for a moment, the electric charge of the capacitor C1 is discharged through the primary winding L11 of the output transformer T1, and a voltage is induced in each of the windings L21, L22; L13, L12 of the output transformer T1. The induced voltage of the secondary windings L21 and L22 is first rectified and smoothed by the diode D3 or D4 and the capacitor C31. However, in the present invention, the capacitor C31 has a degree that does not hinder the startability as described above. The capacitance value is set, and the original second capacitor C32 is connected to the output side via the impedance circuit Z. The secondary voltage induced by the start pulse is less likely to be absorbed by the second capacitor C32 by the amount corresponding to the impedance circuit Z, and a feedback signal from the auxiliary windings L13 and L12 is also likely to be generated. As a result, it becomes easy to turn on the power MOSFET, and in the voltage feedback type self-excited complex resonance series converter, the circuit is greatly simplified compared to the other excitation type while maintaining the low loss and low noise which are the original features. In addition, the cost can be reduced and the startability can be improved.

  2 to 6 are diagrams showing a specific configuration example of the impedance circuit Z shown in FIG. 1, and a switch Q31 (indicated by a MOSFET) is used. In FIG. 2, the switch Q31 is turned on after the voltage of the first capacitor C31 becomes equal to or higher than a certain value by the Zener diode ZD and the resistor R31, and the smoothing by the second capacitor C32 and the direct current output to the load Load are performed. Is configured to do.

  FIG. 3 shows an example in which the switch Q31 is turned on by the divided voltage of the impedance elements Z1 and Z2 instead of the Zener diode ZD of FIG. Furthermore, FIG. 4 shows an example in which a series time constant circuit composed of a resistor R32 and a capacitor C33 is used, and the switch Q31 is quickly turned on after startup when the charging voltage of the capacitor C33 exceeds a certain value. It is. FIG. 5 shows an example in which an impedance element Z1 is provided in parallel with the switch Q31, and the impedance circuit Z shown in FIG. Further, FIG. 6 shows an example in which when the abnormality detection circuit 11 detects an abnormality in the load load, the switch Q31 is turned OFF and the abnormality protection for disconnecting the load load is also used as the switch Q31.

  On the other hand, FIG. 7 shows an example in which an impedance element Z1 is provided in series with the second capacitor C32 having a larger capacitance, and FIG. 8 shows a switch Q31 provided in series with the second capacitor C32. This is an example, and the same effect as in the case of FIGS.

  FIG. 9 is a diagram illustrating an entire circuit when an LED is used as the load load. In this configuration, the output of the output transformer T1 is rectified and smoothed by the diodes D3 and D4 and the capacitor C31, the DC output is supplied to the series circuit of the LED load Load and the current detection resistor R15, and the first capacitor C31 is provided. A series circuit of a second capacitor C32 and a switch Q31 (described as a MOSFET) is formed in parallel with the reference voltage source Vref and the comparator comp1, and the voltage drop at the current detection resistor R15 becomes a certain value or more. In this case, the switch Q31 is turned on to ensure the smoothness of the output voltage by the second capacitor C32.

  In such a configuration, a current does not flow until a voltage equal to or higher than the forward voltage Vf of the LED (n times in the case of a series configuration) is applied, and a so-called no-load state is established at the time of startup. The self-excited composite resonance series converter has an effect of suppressing the rush current to the LED generated when the LED load is attached and detached.

In contrast, FIGS. 10 and 11, in the self-excited composite resonance series converter of the voltage feedback, using the secondary side switch Q 3, output transformer and voltage, current, power to the load Load An example of control on the secondary side of T1 is shown. 10, the switch Q 3 AND ITS control circuit cont3, a choke coil L2, an example in which the buck chopper diode D 5, 11 constituted the boost chopper by changing the arrangement of similar devices Example It is. According to such a configuration, there is no need to feed back the secondary side signal of the transformer T1 to the primary side for constant voltage and constant current control, and the operation waveform of the self-excited composite resonant series converter is kept optimal. It becomes possible.

  Here, Japanese Patent Laid-Open No. 2006-333555 discloses a configuration in which a choke coil and a capacitor are provided after the rectifier diode and the smoothing capacitor on the secondary side of the output transformer, but spike noise is removed. Contrary to the present invention, the capacity of the smoothing capacitor on the output transformer side is large, the capacity of the capacitor on the output end side is small, and the impedance of the choke coil is always small.

1 is a block diagram showing an electrical configuration of a switching power supply device according to an embodiment of the present invention. It is a figure which shows one specific structural example of the impedance circuit of the switching power supply device shown in FIG. It is a figure which shows one specific structural example of the impedance circuit of the switching power supply device shown in FIG. It is a figure which shows one specific structural example of the impedance circuit of the switching power supply device shown in FIG. It is a figure which shows one specific structural example of the impedance circuit of the switching power supply device shown in FIG. It is a figure which shows one specific structural example of the impedance circuit of the switching power supply device shown in FIG. It is a figure which shows one specific structural example of the impedance circuit of the switching power supply device shown in FIG. It is a figure which shows one specific structural example of the impedance circuit of the switching power supply device shown in FIG. FIG. 1 is a diagram showing an entire circuit when an LED is used as a load. It is a block diagram which shows the example which comprised the pressure | voltage fall type | mold chopper using the switch which will be interposed by the secondary side by this invention. It is a block diagram which shows the example which comprised the pressure | voltage rise type chopper using the switch which will be interposed by the secondary side by this invention. It is a circuit diagram which shows the basic composition of a separately excited composite resonance series converter. FIG. 13 is an operation waveform diagram of FIG. 12. It is a circuit diagram which shows the example of the conventional self-excitation (current feedback) type | mold series converter. It is a circuit diagram which shows the example of the conventional self-excitation (voltage feedback) type | mold series converter. 1 is a circuit diagram of a self-excited (voltage feedback) series converter that is a premise of the present invention. FIG. FIG. 17 is an operation waveform diagram of FIG. 16. It is an electric circuit diagram which shows the specific circuit structure of a control circuit. It is a figure which shows the specific circuit structure of FIG. 16 including a starting circuit. It is a circuit diagram of a conventional self-excited (voltage feedback) type series converter with improved startability.

Explanation of symbols

11 Abnormality detection circuit C0, C1, C2 Capacitor C31 First capacitor C32 Second capacitor C33 Capacitor Comp Comparator Comp1 Comparator Cont1, Cont2, Cont3; Conta, Contb Control circuit D1, D2, D3, D4, D20 Diode E DC Input power source IGN Start circuit L Inductor L2 Choke coil L11 Primary winding L12, L13 Auxiliary winding L21, L22 Secondary winding Load DC load Q1, Q2 Switching element (power MOSFET)
Q31 switch R0 start resistor R11, R12 gate resistor R15 current detection resistor T1 output transformer Vref reference voltage source Z impedance circuit Z1, Z2 impedance element ZD Zener diode

Claims (10)

A series circuit composed of first and second switching elements is connected between both terminals of the DC input power supply, and between the connection point of the first and second switching elements and one terminal of the DC input power supply, A series circuit composed of an inductor, a capacitor, and a primary winding of a transformer is connected, and the start circuit starts switching by giving a start pulse to one of the first and second switching elements, and is obtained by the switching. The first and second control circuits indicate that the secondary induced current of the transformer has been rectified and smoothed by a diode and a smoothing capacitor, and the secondary induced current has stopped flowing due to the completion of charging of the smoothing capacitor. One of the first and second switching elements detected from a decrease in the voltage of one auxiliary winding of the transformer and the ON side A voltage feedback type self-excited composite that is driven to be turned off and that switching is continued by turning on the off-side of the first and second switching elements by increasing the other auxiliary winding voltage. A switching power supply device comprising a resonant series converter,
The smoothing capacitor is divided into a first capacitor having a relatively small capacity and a second capacitor having a large capacity in parallel with each other;
The first capacitor is provided between the first capacitor and the second capacitor, and the first capacitor is used as a secondary winding side and the second capacitor is used as an output end side. A switching power supply comprising an impedance circuit that maintains a high impedance until the amount of charge of the battery becomes equal to or higher than a predetermined level.   The impedance circuit is interposed between a series circuit of a Zener diode and a resistor provided in parallel with the first capacitor, and one terminal of the first and second capacitors, and is connected to the Zener diode and the resistor. 2. The switching power supply device according to claim 1, further comprising a switch that is turned on when the voltage at the point exceeds a certain value.   The impedance circuit is interposed between a series circuit of first and second impedance elements provided in parallel with the first capacitor and one terminal of the first and second capacitors, and the first and second capacitors 2. The switching power supply device according to claim 1, further comprising a switch that is turned on when a divided voltage by the impedance element of 2 becomes a certain value or more.   The impedance circuit is interposed between a series time constant circuit of a resistor and a capacitor provided in parallel to the first capacitor and one terminal of the first and second capacitors, and a charging voltage of the capacitor is constant. The switching power supply device according to claim 1, further comprising a switch that is turned on when the value is exceeded.   An impedance element is further provided in parallel with the switch, and the impedance element is inserted between the first and second capacitors immediately after startup, and when the charging voltage of the capacitor exceeds a certain value, the impedance element is The switching power supply according to claim 4, wherein the switching power supply is short-circuited.   2. The switching power supply device according to claim 1, wherein the impedance circuit includes an impedance element provided in series with the second capacitor.   The impedance circuit is interposed in series with a series circuit of a Zener diode and a resistor provided in parallel to the first capacitor, and the second capacitor, and a voltage at a connection point between the Zener diode and the resistor is a constant value. The switching power supply device according to claim 1, comprising a switch that is turned on when the above is reached. The switching power supply device according to any one of claims 2-5, characterized by further comprising an abnormality detection circuit for OFF the switch and detects an abnormality of the load.   The impedance circuit includes a switch for separating the first capacitor and the second capacitor, and the switch further includes a control circuit, a choke coil, and a diode to form a step-down or step-up chopper circuit. The switching power supply device according to claim 1, wherein: The switching power supply according to any one of claims 1 to 9, wherein the load is an LED.

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