JP4151016B2 - Isolated switching DC / DC converter - Google Patents

Description

  The present invention relates to a switching power supply device, and more particularly to a technique for realizing high efficiency and miniaturization of an insulating switching DC / DC converter.

  Conventionally, a switching power supply circuit that employs a partial resonance circuit to realize zero volt switching has been proposed (see Non-Patent Document 1 and Non-Patent Document 2). FIG. 16 shows a conventional circuit configuration example. In FIG. 16, Vi is an input voltage source, C1 is a clamp capacitor, and Q1 'is a switch element (eg, MOSFET). Lr1 is a leakage inductance and functions as an inductance for resonance operation. Tr1 is a forward transformer (number of primary coil turns N, number of secondary coil turns n, where N and n are natural numbers), and Tr2 is a flyback transformer (number of primary coil turns N, number of secondary coil turns n). Q1 is a switch element (for example, MOSFET), Q2 and Q2 'are switch elements (for example, MOSFET) that function as synchronous rectifier elements, Co is an output smoothing capacitor, and Vo is an output DC voltage. The switching terminals Q2 and Q2 'for synchronous rectification have their gate terminals driven by voltages obtained from the secondary coils of the transformers Tr2 and Tr1, respectively.

  As a modification of FIG. 16, a configuration in which a series circuit of a clamp capacitor C1 and a switch element Q1 ′ is connected between the terminals of the switch element Q1 as shown in FIG.

  Next, the operation of the conventional circuit shown in FIG. 16 will be described. FIG. 18 is an operation waveform of each part of the circuit shown in FIG. In FIG. 18, VGS (Q1) and VGS (Q1 ') are the gate voltages of the switching elements Q1 and Q1', respectively. These two switch elements Q1 and Q1 'are alternately turned on / off by a control circuit (not shown) so that one of the two switch elements Q1 and Q1' is turned off while the other is turned on except for a dead time period in which both are turned off. . The output DC voltage Vo can be controlled by changing the ratio (ON duty ratio D) of the ON period DT of the switch element Q1 to the operating period T. That is, the output DC voltage Vo satisfies the relationship of the following equation.

[Equation 1]
Vo = (n / N) x D x Vi
VDS (Q1) and VDS (Q1 ′) in FIG. 18 are waveforms of drain-source voltages of the switch elements Q1 and Q1 ′, respectively. ID (Q1) is a sum of currents flowing through the switching element Q1, the body diode D1, and the output junction capacitor C11. ID (Q1 ′) is the sum of currents flowing through the switching element Q1 ′, the body diode D1 ′, and the output junction capacitor C12. ILr1 is the primary current of the transformer Tr1. ILm1 and ILm2 are exciting currents of the transformers Tr1 and Tr2, respectively.

  This DC / DC converter circuit can be divided into eight operating states of modes 1-8. 19 to 26 show equivalent circuits according to modes. The operation will be outlined with reference to an equivalent circuit corresponding to each mode.

[Mode 1; t0 ≤ t ≤ t1]
In the mode 1 period, as shown in FIG. 19, the switch elements Q1 and Q2 are on, and energy is transmitted to the output side via the transformer Tr1. At this time, since the current of the transformer Tr2 is blocked, the transformer Tr2 operates as a choke coil that stores energy. In FIG. 19, Lm1 and Lm2 indicate exciting inductances of the transformers Tr1 and Tr2, respectively, and RL indicates a load resistance.

[Mode 2; t1 ≤ t ≤ t2]
During this period, when the switch element Q1 is turned off at t = t1, the current continues to flow through the leakage inductance Lr1, so that the voltage of the output junction capacitor C11 of the switch element Q1 is charged until it becomes equal to the input voltage Vi (FIG. 20). Further, the voltage of the output junction capacitance C12 of the switch element Q1 ′ is discharged until it becomes equal to the voltage Vc1 of the clamp capacitor C1.

[Mode 3; t2 ≤ t ≤ t3]
When the voltages of the output junction capacitance C12 and the clamp capacitor C1 become equal at t = t2, the body diodes D2 and D2 'of the secondary side synchronous rectification switch elements Q2 and Q2' are both turned on, and the transformer secondary side It will be in a short circuit state (refer to Drawing 21). In addition, only the leakage inductance Lr1 becomes an inductance and performs a resonance operation.

[Mode 4; t3 ≤ t ≤ t4]
During the period of mode 4, at t = t3, the body diode D1 'of the switch element Q1' is forward biased as shown in FIG. 22, the body diodes D2 and D2 'remain on, and the current ID2 flowing through the body diode D2 is The current ID2 'flowing through the body diode D2' increases, and eventually the current ID2 becomes zero at t = t4.

[Mode 5; t4 ≤ t ≤ t5]
The period of mode 5 is a period in which the switch elements Q1 ′ and Q2 ′ are on as shown in FIG. 23, and the energy stored in the exciting inductance Lm2 of the transformer Tr2 is transmitted to the secondary side. During this time, the direction of the current flowing through the switching element Q1 ′ changes from negative (dotted arrow in FIG. 23) to positive (solid arrow in FIG. 23).

[Mode 6; t5 ≤ t ≤ t6]
During the period of mode 6, when the switch element Q1 'is turned off at t = t5, the current continues to flow through the leakage reactance Lr1, whereby the output junction capacitor C11 is discharged to the input voltage Vi, and the output junction capacitor C12 is clamped. The capacitor C1 is charged up to the voltage Vc1 (see FIG. 24).

[Mode 7; t6≤t≤t7]
When the voltage Vc12 of the output junction capacitor C12 becomes equal to the voltage Vc1 of the clamp capacitor C1 at t = t6 (Vc12 = Vc1), the body diodes D2 and D2 'are both turned on, and the transformer secondary side is short-circuited (see FIG. 25). For this reason, only the leakage inductance Lr1 functions as an inductance and performs a resonance operation.

[Mode 8; t7 ≤ t ≤ t8]
At t = t7, the body diode D1 of the switch element Q1 is forward biased and turned on (see FIG. 26). Both body diodes D2 and D2 'remain on, current ID2' decreases, and current ID2 increases. When the current ID2 'becomes zero at t = t8, the mode 1 is entered. During the mode 8 period, the current direction of the switching element Q1 changes from negative (dotted arrow in FIG. 26) to positive (solid arrow in FIG. 26).
Satoshi Tomioka, “Capacity Technology and Challenges for On-Board Power Supply”, 2000 Switching Power Supply System Symposium, Japan Management Association, B4-2 Kosuke Harada, “High Frequency Countermeasures for Switching Power Supplies”, first edition, Nikkan Kogyo Shimbun, February 25, 1997, p. 232-237

  However, the above-described conventional circuit configuration is a so-called double transformer system, and the output ripple voltage is small, but the total DC bias of the magnetic component core is very large, and the core volume is significantly large. Also, the core loss is greatly increased, and the efficiency of the entire apparatus is poor. Further, the conventional circuit configuration has a drawback that the number of parts is large.

  The present invention has been made in view of such circumstances, and the number of parts can be reduced by reducing the core volume and core loss of magnetic parts, improving the efficiency of the entire apparatus, and devising the circuit configuration. It is an object to provide a switching DC / DC converter that can achieve the above.

  In order to achieve the above object, according to the present invention, a DC voltage source is connected to the primary side of an isolation transformer, and both the first switch element and the second switch element connected to the primary side are simultaneously connected. In an isolated switching DC / DC converter that performs voltage conversion by alternately turning on and off so as not to be in an on state, and obtains a DC voltage output via the secondary side rectifier circuit of the insulation transformer, The primary coil of the insulating transformer, the secondary coil of the insulating transformer, and the output choke coil are wound around a common core, and the DC magnetic flux generated by the windings of these coils cancels each other out. The first switching element connected in series with the primary coil and the primary coil are arranged on the primary side of the insulating transformer. A primary side circuit including the input choke coil connected between the terminals, a series circuit of a clamp capacitor connected between the terminals of the primary coil and the second switch element is formed; On the secondary side of the transformer, a first rectifying element and a second rectifying element connected to the secondary coil, and the output choke coil connected to the first and second rectifying elements, And a secondary side circuit including an output smoothing capacitor connected to the output choke coil.

  According to the present invention, in a DC / DC converter using an insulating transformer, an input choke coil and an output choke coil are integrated with the insulating transformer, and the primary coil, secondary coil, and input / output coils of the transformer are integrated into the same core (magnetic core). By winding the output choke coil and designing the number of turns and the direction of winding in a direction that cancels the DC magnetic flux generated by each coil, the amount of DC bias in the core is made extremely small. As a result, the volume of the core can be significantly reduced as compared with the conventional case, and the core loss can be reduced, so that a highly efficient device can be realized. Further, since the circuit configuration can be realized with a single transformer, the number of parts can be reduced as compared with the conventional double transformer system.

  As another aspect of the present invention, the connection location of the series circuit of the clamp capacitor and the second switch element in the primary circuit is changed, and the clamp capacitor and the second switch are connected between the terminals of the first switch element. There is also an aspect in which a primary circuit having a configuration in which a series circuit with an element is connected is used.

  Further, in the circuit configuration of the present invention described above, a configuration in which a capacitor is inserted in series with the primary coil (transformer primary winding) of the insulation transformer is also preferable in order to prevent DC demagnetization of the insulation transformer.

  In another aspect of the present invention, in order to preliminarily distinguish between the input choke coil (primary choke winding) and the primary coil of the insulation transformer (transformer primary winding), the secondary of the insulation transformer By providing a difference in the degree of coupling with the coil (transformer secondary winding) in advance, the higher degree of coupling becomes the primary coil of the insulating transformer, and the lower degree of coupling becomes the input choke coil. Features.

  According to the present invention, the input choke coil and the output choke coil are integrated with the insulating transformer, and the primary coil, the secondary coil, and the input / output choke coil of the transformer are wound around the same core, and each coil is made. Since the amount of DC bias in the core has been made extremely small in the direction that cancels the DC magnetic flux, the core volume can be significantly reduced compared to the previous model, and the core loss can be reduced, resulting in high efficiency of the device. Can be achieved. In addition, since it can be realized by a single transformer in terms of circuit configuration, there is an advantage that the number of parts is small as compared with the conventional double transformer system.

  Hereinafter, preferred embodiments of the present invention will be described in detail with reference to the accompanying drawings.

  FIG. 1 is a circuit diagram of a DC / DC converter (step-up active clamp forward converter) according to an embodiment of the present invention. In FIG. 1, Vi is an input voltage source, C1 is a clamp capacitor, Q1 'is a switching element using an FET, Lr1 is a leakage inductance, L1 is an input choke coil (number of turns N, N is a natural number), and Lr2 is a leakage inductance. , Tr is an insulating transformer (the number of turns of the primary coil is N, the number of turns of the secondary coil is n, n is a natural number), Q1 is a switching element using an FET, Q2 and Q2 'are switching elements that function as synchronous rectifying elements, and Lr3 is The leakage inductance or the sum of the external insertion inductance and the leakage inductance, L3 is an output choke coil (number of turns n), Co is an output smoothing capacitor, and Vo is an output DC voltage. In this example, MOSFETs are used as the switching elements Q1, Q1 ', Q2, Q2', but other semiconductor elements may be used in the practice of the present invention.

  The input / output choke coils L1 and L3 are integrated with the transformer Tr, and the primary and secondary coils of the input / output choke coils L1 and L3 and the transformer Tr are wound around a common core (for example, EI type core). In other words, the DC magnetic fluxes generated by the windings of the respective coils are configured to cancel each other.

  In order to distinguish the input choke coil L1 from the primary coil of the transformer Tr in advance, the degree of coupling between the transformer Tr and the secondary coil is designed so that the degree of coupling between the transformer Tr and the secondary coil is different. The higher one is the input choke coil L1 with the lower degree of coupling with the primary coil of the transformer Tr and the lower degree of coupling with the secondary coil of the transformer Tr. An example of the transformer / choke coil winding is the configuration shown in FIG.

  FIG. 2 shows an example using an EI type core. As shown in the figure, the EI core 20 has a structure in which an E core 21 and an I core 22 are combined. A gap 23 is provided on the connection surface between the three magnetic legs 21 </ b> A, 21 </ b> B, 21 </ b> C of the E-shaped core 21 and the I-shaped core 22.

  One outer magnetic leg 21A of the E-shaped core 21 is wound with an input choke coil L1, a primary coil Np (number of turns N) and a secondary coil Ns (number of turns n) of a transformer Tr as shown in FIG. An output choke coil L3 having n turns is wound around the magnetic leg 21B. The right outer magnetic leg 21C in FIG. 2 is a magnetic leakage leg into which leakage magnetic flux flows.

  The direction of winding of each coil wound around the outer magnetic leg 21A and the central magnetic leg 21B is generated in each coil when a current flows through each winding, as indicated by a dot (·) in the figure. Magnetomotive forces are in a direction to cancel each other. In this way, by configuring a plurality of windings in a direction that cancels the DC magnetic flux, the amount of DC bias of the core can be made extremely small. Ideally, the DC magnetic flux generated in the core can be made zero, and a UI core without the magnetic leakage leg 21C and a core without the gap 23 can be used. In practice, however, magnetic leaking legs are used for slight AC voltage imbalance.

  Thus, the volume of the entire core can be significantly reduced, and the core loss can be reduced and the efficiency of the entire apparatus can be improved. In the implementation of the present invention, the shape of the core is not limited to the EI type core, and it is possible to use a core having another shape such as an EE type core.

  Next, the operation of the circuit shown in FIG. 1 will be described.

  FIG. 3 is an operation waveform of each part of the circuit shown in FIG. In FIG. 3, periods other than mode 1 and mode 5 are shown longer than actual.

  In FIG. 3, VGS (Q1) and VGS (Q1 ') are the gate voltages of the switch elements Q1 and Q1', respectively. These two switching elements Q1 and Q1 'are alternately turned on / off by a control circuit (not shown) so that one of the two switching elements Q1 and Q1' is turned off while the other is turned on except for the periods of the dead times Td1 and Td2. The output DC voltage Vo can be controlled by changing the ratio (ON duty ratio D) of the ON period DT of the switch element Q1 to the operating period T. The output DC voltage Vo satisfies the relationship of the following equation.

[Equation 2]
Vo = (n / N) x D x Vi
VDS (Q1) in FIG. 3 is a waveform of the drain-source voltage of the switch element Q1. ID (Q1) is a sum of currents flowing through the switching element Q1, the body diode D1, and the output junction capacitor C11. ID (Q1 ′) is the sum of currents flowing through the switching element Q1 ′, the body diode D1 ′, and the output junction capacitor C12. ID (Q2) is a sum of currents flowing through the switching element Q2 and the body diode D2. ID (Q2 ′) is the sum of currents flowing through the switching element Q2 ′ and the body diode D2 ′.

  The DC / DC converter circuit of this example can be divided into eight operating states of modes 1-8. 4 to 11 are equivalent circuits of the modes 1 to 8. The operation will be outlined with reference to an equivalent circuit corresponding to each mode.

[1] Mode 1; t0 ≤ t ≤ t1
In the mode 1 period, as shown in FIG. 4, the switch elements Q1 and Q2 are turned on, and energy is transmitted to the output side via the output choke coil L3. At this time, since the current of the transformer Tr is blocked, only the exciting current flows. In FIG. 4, Lm represents the excitation inductance of the transformer Tr, and RL represents the load resistance.

[2] Mode 2; t1 ≤ t ≤ t2
When the switch element Q1 is turned off at t = t1, as shown in FIG. 5, the current continues to flow through the leakage inductances Lr1 and Lr2, thereby charging the output junction capacitor C11 of the switch element Q1 and the output of the switch element Q1 '. The junction capacitance C12 is discharged. The voltage of the output junction capacitor C11 is charged to the input voltage Vi, and the voltage of the output junction capacitor C12 is discharged until it becomes equal to the voltage of the clamp capacitor C1.

[3] Mode 3; t2 ≤ t ≤ t3
When the voltages of the output junction capacitance C12 and the clamp capacitor C1 become equal at t = t2, both the body diodes D2 and D2 'are turned on, and the secondary side of the transformer Tr is short-circuited (see FIG. 6). Also, during this period, only the leakage inductances Lr1 and Lr2 become inductances and perform a resonance operation.

[4] Mode 4; t3 ≤ t ≤ t4
During the period of mode 4, at t = t3, the body diode D1 'of the switch element Q1' is forward-biased as shown in FIG. 'Is the period until the current ID2 increases to zero.

[5] Mode 5; t4 ≤ t ≤ t5
The period of mode 5 is a period from when the body diode D2 is turned off at t = t4 to when the switch element Q1 'is turned off at t = t5 as shown in FIG. In this mode 5, the switch elements Q1 ′ and Q2 ′ are both turned on, and the direction of the current of the switch element Q1 ′ changes from negative (dotted arrow in FIG. 8) to positive (solid arrow in FIG. 8). At t = t4, the winding of the transformer Tr is released from the short-circuited state, and a winding voltage is generated.

[6] Mode 6; t5 ≤ t ≤ t6
When the switch element Q1 'is turned off at t = t5, as shown in FIG. 9, the current continues to flow through the leakage inductances Lr1 and Lr2, thereby supplying the load current and the output junction capacitance C11. Is discharged to the input voltage Vi, and the output junction capacitance C12 is charged to the voltage Vc1 of the clamp capacitor C1.

[7] Mode 7; t6 ≤ t ≤ t7
When the voltage Vc12 of the output junction capacitor C12 becomes equal to Vc1 at t = t6 (Vc12 = Vc1), both the body diodes D2 and D2 'are turned on as shown in FIG. 10, and the secondary side of the transformer Tr is in a short-circuited state. Become. For this reason, only the leakage inductances Lr1 and Lr2 function as inductances and perform a resonance operation.

[8] Mode 8; t7 ≤ t ≤ t8
At t = t7, the body diode D1 of the switch element Q1 is forward biased and turned on (see FIG. 11). Both body diodes D2 and D2 'remain on, current ID2' decreases, and current ID2 increases. When the current ID2 'becomes zero, the mode 1 is entered. During the mode 8 period, the current direction of the switching element Q1 changes from negative (dotted arrow in FIG. 11) to positive (solid arrow in FIG. 11).

  According to the switching DC / DC converter according to the above-described embodiment of the present invention, the input choke coil L1, the output choke coil L3, and the transformer Tr are integrated so that the magnetic flux generated by each winding is canceled and the DC dc bias amount of the core. Since the core is made very small, the core can be greatly reduced in size and the core loss can be greatly reduced as compared with the conventional circuit configuration. As a result, the efficiency of the entire apparatus can be dramatically improved and the number of parts can be reduced.

  The application range of the present invention is not limited to the circuit configuration shown in FIG. 1, and various modifications are possible. 12 to 14 show circuit modifications. For each circuit, the main differences from the circuit of FIG. 1 are pointed out, and the description of the circuit operation is omitted. 1, FIG. 12 and FIG. 13 have the switching element Q1 inserted below the DC bus, but the same is true if the switching element Q1 is inserted above the DC bus.

  The circuit shown in FIG. 12 is different from the circuit of FIG. 1 in the connection place of the series circuit of the switch element Q1 ′ and the capacitor C1. That is, in FIG. 12, a series circuit of a switch element Q1 'and a capacitor C1 is connected between the terminals of the switch element Q1.

  The circuit shown in FIG. 13 is different from the circuit shown in FIG. 12 in that the connection relationship between the capacitor C1 and the switch element Q1 ′ is switched and a P-channel MOSFET is used for the switch element Q1 ′.

  In addition, a configuration shown in FIG. 14 is possible as a secondary side circuit of the transformer Tr. In FIG. 14, the primary side circuit of the transformer Tr is not shown, but the primary side circuit of the transformer Tr which is omitted in the figure is shown in any one of FIGS. The circuit configuration shown can be used. The circuit shown in FIG. 14 is different from the circuit of FIG. 1 in the connection place of the switch element Q2 ′.

  FIG. 15 is a circuit diagram showing still another embodiment of the present invention. In the figure, parts common to those in FIG. According to FIG. 15, a capacitor C2 is inserted in series with the transformer primary winding (primary coil Np) in order to prevent direct current bias of the transformer Tr.

  FIG. 15 shows an example in which the capacitor C2 is added to the circuit diagram of FIG. 1, but the capacitor C2 is also connected in series with the transformer primary winding (primary coil Np) in the circuit diagrams shown in FIGS. Can be inserted.

  The present invention is particularly useful in the case of a wide range, a high input voltage, a large capacity, and a low output voltage (for example, an input voltage of DC 200 to 400 V, an output of 15 V), such as for vehicles such as electric vehicles and hybrid vehicles. Of course, the application range of the present invention is not limited to this, and can be applied to power supplies of various uses and specifications.

1 is a circuit diagram of a DC / DC converter according to a first embodiment of the present invention. The figure which shows the example which utilizes EI type core The figure which shows the operation waveform of each part of the circuit shown in FIG. Equivalent circuit diagram showing operation of mode 1 in the DC / DC converter of this example Equivalent circuit diagram showing operation of mode 2 in DC / DC converter of this example Equivalent circuit diagram showing operation of mode 3 in DC / DC converter of this example Equivalent circuit diagram showing operation of mode 4 in DC / DC converter of this example Equivalent circuit diagram showing operation of mode 5 in DC / DC converter of this example Equivalent circuit diagram showing operation of mode 6 in the DC / DC converter of this example Equivalent circuit diagram showing operation of mode 7 in DC / DC converter of this example Equivalent circuit diagram showing operation of mode 8 in the DC / DC converter of this example The circuit diagram which shows the 2nd Embodiment of this invention Circuit diagram showing a third embodiment of the present invention Circuit diagram showing a fourth embodiment of the present invention Circuit diagram showing a fifth embodiment of the present invention Circuit diagram showing an example of a conventional DC / DC converter Circuit diagram showing another example of a conventional DC / DC converter The figure which shows the operation waveform of each part of the conventional circuit shown in FIG. Equivalent circuit diagram showing operation of mode 1 in the conventional DC / DC converter shown in FIG. Equivalent circuit diagram showing operation of mode 2 in the conventional DC / DC converter shown in FIG. Equivalent circuit diagram showing operation of mode 3 in the conventional DC / DC converter shown in FIG. Equivalent circuit diagram showing operation of mode 4 in the conventional DC / DC converter shown in FIG. Equivalent circuit diagram showing operation of mode 5 in the conventional DC / DC converter shown in FIG. Equivalent circuit diagram showing operation of mode 6 in the conventional DC / DC converter shown in FIG. Equivalent circuit diagram showing the operation of mode 7 in the conventional DC / DC converter shown in FIG. Equivalent circuit diagram showing operation of mode 8 in the conventional DC / DC converter shown in FIG.

Explanation of symbols

  Vi: input voltage source, L1: input choke coil, Q1, Q1 ': switch element, C1: clamp capacitor, C2: capacitor, Tr: transformer, Q2, Q2': switch element, L3: output choke coil, Co: output Smoothing capacitor, 20 ... EI type core

Claims (6)

A DC voltage source is connected to the primary side of the isolation transformer, and the first switch element and the second switch element connected to the primary side are alternately turned on / off so that they are not simultaneously turned on. In the insulation type switching DC / DC converter that performs voltage conversion and obtains a DC voltage output through the secondary side rectifier circuit of the insulation transformer,
An input choke coil, a primary coil of the insulation transformer, a secondary coil of the insulation transformer, and an output choke coil are wound around a common core, and a DC magnetic flux generated by the windings of these coils. Are configured to cancel each other out,
The primary side of the isolation transformer includes the first switch element connected in series with the primary coil, the input choke coil connected between terminals of the primary coil, and the primary coil. A primary circuit including a clamp capacitor connected between the terminals and a series circuit of the second switch element is formed;
A secondary side of the isolation transformer includes a first rectifying element and a second rectifying element connected to the secondary coil, and the output choke connected to the first and second rectifying elements. An insulating switching DC / DC converter, wherein a secondary circuit including a coil and an output smoothing capacitor connected to the output choke coil is formed. A DC voltage source is connected to the primary side of the isolation transformer, and the first switch element and the second switch element connected to the primary side are alternately turned on / off so that they are not simultaneously turned on. In the insulation type switching DC / DC converter that performs voltage conversion and obtains a DC voltage output through the secondary side rectifier circuit of the insulation transformer,
An input choke coil, a primary coil of the insulation transformer, a secondary coil of the insulation transformer, and an output choke coil are wound around a common core, and a DC magnetic flux generated by the windings of these coils. Are configured to cancel each other out,
On the primary side of the isolation transformer, the first switch element connected in series with the primary coil, the input choke coil connected between terminals of the primary coil, and the first switch A primary circuit including a clamp capacitor connected between terminals of the element and a series circuit of the second switch element is formed;
A secondary side of the isolation transformer includes a first rectifying element and a second rectifying element connected to the secondary coil, and the output choke connected to the first and second rectifying elements. An insulating switching DC / DC converter, wherein a secondary circuit including a coil and an output smoothing capacitor connected to the output choke coil is formed.   The number of turns of the primary choke of the input choke coil and the insulating transformer is N (N is a natural number), and the number of turns of the secondary coil of the insulating transformer and the output choke coil is n (n is a natural number). The isolated switching DC / DC converter according to claim 1 or 2. 4. The isolated switching DC / DC converter according to claim 1, wherein MOSFETs are used for the first and second switch elements and the first and second rectifying elements.   5. The isolated switching DC / DC according to claim 1, wherein a capacitor is connected in series with a primary coil of the insulating transformer in order to prevent direct current demagnetization of the insulating transformer. DC converter.   In order to make a distinction between the input choke coil and the primary coil of the insulation transformer in advance, by providing a difference in the degree of coupling with the secondary coil of the insulation transformer in advance, the higher the degree of coupling, the more the insulation transformer 6. The isolated switching DC / DC converter according to claim 1, wherein the input choke coil having the lower coupling degree is the input choke coil.

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