JP3959617B2 - AC motor constant measuring method and control device - Google Patents

Description

より、漏れインダクタンスｌ＝ｌ１＋ｌ２を演算測定する交流電動機の定数測定方法において、出力電圧誤差のうち少なくともパワー半導体素子のオン電圧降下による振幅および位相差誤差を、測定した電圧指令の絶対値の平均値Ｖ＿ａｖｅおよび電圧指令と出力電流の位相差θ＿ｄｉｆに補正した後、一次および二次の合成抵抗（Ｒ１＋Ｒ２）と漏れインダクタンスｌ＝ｌ１＋ｌ２を演算測定することを特徴としている。
また、請求項２記載の発明は、請求項１記載の交流電動機の定数測定方法において、電流の関数としたパワー半導体素子のオン電圧降下量を内蔵し、少なくとも測定交流電流のピーク値におけるパワー半導体素子のオン電圧降下量を用いて前記振幅誤差と前記位相差誤差を補正することを特徴としている。
また、請求項３記載の発明は、交流電動機の制御装置に係り、出力電圧の大きさ、周波数および位相の制御が可能なパワー半導体素子から構成される電力変換器を介して供給される一次電流を、該交流電動機の磁束と平行な成分（励磁電流）とこれに直交する成分（トルク電流）とに分離して各々を独立に調整し、交流電動機の各相に流れる出力電流を検出する電流検出器を有し、インバータ装置は、前記交流電動機の回転を停止させた状態で、正弦波発生器により交番磁束が発生するように電圧位相を予め設定された任意の値に固定し、周波数ｆｈの正弦波の電圧指令を与える手段と、電圧指令値は出力電流の大きさが所定値になるように調整した後、所定時間経過後に前記電圧指令の絶対値の平均値Ｖ＿ａｖｅおよび出力電流の絶対値の平均値Ｉ＿ａｖｅおよび前記電圧指令と出力電流の位相差θ＿ｄｉｆを測定する手段を具備し、前記Ｖ＿ａｖｅとＩ＿ａｖｅとの比のｃｏｓ成分から一次および二次の合成抵抗（Ｒ１＋Ｒ２）を演算測定し、前記Ｖ＿ａｖｅとＩ＿ａｖｅとの比のｓｉｎ成分から演算測定するか、
あるいは次式、
【数２】

より漏れインダクタンスｌ＝ｌ１＋ｌ２を演算測定する交流電動機の制御装置において、出力電圧誤差のうち少なくともパワー半導体素子のオン電圧降下による振幅および位相差誤差を、測定した電圧指令の絶対値の平均値Ｖ＿ａｖｅおよび電圧指令と出力電流の位相 差θ＿ｄｉｆに補正する手段を具備し、これら補正したＶ＿ａｖｅおよびθ＿ａｖｅを用いて一次および二次の合成抵抗（Ｒ１＋Ｒ２）と漏れインダクタンスｌ＝ｌ１＋ｌ２を演算測定し、これらの測定値を基に交流電動機を制御することを特徴としている。
また、請求項４記載の発明は、請求項３記載の交流電動機の制御装置において、前記測定した電圧指令の絶対値の平均値Ｖ＿ａｖｅおよび電圧指令と出力電流の位相差θ＿ｄｉｆに補正する手段は、電流の関数としたパワー半導体素子のオン電圧降下量を内蔵し、少なくとも測定交流電流のピーク値におけるパワー半導体素子のオン電圧降下量を用いて前記振幅誤差と前記位相差誤差を補正することを特徴としている。
【０００７】
【発明の実施の形態】
以下、本発明の第１の実施の形態について図を参照して説明する。
図１は本発明の第１の実施の形態に係る電圧形インバータの回路構成図である。
図２は図１に示すインバータ制御装置の詳細回路図である。
図３は図１に示す平均値・位相差演算器の詳細回路図である。
図４は図１に示す電動機の等価回路図である。
図５は図１に示す電圧形インバータの動作のフローチャーチャートである。
図６は図４に示す等価回路のインピーダンスのベクトル図である。
図７は図３に示す平均的・位相差演算器の電圧指令値・電流検出値のタイムチャートである。
図１において、１は電圧形インバータ、２は交流電動機、３は電流検出器、４は比較器、５は発振器、６は加算器、７はゲート回路、８はインバータ制御装置、 ９はデッドバンド補償器、１０は直流電源、１１は速度指令回路、１４は速度検出器、２７は平均値・位相演算器である。
同図において、電圧形インバータ１は直流電源１０から加えられる直流電圧をＰＷＭ制御方式により任意の周波数と電圧の交流に変換する。電圧形インバータ１はトランジスタやＩＧＢＴ等のパワー半導体素子からなるスイッチング素子ＴＵＰ、ＴＶＰ、…ＴＷＮと各スイッチング素子に逆並列接続された帰還ダイオードＤＵＰ、ＤＶＰ、…ＤＷＮとから構成される。
【０００８】
電圧形インバータ１の各相Ｕ、Ｖ、Ｗの交流出力端に交流電動機２が接続されている。交流電動機２のＵ相、Ｖ相、Ｗ相の１次電流Ｉｕ、Ｉｖ、Ｉｗは電流検出器３ｕ、３ｖ、３ｗによって検出される。また、速度検出器１４により交流電動機２の回転速度を検出する。
インバータ制御装置８には、速度指令回路１１で作成された速度指令値ωｒ＊と、電流検出器３ｕ、３ｖ、３ｗにより検出した交流電動機２のＵ相、Ｖ相、Ｗ相の１次電流Ｉｕ、Ｉｖ、Ｉｗと、速度検出器１４からの速度検出値ωｒが加えられ、１２０°位相差のＵ、Ｖ、Ｗの各相の電圧指令パターン信号（Ｖｕ^*、Ｖｖ^*、Ｖｗ^*）と、デッドバンドによる電圧誤差を補償するデッドバンド補償器９ｕ、９ｖ、９ｗに位相γを出力する。デッドバンド補償器９ｕ、９ｖ、９Ｗは位相γを引数として電圧誤差補償値を加算器６ｕ、６ｖ、６ｗに出力する。
【０００９】
また、加算器６ｕ、６ｖ、６ｗは、電圧パターン信号Ｖｕ^*、Ｖｖ^*、Ｖｗ^*と、デッドバンド補償器９ｕ、９ｖ、９ｗの出力値を加算し、加算値を比較器４ｕ、４ｖ、４ｗに送出する。ＰＷＭ制御のための搬送波信号を発生する発振器５の出力信号も比較器４ｕ、４ｖ、４ｗへ送出され、比較器４ｕ、４ｖ、４ｗは加算器６ｕ、６ｖ、６ｗの出力信号と搬送波信号を比較し、電圧形インバータ１を構成するスイッチング素子ＴＵＰ、ＴＶＰ、ＴＷＮをオン／オフするためのＰＷＭパルスを発生する。
ゲート回路７は比較器４ｕ、４ｖ、４ｗの出力するＰＷＭパルスに応じてスイッチング素子ＴＵＰ、ＴＶＰ、…ＴＷＮにゲート信号を与える。
【００１０】
図２において、８はインバータ制御装置、１２は励磁電流指令回路、１５は３相／２相変換器、１６は２相／３相変換器、１７は一次角周波数演算回路、１８は速度制御回路、１９はトルク電流制御回路、２０は励磁電流制御回路、２１は電圧指令補償回路、２２は加算器、２３は積算器、２４は位相計算器、２５はチューニング処理部、２６は電流振幅演算器である。
同図において、インバータ制御装置８には、交流電動機２への一次電流Ｉｕ、Ｉｖ、Ｉｗを検出して座標変換を行ったトルク電流帰還値Ｉｑｆｂと、励磁電流帰還値Ｉｄｆｂを送出する３相／２相変換器１５が設けられている。速度指令回路１１からの速度指令値ωｒ^*と、速度検出器１４からの速度検出値ωｒが一致するように設けられた速度制御回路ＡＳＲ１８の出力値をトルク電流指令値Ｉｑｒｅｆとし、このＩｑｒｅｆと３相／２相変換器１５が出力するトルク電流帰還値Ｉｑｆｂとが一致するように制御するためのトルク電流制御回路ＡＣＲｑ１９と、励磁電流指令回路１２からの励磁電流指令値Ｉｄｒｅｆと、３相／２相変換器１５からの励磁電流帰還値Ｉｄｆｂとが一致するように、励磁電流方向電圧を制御する励磁電流制御回路ＡＣＲｄ２０が設けられている。
【００１１】
また、交流電動機２の磁束で発生し、誘導起電力係数による誘起電圧と一次抵抗Ｒ１による逆起電力のトルク電流方向成分の電圧と、交流電動機２の漏れインダクタンス（ｌ＝ｌ１＋ｌ２）や一次抵抗Ｒ１による逆起電力の励磁電流方向成分の電圧を出力する電圧指令補償回路２１を有している。電圧指令補償回路２１の出力のうちトルク電流方向成分の電圧は、トルク電流制御回路１９の出力と、加算器２２Ａで加算されトルク電流方向電圧指令値Ｖｑｒｅｆを生成し、励磁電流方向成分の電圧は、励磁電流制御回路２０の出力と加算器２２Ｂで加算され励磁電流方向電圧指令値Ｖｄｒｅｆを生成する。
更に、トルク電流方向電圧指令値Ｖｑｒｅｆと、励磁電流方向電圧指令値Ｖｄｒｅｆとから、１２０°位相差のＵ、Ｖ、Ｗの各相の電圧指令パターン信号（Ｖｕ^*、Ｖｖ^*、Ｖｗ^*）を生成して出力する２相／３相変換器１６が設けられている。
【００１２】
なお、３相／２相変換器１５、２相／３相変換器１６は、夫々次の（１）式、（２）式で演算される。
【数５】

また、インバータ制御装置８は、Ｉｄｒｅｆ、Ｉｑｒｅｆと設定された二次抵抗Ｒ２から、すべり周波数指令値ωｓ^*を求め、速度検出器１４からの速度検出値ωｒとから一次角周波数ω１^*を演算して出力する一次角周波数指令演算回路１７を有し、一次角周波数指令演算回路１７からの一次角周波数ω１^*は、積算器２３により積算され、３相／２相変換器１５と２相／３相変換器１６へ、位相θとして出力される。
また、位相θは位相計算回路２４の出力であるｔａｎ^-1（Ｉｑｆｂ／Ｉｄｆｂ）と加算器２２ｃで加算され、位相γとしてデッドバンド補償器９ｕ、９ｖ、９ｗへ出力される。
【００１３】
図２のインバータ制御装置中、本発明の中核部分を構成するチューニング処理は、実運転前の合成抵抗、漏れインダクタンス測定動作をコントロールするチューニング処理部２５と、チューニング処理部２５からの切替信号Ｃｓｗにより実運転と定数測定動作を切替えられるスイッチ回路１３Ａ、１３Ｂ、１３Ｃを設けて、実運転前の定数測定時にはｂ端子側へ切替え、スイッチ回路１３Ａではトルク電流方向電圧指令値Ｖｑｒｅｆは０に切替え、スイッチ回路１３Ｂでは励磁電流方向電圧指令値Ｖｄｒｅｆは、チューニング処理部２５で定数測定用に設定したチューニング電圧指令値Ｖ＿ｒｅｆに切替え、スイッチ回路１３Ｃでは位相θを設定値（固定値）に切替える処理を行う。
従って、定数測定時のチューニング処理部２５の出力としては、スイッチ切替信号Ｃｓｗ、定数測定用信号として設定されたチューニング電圧指令値Ｖ ｒｅｆ、および同じ位相θｈを出力し、スイッチ１３Ｃにより位相θに任意の固定値に設定する処理を行う。
また、電流振幅演算器２６はトルク電流帰還値Ｉｑｆｂと励磁電流帰還値Ｉｄｆｂを入力として、出力電流の振幅値Ｉ＿ｆｂを演算出力する。
【００１４】
図３において、２７は平均値・位相演算器、２８はＬＰＦ、２９はＨＰＦ、３０は絶対値回路、３１は乗算器、３２は減算器、３３は正弦波発生器である。
同図において、平均値・位相差演算器２７は、インバータ制御装置８からの定数測定時の信号Ｖ＿ｒｅｆ、Ｉ＿ｆｂ、位相θｈを入力して、両信号の位相差θ＿ｄｉｆと、各信号の絶対値の平均値Ｖ＿ａｖｅ、Ｉ＿ａｖｅを演算するものであり、直流分を除去するハイパスフィルターＨＰＦ２９Ａ、２９Ｂと、ｓｉｎ、ｃｏｓを出力する正弦波発生器３３と、ＨＰＦ２９の出力とｓｉｎθｈ、ｃｏｓθｈを乗算する乗算回路３１Ａ〜Ｄと、平均値処理を行うローパスフィルタＬＰＦ２８Ａ〜Ｆと、絶対値演算回路（ＡＢＳ）３０Ａ、３０Ｂと、位相計算回路２４Ｂ、２４Ｃを有している。
【００１５】
以上の図１に示した電圧形インバータ１と、図２に示したインバータ制御装置８と、図３に示した平均値・位相差演算器２７による、定数測定処理の概略は、先ず、図２に示すインバータ制御装置８において、実運転前に、チューニング処理部２５より切替信号Ｃｓｗによりスイッチ１３Ａ、１３Ｂ、１３Ｃを測定用のｂ端子側へ切替えて、Ｖｑｒｅｆは０に、Ｖｄｒｅｆは定数測定用の電圧指令値Ｖ＿ｒｅｆに、位相θは固定値に設定して、定数測定時の電動機停止状態での電圧指令パターン信号Ｖｕ^*〜Ｖｗ^*を出力して、図１に示す電圧形インバータにより交流電動機２を駆動し、定数測定時の一次電流Ｉｕ、Ｉｖ、Ｉｗを検出する。定数測定用電圧指令Ｖ＿ｒｅｆと同位相θｈと、電流振幅Ｉｆｂを図３の平均値・位相差演算器２７へ出力する。
平均値・位相差演算器２７では、Ｖ＿ｒｅｆ、θｈ、Ｉｆｂ信号より位相差θ＿ｄｉｆと、平均値Ｖ＿ａｖｅ、Ｉ＿ａｖｅを演算し、定数演算器（図示していない）により合成抵抗（Ｒ１＋Ｒ２）、漏れインダクタンス（ｌ１＋ｌ２）を求めて記憶する。
定数測定が終了したら、図２のスイッチ１３Ａ、１３Ｂ、１３Ｃを切替え、実運転用のａ端子側に戻し、求めた電動機定数を用いて、図１に示すインバータ１を制御し実運転を行うものである。
【００１６】
つぎに図５を参照して本発明の定数測定方法について説明する。
定数測定時（チューニング時）には、インバータ制御装置８のチューニング処理部２５において、チューニング時に流す定数測定用の交流電流の大きさと周波数ｆｈを決める（Ｓ１００）。
この場合の交流電流の大きさは、電圧形インバータ１と交流電動機２の定格電流値を基に、例えば、２つの定格電流値のうち小さい方の５０％〜１００％程度とし、周波数ｆｈは、２π・ｆh ・Ｍ≫Ｒ２が成立する周波数とする。
図４（ａ）は交流電動機のＴ−１型等価回路であり、Ｒ１、Ｒ２、ｌ＝ｌ１＋ｌ２、Ｍはそれぞれ１次抵抗、２次抵抗、漏れインダクタンス、相互インダクタンスであるが、定数測定用ｆｈのように印加する運転周波数が高いと、ωＭ≫Ｒ２となるので、Ｍには殆ど電流は流れず等価回路は図４（ｂ）のようになる。
また、この時の電圧と電流の位相差θ＿ｄｉｆと、（Ｒ１＋Ｒ２）、（ｌ１＋ｌ２）×２πｆｈ、Ｚ、の関係は図６のようになる。
【００１７】
次に、チューニング処理部２５は、定数測定時には切替信号Ｃｓｗにより、スイッチ回路１３ＡではＶｑｒｅｆを０に、１３Ｂではチューニング電圧指令値Ｖ＿ｒｅｆに、１３Ｃでは位相θを固定値に設定する（Ｓ１０１）。
ここでＶ＿ｒｅｆ＝Ｖａｍｐ・ｓｉｎ（２πｆｈ・ｔ）、であり、Ｖａｍｐの初期値は０として測定運転を開始する。
その後、電流検出値の絶対値の平均値Ｉ＿ａｖｅが、Ｓ１００で決めた所定の電流値になるように、Ｖａｍｐ（Ｖ＿ｒｅｆの振幅）を増加させる。そしてＩ＿ａｖｅの値が所定値に達したら、平均値・位相差演算器２７内のＬＰＦ２８Ａ〜２８Ｆの出力が安定するまで、所定時間の経過を待つ（Ｓ１０２）。
【００１８】
なお、電流平均値Ｉ＿ａｖｅは、図３に示す平均値・位相差演算器２７において、Ｉ＿ｆｂ入力から絶対値回路３０Ｂと平均値処理回路ＬＰＦ２８Ｆにより演算している。Ｖ＿ａｖｅは絶対値回路３０ＡとＬＰＦ２８Ｅにより求めている。また、基本信号と各信号の位相差θｖ´、θｉ´を位相計算回路２４Ｂ、２４Ｃの出力として求め、その差θｖ´−θｉ´を減算器３２出力θ＿ｄｉｆとして求めている。
次に、求めた電圧平均値Ｖ＿ａｖｅ、電流平均値Ｉ＿ａｖｅ、位相差θ＿ｄｉｆの値をそれぞれメモリに保存して、運転を停止する（Ｓ１０３）。
この時の電圧指令Ｖ＿ｒｅｆと、電流検出値Ｉ＿ｆｂの変化の様子を図７に示す。図７（ａ）には、ＬＰＦの出力が時間ｔの経過と共に安定する様子が、図７（ｂ）にはその時のＶ＿ｒｅｆとＩ＿ｆｂと位相差θ＿ｄｉｆの関係図が示されている。
【００１９】
Ｖ＿ａｖｅ、Ｉ＿ａｖｅ、θ＿ｄｉｆが求められたら、次の（３）式、（４）式、（５）式により、
【数６】

（Ｒ１＋Ｒ２）、（ｌ１＋ｌ２）を求める（Ｓ１０４）。
また、（３）式、（４）式、（５）式は、図６から、回路のインピーダンスをＺ＝（Ｖ＿ａｖｅ／√３）／Ｉ＿ａｖｅ、Ｚと実軸Ｒｅとのなす角をθ＿ｄｉｆとすれば、図示の関係から求められる。
【００２０】
以上の第１の実施の形態によれば、ここまでＶａｍｐの初期値を０として測定開始するように説明したが、実際運転時には、流れる電流値を交流電動機のＶ／ｆ特性値から予測し、予めいくらかの値を設定して、そこから加減することにより時間短縮することも可能である。
また、図３において各信号の平均値Ｖ＿ａｖｅ、Ｉ＿ａｖｅをＬＰＦ２８Ｅ、２８Ｆによって求めているが、移動平均によって求めてもよいし、Ｖ＿ａｖｅ、位相θＶ´は、チューニング電圧指令値Ｖ＿ｒｅｆの振幅、周波数、及びＬＰＦ２８Ｅの時定数が自明なので、演算で求めることもできる。
更に、チューニング電圧指令値Ｖ＿ａｖｅにオフセット値Ｖ＿ｒｅｆ＿ｏｆｓを加えたものを電圧指令値とすると、オフセット分の電圧は直流として出力されるので、この直流分から同時に一次抵抗Ｒ１を求めることもできる。Ｒ１が求まれば、（Ｒ１＋Ｒ２）よりＲ２を求めることは容易である。
また、本実施の形態では、１種類の振幅、周波数の交流信号Ｖ＿ｒｅｆで合成抵抗、漏れインダクタンスを測定したが、複数の条件で測定を行い、測定結果の平均を取ったり、オフセット量の影響を避けるために差分量を用いて演算する等しても本発明は実施できる。
【００２１】
次に、本発明の第２の実施の形態について図を参照して説明する。
図８は本発明の第２の実施の形態に係る電圧形インバータの動作のフローチャートである。
図９は図８に示す電圧形インバータのオン電圧降下量を示す図である。
図１０は図８に示すオン電圧降下量による補正処理の説明図である。
第１の実施の形態では、パワー半導体素子の電圧降下の補正を行っていないために、測定したＶ＿ａｖｅ、θ＿ｄｉｆに誤差が生じる場合があるので、第２の実施の形態では、Ｖ＿ａｖｅとθ＿ｄｉｆの値の補正手段（図示していない）を平均値・位相差演算器２７内に設けて、図８に示すように、第１の実施の形態の処理に、Ｓ１０５、Ｓ１０６の補正処理を追加している。
【００２２】
なお、図８において、Ｓ１００〜Ｓ１０４の処理は第１の実施の形態そのままなので、重複する説明は省略する。
先ず、Ｓ１００〜Ｓ１０３の処理により測定したＶ＿ａｖｅの補正を行う（Ｓ１０５）。
具体的には、測定交流電流のピーク値Ｉｐｋを電流の平均値×π／２として、電流の位相がθ＿ｄｉｆの時の瞬時電流値をＩｐｋ×ｓｉｎθ＿ｄｉｆ、として求める。次に、各電流値でのパワー半導体素子のオン電圧降下量の平均値Ｖｏｎ＿ａｖｅを次の（６）式で演算し、Ｖ＿ａｖｅからＶｏｎ＿ａｖｅを減じて、新たなＶ＿ａｖｅとする。
【００２３】

なお、Ｖｏｎ（Ｉ）は、電流Ｉが流れている時のパワー半導体素子のオン電圧降下量を示し、図９に示すようにスイッチング素子による降下量とダイオードによる降下量との平均値として求め、電流Ｉの関数として平均値・位相演算器２７に内蔵されている。
【００２４】
次に、位相差θ＿ｄｉｆの補正を行う（Ｓ１０６）。具体的には、位相補正量θｃｍｐを次の（７）式で演算し、θ＿ｄｉｆにθｃｍｐを加算して新たなθ＿ｄｉｆとする。
θｃｍｐ＝Ｋ×４／３×Ｖｏｎ（Ｉｐｋ）
／Ｖ＿ａｖｅ×θ＿ｄｉｆ／９０ …（７）
なお、（７）式中のＫは、電圧指令値Ｖ＿ａｖｅとピーク電流値Ｉｐｋの時のオン電圧降下量４／３×Ｖｏｎ（Ｉｐｋ）との比で決定される、位相角度の補正のための比例係数である。
【００２５】
上の（６）式、（７）式の、（９０−θ＿ｄｉｆ）／９０、とθ＿ｄｉｆ／９０は、パワー半導体素子のオン電圧降下の影響が平均することで相殺される残りの部分を位相９０°単位で平均していることを示している。
図１０にこの関係を示している。図１０（ａ）のＡ１とＡ２、図１０（ｂ）のＢ１とＢ２の領域での影響は相殺されること、Ａ１の領域はθ＿ｄｉｆと一致すること、Ｂ１の領域は９０−θ＿ｄｉｆと一致することが分かる。
また、図１０（ａ）から分かるように、電流０付近の電圧誤差の影響は相殺されてしまうために、微小電流でのオン電圧降下量の特性を知る必要が無く、オン電圧分の影響を補正することが可能になる。
最後に、Ｓ１０４でオン電圧量により補正された制御定数を演算して、測定を終了する（Ｓ１０４）。
【００２６】
このように本発明によれば、測定した合成抵抗、漏れインダクタンス測定値と、事前に設定された１次抵抗Ｒ１の値から、Ｒ２及び（ｌ＝ｌ１＋ｌ２）を求め、一次角周波数演算回路１７と、電圧指令補償回路２１に設定して、実運転時にはスイッチ回路１３Ａ〜１３Ｃをａ端子側に戻し切替えることで、測定した抵抗、漏れインダクタンス等の制御定数によって、ベクトル制御を行うことができる。
また、ここまでは実施対象を速度検出器付きの誘導電動機により説明したが、速度検出器無しの誘導電動機や、同期機を用いても適用可能である。すなわち、速度検出器の代りに速度推定器や速度指令値から周波数指令を作成したり、すべり周波数指令値ωｓ＊の補償をしない等で実施可能であり、速度推定器に測定した抵抗、漏れインダクタンスの制御定数を設定することもできる。なお、本発明により開示された合成抵抗、漏れインダクタンス測定方法は、実運転時の制御方法が異なっても、何等問題なく使用できることは言うまでもない。
【００２７】
【発明の効果】
以上説明したように、本発明によれば、合成抵抗（Ｒ１＋Ｒ２）、漏れインダクタンス（ｌ＝ｌ１＋ｌ２）等の制御定数を測定する場合に、インバータの出力電圧検出器が不要で、且つ、電動機を停止した状態で合成抵抗及び漏れインダクタンスの測定演算が可能であって、特に、パワー半導体素子のオン電圧降下の影響を電圧指令の平均値Ｖ＿ａｖｅ、及び電圧指令と出力電流の位相差θ＿ｄｉｆに補正することで、制御周期毎のオン電圧降下の瞬時補正、精度を考慮したパワー素子のオン電圧降下特性の設定なしでも、簡単に高精度な制御定数の測定演算が可能になるという効果がある。
【図面の簡単な説明】
【図１】本発明の第１の実施の形態に係る電圧形インバータの回路構成図である。
【図２】図１に示すインバータ制御装置の詳細回路図である。
【図３】図１に示す平均値・位相差演算器の詳細回路図である。
【図４】図１に示す交流電動機の等価回路図である。
【図５】図１に示す電圧形インバータの動作のフローチャートである。
【図６】図４に示す等価回路のインピーダンスのベクトル図である。
【図７】図３に示す平均値・位相差演算器の電圧指令値・電流検出値のタイムチャートである。
【図８】本発明の第２の実施の形態に係る電圧形インバータの動作のフローチャートである。
【図９】図８に示す電圧形インバータのオン電圧降下量を示す図である。
【図１０】図８に示すオン電圧降下量による補正処理の説明図である。
【図１１】従来の交流電動機の定数測定装置の構成図である。
【符号の説明】
１ 電圧形インバータ
２ 交流電動機
３ 電流検出器
４ 比較器
５ 発振器
６ 加算器
７ ゲート回路
８ インバータ制御装置
９ デッドバンド補償器
１０ 直流電源
１１ 速度指令回路
１２ 励磁電流指令回路
１３ スイッチ回路
１４ 速度検出器
１５ ３相／２相変換器
１６ ２相／３相変換器
１７ 一次角周波数演算回路
１８ 速度制御回路
１９ トルク電流制御回路
２０ 励磁電流制御回路
２１ 電圧指令補償回路
２２ 加算器
２３ 積算器
２４ 位相計算器
２５ チューニング処理部
２６ 電流振幅演算器
２７ 平均値・位相演算器
２８ ＬＰＦ
２９ ＨＰＦ
３０ 絶対値回路
３１ 乗算器
３２ 減算器
３３ 正弦波発生器[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a method for measuring a primary and secondary combined resistance (R1 + R2) and leakage inductance (l = l1 + l2) of an AC motor used as a control constant in vector control or the like, and more specifically, an inverter that variably controls the AC motor. The present invention relates to a method for measuring a combined resistance (R1 + R2) and a leakage inductance (l = l1 + l2) of an AC motor and a control device for the AC motor.
[0002]
[Prior art]
As a conventional technique, a method for measuring the combined resistance (R1 + R2) and leakage inductance (l = l1 + l2) of an AC motor while the AC motor is stopped is disclosed in, for example, Japanese Patent Laid-Open No. 06-98595. An AC motor constant measuring method and control apparatus "are known.
FIG. 11 is a block diagram of a conventional AC motor constant measuring device. During normal operation, the inverter input voltage Vdc is PWM controlled by the inverter 104 to generate an AC voltage, and the induction motor 105 is controlled at a variable speed. Yes.
In addition, the control circuit 106 using a one-chip microcomputer or the like performs a speed sensorless vector control process 107 so as to follow the speed command ωr during normal operation, and generates a PWM signal in the gate circuit 108. In this case, speed and torque control is performed based on the primary resistance measurement value r1, the secondary resistance value r2, the leakage inductance measurement value (l1 + l2), other motor constant setting values, and the output of the motor current detector 109. Yes.
[0003]
Accordingly, constant measurements such as the combined resistance value (r1 + r2) and the leakage inductance (l1 + l2) are performed before operation. First, a sine wave modulation signal for measurement is generated by the single-phase AC excitation processing 110, and the inverter 104 is operated via the gate circuit 108, and an AC current is supplied to the motor 105 by the AC excitation voltage for measurement. In the effective power current Iq reactive power current Id calculation processing circuit 111 while the motor 105 is stopped, if the rotational phase of the AC excitation voltage vector obtained by integrating the primary frequency command ω1 for measurement is θ, sin θ, Based on −cos θ and the U-phase motor current iu, Iq and Id are calculated.
Next, the resistance / inductance calculation processing circuit 112 calculates (r1 + r2) and (l1 + l2) from the Iq and Id calculation values and the magnitude Vc1 of the excitation voltage command.
[0004]
[Problems to be solved by the invention]
However, in the above conventional example, by correcting the inverter output voltage error due to the on-voltage drop of the power element, the combined resistance (r1 + r2) and leakage inductance (l1 + l2) are calculated from only the magnitude of the excitation voltage command and the instantaneous current detection value of the motor. ) Is calculated and measured with high accuracy, and the output voltage error due to the drop in the on-voltage of the power element varies depending on the magnitude of the output current. Therefore, the on-voltage drop with respect to the instantaneous value of the output current every control cycle. It was necessary to correct. For this reason, there are problems that the constant measurement software becomes complicated and that it is necessary to set the on-voltage drop of the power element with high accuracy in advance.
[0005]
Therefore, the present invention corrects the amplitude error and phase error due to the ON voltage drop of the power semiconductor element to the average value V_ave of the absolute value of the voltage command after measurement and the phase difference θ_dif between the voltage command and the output current, thereby instantaneously. A highly accurate calculation method of the combined resistance (R1 + R2) and the leakage inductance (l = l1 + l2) that eliminates the need to correct the on-voltage drop for each control cycle with respect to the value and to set the on-voltage drop characteristic of the power element with high accuracy; It aims at providing the control apparatus of the alternating current motor using the measured value.
[0006]
[Means for Solving the Problems]
  In order to achieve the above object, an invention according to claim 1 relates to a constant measuring method of an AC motor, and through a power converter composed of a power semiconductor element capable of controlling the magnitude, frequency and phase of an output voltage. The primary current supplied is separated into a component parallel to the magnetic flux of the AC motor (excitation current) and a component orthogonal to the component (torque current), and each is adjusted independently to flow through each phase of the AC motor Has a current detector to detect the output currentShiThe inverter device fixes the voltage phase to an arbitrary value set in advance so that an alternating magnetic flux is generated by the sine wave generator while the rotation of the AC motor is stopped, and a voltage command of a sine wave of frequency fh And the voltage command value is adjusted so that the magnitude of the output current becomes a predetermined value, and then the average value V_ave of the absolute value of the voltage command, the average value I_ave of the absolute value of the output current and the voltage after a predetermined time elapses The phase difference θ_dif between the command and the output current is measured, the primary and secondary combined resistance (R1 + R2) is calculated from the cos component of the ratio of V_ave and I_ave, and is calculated from the sin component of the ratio of V_ave and I_ave Measure or
  Or
[Expression 1]

From the above, in the constant measuring method of the AC motor for calculating and measuring the leakage inductance l = l1 + l2,Among the output voltage errors, at least the amplitude and phase difference error due to the ON voltage drop of the power semiconductor element are corrected to the average value V_ave of the absolute value of the measured voltage command and the phase difference θ_dif between the voltage command and the output current, and then the primary and second Calculation measurement of the next combined resistance (R1 + R2) and leakage inductance l = l1 + l2It is characterized by doing.
  The invention described in claim 2 is the constant measuring method for an AC motor according to claim 1,An on-voltage drop amount of the power semiconductor element as a function of current is incorporated, and the amplitude error and the phase difference error are corrected using at least the on-voltage drop amount of the power semiconductor element at the peak value of the measured alternating current. It is said.
  The invention according to claim 3 relates to an AC motor control device,The primary current supplied via a power converter composed of power semiconductor elements capable of controlling the magnitude, frequency and phase of the output voltage is divided into a component (excitation current) parallel to the magnetic flux of the AC motor. The inverter device has a current detector that detects the output current flowing in each phase of the AC motor by independently adjusting each component separately from the orthogonal component (torque current), and the inverter device stops the rotation of the AC motor. In this state, the voltage phase is fixed to an arbitrary value set in advance so that an alternating magnetic flux is generated by the sine wave generator, and a voltage command value of a sine wave having a frequency fh is provided. After adjusting the magnitude to be a predetermined value, the absolute value V_ave of the absolute value of the voltage command, the average value I_ave of the absolute value of the output current, and the phase difference θ_ between the voltage command and the output current after a predetermined time elapses. comprising means for measuring if, the primary and secondary combined resistance from cos component of the ratio (R1 + R2) of the V_ave and I_ave calculated measure, computed measuring the sin component of the ratio of the V_ave and I_aveOr,
Or
[Expression 2]

In the control apparatus for an AC motor that calculates and measures the leakage inductance l = l1 + l2, at least the amplitude and the phase difference error due to the on-voltage drop of the power semiconductor element among the output voltage errors are measured as the average value V_ave of the absolute values of the measured voltage commands and Voltage command and output current phase Means for correcting the difference θ_dif, and using these corrected V_ave and θ_ave, the primary and secondary combined resistance (R1 + R2) and the leakage inductance l = l1 + l2 are calculated and measured, and the AC motor is configured based on these measured values. ControlIt is characterized by that.
  According to a fourth aspect of the present invention, there is provided the control apparatus for an AC motor according to the third aspect,The means for correcting the absolute value V_ave of the measured voltage command and the phase difference θ_dif between the voltage command and the output current incorporates an on-voltage drop amount of the power semiconductor element as a function of the current, and at least the measured AC current The amplitude error and the phase difference error are corrected using the on-voltage drop amount of the power semiconductor element at the peak value.It is characterized by that.
[0007]
DETAILED DESCRIPTION OF THE INVENTION
Hereinafter, a first embodiment of the present invention will be described with reference to the drawings.
FIG. 1 is a circuit configuration diagram of a voltage source inverter according to a first embodiment of the present invention.
FIG. 2 is a detailed circuit diagram of the inverter control device shown in FIG.
FIG. 3 is a detailed circuit diagram of the average value / phase difference calculator shown in FIG.
FIG. 4 is an equivalent circuit diagram of the electric motor shown in FIG.
FIG. 5 is a flowchart of the operation of the voltage source inverter shown in FIG.
FIG. 6 is a vector diagram of the impedance of the equivalent circuit shown in FIG.
FIG. 7 is a time chart of voltage command values and current detection values of the average / phase difference calculator shown in FIG.
In FIG. 1, 1 is a voltage source inverter, 2 is an AC motor, 3 is a current detector, 4 is a comparator, 5 is an oscillator, 6 is an adder, 7 is a gate circuit, 8 is an inverter control device, and 9 is a dead band. Compensator 10 is a DC power supply, 11 is a speed command circuit, 14 is a speed detector, and 27 is an average value / phase calculator.
In the figure, a voltage source inverter 1 converts a DC voltage applied from a DC power source 10 into an AC having an arbitrary frequency and voltage by a PWM control method. The voltage source inverter 1 includes switching elements TUP, TVP,... TWN made of power semiconductor elements such as transistors and IGBTs, and feedback diodes DUP, DVP,... DWN connected in antiparallel to the switching elements.
[0008]
The AC motor 2 is connected to the AC output terminals of the phases U, V, and W of the voltage source inverter 1. The primary currents Iu, Iv, and Iw of the U-phase, V-phase, and W-phase of the AC motor 2 are detected by current detectors 3u, 3v, and 3w. Further, the rotational speed of the AC motor 2 is detected by the speed detector 14.
The inverter control device 8 includes the speed command value ωr * created by the speed command circuit 11 and the U-phase, V-phase, and W-phase primary currents Iu detected by the current detectors 3u, 3v, and 3w. , Iv, Iw and the speed detection value ωr from the speed detector 14 are added, and the voltage command pattern signal (Vu) of each phase of U, V, W with 120 ° phase difference is added.^*, Vv^*, Vw^*) And the phase γ is output to the dead band compensators 9u, 9v, and 9w that compensate the voltage error due to the dead band. The dead band compensators 9u, 9v, 9W output the voltage error compensation values to the adders 6u, 6v, 6w with the phase γ as an argument.
[0009]
The adders 6u, 6v, 6w are connected to the voltage pattern signal Vu.^*, Vv^*, Vw^*Then, the output values of the dead band compensators 9u, 9v, 9w are added, and the added values are sent to the comparators 4u, 4v, 4w. The output signal of the oscillator 5 that generates a carrier wave signal for PWM control is also sent to the comparators 4u, 4v, and 4w. The comparators 4u, 4v, and 4w compare the output signals of the adders 6u, 6v, and 6w with the carrier wave signal. Then, a PWM pulse for turning on / off the switching elements TUP, TVP, TWN constituting the voltage source inverter 1 is generated.
The gate circuit 7 gives a gate signal to the switching elements TUP, TVP,... TWN according to the PWM pulse output from the comparators 4u, 4v, 4w.
[0010]
In FIG. 2, 8 is an inverter control device, 12 is an excitation current command circuit, 15 is a 3-phase / 2-phase converter, 16 is a 2-phase / 3-phase converter, 17 is a primary angular frequency calculation circuit, and 18 is a speed control circuit. , 19 is a torque current control circuit, 20 is an excitation current control circuit, 21 is a voltage command compensation circuit, 22 is an adder, 23 is an integrator, 24 is a phase calculator, 25 is a tuning processing unit, and 26 is a current amplitude calculator. It is.
In the figure, the inverter control device 8 detects the primary currents Iu, Iv, Iw to the AC motor 2 and performs coordinate transformation to send the torque current feedback value Iqfb and the excitation current feedback value Idfb. A two-phase converter 15 is provided. Speed command value ωr from speed command circuit 11^*And the output value of the speed control circuit ASR18 provided so that the speed detection value ωr from the speed detector 14 coincides with the torque current command value Iqref, and this Iqref and the torque output from the three-phase / 2-phase converter 15 Torque current control circuit ACRq19 for controlling the current feedback value Iqfb to match, the excitation current command value Idref from the excitation current command circuit 12, and the excitation current feedback value Idfb from the three-phase / two-phase converter 15 Is provided with an exciting current control circuit ACRd20 for controlling the exciting current direction voltage.
[0011]
Further, an induced voltage generated by the magnetic flux of the AC motor 2, an induced voltage due to the induced electromotive force coefficient, a voltage of the counter electromotive force due to the primary resistance R1, a torque current direction component voltage, a leakage inductance (l = l1 + l2) of the AC motor 2, and a primary resistance R1. The voltage command compensation circuit 21 outputs the voltage of the excitation current direction component of the back electromotive force due to. The voltage of the torque current direction component of the output of the voltage command compensation circuit 21 is added to the output of the torque current control circuit 19 and the adder 22A to generate a torque current direction voltage command value Vqref, and the voltage of the excitation current direction component is The output of the excitation current control circuit 20 and the adder 22B are added to generate an excitation current direction voltage command value Vdref.
Further, from the torque current direction voltage command value Vqref and the excitation current direction voltage command value Vdref, voltage command pattern signals (Vu) for each phase of U, V, and W with a phase difference of 120 ° are obtained.^*, Vv^*, Vw^*) Is generated and output, a two-phase / three-phase converter 16 is provided.
[0012]
The 3-phase / 2-phase converter 15 and the 2-phase / 3-phase converter 16 are calculated by the following equations (1) and (2), respectively.
[Equation 5]

Further, the inverter control device 8 determines the slip frequency command value ωs from the secondary resistance R2 set as Idref and Iqref.^*And the primary angular frequency ω1 from the detected speed value ωr from the speed detector 14.^*The primary angular frequency ω1 from the primary angular frequency command computing circuit 17 is provided.^*Are integrated by the integrator 23 and output to the 3-phase / 2-phase converter 15 and the 2-phase / 3-phase converter 16 as the phase θ.
Further, the phase θ is an output of the phase calculation circuit 24 tan^-1(Iqfb / Idfb) is added by the adder 22c, and is output to the dead band compensators 9u, 9v, 9w as the phase γ.
[0013]
In the inverter control device of FIG. 2, the tuning process constituting the core part of the present invention is performed by a tuning processing unit 25 that controls the combined resistance and leakage inductance measurement operation before actual operation, and a switching signal Csw from the tuning processing unit 25. Switch circuits 13A, 13B, and 13C that can switch between actual operation and constant measurement operation are provided. When the constant is measured before actual operation, the circuit is switched to the b terminal side. In the switch circuit 13A, the torque current direction voltage command value Vqref is switched to 0. In the circuit 13B, the excitation current direction voltage command value Vdref is switched to the tuning voltage command value V_ref set for constant measurement by the tuning processing unit 25, and in the switch circuit 13C, processing for switching the phase θ to a set value (fixed value) is performed.
Accordingly, as the output of the tuning processing unit 25 at the time of constant measurement, the switch switching signal Csw, the tuning voltage command value V ref set as a constant measurement signal, and the same phase θh are output, and the phase θ is arbitrarily set by the switch 13C. Perform processing to set to a fixed value.
The current amplitude calculator 26 receives the torque current feedback value Iqfb and the excitation current feedback value Idfb, and calculates and outputs the output current amplitude value I_fb.
[0014]
In FIG. 3, 27 is an average value / phase calculator, 28 is an LPF, 29 is an HPF, 30 is an absolute value circuit, 31 is a multiplier, 32 is a subtractor, and 33 is a sine wave generator.
In the figure, an average value / phase difference calculator 27 receives signals V_ref, I_fb and phase θh at the time of constant measurement from the inverter control device 8, and calculates the phase difference θ_dif of both signals and the absolute value of each signal. Average values V_ave and I_ave are calculated, high-pass filters HPA 29A and 29B for removing DC components, a sine wave generator 33 for outputting sin and cos, and a multiplier circuit 31A for multiplying the output of HPF 29 by sin θh and cos θh. To D, low-pass filters LPF 28A to F for performing average value processing, absolute value calculation circuits (ABS) 30A and 30B, and phase calculation circuits 24B and 24C.
[0015]
The outline of the constant measurement process by the voltage source inverter 1 shown in FIG. 1, the inverter control device 8 shown in FIG. 2, and the average value / phase difference calculator 27 shown in FIG. In the inverter control device 8 shown in FIG. 4, before the actual operation, the tuning processor 25 switches the switches 13A, 13B, and 13C to the b terminal side for measurement by the switching signal Csw, Vqref is 0, and Vdref is for constant measurement. The voltage θ is set to a fixed value in the voltage command value V_ref, and the voltage command pattern signal Vu in the motor stop state at the time of constant measurement^*~ Vw^*, And the AC motor 2 is driven by the voltage source inverter shown in FIG. 1 to detect the primary currents Iu, Iv, Iw at the time of constant measurement. The constant measurement voltage command V_ref and the same phase θh and the current amplitude Ifb are output to the average value / phase difference calculator 27 of FIG.
The average value / phase difference calculator 27 calculates the phase difference θ_dif and the average values V_ave and I_ave from the V_ref, θh, and Ifb signals, and a constant calculator (not shown) for the combined resistance (R1 + R2), leakage inductance ( l1 + l2) is determined and stored.
When the constant measurement is completed, switches 13A, 13B, and 13C in FIG. 2 are switched and returned to the a terminal side for actual operation, and the inverter 1 shown in FIG. 1 is controlled to perform actual operation using the obtained motor constant. It is.
[0016]
Next, the constant measuring method of the present invention will be described with reference to FIG.
At the time of constant measurement (at the time of tuning), the tuning processing unit 25 of the inverter control device 8 determines the magnitude and frequency fh of the alternating current for constant measurement that flows during tuning (S100).
The magnitude of the alternating current in this case is, for example, about 50% to 100% of the smaller of the two rated current values based on the rated current values of the voltage source inverter 1 and the AC motor 2, and the frequency fh is The frequency at which 2π · fh · M >> R2 is established.
FIG. 4A is a T-1 type equivalent circuit of an AC motor, where R1, R2, l = l1 + l2, and M are primary resistance, secondary resistance, leakage inductance, and mutual inductance, respectively. When the operating frequency to be applied is high, ωM >> R2, so that almost no current flows through M and the equivalent circuit is as shown in FIG.
Further, the relationship between the phase difference θ_dif between the voltage and current at this time and (R1 + R2), (l1 + l2) × 2πfh, Z is as shown in FIG.
[0017]
Next, the tuning processing unit 25 sets Vqref to 0 in the switch circuit 13A, to the tuning voltage command value V_ref in 13B, and to the phase θ as a fixed value in 13C by the switching signal Csw during constant measurement (S101).
Here, V_ref = Vamp · sin (2πfh · t), and the initial value of Vamp is set to 0, and the measurement operation is started.
Thereafter, Vamp (the amplitude of V_ref) is increased so that the average value I_ave of the absolute values of the current detection values becomes the predetermined current value determined in S100. When the value of I_ave reaches a predetermined value, the passage of a predetermined time is waited until the outputs of the LPFs 28A to 28F in the average value / phase difference calculator 27 are stabilized (S102).
[0018]
The average current value I_ave is calculated by the absolute value circuit 30B and the average value processing circuit LPF28F from the I_fb input in the average value / phase difference calculator 27 shown in FIG. V_ave is obtained by the absolute value circuit 30A and the LPF 28E. Further, the phase differences θv ′ and θi ′ between the basic signal and each signal are obtained as outputs of the phase calculation circuits 24B and 24C, and the difference θv′−θi ′ is obtained as the subtractor 32 output θ_dif.
Next, the obtained average voltage value V_ave, average current value I_ave, and phase difference θ_dif are stored in the memory, and the operation is stopped (S103).
FIG. 7 shows how the voltage command V_ref and the current detection value I_fb change at this time. FIG. 7A shows how the output of the LPF stabilizes with the lapse of time t, and FIG. 7B shows a relationship diagram of V_ref, I_fb, and phase difference θ_dif at that time.
[0019]
When V_ave, I_ave, and θ_dif are obtained, the following equations (3), (4), and (5) are used.
[Formula 6]

(R1 + R2) and (l1 + l2) are obtained (S104).
In addition, from equations (3), (4) and (5), the impedance of the circuit is Z = (V_ave / √3) / I_ave, and the angle between Z and the real axis Re is θ_dif. For example, it is obtained from the illustrated relationship.
[0020]
According to the first embodiment described above, it has been described so far that the measurement is started with the initial value of Vamp set to 0. However, during actual operation, the flowing current value is predicted from the V / f characteristic value of the AC motor, It is also possible to shorten the time by setting some value in advance and adjusting it from there.
Further, in FIG. 3, the average values V_ave and I_ave of the respective signals are obtained by the LPFs 28E and 28F. However, the average values V_ave and the phase θV ′ may be obtained by a moving average. Since the time constant of the LPF 28E is self-explanatory, it can be obtained by calculation.
Furthermore, if the voltage command value is obtained by adding the offset value V_ref_ofs to the tuning voltage command value V_ave, the offset voltage is output as a direct current, and the primary resistance R1 can be obtained simultaneously from this direct current component. If R1 is obtained, it is easy to obtain R2 from (R1 + R2).
In the present embodiment, the combined resistance and the leakage inductance are measured with the AC signal V_ref having one kind of amplitude and frequency. However, the measurement is performed under a plurality of conditions, the measurement result is averaged, and the influence of the offset amount is affected. In order to avoid this, the present invention can be implemented even by calculating using the difference amount.
[0021]
Next, a second embodiment of the present invention will be described with reference to the drawings.
FIG. 8 is a flowchart of the operation of the voltage source inverter according to the second embodiment of the present invention.
FIG. 9 is a diagram showing an on-voltage drop amount of the voltage source inverter shown in FIG.
FIG. 10 is an explanatory diagram of the correction processing based on the on-voltage drop amount shown in FIG.
In the first embodiment, since the voltage drop of the power semiconductor element is not corrected, an error may occur in the measured V_ave and θ_dif. Therefore, in the second embodiment, the values of V_ave and θ_dif are The correction means (not shown) is provided in the average value / phase difference calculator 27, and the correction processing of S105 and S106 is added to the processing of the first embodiment as shown in FIG. Yes.
[0022]
In FIG. 8, the processing of S100 to S104 is the same as that of the first embodiment, and a duplicate description is omitted.
First, the V_ave measured by the processes of S100 to S103 is corrected (S105).
Specifically, the peak value Ipk of the measured alternating current is determined as the average value of current × π / 2, and the instantaneous current value when the current phase is θ_dif is determined as Ipk × sin θ_dif. Next, the average value Von_ave of the ON voltage drop amount of the power semiconductor element at each current value is calculated by the following equation (6), and Von_ave is subtracted from V_ave to obtain new V_ave.
[0023]

Von (I) represents the amount of on-voltage drop of the power semiconductor element when the current I is flowing, and is obtained as an average value of the amount of drop by the switching element and the amount of drop by the diode as shown in FIG. The average value / phase calculator 27 is built in as a function of the current I.
[0024]
  Next, the phase difference θ_dif is corrected (S106). Specifically, the phase correction amount θcmp is calculated by the following equation (7), and θcmp is added to θ_dif to obtain a new θ_dif.
  θcmp= K × 4/3 × Von (Ipk)
            / V_ave × θ_dif / 90 (7)
  Note that K in the equation (7) is determined by a ratio of the voltage command value V_ave and the on-voltage drop amount 4/3 × Von (Ipk) at the time of the peak current value Ipk for correcting the phase angle. Proportional coefficient.
[0025]
In the above formulas (6) and (7), (90−θ_dif) / 90 and θ_dif / 90 represent the remaining portion which is canceled out by averaging the influence of the on-voltage drop of the power semiconductor element. It shows that it is averaged by °.
FIG. 10 shows this relationship. The effects in the areas A1 and A2 in FIG. 10A and the areas B1 and B2 in FIG. 10B are canceled out, the area A1 matches θ_dif, and the area B1 matches 90−θ_dif. I understand that.
Further, as can be seen from FIG. 10A, since the influence of the voltage error near the current 0 is canceled out, it is not necessary to know the characteristics of the on-voltage drop amount at a minute current, and the influence of the on-voltage is reduced. It becomes possible to correct.
Finally, the control constant corrected by the on-voltage amount in S104 is calculated, and the measurement ends (S104).
[0026]
As described above, according to the present invention, R2 and (l = l1 + l2) are obtained from the measured combined resistance and leakage inductance measured value and the value of the primary resistance R1 set in advance, and the primary angular frequency calculation circuit 17 By setting the voltage command compensation circuit 21 and switching the switch circuits 13A to 13C back to the a terminal side during actual operation, vector control can be performed based on control constants such as measured resistance and leakage inductance.
In addition, although the implementation target has been described with an induction motor with a speed detector, the present invention can also be applied using an induction motor without a speed detector or a synchronous machine. That is, it can be implemented without creating a frequency command from the speed estimator or speed command value instead of the speed detector, or without compensating the slip frequency command value ωs *, and the resistance and leakage inductance measured by the speed estimator You can also set the control constant. Needless to say, the combined resistance and leakage inductance measuring method disclosed by the present invention can be used without any problems even if the control method during actual operation is different.
[0027]
【The invention's effect】
As described above, according to the present invention, when measuring control constants such as the combined resistance (R1 + R2) and leakage inductance (l = l1 + l2), the output voltage detector of the inverter is unnecessary and the motor is stopped. It is possible to measure and calculate the combined resistance and leakage inductance in this state. In particular, the influence of the ON voltage drop of the power semiconductor element is corrected to the average value V_ave of the voltage command and the phase difference θ_dif between the voltage command and the output current. Thus, there is an effect that it is possible to easily perform highly accurate control constant measurement calculation without instantaneous correction of the on-voltage drop for each control cycle and without setting the on-voltage drop characteristic of the power element in consideration of accuracy.
[Brief description of the drawings]
FIG. 1 is a circuit configuration diagram of a voltage source inverter according to a first embodiment of the present invention.
FIG. 2 is a detailed circuit diagram of the inverter control device shown in FIG. 1;
FIG. 3 is a detailed circuit diagram of the average value / phase difference calculator shown in FIG. 1;
4 is an equivalent circuit diagram of the AC motor shown in FIG. 1. FIG.
FIG. 5 is a flowchart of the operation of the voltage source inverter shown in FIG. 1;
6 is a vector diagram of impedance of the equivalent circuit shown in FIG. 4;
7 is a time chart of a voltage command value and a current detection value of the average value / phase difference calculator shown in FIG. 3; FIG.
FIG. 8 is a flowchart of the operation of the voltage source inverter according to the second embodiment of the present invention.
FIG. 9 is a diagram showing an on-voltage drop amount of the voltage source inverter shown in FIG. 8;
10 is an explanatory diagram of a correction process using an on-voltage drop amount shown in FIG. 8. FIG.
FIG. 11 is a configuration diagram of a constant measuring apparatus for a conventional AC motor.
[Explanation of symbols]
1 Voltage type inverter
2 AC motor
3 Current detector
4 comparator
5 Oscillator
6 Adder
7 Gate circuit
8 Inverter controller
9 Deadband compensator
10 DC power supply
11 Speed command circuit
12 Excitation current command circuit
13 Switch circuit
14 Speed detector
15 3-phase / 2-phase converter
16 2-phase / 3-phase converter
17 Primary angular frequency arithmetic circuit
18 Speed control circuit
19 Torque current control circuit
20 Excitation current control circuit
21 Voltage command compensation circuit
22 Adder
23 Accumulator
24 Phase calculator
25 Tuning processing section
26 Current amplitude calculator
27 Average / phase calculator
28 LPF
29 HPF
30 Absolute value circuit
31 multiplier
32 Subtractor
33 Sine wave generator

Claims (4)

The primary current supplied via a power converter composed of power semiconductor elements capable of controlling the magnitude, frequency and phase of the output voltage is divided into a component (excitation current) parallel to the magnetic flux of the AC motor. is separated into the orthogonal components (torque current) was adjusted independently, have a current detector for detecting an output current flowing through each phase of AC motor and the inverter device stops the rotation of the AC motor In this state, the voltage phase is fixed to an arbitrary value set in advance so that an alternating magnetic flux is generated by the sine wave generator, and a sine wave voltage command with a frequency fh is given. After adjusting to a predetermined value, the average value V_ave of the absolute value of the voltage command, the average value I_ave of the absolute value of the output current, and the phase difference θ_dif between the voltage command and the output current are measured after a predetermined time has elapsed. Or to, said primary from cos component of the ratio of the V_ave and I_ave and secondary combined resistance of (R1 + R2) calculated measured, calculates measured from sin component of the ratio of the V_ave and I_ave,
Or

From the above, in the constant measuring method of the AC motor for calculating and measuring the leakage inductance l = l1 + l2 ,
Among the output voltage errors, at least the amplitude and phase difference error due to the ON voltage drop of the power semiconductor element are corrected to the average value V_ave of the absolute value of the measured voltage command and the phase difference θ_dif between the voltage command and the output current, and then the primary and second A constant measuring method for an AC motor, characterized in that the following combined resistance (R1 + R2) and leakage inductance l = l1 + l2 are calculated and measured . In the alternating current motor constant measurement method according to claim 1,
Built-in power semiconductor element on-voltage drop as a function of current,
A constant measuring method for an AC motor, wherein the amplitude error and the phase difference error are corrected using at least an on-voltage drop amount of a power semiconductor element at a peak value of a measured AC current . The primary current supplied via a power converter composed of power semiconductor elements capable of controlling the magnitude, frequency and phase of the output voltage is divided into a component (excitation current) parallel to the magnetic flux of the AC motor. The inverter device has a current detector that detects the output current flowing in each phase of the AC motor by independently adjusting each component separately from the orthogonal component (torque current), and the inverter device stops the rotation of the AC motor. In this state, the voltage phase is fixed to an arbitrary value set in advance so that an alternating magnetic flux is generated by the sine wave generator, and a voltage command value of a sine wave having a frequency fh is provided. After adjusting the magnitude to be a predetermined value, the absolute value V_ave of the absolute value of the voltage command, the average value I_ave of the absolute value of the output current, and the phase difference θ_ between the voltage command and the output current after a predetermined time elapses. a means for measuring if, wherein the primary and secondary combined resistance (R1 + R2) is calculated from the cos component of the ratio of V_ave and I_ave, and is calculated from the sin component of the ratio of V_ave and I_ave. ,
Or

In the control device for an AC motor that calculates and measures the leakage inductance l = l1 + l2,
Means for correcting at least the amplitude and phase difference error due to the on-voltage drop of the power semiconductor element among the output voltage errors to the average value V_ave of the absolute value of the measured voltage command and the phase difference θ_dif between the voltage command and the output current; Using the corrected V_ave and θ_ave, primary and secondary combined resistance (R1 + R2) and leakage inductance l = l1 + l2 are calculated and measured, and the AC motor is controlled based on these measured values . Control device. In the control apparatus for an AC motor according to claim 3,
The means for correcting the average value V_ave of the absolute value of the measured voltage command and the phase difference θ_dif between the voltage command and the output current incorporates an on-voltage drop amount of the power semiconductor element as a function of current,
A control apparatus for an AC motor, wherein the amplitude error and the phase difference error are corrected using at least an on-voltage drop amount of a power semiconductor element at a peak value of a measured AC current .

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