JP3658769B2 - Receiver - Google Patents

Description

[0001]
[Industrial application fields]
The present invention relates to a receiving apparatus mainly used for wireless communication.
[0002]
[Prior art]
In general, a single super heterodyne system or a double super heterodyne system is used as a reception system in wireless communication. However, the conventional heterodyne method requires a band filter for removing image frequencies and a band filter for removing adjacent channel signals. A mechanical filter using mechanical vibration characteristics of quartz or ceramic is used as the band filter. Therefore, there are various problems such as large shape and high price. Therefore, in recent years, a direct conversion reception method has been studied as a new reception method. FIG. 9 shows a block diagram of a conventional direct conversion reception system. 1 is an antenna, 2 is high-frequency amplification means, 3 is first mixing means, 4 is a first low-pass filter for removing adjacent channel signals, 5 is first low-frequency amplification means, and 6 is signal generation means. , 7 is a 90 ° phase shifter, 8 is a second mixing means, 9 is a second low-pass filter for removing adjacent channel signals, and 10 is a second low-frequency amplification means. Reference numeral 11 denotes a phase difference detection means, which includes a first waveform shaping means 12, a second waveform shaping means 13, and a D-flip flop 14. Terminal a is an output terminal of the first low-frequency amplification means 5, terminal b is an output terminal of the first waveform shaping means 12, terminal c is an output terminal of the second low-frequency amplification means 10, and terminal d is a second output terminal. The output terminal of the waveform shaping means 13 and the terminal e are the output terminals of the D-flip flop 14. Now to antenna 1
S = cos {ω + P (t) · Δω} · t P (t): code sequence of 1 or −1
ω: carrier wave angular frequency Δω: angular frequency deviation, polarity is positive
Consider the case where the FSK signal S represented by The FSK signal S is amplified by the high frequency amplification means 2 and then input to the first and second mixing means 3 and 8. In the signal generating means 6
Q = COS {ω + x} · t x: Angular frequency error from carrier angular frequency ω
A signal Q represented by In the 90 ° phase shifter, the signal Q from the signal generating means 6 is phase-shifted by 90 ° so that Q ′ = SIN {ω + x} · t. In the first mixing means 3, the signal Q from the signal generating means 6 and the FSK signal S are multiplied. In the second mixing means 8, the signal Q ′ from the 90 ° phase shifter 7 and the FSK signal are multiplied. High-frequency components other than the desired signal are removed by the first and second low-pass filters 4 and 9, and the desired signal is amplified by the first and second low-frequency amplifiers 5 and 10. Therefore, the following signals are output to the terminals a and c.
[0003]
Terminal a: S × Q = COS {P (t) · Δω−x} · t
Terminal c: S × Q ′ = SIN {P (t) · Δω−x} · t
As the signal generating means 6, a crystal having high oscillation frequency stability is used, and Δω >> x is selected. In order to simplify the description, the following description will be made assuming that x = 0. The relationship between the code string P (t) and the signal waveforms of the terminals a, b, c, d, e is shown in FIG. 10 and will be described with reference to FIG. As apparent from FIG. 10, when the code string P (t) is −1, the signal of the terminal c is advanced by 90 ° compared to the signal of the terminal a. On the other hand, when the code string P (t) is 1, the signal at the terminal c is delayed by 90 ° compared to the signal at the terminal a. Therefore, by detecting the phase difference between the signal at the terminal a and the signal at the terminal c in the phase difference detection means 11, the original code string P (t) can be reproduced. As a phase advance / delay detection method of the phase difference detection means 11, an output waveform of the terminal e can be obtained from the waveforms shown in the terminals b and c of FIG. In FIG. 10, the level (circle) of the terminal d is latched at the falling edge of the terminal b and output to the terminal e.
[0004]
[Problems to be solved by the invention]
However, in the above-described conventional configuration, when the error x from the carrier frequency of the oscillation frequency of the signal generating means 6 is larger than the angular frequency deviation Δω, or the code change rate of the code sequence P (t) is compared with the angular frequency deviation Δω. In the case of a size that cannot be ignored, it had the following problems.
[0005]
(1) When the error x is not negligible compared to the angular frequency deviation Δω
Even if the code string P (t) changes, the phase advances between the signals at the terminals a and c, and the delay does not change. For this reason, the code string P (t) cannot be reproduced.
[0006]
(2) When the sign change rate is not negligible compared to the angular frequency deviation Δω
Within one bit transmission time, the signals at terminals a and c are less than one cycle. Therefore, it becomes difficult to determine the phase advance and delay, and the code string P (t) cannot be accurately reproduced.
[0007]
Further, in the conventional configuration, only the FSK signal can be demodulated. That is, there has been a problem that FM signals and AM signals modulated with analog signals such as audio signals cannot be demodulated.
[0008]
The present invention solves the above-described problems, and not only eliminates the influence of the error x by using a simple switching means, but also enables accurate data demodulation, as well as an FM modulated with an analog signal such as an audio signal. The object is to realize a receiving apparatus capable of demodulating signals and AM signals.
[0011]
[Means for Solving the Problems]
In order to achieve the above object, the receiving apparatus of the present invention provides:First signal generating means for outputting a signal having a frequency close to the carrier signal frequency to be received, and first mixing means for extracting a signal having a frequency difference between the signal from the first signal generating means and the received signal; A second mixing means for extracting a signal having a frequency that is a difference between the signal obtained by phase-shifting the signal from the first signal generating means and the received signal; and a second mixing means for generating a temporally continuous rectangular wave signal. A signal generating means, a first switch means for switching a signal from the first mixing means by a rectangular wave signal from the second signal generating means, and a rectangular wave signal from the second signal generating means. A second switch means for switching a signal from the second mixing means by a phase-shifted rectangular wave signal; an output signal of the first switch means; and a second switch means. An arithmetic means for adding or subtracting a force signal; and the first switch means and the second switch means provided in the front stage or the rear stage of the first switch means and the second switch means. A filter for attenuating a harmonic component of a signal from one mixing means and a signal from the second mixing means, a frequency-voltage converting means for generating a voltage corresponding to the frequency of the output signal of the computing means, and the frequency -A frequency correction means for controlling the output frequency of the first signal generating means in a direction equal to the carrier frequency of the received signal in accordance with the output voltage of the voltage converting means;
[0012]
A first signal generating means for outputting a signal having a frequency close to the carrier signal frequency to be received; and a first mixing for extracting a signal having a frequency difference between the signal from the first signal generating means and the received signal. And second mixing means for extracting a signal having a frequency that is a difference between the signal obtained by phase-shifting the signal from the first signal generating means and the received signal, and generating a rectangular wave signal continuous in time. Two signal generation means, a first switch means for switching a signal from the first mixing means by a rectangular wave signal from the second signal generation means, and a rectangular wave from the second signal generation means Second switch means for switching a signal from the second mixing means by a rectangular wave signal obtained by phase-shifting the signal; an output signal of the first switch means; and the second switch Included in the output signals of the first and second switch means provided in the preceding stage or the subsequent stage of the first switch means and the second switch means, and arithmetic means for adding or subtracting the stage output signal A filter for attenuating the harmonic component of the signal from the first mixing means and the signal from the second mixing means, a frequency-voltage converting means for generating a voltage corresponding to the frequency of the output signal of the computing means, And a noise removal means for removing pulse noise generated at the output of the frequency-voltage conversion means, and the output voltage of the noise removal means is used as a demodulated output.
[0020]
The noise removing means includes a pulse output means for generating a pulse when the interval between zero cross points of the signal input to the frequency-voltage converting means is a certain value or more, and the pulse signal from the pulse output means is converted to the frequency-voltage. It is also composed of pulse insertion means for inserting the signal input to the conversion means at intervals of zero cross points.
[0021]
[Action]
With the above configuration, the present invention sets the signal frequency from the second signal generating means to be larger than the angular frequency error x between the carrier frequency to be received and the oscillation frequency of the first signal generating means. It is not affected by x. Further, the code string P (t) can be accurately reproduced by attenuating the harmonic component generated by the signal from the first mixing means and the signal from the second mixing means passing through the nonlinear circuit. It becomes.
[0022]
【Example】
An embodiment of the present invention will be described below with reference to FIG. The same functional blocks as those in the conventional example of FIG. 1 is an antenna, 2 is high frequency amplification means, 3 is first mixing means, 4 is a first low-pass filter for removing adjacent channel signals, 101 is a first limiter amplifier, and 102 is a first harmonic. A removal filter, 6 is a first signal generating means, 7 is a 90 ° phase shifter, 8 is a second mixing means, 9 is a second low-pass filter for removing adjacent channel signals, and 103 is a second limiter. Amplifier 104, second harmonic elimination filter, 15 first switch means, 16 second switch means, 17 second signal generation means, 18 90 ° phase shift means, 19 add / subtract Arithmetic means, 20 a band filter, 21 a frequency-voltage converting means, 22 a noise removing means, and 23 a frequency correcting means. Now, as the signal S input to the antenna 1,
S = cos {ω + Δω} · t
ω: Angular frequency of carrier wave Δω: Angular frequency shift with both positive and negative polarities
think of. Here, the angular frequency deviation Δω varies with time depending on data or voice. That is, the signal S is a signal subjected to frequency modulation. The first signal generating means 6 is the same as the conventional example.
Q = COS {ω + x} · t x: Angular frequency error from carrier angular frequency ω
A signal Q represented by In the 90 ° phase shifter 7, the signal Q from the signal generating means 6 is phase-shifted by 90 ° so that Q ′ = SIN {ω + x} t. Therefore, as in the conventional example, the output terminals a and c of the first low-pass filter 4 and the second low-pass filter 9 are connected to the output terminals a and c.
Terminal a: S × Q = COS {Δω−x} · t
Terminal c: S × Q ′ = SIN {Δω−x} · t
A signal is generated. The signal is amplified by the first limiter amplifier 101 and the second limiter amplifier 103, and when the signal is large, the signal is clipped to the upper and lower targets. That is, it is converted into a rectangular wave signal, and an angular frequency component that is an integral multiple of the angular frequency {Δω−x} is generated. On the other hand, the second signal generating means 17
R = COS {r ・ t}-(1/3) ・ COS {3 ・ r ・ t} + (1/5) ・ COS {5 ・ r ・ t)
A rectangular wave signal R represented by Here, r> (Δω−x) is set.
[0023]
For example, assuming that the maximum angular frequency shift = 2.4 kHz / (2π) and the maximum frequency error = −4 kHz, the maximum value of (Δω−x) is 6.4 kHz, so r / (2π) = 16 kHz.
[0024]
First, in order to simplify the description, let us consider a case in which the signal is not clipped in the limiter amplifiers 101 and 103 but is amplified linearly. The signal at the terminal a is switched in the first switch means 15 by the rectangular wave signal R generated by the second signal generating means. On the other hand, the output of the 90 ° phase shift means 18
R ’= SIN {r · t} + (1/3) · SIN {3 · r · t} + (1/5) · SIN {5 · r · t)
The rectangular wave signal R ′ is output. Therefore, the signal at the terminal c is switched by the second switch means 16 by the rectangular wave signal R ′. Therefore, the output terminal a 'of the first switch means 15 and the output terminal c' of the second switch means 16 are
Terminal a ′: {COS {(Δω−x) · t}} · {COS {r · t} − (1/3) · COS {3 · r · t} +.
Terminal c ′: {SIN {(Δω−x) · t}} · {SIN {r · t} + (1/3) · SIN {3 · r · t} +
Occurs.
[0025]
The signals at the terminals a ′ and c ′ are added by the computing means 19. Accordingly, the output terminal f of the computing means 19 is connected to the output terminal f.
Terminal f: COS {n · {{r + x} −Δω} · t} − (1/3) · COS {{{3 · r−x} + Δω} · t} + (1)
Is output. The band-pass filter 20 is for removing a term related to the harmonic component of the angular frequency r generated by the first switch means 17 and the second switch means 18, that is, the second term or more of the equation (1). is there. Accordingly, the output terminal f ′ of the bandpass filter 20 is connected to the output terminal f ′.
Terminal f ′: COS {{{r + x} −Δω} · t} Equation (2)
Is output. Here, since r> (Δω−x) is set, the phase of the equation (2) is always positive at the positive time. That is, no negative frequency occurs. Therefore, as apparent from the equation (2), the output signal generated at the terminal f ′ can be regarded as a frequency modulation signal in which a carrier signal having an angular frequency of {r + x} undergoes a frequency shift of Δω. Therefore, the frequency modulation signal generated at the terminal f 'can be demodulated by the frequency-voltage conversion means 21 that generates an output voltage proportional to the frequency. Further, the demodulated signal generated at the terminal f 'is subjected to noise removal means 22 to remove pulsed noise peculiar to the FM demodulation generated in the FM demodulation and output to the terminal g. Since the center frequency of the band-pass filter 20 is as low as about 16 kHz, it can be constituted by a monolithic IC. Depending on the switch configuration of the first switch means 15 and the second switch means 16, an angular frequency r component may occur at the outputs of the switch means 15 and 16. At this time, the rectangular wave signal from the second signal generating means 17 is added to the output of the first switch means 15 or the second switch means 16 to cancel the component of the angular frequency r. Depending on the switch configuration of the first switch means 15 and the second switch means 16, a signal at the terminal a or terminal c may be generated at the first switch means 15 or the second switch means 16. The signal at the terminal a or terminal c can be removed by the band filter 20, but when a low-pass filter is used instead of the band filter 20, the signal at the terminal a or terminal c is changed to the first switch means 15 or In addition to the output of the second switch means 16, the signal at the terminal a or terminal c may be canceled.
[0026]
Next, consider a case where the signal is clipped and converted into a rectangular wave signal in the limiter amplifiers 101 and 103. The phase of the rectangular wave signal changes discontinuously at the rising and falling change points. Therefore, when the harmonic elimination filters 102 and 104 are not provided, the frequency of the output signal of the computing means 19 changes in an impulse manner at the rising and falling points of the rectangular wave signals at the terminals a and c. That is, the output signal of the computing means 19 has a wide frequency component. Therefore, when the band is limited in the band filter 20 in the next stage, the frequency is demodulated in the frequency-voltage conversion means 21 by causing a phenomenon that the amplitude greatly decreases at the timing of the rising and falling points of the rectangular wave signals at the terminals a and c. This will cause distortion in the output. In data transmission, the bit error rate is deteriorated. Therefore, in the present invention, the first harmonic elimination filter 102 and the second harmonic elimination filter 104 are used to remove the harmonic components of the rectangular wave signals at the terminals a and c, thereby rising and rising of the rectangular wave signal. The fall is smooth. As a result, since the frequency of the output signal of the computing means 19 does not change in an impulse manner, the amplitude of the output signal does not drop greatly even if the band limitation is performed by the band filter 20. Therefore, no distortion occurs in the frequency demodulated output in the frequency-voltage converting means 21.
[0027]
As described above, when the configuration of the present invention is used, even if the frequency stability of the signal generated by the first signal generating means 6 is poor and the error angular frequency x with respect to the frequency of the antenna input signal is large, the second signal generating means 17. The original data or audio signal can be demodulated without being affected by the error angular frequency x by selecting the angular frequency r of the rectangular wave signal generated in step r> (maximum value of Δω ± x). If the configuration of the present invention is used, the signal can be demodulated without using the frequency correction means 23, but in order to further improve the stability in reception, the DC voltage of the signal at the terminal g is detected by the frequency correction means 23, It is still effective if the signal frequency of the first signal generating means 6 is controlled so that the DC voltage at the terminal g becomes a certain reference value K. The reference value K is set to be equal to the voltage value output from the terminal g when the error angular frequency x = 0. The frequency correction means 23 may be configured to use a low-pass filter composed of a resistor and a capacitor, and control the frequency generated by the first signal generation means 6 by the output of the low-pass filter. Although not shown in FIG. When it is detected that a signal is received from the demodulated output of g, the voltage at the terminal g is A / D converted, and then the DC voltage for controlling the generated frequency of the first signal generating means 6 is converted by D / A conversion by microcomputer processing. You may make it the structure which generate | occur | produces. Further, although not shown in FIG. 1, when it is detected that a signal is received from the demodulated output of the terminal g, the first low-pass filter 4 and the second low-pass filter 9 are switched so as to lower the cutoff frequency. In this way, the S / N characteristic can be improved.
[0028]
1 has been described as an addition operation, a subtraction operation may be performed. In this case, the signal at the terminal f ′ is COS {{{r−x} + Δω} · t}.
[0029]
The first harmonic removing filter 102 and the second harmonic filter 104 are inserted in front of the first switch means 15 and the second switch means 16, but the first switch means 15 and the second harmonic filter 104 are inserted. You may insert in the back | latter stage of the switch means 16, and the front | former stage of a calculating means.
[0030]
FIG. 2 shows a configuration of switch means applicable to the first switch means 15 and the second switch means 16 in FIG. In FIG. 2, 24 is an input terminal for receiving a signal at terminal a or a signal at terminal c, 25 is an input terminal for receiving a rectangular wave signal R or R ′ from the second signal generating means 17, 26 is an output terminal, Reference numeral 27 denotes an inverting circuit having an amplification factor of 1, and 28 denotes an electronic switch. In the electronic switch 28, the connection between the output terminal and the input terminal is switched depending on whether the phase of the rectangular wave signal R or the rectangular wave signal R 'input to the input terminal 25 is positive or negative. Such an electronic switch 28 can be easily realized by CMOS as an analog switch, and can also be easily configured by using a bipolar transistor.
[0031]
FIG. 3 is a diagram and a waveform diagram showing the configuration of the frequency-voltage converting means 21 and the noise removing means 22 in FIG. In FIG. 1, the noise removal unit 22 is arranged at the subsequent stage of the frequency-voltage conversion unit 21, but in the example of FIG. 3, the noise removal unit 22 is incorporated in the frequency-voltage conversion unit 21. In FIG. 3A, 40 is an input terminal for inputting a frequency modulation signal shown at terminal f ', 29 is an amplifying means, 30 is a binarizing means composed of a comparator, 31 is a first delay means, and a resistor 32 and a capacitor 33 and a comparator 34. 35 is exclusive OR means, 36 is second delay means, 37 is pulse insertion means, 38 is pulse generation means, 39 is a low pass filter, and 41 is a demodulation output terminal. When the frequency of the signal input to the input terminal 40 increases, the output pulse interval of the exclusive OR unit 35 becomes narrower, and when the input frequency decreases, the output pulse interval of the exclusive OR unit 35 becomes wider. Accordingly, if unnecessary high-frequency components are removed by the low-pass filter 39, a voltage change corresponding to the output pulse interval of the exclusive OR means 35 can be taken out. In order to increase the demodulation sensitivity, the delay amount in the first delay means 31 may be increased. Next, the noise removal method will be described with reference to the waveform diagram of FIG. 3B. h and i are inputs to the exclusive OR means 35. Therefore, the output of the exclusive OR 35 is a pulse train as shown in j. When noise is included in the signal input to the input terminal 40, the pulse train of the signals h and i may become irregular due to the influence of noise, and one pulse may be lost. Thus, if the pulses of the signals h and i are lost, the pulse of the output j of the exclusive OR means 35 is also lost. Accordingly, when the signal j is passed through the low-pass filter, a large pulse noise is generated at the portion where the pulse is missing. The pulse generation means 38 generates a pulse output l that repeats HIGH / LOW every T1 if there is no pulse in the signal j for a time T2 (for example, T2 = 2 · T1) sufficiently longer than the normal pulse interval T1. The pulse output l stops when a pulse occurs at j. The signal j is delayed by the second delay means 36 to become a signal k. In the pulse insertion means 37, a pulse equal to the pulse width of the signal k is inserted into the signal k in accordance with the rising and falling edges of the pulse output l. It is a pulse in which a pulse marked with a circle in the signal m in the waveform diagram of FIG. 3B is inserted. The signal m is passed through a low-pass filter to obtain a demodulated signal. By inserting a pulse in this way, there is no missing pulse, and no pulse-like noise is generated from the demodulated output. Although a pulse is inserted into the signal j, a signal i is generated by the first delay means 31 after a pulse is inserted into the signal h, and a signal j having no missing pulse is generated by the exclusive OR means 35. Also good.
[0032]
FIG. 4 is a diagram showing another configuration of the frequency-voltage converting means 21 in FIG. 4, the same functional blocks as those in FIG. 3 are given the same numbers. 42 is an edge detection means, and 43 is a monostable multivibrator. The edge detection means 42 includes the delay means 31 and the exclusive OR means 35 in FIG. The output of the monostable multivibrator 43 has a narrower or wider pulse interval depending on the frequency of the signal input to the terminal 40. Accordingly, when unnecessary high frequency components are removed by the low-pass filter 39 as in the case of FIG. 3, a voltage change corresponding to the output pulse interval of the monostable multivibrator 43 can be taken out. The advantage of the configuration of FIG. 4 is that there is no need to increase the delay amount in the delay means 31. For this reason, it is not necessary to increase the time constant of the circuit composed of the capacitor 33 and the resistor 32. The demodulation sensitivity can be optimally set by appropriately selecting the pulse width of the output of the monostable multivibrator 43. The method for removing noise is the same as in FIG.
[0033]
FIG. 5 shows another configuration and waveform diagram of the noise removing means 22. In FIG. 5A, 44 is an input terminal to which the demodulated output of the frequency-voltage converting means 21 is input, 45 is a high-pass filter that removes a desired signal such as a data signal and extracts only noise components, 46 is a comparator that extracts noise of a certain level or more Reference numeral 47 denotes a pulse width extending means, and 45, 46 and 47 constitute a pulse generating means 48. Reference numeral 49 is a delay means, 50 is a holding means, and 51 is an output terminal. The operation of the noise removing unit 22 will be described with reference to the waveform diagram of FIG. 5B. The signal v input to the input terminal 44 becomes a signal w by the high pass filter 45. Only pulsed noise is taken out by the comparator 46 and becomes a signal x. The pulse width of the signal x is expanded by the pulse width extension means 47 to become a signal y. On the other hand, the signal v is delayed by the delay means 49, and the holding means 50 samples and holds the value of the delayed signal v during the pulse output period of the signal y. Therefore, the output of the output terminal 51 is the signal z. Although this signal z is not shown, it can be changed smoothly by passing through a low-pass filter. Further, subtraction means may be used in place of the holding means 50, and noise may be removed by subtracting the signal z and the signal y. Note that the signal 1 in FIG. 3 may be used as the output signal y of the pulse generating means.
[0034]
FIG. 6 shows the configuration of another embodiment of the frequency-voltage conversion means. In FIG. 6, 60 is an input terminal to which the output signal of the computing means 19 in FIG. 1 is input, 61 is a first filter composed of a bandpass filter having a center frequency near 11 kHz, and 62 is a center frequency near 21 kHz. A second filter composed of a band-pass filter, 63 a first signal output means composed of a rectifier circuit composed of a diode, and 64 a second signal output means composed of a rectifier circuit composed of a diode, 65 Is a subtracting means, 39 is a low-pass filter, and 66 is an output terminal. FIG. 7 is a characteristic diagram showing the characteristics of the output level with respect to the frequency for explaining the operation of FIG. In FIG. 7A, characteristic 1 is the output characteristic of the first filter 61, and characteristic 2 is the output characteristic of the second filter 62. FIG. 7B shows the output characteristics of the subtracting means 65. 7B, the frequency change of the signal input to the input terminal 60 can be output from the output terminal 66 as a voltage change. A low-pass filter 39 is a filter for removing the vicinity of 16 kHz and its harmonic components. If the frequency-voltage conversion means of FIG. 6 is used, the band filter 20 in FIG. 1 can be omitted. That is, the first filter 61 and the second filter 62 also serve as the band-pass filter 20 that removes unnecessary harmonic components of 16 kHz. Further, since the pulse-like noise generated by the configuration of the frequency-voltage converting means shown in FIGS. 3 and 4 is difficult to generate, there is an advantage that data demodulation with less error occurrence can be performed without using the noise removing means 22. is there. The first filter 61 and the second filter 62 can be configured by active filters using resistors, capacitors, and transistors.
[0035]
FIG. 8 shows the configuration of the second signal generating means 17 and the 90 ° phase shifting means 18. 52 is an input terminal for receiving a clock signal generated from a reference clock of the microcomputer, 53, 54 and 55 are D-flip flops, each of which constitutes a 1/2 frequency divider. Reference numerals 56 and 57 denote output terminals, which output a rectangular wave signal R from the terminal 56 and a rectangular wave signal R ′ orthogonal to the rectangular wave signal R from the terminal 57. If the circuit of FIG. 8 is used, an IC can be easily formed. Note that a signal oscillated by a bistable multivibrator or the like may be used as a signal input to the input terminal 52.
[0036]
In the embodiment of FIG. 1, the demodulation of the frequency modulation signal has been described. However, not only the frequency modulation signal but also the amplitude modulation signal and the phase modulation signal are demodulated in place of the frequency-voltage conversion means 21 shown in FIG. This is possible by using corresponding demodulation means.
[0037]
Although the signal from the high frequency amplifying means 2 is used as the input signal of the constituent device of the present invention, an intermediate frequency signal obtained by frequency conversion of the output signal of the high frequency amplifying means 2 may be used as the input signal of the constituent device of the present invention.
[0038]
【The invention's effect】
As described above, according to the receiving apparatus of the present invention, accurate demodulation can be performed even if the frequency stability of the first signal generating means is poor by using an electronic switch having a simple configuration. Therefore, it is not necessary to use an expensive crystal oscillator with good frequency stability for the first signal generating means, and an expensive mechanical filter is not necessary, and it is easy to make an IC, so that a receiving device can be realized at low cost. In addition, there is an effect that an AGC circuit is not required because a large level signal is input to the antenna and can be demodulated even when it is clipped by the receiving circuit.
[0039]
If the frequency correction means is used, the frequency modulation signal can be accurately demodulated even if the frequency stability is worse.
[0040]
Further, if noise removing means is used, pulse-like noise can be removed in the demodulation of the frequency modulation signal, so that the S / N characteristics can be improved.
[Brief description of the drawings]
FIG. 1 is a block diagram of a receiving apparatus according to an embodiment of the present invention.
FIG. 2 is a block diagram of switch means in one embodiment of the present invention.
FIG. 3A is a block diagram of frequency-voltage conversion means and noise removal means in an embodiment of the present invention.
(B) Waveform diagram
FIG. 4 is another block diagram of frequency-voltage converting means and noise removing means in one embodiment of the present invention.
FIG. 5A is a block diagram of noise removing means in one embodiment of the present invention, and FIG.
FIG. 6 is another block diagram of frequency-voltage conversion means in one embodiment of the present invention.
7 is a characteristic diagram of frequency-voltage conversion means in FIG.
FIG. 8 is a block diagram of second signal generating means and 90 ° phase shifting means in an embodiment of the present invention.
FIG. 9 is a block diagram of a conventional receiving apparatus.
FIG. 10 is an output diagram of each output terminal in a conventional receiver.
[Explanation of symbols]
1 Antenna
2 High frequency amplification means
3 First mixing means
4 First low-pass filtering means
6 First signal generating means
7 90 ° shifter
8 Second mixing means
9 Second low-pass filtering means
15 First switch means
16 Second switch means
17 Second signal generating means
18 90 ° phase shift means
19 Calculation means
21 Frequency-voltage conversion means
22 Frequency correction means
22 Noise removal means
102 Harmonic rejection filter
104 Harmonic rejection filter

Claims (2)

  First signal generating means for outputting a signal having a frequency close to the carrier signal frequency to be received, and first mixing means for extracting a signal having a frequency difference between the signal from the first signal generating means and the received signal; A second mixing means for extracting a signal having a frequency that is a difference between the signal obtained by phase-shifting the signal from the first signal generating means and the received signal; and a second mixing means for generating a temporally continuous rectangular wave signal. A signal generating means, a first switch means for switching a signal from the first mixing means by a rectangular wave signal from the second signal generating means, and a rectangular wave signal from the second signal generating means. A second switch means for switching a signal from the second mixing means by a phase-shifted rectangular wave signal; an output signal of the first switch means; and a second switch means. An arithmetic means for adding or subtracting a force signal; and the first switch means and the second switch means provided in the front stage or the rear stage of the first switch means and the second switch means. A filter for attenuating a harmonic component of a signal from one mixing means and a signal from the second mixing means, a frequency-voltage converting means for generating a voltage corresponding to the frequency of the output signal of the computing means, and the frequency A receiving device comprising frequency correcting means for controlling the output frequency of the first signal generating means in a direction equal to the carrier frequency of the received signal in accordance with the output voltage of the voltage converting means. First signal generating means for outputting a signal having a frequency close to the carrier signal frequency to be received, and first mixing means for extracting a signal having a frequency difference between the signal from the first signal generating means and the received signal; A second mixing means for extracting a signal having a frequency that is a difference between the signal obtained by phase-shifting the signal from the first signal generating means and the received signal; and a second mixing means for generating a temporally continuous rectangular wave signal. A signal generating means, a first switch means for switching a signal from the first mixing means by a rectangular wave signal from the second signal generating means, and a rectangular wave signal from the second signal generating means. A second switch means for switching a signal from the second mixing means by a phase-shifted rectangular wave signal; an output signal of the first switch means; and a second switch means. An arithmetic means for adding or subtracting a force signal; and the first switch means and the second switch means provided in the front stage or the rear stage of the first switch means and the second switch means. A filter for attenuating a harmonic component of a signal from one mixing means and a signal from the second mixing means, a frequency-voltage converting means for generating a voltage corresponding to the frequency of the output signal of the computing means, and the frequency A noise removing means for removing pulsed noise generated at the output of the voltage converting means , wherein the noise removing means has a zero cross point interval of a signal input to the frequency-voltage converting means when the interval is greater than a certain value. Pulse output means for generating a pulse, and a pulse for inserting a pulse signal from the pulse output means into an interval between zero cross points of a signal input to the frequency-voltage conversion means Constituted by the input means, the receiving apparatus to demodulate an output voltage of said noise removing means.

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