JP3569995B2 - TM dual mode dielectric resonator and high frequency band pass filter device - Google Patents

Description

【００４５】
ここで、上記誘電体１，２は、誘電体セラミクス（Ｚｒ，Ｓｎ）ＴｉＯ_４からなり、平板電極３と端面電極６，７は、上記誘電体１，２の上端面と下端面にそれぞれ銀ペーストを焼き付けて形成した。
【００４６】
【表２】

【００４７】
以上詳述した第１の実施例のＴＭ２重モード誘電体共振器によれば、当該ＴＭ２重モード誘電体共振器の無負荷Ｑを３０００から３５００の間の値に設定したとき、当該ＴＭ２重モード誘電体共振器の厚さを２３．５ｍｍにでき、無負荷Ｑを７０００から８０００の間の値に設定したときに厚さが５０．０ｍｍ以上になる従来例のＴＭ２重モード共振器の比較して薄くすることができる。
【００４８】
以上の第１の実施例のＴＭ２重モード誘電体共振器によれば、共振周波数ｆ_０をほとんど変化させずに、誘電体１，２の厚さｈを所定の値に設定することによって所望の無負荷Ｑに設定することができる。
【００４９】
以上の第１の実施例のＴＭ２重モード誘電体共振器によれば、上記誘電体１，２を比誘電率ε_ｒの比較的大きな材料を用いて構成することにより、比誘電率ε_ｒの比較的小さい材料を用いて構成したときに比べてほとんど無負荷Ｑを劣化させないで直径方向の小型化ができる。
【００５０】
＜第２の実施例＞
図１０は第２の実施例のＴＭ２重モード誘電体共振器の横断面図であり、図２５は、図１０におけるＢ−Ｂ’線についての縦断面図である。上記第２の実施例のＴＭ２重モード誘電体共振器は、第１の実施例のＴＭ２重モード誘電体共振器において、上記誘電体１，２の外周に切り欠き１１，１２，１３，１４を設けたことを特徴とする。
【００５１】
第２の実施例のＴＭ２重モード誘電体共振器において、図１０に示すように、上記誘電体２の軸を中心としてｘ軸と４５度だけ隔てた上記誘電体２の外周の互いに対向する位置に、それぞれ上記誘電体２の上端面から下端面まで半円形の溝状に誘電体を除去して形成した切り欠き１３と切り欠き１４を設ける。また、図２５に示すように、上記誘電体１の外周の互いに対向する位置に、それぞれ上記誘電体１の上端面から下端面まで半円形の溝状に誘電体を除去して形成した切り欠き１１と切り欠き１２を、それぞれ上記切り欠き１３，１４と同軸になるように設ける。これによって、上記第２の実施例のＴＭ２重モード誘電体共振器を、ｘ軸，ｙ軸に対してそれぞれ非対称になるように構成している。
【００５２】
以上のように構成した第２の実施例のＴＭ２重モード誘電体共振器において、第１の実施例の説明において詳述したｘ軸共振器とｙ軸共振器とは、上記誘電体１に設けられた切り欠き１１，１２と上記誘電体２に設けられた切り欠き１３，１４によって互いに結合して、互いに共振周波数が異なる２つの独立したモードである偶モードと奇モードを発生する。すなわち、上記切り欠き１１，１２，１３，１４は、２重に縮退したｘモードとｙモードの縮退を分離して互いに共振周波数が異なる２つの独立したモードである偶モードと奇モードを発生する縮退分離手段を構成する。
【００５３】
また、図１１（ａ）に、上記奇モードの上記平板電極３の上面における電流分布を示す。上記奇モードの上記平板電極３の上面における電流は、上記切り欠き１２に対向する上記平板電極３の外周上の縁端部３４から上記切り欠き１１に対向する上記平板電極３の外周上の縁端部３３に向かう方向に分布する。図１１（ｂ）に、上記偶モードの上記平板電極３の上面における電流分布を示す。上記偶モードの上記平板電極３の上面における電流は、上記縁端部３３，３４から互いに上記平板電極３の中心を中心として互いに９０度だけ隔てた上記外周上の対向する２つの縁端部のうちの１つである縁端部３５から他方の縁端部３６に向かう方向に分布する。
【００５４】
図１２は、第２の実施例のＴＭ２重モード誘電体共振器の等価回路を示す回路図である。図１２の等価回路は、分布定数線路ＬＮ１乃至ＬＮ８をリング状に直列接続して構成される回転対称のリング分布定数線路を備える。ここで、上記分布定数線路ＬＮ１乃至ＬＮ８は、それぞれ共振周波数における１／４波長の長さに設定される。従って、上記リング分布定数線路は２πの電気長を有する。上記分布定数線路ＬＮ１と分布定数線路ＬＮ２との接続点が内部結合キャパシタＣ３を介して接地され、上記分布定数線路ＬＮ５と分布定数線路ＬＮ６との接続点が内部結合キャパシタＣ４を介して接地されて構成される。ここで、上記内部結合キャパシタＣ３，Ｃ４は、上記ｘ軸共振器と上記ｙ軸共振器を結合させる為のキャパシタであって、上記縮退分離手段に対応する。
【００５５】
図１２の等価回路において、ｘ軸上に位置する入出力端子Ｔ１，Ｔ３は、それぞれｘ軸共振器の励振点と一致し、それぞれｘ軸共振器の入出力端子であり、ｙ軸上に位置する入出力端子Ｔ２，Ｔ４は、それぞれｙ軸共振器の励振点と一致し、それぞれｙ軸共振器の入出力端子である。すなわち、ｘ軸共振器は、分布定数線路ＬＮ１乃至ＬＮ４を直列に接続してなりかつ電気長がπの分布定数線路と、分布定数線路ＬＮ４乃至ＬＮ８を直列に接続してなりかつ電気長がπの分布定数線路とを並列接続した半波長共振器として構成される。また、ｙ軸共振器は、分布定数線路ＬＮ３，ＬＮ４，ＬＮ５，ＬＮ６を直列に接続してなりかつ電気長がπの分布定数線路と分布定数線路ＬＮ７，ＬＮ８，ＬＮ１，ＬＮ２を直列に接続してなりかつ電気長がπの分布定数線路とを並列接続した半波長共振器として構成される。
【００５６】
ここで、図１２の等価回路の内部結合キャパシタＣ３，Ｃ４の静電容量は、互いに等しい負の静電容量を有し、次に示す数１で与えられる静電容量ΔＣを用いて、−ΔＣで与えられる。ここで、数１中のω_０は、上記２重モード誘電体共振器の共振周波数ｆ_０に対応する角周波数であって、数２で与えられる。また、各分布定数線路ＬＮ１乃至ＬＮ８の電気長θは、以下に示す数３で与えられる。さらに、数１，数３におけるｋは、所定の通過帯域特性と所帯の阻止帯域特性を有する高周波帯域通過フィルタを設計するときに与えられる結合係数であって、次の数４で与えられる。数２，数４中のｆ_ｏｄｄとｆ_ｅｖｅｎはそれぞれ、奇モードと偶モードの共振周波数である。
【００５７】
【数１】
ΔＣ＝（２Ｙ_ａ／ω_０）・ｔａｎ｛πｋ／（２−ｋ）｝
【数２】
ω_０＝２πｆ_０＝π（ｆ_ｅｖｅｎ＋ｆ_ｏｄｄ）
【数３】
θ＝（π／４）・（１ ＋ ｋ／２）
【数４】
ｋ＝２（ｆ_ｅｖｅｎ−ｆ_ｏｄｄ）／（ｆ_ｅｖｅｎ＋ｆ_ｏｄｄ）＝１．４１８（２ΔＣ／Ｃ_０）
【００５８】
ここで、数４におけるＣ_０は、上記ＴＭ２重モード誘電体共振器において、端面電極６と平板電極３とによって挟設された誘電体１の静電容量である。また、誘電体２は、上記誘電体１と同じ形状に形成されているので、上記端面電極７と上記平板電極３とによって挟設された誘電体２の静電容量もＣ_０である。当該ＴＭ２重モード誘電体共振器では誘電体１，２を有することを考慮すると次の数５で与えられる。数５におけるε_０は、真空中の誘電率であり、ａは誘電体１，２の半径であり、ａ＝ｄ／２である。
【００５９】
【数５】
Ｃ_０＝（２ ε_０ε_ｒ・πａ^２）／ｈ
【００６０】
図１３のグラフに数４で与えられるΔＣとｋの関係を実線で示した。また、図１３のグラフには、理想的な開放モデルを用いて２次元有限要素法によって計算した結果と測定結果とを合わせて示している。図１３から明らかなように、数４で与えられるΔＣとｋの関係と上記２次元有限要素法によって計算した結果と測定結果とはよく一致している。ここで、上記測定に用いたＴＭ２重モード誘電体共振器は、上述したように誘電体１，２の外周の低誘電率領域は誘電体の一部を切削することによって低誘電率底領域を形成した。
【００６１】
数３で与えられる電気長θは結合後の偶モードと奇モードの共振周波数の平均値を結合前の共振周波数ｆ_０に一致させるための補正係数（π／４）×（ ｋ／２）を含んでいる。また、線路の特性アドミタンスＹ_ａは、上記ＴＭ２重モード誘電体共振器のサセプタンスを周波数で微分したサセプタンススロープｂを用いて次の数６で与えられる。
【００６２】
【数６】
Ｙ_ａ＝ｂ／π
【００６３】
ここで、ＴＭ２重モード誘電体共振器のサセプタンススロープｂは、誘電体１，２内部への電気エネルギーの集中度が９６％程度と高いことから理想的な開放条件を満たす解析解を用いた摂動計算により、近似的に次の数７で与えられる。また、上記摂動計算は、文献「Ｓｈｕｈ−Ｈａｎ Ｃｈａｏ，“Ｍｅａｓｕｒｅｍｅｎｔｓ ｏｆ ｍｉｃｒｏｗａｖｅ ｃｏｎｄｕｃｔｉｖｉｔｙ ａｎｄ ｄｉｅｌｅｃｔｒｉｃ ｃｏｎｓｔａｎｔ ｂｙ ｔｈｅ ｃａｖｉｔｙ ｐｅｒｔｕｒｂａｔｉｏｎ ｍｅｔｈｏｄ ａｎｄ ｔｈｅｉｒ ｅｒｒｏｒｓ”， ＩＥＥＥ Ｔｒａｎｓａｃｔｉｏｎ ｏｎ Ｍｉｃｒｏｗａｖｅ Ｔｈｅｏｒｙ Ｔｅｃｈｎｏｌｏｇｙ，ｖｏｌ．ＭＴＴ−３３，Ｎｏ．６， ｐｐ． ５１９−５２６ １９８５年」を参照して行った。ここで、Ｊ_０（・）は、０次の第１種ベッセル関数であり、Ｊ_１（・）は、１次の第１種ベッセル関数である。また、ｋ_ｃは、動径方向の境界条件から決まる固有値である。
【００６４】
【数７】

【００６５】
次に、上記ＴＭ２重モード誘電体共振器の無負荷Ｑ_０は、上記ＴＭ２重モード誘電体共振器の無負荷Ｑを上記分布定数線路ＬＮ１乃至ＬＮ８の伝送Ｑと対応させることにより、上記分布定数線路ＬＮ１乃至ＬＮ８の減衰定数αと位相定数βを用いて次の数８で与られる。
【００６６】
【数８】
Ｑ_０＝β／（２α）
【００６７】
以上のように構成した第２の実施例のＴＭ２重モード誘電体共振器は、誘電体１，２の側面に上記切り欠き１１，１２，１３，１４を形成しているので、上記ｘ軸共振器と上記ｙ軸共振器を結合させることができ、２つの独立したモードである偶モードと奇モードを発生することができる。
【００６８】
＜第３の実施例＞
図１５は本発明に係る第３の実施例の高周波帯域通過フィルタ装置の縦断面図である。また、図１６は、上記第３の実施例の高周波帯域通過フィルタ装置の図１５におけるＣ−Ｃ’線での横断面図である。
【００６９】
上記第３の実施例の高周波帯域通過フィルタ装置は、第２の実施例のＴＭ２重モード誘電体共振器と同様に構成された３つのＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３と、上記ＴＭ２重モード誘電体共振器Ｒ１と上記ＴＭ２重モード誘電体共振器Ｒ２とを誘導結合させる結合ループインダクタ２１と、上記ＴＭ２重モード誘電体共振器Ｒ２と上記ＴＭ２重モード誘電体共振器Ｒ３とを容量結合させる結合キャパシタ２３，２４とを備えたことを特徴とする。
【００７０】
以下、図面を用いて第３の実施例の高周波帯域通過フィルタ装置の構成を詳細に説明する。上記第３の実施例の高周波帯域通過フィルタ装置において、図１５，１６に示すように、上記ＴＭ２重モード誘電体共振器Ｒ１は、円柱形状の誘電体１ａと、円柱形状の誘電体２ａと、平板電極３ａと、トーラス電極４ａと、導体ケース８０に形成されたキャビティー８０ａとを備えて、上記第２の実施例のＴＭ２重モード誘電体共振器と同様に構成される。ここで、上記誘電体１ａは、上端面に形成された端面電極６ａと、外周の互いに対向する位置に形成された切り欠き１１ａ，１２ａを備え、上記誘電体２ａは、下端面に形成された端面電極７ａと、外周の互いに対向する位置に形成された切り欠き１３ａ，１４ａを備える。図１６において、上記切り欠き１３ａ，１４ａの符号は、それぞれ切り欠き１１ａ，１２ａの符号の上又は下の括弧内に示している。
【００７１】
上記ＴＭ２重モード誘電体共振器Ｒ２は、同様に、誘電体１ｂと、誘電体２ｂと、平板電極３ｂと、トーラス電極４ｂと、導体ケース８０に形成されたキャビティー８０ｂとを備えて構成され、上記ＴＭ２重モード誘電体共振器Ｒ３は、同様に、誘電体１ｃと、誘電体２ｃと、平板電極３ｃと、トーラス電極４ｃと、導体ケース８０に形成されたキャビティー８０ｃとを備えて構成される。また、上記誘電体１ｂは、上端面に形成された端面電極６ｂと、外周の互いに対向する位置に形成された切り欠き１１ｂ，１２ｂを備え、上記誘電体２ｂは、下端面に形成された端面電極７ｂと、外周の互いに対向する位置に形成された切り欠き１３ｂ，１４ｂを備える。図１６において、上記切り欠き１３ｂ，１４ｂの符号は、それぞれ切り欠き１１ｂ，１２ｂの符号の上又は下の括弧内に示している。上記誘電体１ｃは、上端面に形成された端面電極６ｃと、外周の互いに対向する位置に形成された切り欠き１１ｃ，１２ｃを備え、上記誘電体２ｃは、下端面に形成された端面電極７ｃと、外周の互いに対向する位置に形成された切り欠き１３ｃ，１４ｃを備える。図１６において、上記切り欠き１３ｃ，１４ｃの符号は、それぞれ切り欠き１１ｃ，１２ｃの符号の上又は下の括弧内に示している。ここで、上記キャビティー８０ａ，８０ｂ，８０ｃは、上記導体ケース８０に所定の間隔だけ隔てて並置して形成される。従って、上記ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３は、上記導体ケース８０の両端面の間に、所定の間隔だけ隔てて並置して設けられる。
【００７２】
上記導体ケース８０の長手方向の一方の端面の中央部には、上記高周波帯域通過フィルタ装置を外部回路と接続するための入力コネクタ４１が設けられる。上記入力コネクタ４１の中心導体は、上記端面と電気的に絶縁されて上記端面を貫通して入力キャパシタ２２の一方の電極に接続される。また、上記入力キャパシタ２２の他方の電極は、上記切り欠き１１ａから４５度隔てた上記トーラス電極４ａのｘ軸上の外周（外側の面）に接続される。以上のように接続されて、上記入力コネクタ４１は、入力キャパシタ２２を介して上記ＴＭ２重モード誘電体共振器Ｒ１のｘ軸共振器と容量結合する。ここで、上記入力キャパシタ２２は、円柱形状のセラミック誘電体の両端面に電極が形成されて構成される。
【００７３】
また、上記結合ループインダクタ２１は、５つの導体線２１１，２１２，２１３，２１４，２１５からなり、上記導体線２１１，２１２，２１３，２１４が略四角形のループを形成するように直列に接続され、上記導体線２１５の両端がそれぞれ導体線２１２と導体線２１４の略中央点に接続されて構成される。そして、上記結合ループインダクタ２１は、上記平板電極３ａ，３ｂと略同一平面上に位置するようにかつ上記導体線２１１と上記導体線２１３と上記導体線２１５とが上記キャビティー８０ａと上記キャビティー８０ｂとを隔てる上記導体ケース８０の導体壁の中央部を長手方向に貫通するように設けられる。ここで、上記導体線２１１，２１３は、上記導体壁を上記導体ケース８０の長手方向に貫通するように設けられた貫通孔（図示せず。）によって上記導体壁と電気的に絶縁されて貫通する。また、上記導体線２１５は、上記壁と電気的に導通するように上記導体壁を貫通し、これによって、上記導体線２１５はその中央部で接地される。以上のように構成することにより、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸共振器の磁界の一部は、上記導体線２１２を廻って上記結合ループインダクタ２１のループ内部を横切り、上記ＴＭ２重モード誘電体共振器Ｒ２のｙ軸共振器の磁界の一部は、上記導体線２１４を廻って上記結合ループインダクタ２１のループ内部を横切る。これによって、上記結合ループインダクタ２１を介して、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸共振器と上記ＴＭ２重モード誘電体共振器Ｒ２のｙ軸共振器とは誘導結合する。
【００７４】
さらに、上記結合ループインダクタ２１の反対側に位置するｘ軸上のトーラス電極４ｂの外周表面に結合キャパシタ２３が接続される。ここで、上記結合キャパシタ２３は、円柱形状の誘電体の両端面に電極が形成されて構成され、一端面に形成された電極と上記トーラス電極４ｂが接続される。また、上記結合キャパシタ２３と対向するｘ軸上のトーラス電極４ｃの外周表面に結合キャパシタ２４が接続される。ここで、上記結合キャパシタ２４は、上記結合キャパシタ２３と同様に、円柱形状の誘電体の両端面に電極が形成されて構成され、一端面に形成された電極と上記トーラス電極４ｃが接続される。また、上記結合キャパシタ２３の他方の端面に形成された電極と上記結合キャパシタ２４の他方の端面に形成された電極は互いに導体線２３４によって接続される。そして、上記導体線２３４は、上記キャビティー８０ｂと上記キャビティー８０ｃを隔てる導体壁を当該導体壁に設けられた貫通孔（図示せず。）を介して電気的に絶縁されて当該導体壁を貫通する。以上のように構成された上記結合キャパシタ２３，２４と上記導体線２３４とによって、上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器と上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸共振器とは容量結合する。
【００７５】
上記導体ケース８０の幅方向の一方の側面には、中心導体と接地導体とを有する出力コネクタ４２が、上記中心導体が上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸と一致するように設けられる。そして、上記出力コネクタ４２の中心導体は、電気的に絶縁されて上記導体ケース８０の側面を貫通して出力キャパシタ２５の一方の電極に接続される。また、上記出力キャパシタ２５の他方の電極は、上記出力コネクタ４２と対向するｙ軸上のトーラス電極４ｃの外周上に接続される。以上のように接続されて、上記出力コネクタ４２は、出力キャパシタ２５を介して上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸共振器と容量結合する。
【００７６】
上記ＴＭ２重モード誘電体共振器Ｒ１のｘ軸と４５度だけ隔てかつ上記切り欠き１３ａに近接して、上記ＴＭ２重モード誘電体共振器Ｒ１のｘ軸共振器とｙ軸共振器との結合量を徴調整するための結合調整ねじ２６ａが導体ケース８０の下底面に設けられる。上記結合調整ねじ２６ａは、上記切り欠き１３ａの形成方向と平行になるように設けられ、上記結合調整ねじ２６ａのキャビティー８１ａ内の突出長が変化すると上記ＴＭ２重モード誘電体共振器Ｒ１のｘ軸共振器とｙ軸共振器との結合量が変化するので、上記結合調整ねじ２６ａの突出長を変化させることによって上記結合量を調整することができる。また、同様に上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸と４５度だけ隔てかつ上記切り欠き１４ｂに近接して、上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器とｙ軸共振器との結合量を徴調整するための結合調整ねじ２６ｂが上記切り欠き１４ｂの形成方向と平行になるように下底面に設けられ、上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸と４５度だけ隔てかつ上記切り欠き１４ｃに近接して、上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸共振器とｙ軸共振器との結合量を徴調整するための結合調整ねじ２６ｃが上記切り欠き１４ｃの形成方向と平行になるように下底面に設けられる。
【００７７】
また、上記導体ケース８０の幅方向の他方の側面には、それぞれ上記ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３の各ｘ軸共振器の共振周波数を徴調整するための周波数調整ねじ５１，５２，５３が、それぞれのねじの軸が上記ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３の各ｙ軸に一致するように設けられる。上記周波数調整ねじ５１，５２，５３のキャビティー８１ａ，８１ｂ，８１ｃ内の突出長が変化すると上記ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３の各ｘ軸共振器の共振周波数が変化するので、上記周波数調整ねじ５１，５２，５３の突出長を変えることによって各ｘ軸共振器の共振周波数を調整することができる。さらに、上記導体ケース８０の他方の端面には、上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸共振器の共振周波数を徴調整するための周波数調整ねじ５４が、周波数調整ねじ５４の軸が上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸に一致するように設けられる。上記周波数調整ねじ５４のキャビティー８１ｃ内の突出長が変化すると上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸共振器の共振周波数が変化するので、上記周波数調整ねじ５４の突出長を変えることによって当該共振周波数を調整することができる。
【００７８】
以上のように、第３の実施例の高周波帯域通過フィルタ装置は、３つのＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３を備えている。そして、当該高周波帯域通過フィルタ装置において、上記入力コネクタ４１は、入力キャパシタ２２によって、上記ＴＭ２重モード誘電体共振器Ｒ１のｘ軸共振器と容量結合する。そして、上記ＴＭ２重モード誘電体共振器Ｒ１のｘ軸共振器は、上記誘電体１ａに設けられた切り欠き１１ａ，１２ａと上記誘電体２ａに設けられた切り欠き１３ａ，１４ａとによって、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸共振器と電磁的に結合する。また、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸共振器は、上記結合ループインダクタ２１によって、上記ＴＭ２重モード誘電体共振器Ｒ２のｙ軸共振器と誘導結合する。上記ＴＭ２重モード誘電体共振器Ｒ２のｙ軸共振器は、上記誘電体１ｂに設けられた切り欠き１１ｂ，１２ｂと上記誘電体２ｂに設けられた切り欠き１３ｂ，１４ｂとによって、上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器と電磁的に結合する。さらに、上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器は、結合キャパシタ２３，２４によって、上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸共振器と容量結合する。上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸共振器は、上記誘電体１ｃに設けられた切り欠き１１ｃ，１２ｃと上記誘電体２ｃに設けられた切り欠き１３ｃ，１４ｃとによって、上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸共振器と電磁的に結合する。またさらに、上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸共振器は、上記出力キャパシタ２５によって、上記出力コネクタ４２と容量結合する。以上のように構成することにより、入力コネクタ４１と出力コネクタ４２との間に、ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３を備えた６段の高周波帯域通過フィルタ装置が構成される。
【００７９】
次に以上のように構成された高周波帯域通過フィルタ装置の動作を等価回路を用いて説明する。
【００８０】
図１７は高周波帯域通過フィルタ装置の等価回路を示す回路図である。当該等価回路において、各ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３は、図１２の第２の実施例のＴＭ２重モード誘電体共振器の等価回路と同様に表わすことができる。すなわち各ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３の等価回路は、それぞれｘ軸共振器とｙ軸共振器が結合した２重モードの等価回路であって、それぞれ各共振周波数における１／４波長の長さを有する分布定数線路ＬＮ１ａ乃至ＬＮ８ａ，分布定数線路ＬＮ１ｂ乃至ＬＮ８ｂ，分布定数線路ＬＮ１ｃ乃至ＬＮ８ｃが直列に接続された２πの電気長を有する回転対称の各リング分布定数線路を備える。そして、上記ＴＭ２重モード誘電体共振器Ｒ１の等価回路においては、上記分布定数線路ＬＮ１ａと分布定数線路ＬＮ２ａとの接続点が内部結合キャパシタＣ３ａを介して接地され、上記分布定数線路ＬＮ５ａと分布定数線路ＬＮ６ａとの接続点が内部結合キャパシタＣ４ａを介して接地される。
【００８１】
また、上記ＴＭ２重モード誘電体共振器Ｒ２の等価回路においては、上記分布定数線路ＬＮ１ｂと分布定数線路ＬＮ２ｂとの接続点が内部結合キャパシタＣ３ｂを介して接地され、上記分布定数線路ＬＮ５ｂと分布定数線路ＬＮ６ｂとの接続点が内部結合キャパシタＣ４ｂを介して接地される。さらに、上記ＴＭ２重モード誘電体共振器Ｒ３の等価回路においては、上記分布定数線路ＬＮ１ｃと分布定数線路ＬＮ２ｃとの接続点が内部結合キャパシタＣ３ｃを介して接地され、上記分布定数線路ＬＮ５ｃと分布定数線路ＬＮ６ｃとの接続点が内部結合キャパシタＣ４ｃを介して接地される。以上のようにして各ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３の等価回路は、構成される。
【００８２】
また、上記ＴＭ２重モード誘電体共振器Ｒ１のｘ軸上に位置する分布定数線路ＬＮ１ａと分布定数線路ＬＮ８ａの接続点は、一端が入力端子Ｔ１１である入力キャパシタ２２の他端に接続されるとともに、キャパシタＣ２ａを介して接地される。ここで、キャパシタＣ２ａは、入力キャパシタ２２が上記ＴＭ２重モード誘電体共振器Ｒ１のｘ軸共振器と結合することにより変化する当該ｘ軸共振器の共振周波数を補正するための静電容量であって、負の静電容量値を有する。分布定数線路ＬＮ２ａと分布定数線路ＬＮ３ａの接続点と、分布定数線路ＬＮ６ａと分布定数線路ＬＮ７ａの接続点は、それぞれキャパシタＣ５ａ，Ｃ６ａを介して接地される。ここで、上記キャパシタＣ５ａ，Ｃ６ａは、後述するように結合ループインダクタ２１が上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸共振器と結合することにより変化する当該ｙ軸共振器の共振周波数を補正するための静電容量であって、それぞれ負の静電容量値を有する。
【００８３】
さらに、上記ＴＭ２重モード誘電体共振器Ｒ１の分布定数線路ＬＮ４ａと分布定数線路ＬＮ５ａとの間にはインダクタＬ１が接続され、上記インダクタＬ１はインダクタＬ１１と誘導結合する。上記インダクタＬ１は、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸共振器によって発生する磁界のうちの上記結合ループインダクタ２１に誘導結合する磁界を発生するための等価的なインダクタを表わす。また、上記インダクタＬ１１とインダクタＬ１２とインダクタＬ１３は、並列に接続されて、結合ループインダクタ２１を構成し、上記結合ループインダクタ２１の一端は接地される。そして、上記インダクタＬ１３は、上記ＴＭ２重モード誘電体共振器Ｒ２の分布定数線路ＬＮ１ｂと分布定数線路ＬＮ８ｂとの間に接続されたインダクタＬ２と誘導結合する。ここで、上記インダクタＬ２は、上記ＴＭ２重モード誘電体共振器Ｒ２のｙ軸共振器によって発生する磁界のうちの上記結合ループインダクタ２１に誘導結合する磁界を発生するための等価的なインダクタを表わす。これによって、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸共振器と上記ＴＭ２重モード誘電体共振器Ｒ２のｙ軸共振器は誘導結合する。
【００８４】
上記ＴＭ２重モード誘電体共振器Ｒ２の分布定数線路ＬＮ２ｂと分布定数線路ＬＮ３ｂの接続点と、分布定数線路ＬＮ６ｂと分布定数線路ＬＮ７ｂの接続点は、それぞれキャパシタ５ｂ，６ｂを介して接地される。ここで、上記キャパシタ５ｂ，６ｂは、結合ループインダクタ２１が上記ＴＭ２重モード誘電体共振器Ｒ２のｙ軸共振器と結合することにより変化する当該ｙ軸共振器の共振周波数を補正するための静電容量であって、それぞれ負の静電容量値を有する。
【００８５】
上記ＴＭ２重モード誘電体共振器Ｒ２の分布結合線路ＬＮ４ｂと分布定数結合線路ＬＮ５ｂの接続点は、キャパシタＣ７ｂを介して接地されるとともに、結合キャパシタＣ８の一端に接続される。ここで、上記結合キャパシタＣ８は、図１６の結合キャパシタ２３，２４とが直列に接続されて構成されるキャパシタに対応する。また、上記結合キャパシタＣ８の他端は、キャパシタ７ｃを介して接地されるとともに、上記ＴＭ２重モード誘電体共振器Ｒ３の分布定数線路ＬＮ１ｃと分布定数線路ＬＮ８ｃとの接続点に接続される。これによって、上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器と上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸共振器とは容量結合する。ここで、上記キャパシタＣ７ｂは、上記結合キャパシタＣ８が上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器と結合することによって変化する当該ｘ軸共振器の共振周波数を補正するための静電容量であって、負の静電容量値を有する。また、同様に上記キャパシタＣ７ｃは、上記結合キャパシタＣ８が上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸共振器と結合することによって変化する当該ｘ軸共振器の共振周波数を補正するための静電容量であって、負の静電容量値を有する。
【００８６】
上記ＴＭ２重モード誘電体共振器Ｒ３の上記分布結合線路ＬＮ６ｃと上記分布定数結合線路ＬＮ７ｃの接続点は、キャパシタＣ２ｃを介して接地されるとともに、一端が出力端子Ｔ１２である出力キャパシタ２５の他端に接続される。これによって、上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸共振器を外部回路と容量結合させる。
【００８７】
以上のようにして第３の実施例の高周波帯域通過フィルタ装置の等価回路は構成される。すなわち、第３の実施例の高周波帯域通過フィルタ装置において、入力端子Ｔ１１に、入力された所定の周波数を有する高周波信号は、入力キャパシタ２２を介して上記２重モード誘電体共振器Ｒ１のｘ軸共振器に入力され、上記高周波信号は、内部結合キャパシタＣ３ａ，Ｃ４ａによって上記２重モード誘電体共振器Ｒ１のｙ軸共振器に伝送される。上記２重モード誘電体共振器Ｒ１のｙ軸共振器に伝送された高周波信号は、上記結合ループインダクタ２１を介して上記２重モード誘電体共振器Ｒ２のｙ軸共振器に伝送され、さらに内部結合キャパシタＣ３ｂ，Ｃ４ｂによって上記２重モード誘電体共振器Ｒ２のｘ軸共振器に伝送される。上記２重モード誘電体共振器Ｒ２のｘ軸共振器に伝送された高周波信号は、上記結合キャパシタＣ８を介して上記２重モード誘電体共振器Ｒ３のｘ軸共振器に伝送され、さらに内部結合キャパシタＣ３ｃ，Ｃ４ｃによって上記２重モード誘電体共振器Ｒ３のｙ軸共振器に伝送される。そして、上記２重モード誘電体共振器Ｒ３のｙ軸共振器に伝送された高周波信号は、上記出力キャパシタ２５を介して出力端子Ｔ１２から出力される。以上のようにして、第３の実施例の高周波帯域通過フィルタ装置は、入力された所定の周波数を有する信号を通過させて出力する。
【００８８】
次に、図１７の等価回路における各回路定数の設定方法を示す。
【００８９】
上記第３の実施例の高周波帯域通過フィルタ装置において、上記各ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３におけるｘ軸共振器とｙ軸共振器の内部結合と、上記ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３の間における結合は、それぞれＫインバータおよびＪインバータを用いたフィルタ設計手法と同様の回路によって表わすことができる。従って、図１７に示した等価回路の各回路定数はチェビシェフ設計の外部Ｑ_ｅおよび結合係数ｋを用いて求められる。ここで、図１７の等価回路では、ＫインバータおよびＪインバータはそれぞれ集中定数回路表現を変形したうえで用いている。
【００９０】
これによると、まず、入力キャパシタ２２の静電容量Ｃ_０１と出力キャパシタ２５の静電容量Ｃ_６７は、次の数９で表わすことができ、キャパシタＣ２ａの静電容量Ｃ_１１とキャパシタＣ２ｃの静電容量Ｃ_６６は、次の数１０で表わすことができる。ここで、Ｊは入出力部のアドミタンス・インバータ・パラメータであり次の数１１で表され、Ｚ_Ｌは入出力インピーダンスであり、ここでは、入力インピーダンスと出力インピーダンスとはともにＺ_Ｌ＝５０Ωに設定した。
【００９１】
【数９】
Ｃ_０１＝Ｃ_６７＝（Ｊ／ω_０）・｛１／√（１−Ｊ^２Ｚ_Ｌ ^２）｝
【数１０】
Ｃ_１１＝Ｃ_６６＝（Ｊ／ω_０）・√（１−Ｊ^２Ｚ_Ｌ ^２）
【数１１】
Ｊ＝√｛（πＹ_ａ）／（Ｚ_ＬＱ_ｅ）｝
【００９２】
また、内部結合キャパシタＣ３ａ，Ｃ４ａの静電容量Ｃ_１２と、内部結合キャパシタＣ３ｂ，Ｃ４ｂの静電容量Ｃ_３４と、内部結合キャパシタＣ３ｃ，Ｃ４ｃの静電容量Ｃ_５６は、次の数１２で表される。ここで、数１２と数１３において添え字の（ｉ，ｊ）は、それぞれ静電容量Ｃ_１２，Ｃ_３４，Ｃ_５６の添え字（１，２），（３，４），（５，６）に対応させている。また、ｋ_１２，ｋ_３４，ｋ_５６は、それぞれ上記ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３におけるｘ軸共振器とｙ軸共振器との結合係数である。
【００９３】
【数１２】
Ｃ_ｉｊ＝（−２Ｙ_ａ／ω_０）・ｔａｎ｛（πｋ_ｉｊ）／（２−ｋ_ｉｊ）｝，（ｉ，ｊ）＝（１，２），（３，４），（５，６）
【００９４】
さらに、ＴＭ２重モード誘電体共振器Ｒ１の各分布定数線路ＬＮ１ａ乃至ＬＮ８ａの電気長θ_１２と、ＴＭ２重モード誘電体共振器Ｒ２の各分布定数線路ＬＮ１ｂ乃至ＬＮ８ｂの電気長θ_３４と、ＴＭ２重モード誘電体共振器Ｒ３の各分布定数線路ＬＮ１ｃ乃至ＬＮ８ｃの電気長θ_５６は、次の数１３で表される。
【００９５】
【数１３】
θ_ｉｊ＝（π／４）・（１＋ｋ_ｉｊ／２），（ｉ，ｊ）＝（１，２），（３，４），（５，６）
【００９６】
またさらに、結合ループインダクタ２１のインダクタンスＬ_２３と、結合キャパシタＣ８の静電容量Ｃ_４５は、それぞれ次の数１４，数１５で表される。また、キャパシタ５ａ，５ｂ，６ａ，６ｂの静電容量Ｃ_２３０は、次の数１６で表される静電容量Ｃ_２３を用いて数１７で表される。ここで、数１４乃至数１６におけるｋ_２３とｋ_４５は、それぞれ上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸共振器と上記ＴＭ２重モード誘電体共振器Ｒ２のｙ軸共振器との結合係数と、上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器と上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸共振器との結合係数である。また、Ｚ_ａは、全て等しく設定された分布定数伝送線路ＬＮ１ａ乃至ＬＮ８ａ、分布定数伝送線路ＬＮ１ｂ乃至ＬＮ８ｂ、分布定数伝送線路ＬＮ１ｃ乃至ＬＮ８ｃの特性インピーダンスである。
【００９７】
【数１４】
Ｌ_２３＝（ｋ_２３πＺ_ａ）／ω_０
【数１５】
Ｃ_４５＝（ｋ_４５πＹ_ａ）／ω_０
【数１６】
Ｃ_２３＝（ｋ_２３πＹ_ａ）／ω_０
【数１７】
Ｃ_２３０＝−Ｃ_２３／２
【００９８】
以上のようにして図１７の等価回路の各回路定数は設定される。
【００９９】
次に、上記第３の実施例の高周波帯域通過フィルタ装置の電気設計例を示す。上記ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３を用いたフィルタ設計においてもチェビシェフ型フィルタの設計理論を適用することができる。チェビシェフ型フィルタの設計理論に従うと、表３に示したフィルタ設計仕様より外部Ｑおよび結合係数ｋ_１２，ｋ_２３，ｋ_３４，ｋ_４５，ｋ_５６が表４のように求められる。表４の各パラメータと上述した回路定数の設定方法を用いて、図１７の等価回路の各回路定数は表５のように計算される。表５には、チェビシェフ型フィルタの設計理論を用いて計算した図１７の等価回路の各回路定数と、これらの回路定数を初期値として、チェビシェフ型フィルタの設計理論により求めた出力端伝送係数Ｓ_２１と入力端反射係数Ｓ_１１に一致するように回路シミュレータ（ヒューレット・パッカード社製ＭＤＳ）により最適化した各回路定数を表５に合わせて示す。そのときの出力端伝送係数Ｓ_２１と入力端反射係数Ｓ_１１と、チェビシェフ型フィルタの設計理論により求めた出力端伝送係数Ｓ_２１と入力端反射係数Ｓ_１１を図１８に示す。出力端伝送係数Ｓ_２１は減衰領域で若干異なるが、通過帯域内では良く一致している。入力端反射係数Ｓ_１１は全ての周波数領域にわたって良く一致している。
【０１００】
【表３】

【０１０１】
【表４】

【０１０２】
【表５】

【０１０３】
本発明者は、表３に示したフィルタ設計仕様に従って上記高周波帯域通過フィルタ装置を試作して、その特性をネットワークアナライザ（ＨＰ８７５３Ｃ）を用いて評価した。図１９に上記試作した高周波帯域通過フィルタ装置の特性評価結果と当該高周波帯域通過フィルタ装置の等価回路を用いて行ったシミュレーション結果を示す。当該高周波帯域通過フィルタ装置の中心周波数は９８５ＭＨｚ、通過帯域幅は１２ＭＨｚである。上記測定結果によると中心周波数における挿入損失は１．３ｄＢ、反射損失は２２ｄＢである。また、３ｄＢ減衰点の帯域幅は１３．３ＭＨｚ、中心周波数である９８５ＭＨｚから２０ＭＨｚ離れた点での減衰量は６２．８ｄＢである。図１９から明らかなように、測定結果と等価回路のシミュレーション結果が良く一致していることがわかる。また挿入損失の値より、結合機構や調整機構を実装した状態で各段の共振器の無負荷Ｑは約２７００で動作していることが推察される。試作した高周波帯域通過フィルタ装置の外形寸法は１５８×５４×２３．５ｍｍであり、従来の高周波帯域通過フィルタ装置に比較して小型薄型化が実現できた。
【０１０４】
以上の第３の実施例の高周波帯域通過フィルタ装置は、従来例の高周波帯域通過フィルタ装置に比較して、小型でかつ薄型にできる。
【０１０５】
以上の第３の実施例の高周波帯域通過フィルタ装置では、入力端子Ｔ１１とＴＭ２重モード誘電体共振器Ｒ１のｘ軸共振器との結合と、ＴＭ２重モード誘電体共振器Ｒ３のｙ軸共振器と出力端子Ｔ１２との結合を、それぞれ入力キヤパシタ２２と出力キャパシタ２５による結合損失の小さい容量結合を用いているので、当該高周波帯域通過フィルタ装置の通過帯域における損失を小さくすることができる。
【０１０６】
＜第１の変形例＞
図２０（ａ）は、本発明に係る第１の変形例のＴＭ２重モード誘電体共振器の縦断面図であり、図２０（ｂ）は、当該ＴＭ２重モード誘電体共振器の図２０（ａ）のＥ−Ｅ’線についての横断面図である。当該ＴＭ２重モード誘電体共振器が第１の実施例のＴＭ２重モード誘電体共振器と異なる所は、誘電体１０１，１０２と平板電極１０３と導体ケース１０８が形成するキャビティー１０８ａのそれぞれの横断面形状を正方形にして構成している点である。ここで、誘電体１０１，１０２は、それぞれ一方の端面に端面電極１０６，１０７を備える。また、上記平板電極１０３の外周の全周には、上記平板電極１０３の厚さより大きい直径の円形断面を有するトーラス電極１０４を備える。
【０１０７】
以上のように構成した第１の変形例のＴＭ２重モード誘電体共振器は、第１の実施例のＴＭ２重モード誘電体共振器と同様に動作し、かつ同様の効果を有する。
【０１０８】
＜第２の変形例＞
本発明に係る第２の変形例の高周波帯域通過フィルタ装置は、第３の実施例の高周波帯域通過フィルタ装置の結合キャパシタ２３，２４に代えて結合ループインダクタ２１ａを設けたことを特徴とする。図２６（ａ）は、第２の変形例の高周波帯域通過フィルタ装置の概略平面図である。図２６（ａ）に示すように、当該高周波帯域通過フィルタ装置において、上記導体ケース８１は、互いに同一の正方柱形状を有する３つの導体部分８１ａ，８１ｂ，８１ｃからなり、隣接する２つの導体部分８１ａ，８１ｂが互いに１側面を介して連結されかつ導体部分８１ｂ，８１ｃが互いに１側面を介して連結されることによってＬ字型に一体的に形成される。そして、上記導体部分８１ａには、ＴＭ２重モード誘電体共振器Ｒ１が設けられ、上記導体部分８１ｂには、ＴＭ２重モード誘電体共振器Ｒ２が設けられ、上記導体部分８１ｃには、ＴＭ２重モード誘電体共振器Ｒ３が設けられる。ここで、上記３つのＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３の各ｘ軸は互いに平行になり、ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３のｙ軸は互いに平行になるように、ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３は設けられる。また、上記ＴＭ２重モード誘電体共振器Ｒ１のｘ軸と上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸は、１直線に存在する一方、上記ＴＭ２重モード誘電体共振器Ｒ２のｙ軸と上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸は１直線に存在する。また、入力コネクタ４１は、上記導体ケース８１の端面８１ｔａの中央部に設けられ、出力コネクタ４２は、上記導体ケース８１の端面８１ｔｂの中央部に設けられる。さらに、上記結合ループインダクタ２１は、上記ＴＭ２重モード誘電体共振器Ｒ１と上記ＴＭ２重モード誘電体共振器Ｒ２を隔てる導体ケース８１の導体壁の中央部に設けられ、上記結合ループインダクタ２１ａは、上記ＴＭ２重モード誘電体共振器Ｒ２と上記ＴＭ２重モード誘電体共振器Ｒ３を隔てる導体ケース８１の導体壁の中央部に設けられる。
【０１０９】
図２１は、第２の変形例の高周波帯域通過フィルタ装置の等価回路を示す回路図である。図２１の等価回路は、図１７の等価回路と同様に、ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３の等価回路を備え、以下のように構成される。
【０１１０】
上記ＴＭ２重モード誘電体共振器Ｒ１における分布定数線路ＬＮ１ａ乃至分布定数線路Ｌ８ａの各接続点は、図１７の等価回路と同様な各素子が接続される。また、図１７の等価回路と同様な構成により、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸共振器と上記ＴＭ２重モード誘電体共振器Ｒ２のｙ軸共振器とは、結合ループインダクタ２１を介して誘導結合する。
【０１１１】
上記ＴＭ２重モード誘電体共振器Ｒ２の分布定数線路ＬＮ１ｂの上記インダクタＬ２との接合点は、キャパシタＣ９ｂを介して接地される。分布定数線路ＬＮ２ｂと分布定数線路ＬＮ３ｂの接続点と、分布定数線路ＬＮ４ｂと分布定数線路ＬＮ５ｂの接続点は、それぞれキャパシタ５ｂ，７ｂを介して接地される。分布定数線路ＬＮ６ｂと分布定数線路ＬＮ７ｂの間には、インダクタＬ３が直列に接続され、上記分布定数線路ＬＮ７ｂと上記インダクタＬ３の接続点は、キャパシタＣ６ｂを介して接地される。ここで、上記インダクタＬ３は、上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器によって発生する磁界のうちの上記結合ループインダクタ２１ａに誘導結合する磁界を発生するための等価的なインダクタを表わす。また、上記キャパシタＣ７ｂ、Ｃ９ｂは、結合ループインダクタ２１ａが上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器と結合することにより変化する当該ｘ軸共振器の共振周波数を補正するための静電容量であって、それぞれ負の静電容量値を有する。
【０１１２】
上記インダクタＬ３はインダクタＬ２１と誘導結合する。また、上記インダクタＬ２１とインダクタＬ２２とインダクタＬ２３は、並列に接続されて、結合ループインダクタ２１ａを構成し、上記結合ループインダクタ２１ａの一端は接地される。そして、上記インダクタＬ２３は、上記ＴＭ２重モード誘電体共振器Ｒ３の分布定数線路ＬＮ２ｃと分布定数線路ＬＮ３ｃとの間に接続されたインダクタＬ４と誘導結合する。ここで、上記インダクタＬ４は、上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸共振器によって発生する磁界のうちの上記結合ループインダクタ２１ａに誘導結合する磁界を発生するための等価的なインダクタを表わす。以上のようにして、上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器と上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸共振器は誘導結合する。
【０１１３】
上記ＴＭ２重モード誘電体共振器Ｒ３の分布定数線路ＬＮ１ｃと分布定数線路ＬＮ８ｃの接続点と、分布定数線路ＬＮ４ｃと分布定数線路ＬＮ５ｃの接続点は、それぞれキャパシタＣ９ｃ，Ｃ１０ｃを介して接地される。ここで、上記キャパシタＣ９ｃ，Ｃ１０ｃは、結合ループインダクタ２１ａが上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸共振器と結合することにより変化する当該ｘ軸共振器の共振周波数を補正するための静電容量であって、それぞれ負の静電容量値を有する。
【０１１４】
上記ＴＭ２重モード誘電体共振器Ｒ３の上記分布結合線路ＬＮ６ｃと上記分布定数結合線路ＬＮ７ｃの接続点は、キャパシタＣ２ｃを介して接地されるとともに、一端が出力端子Ｔ１２である出力キャパシタ２５の他端に接続される。これによって、上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸共振器を外部回路と容量結合させることができる。
【０１１５】
以上のように、第２の変形例の高周波帯域通過フィルタ装置は、３つのＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３を備えている。そして、当該高周波帯域通過フィルタ装置において、上記入力コネクタ４１は、入力キャパシタ２２によって、上記ＴＭ２重モード誘電体共振器Ｒ１のｘ軸共振器と容量結合する。そして、上記ＴＭ２重モード誘電体共振器Ｒ１のｘ軸共振器は、上記誘電体１ａに設けられた切り欠き１１ａ，１２ａと上記誘電体２ａに設けられた切り欠き１３ａ，１４ａとによって、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸共振器と電磁的に結合する。また、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸共振器は、上記結合ループインダクタ２１によって、上記ＴＭ２重モード誘電体共振器Ｒ２のｙ軸共振器と誘導結合する。上記ＴＭ２重モード誘電体共振器Ｒ２のｙ軸共振器は、上記誘電体１ｂに設けられた切り欠き１１ｂ，１２ｂと上記誘電体２ｂに設けられた切り欠き１３ｂ，１４ｂとによって、上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器と電磁的に結合する。さらに、上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器は、結合ループインダクタ２１ａによって、上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸共振器と誘導結合する。上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸共振器は、上記誘電体１ｃに設けられた切り欠き１１ｃ，１２ｃと上記誘電体２ｃに設けられた切り欠き１３ｃ，１４ｃとによって、上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸共振器と電磁的に結合する。またさらに、上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸共振器は、上記出力キャパシタ２５によって、上記出力コネクタ４２と容量結合する。以上のように構成することにより、入力コネクタ４１と出力コネクタ４２の間に、ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３を備えた６段の高周波帯域通過フィルタ装置が構成される。これによって、第２の変形例の高周波帯域通過フィルタ装置は、第３の実施例の高周波帯域通過フィルタ装置と同様の動作をして同様の効果を有する。
【０１１６】
＜第３の変形例＞
本発明に係る第３の変形例の高周波帯域通過フィルタ装置は、第３の実施例の高周波帯域通過フィルタ装置の結合ループインダクタ２１に代えてＴＭ２重モード誘電体共振器Ｒ１とＴＭ２重モード誘電体共振器Ｒ２とを容量結合する結合キャパシタ２３ａ，２４ａを設けたことを特徴とする。図２６（ｂ）は、第３の変形例の高周波帯域通過フィルタ装置の概略平面図である。図２６（ｂ）に示すように、当該高周波帯域通過フィルタ装置は、第２の変形例と同様に、Ｌ字型に形成された導体ケース８１を備え、ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３が第２の変形例と同様に構成される。そして、入力コネクタ４１は、入力コネクタ４１の中心導体が上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸上に位置するように上記導体ケース８１の側面８１ｓａに設けられ、出力コネクタ４２は、出力導体４２の中心導体が上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸上に位置するように上記導体ケース８１の側面８１ｓｂに設けられる。さらに、上記結合キャパシタ２３ａ，２４ａはそれぞれ、互いに対向する上記ＴＭ２重モード誘電体共振器Ｒ１のトーラス電極４ａの外周面と上記ＴＭ２重モード誘電体共振器Ｒ２のトーラス電極４ｂの外周面に設けられ、上記ＴＭ２重モード誘電体共振器Ｒ１と上記ＴＭ２重モード誘電体共振器Ｒ２を隔てる導体ケース８１の導体壁の中央部を、電気的に絶縁されて貫通する導体線によって接続される。上記結合キャパシタ２３，２４はそれぞれ、互いに対向する上記ＴＭ２重モード誘電体共振器Ｒ２のトーラス電極４ｂの外周面と上記ＴＭ２重モード誘電体共振器Ｒ３のトーラス電極４ｃの外周面に設けられ、上記ＴＭ２重モード誘電体共振器Ｒ２と上記ＴＭ２重モード誘電体共振器Ｒ３を隔てる導体ケース８１の導体壁の中央部を、電気的に絶縁されて貫通する導体線によって接続される。
【０１１７】
図２２は、第３の変形例の高周波帯域通過フィルタ装置の等価回路を示す回路図である。図２２の等価回路は、図１７の等価回路と同様に、ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３の等価回路を備え、以下のように構成される。
【０１１８】
上記ＴＭ２重モード誘電体共振器Ｒ１の分布定数線路ＬＮ２ａと分布定数線路ＬＮ３ａの接続点は、一端が入力端子Ｔ１１である入力キャパシタ２２の他端が接続されるとともに、キャパシタＣ２ａを介して接地される。
【０１１９】
また、上記ＴＭ２重モード誘電体共振器Ｒ１の分布定数線路ＬＮ４ａと分布定数線路ＬＮ５ａの接続点には、キャパシタＣ１１ａを介して接地されるとともに、結合キャパシタＣ１２の一端が接続される。ここで、上記結合キャパシタＣ１２は、上記結合キャパシタ２３ａ，２４ａが直列に接続されて構成される結合キャパシタに対応して設けられる。そして、上記結合キャパシタＣ１２の他端には、上記ＴＭ２重モード誘電体共振器Ｒ２の分布定数線路ＬＮ１ｂと分布定数線路ＬＮ８ｂとの接続点が接続される。さらに、分布定数線路ＬＮ１ｂと分布定数線路ＬＮ８ｂとの接続点は、キャパシタＣ１１ｂを介して接地される。これによって、上記ＴＭ２重モード誘電体共振器Ｒ１のｘ軸共振器と上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器とは容量結合する。ここで、キャパシタＣ１１ａは、上記ＴＭ２重モード誘電体共振器Ｒ１のｘ軸共振器が上記結合キャパシタＣ１２と結合することにより変化する当該ｘ軸共振器の共振周波数を補正するための静電容量であって、負の静電容量値を有する。また、キャパシタＣ１１ｂは、上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器が上記結合キャパシタＣ１２と結合することにより変化する当該ｘ軸共振器の共振周波数を補正するための静電容量であって、負の静電容量値を有する。
【０１２０】
さらに、上記ＴＭ２重モード誘電体共振器Ｒ２の分布定数線路ＬＮ６ｂと分布定数線路ＬＮ７ｂの接続点は、キャパシタＣ７ｂを介して接地されるとともに、結合キャパシタＣ８の一端が接続される。上記ＴＭ２重モード誘電体共振器Ｒ３の分布定数線路ＬＮ２ｃと分布定数線路ＬＮ３ｃの接続点は、キャパシタＣ７ｃを介して接地されるとともに、結合キャパシタＣ８の他端が接続される。これによって、上記ＴＭ２重モード誘電体共振器Ｒ２のｙ軸共振器と上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸共振器とは容量結合する。
【０１２１】
上記ＴＭ２重モード誘電体共振器Ｒ３の上記分布結合線路ＬＮ４ｃと上記分布定数結合線路ＬＮ５ｃの接続点は、キャパシタＣ２ｃを介して接地されるとともに、一端が出力端子Ｔ１２である出力キャパシタ２５の他端に接続される。これによって、上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸共振器を外部回路と容量結合させることができる。
【０１２２】
以上のように、第３の変形例の高周波帯域通過フィルタ装置は、３つのＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３を備えている。そして、当該高周波帯域通過フィルタ装置において、上記入力コネクタ４１は、入力キャパシタ２２によって、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸共振器と容量結合する。そして、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸共振器は、上記誘電体１ａに設けられた切り欠き１１ａ，１２ａと上記誘電体２ａに設けられた切り欠き１３ａ，１４ａとによって、上記ＴＭ２重モード誘電体共振器Ｒ１のｘ軸共振器と電磁的に結合する。また、上記ＴＭ２重モード誘電体共振器Ｒ１のｘ軸共振器は、上記結合キャパシタＣ１２によって、上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器と容量結合する。上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器は、上記誘電体１ｂに設けられた切り欠き１１ｂ，１２ｂと上記誘電体２ｂに設けられた切り欠き１３ｂ，１４ｂとによって、上記ＴＭ２重モード誘電体共振器Ｒ２のｙ軸共振器と電磁的に結合する。さらに、上記ＴＭ２重モード誘電体共振器Ｒ２のｙ軸共振器は、結合キャパシタＣ８によって、上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸共振器と容量結合する。上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸共振器は、上記誘電体１ｃに設けられた切り欠き１１ｃ，１２ｃと上記誘電体２ｃに設けられた切り欠き１３ｃ，１４ｃとによって、上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸共振器と電磁的に結合する。またさらに、上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸共振器は、上記出力キャパシタ２５によって、上記出力コネクタ４２と容量結合する。以上のように構成することにより、入力コネクタ４１と出力コネクタ４２の間に、ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３を備えた６段の高周波帯域通過フィルタ装置が構成される。これによって、第３の変形例の高周波帯域通過フィルタ装置は、第３の実施例の高周波帯域通過フィルタ装置と同様の動作をして同様の効果を有する。
【０１２３】
＜第４の変形例＞
本発明に係る第４の変形例の高周波帯域通過フィルタ装置は、第３の実施例の高周波帯域通過フィルタ装置の入力キャパシタ２２に代えて入力ループインダクタ３１を備え、かつ出力キャパシタ２５に代えて出力ループインダクタ４３を備えたことを特徴とする。図２７（ａ）は、第４の変形例の高周波帯域通過フィルタ装置の概略平面図である。図２７（ａ）に示すように、当該高周波帯域通過フィルタ装置は、上記第３の実施例の高周波帯域通過フィルタ装置と同様の導体ケース８０を備え、上記ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３が互いに所定の間隔を隔てて並置されて構成される。そして、上記第４の変形例の高周波帯域通過フィルタ装置において、入力コネクタ４１は、上記導体ケース８０の一方の側面に、上記入力コネクタ４１の中心導体が上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸に一致するように設けられ、出力コネクタ４２は、上記導体ケース８０の上記ＴＭ２重モード誘電体共振器Ｒ３に隣接する端面の中央部に設けられる。
【０１２４】
図２３は、第４の変形例の高周波帯域通過フィルタ装置の等価回路を示す回路図である。図２３の等価回路は、図１７の等価回路と同様に、ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３の等価回路を備え、以下のように構成される。
【０１２５】
入力端子Ｔ１１に、一端が接地されたインダクタＬ５の他端が接続される。ここで、上記インダクタＬ５は、例えば上記入力コネクタの中心導体の先端を接地させて構成されるループに対応したインダクタである。そして、上記インダクタＬ５にはインダクタＬ３１が誘導結合する。上記インダクタＬ３１とインダクタＬ３２とインダクタＬ３３は、並列に接続されて、入力ループインダクタ３１を構成し、上記入力ループインダクタ３１の一端は接地される。そして、上記インダクタＬ３３は、上記ＴＭ２重モード誘電体共振器Ｒ１の分布定数線路ＬＮ６ａと分布定数線路ＬＮ７ａとの間に接続されたインダクタＬ６と誘導結合する。ここで、上記インダクタＬ６は、上記ＴＭ２重モード誘電体共振器Ｒ１のｘ軸共振器によって発生する磁界のうちの上記結合ループインダクタ３１に誘導結合する磁界を発生するための等価的なインダクタを表わす。これによって、上記ＴＭ２重モード誘電体共振器Ｒ１のｘ軸共振器と上記入力端子Ｔ１１は誘導結合する。
【０１２６】
上記ＴＭ２重モード誘電体共振器Ｒ１の分布定数線路ＬＮ７ａと上記インダクタＬ６の接続点は、キャパシタＣ６ａを介して接地される。
【０１２７】
また、上記ＴＭ２重モード誘電体共振器Ｒ１の分布定数線路ＬＮ８ａと分布定数線路ＬＮ１ａの接続点は、キャパシタＣ１３ａを介して接地される。分布定数線路ＬＮ２ａと分布定数線路ＬＮ３ａの接続点は、キャパシタＣ５ａを介して接地される。そして、分布定数線路ＬＮ４ａと分布定数線路ＬＮ５ａの間にはインダクタＬ１が直列に接続され、上記分布定数線路ＬＮ４ａとインダクタＬ１の接続点は、キャパシタＣ１４ａを介して接地される。ここで、上記キャパシタＣ１３ａ，Ｃ１４ａは、入力ループインダクタＬ３０が上記ＴＭ２重モード誘電体共振器Ｒ１のｘ軸共振器と結合することにより変化する当該ｘ軸共振器の共振周波数を補正するための静電容量であって、それぞれ負の静電容量値を有する。
【０１２８】
上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸共振器と上記ＴＭ２重モード誘電体共振器Ｒ２のｙ軸共振器は、図１７の等価回路と同様に構成されて、誘導結合する。また、上記ＴＭ２重モード誘電体共振器Ｒ２の分布定数線路ＬＮ１ｂ乃至ＬＮ８ｂの各接続点には、図１７の等価回路と同様の各素子が接続される。
【０１２９】
上記２重モード誘電体共振器Ｒ３の分布定数線路ＬＮ１ｃと分布定数線路ＬＮ８ｃの接続点は、キャパシタＣ７ｃを介して接地されるとともに、上記結合キャパシタＣ８の他端が接続される。これによって、上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器と上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸共振器とは容量結合する。
【０１３０】
さらに、上記ＴＭ２重モード誘電体共振器Ｒ３の分布定数線路ＬＮ２ｃと分布定数線路ＬＮ３ｃの接続点と、分布定数線路ＬＮ６ｃと分布定数線路ＬＮ７ｃの接続点は、それぞれキャパシタＣ１５ｃとキャパシタ１６ｃを介して接地される。そして、分布定数線路ＬＮ４ｃと分布定数線路ＬＮ５ｃの間にはインダクタＬ７が直列に接続される。ここで、上記インダクタＬ７は、上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸共振器によって発生する磁界のうちの上記結合ループインダクタ４３に誘導結合する磁界を発生するための等価的なインダクタを表わす。そして、上記インダクタＬ７にはインダクタＬ４１が誘導結合する。上記インダクタＬ４１とインダクタＬ４２とインダクタＬ４３は、並列に接続されて、出力ループインダクタ４３を構成し、上記出力ループインダクタ４３の一端は接地される。そして、上記インダクタＬ４３は、一端に出力端子Ｔ１２が接続され他端が接地されたインダクタＬ８と誘導結合する。ここで、上記インダクタＬ８は、例えば上記出力コネクタ４２の中心導体の先端を接地させて構成されるループに対応したインダクタである。これによって、上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸共振器は、上記出力ループインダクタ４３を介して出力端子Ｔ１２と誘導結合する。ここで、上記キャパシタＣ１５ｃ，１６ｃは、上記出力ループインダクタ４３が上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸共振器と結合することにより変化する当該ｙ軸共振器の共振周波数を補正するための静電容量であって、それぞれ負の静電容量値を有する。
【０１３１】
以上のように、第４の変形例の高周波帯域通過フィルタ装置は、３つのＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３を備えている。そして、当該高周波帯域通過フィルタ装置において、上記入力コネクタ４１は、入力ループインダクタ３１によって、上記ＴＭ２重モード誘電体共振器Ｒ１のｘ軸共振器と誘導結合する。そして、上記ＴＭ２重モード誘電体共振器Ｒ１のｘ軸共振器は、上記誘電体１ａに設けられた切り欠き１１ａ，１２ａと上記誘電体２ａに設けられた切り欠き１３ａ，１４ａとによって、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸共振器と電磁的に結合する。また、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸共振器は、上記結合ループインダクタ２１によって、上記ＴＭ２重モード誘電体共振器Ｒ２のｙ軸共振器と誘導結合する。上記ＴＭ２重モード誘電体共振器Ｒ２のｙ軸共振器は、上記誘電体１ｂに設けられた切り欠き１１ｂ，１２ｂと上記誘電体２ｂに設けられた切り欠き１３ｂ，１４ｂとによって、上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器と電磁的に結合する。さらに、上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器は、結合キャパシタＣ８によって、上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸共振器と容量結合する。上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸共振器は、上記誘電体１ｃに設けられた切り欠き１１ｃ，１２ｃと上記誘電体２ｃに設けられた切り欠き１３ｃ，１４ｃとによって、上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸共振器と電磁的に結合する。またさらに、上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸共振器は、上記出力ループインダクタ４３によって、上記出力コネクタ４２と誘導結合する。以上のように構成することにより、入力コネクタ４１と出力コネクタ４２の間に、ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３を備えた６段の高周波帯域通過フィルタ装置が構成される。これによって、第４の変形例の高周波帯域通過フィルタ装置は、第３の実施例の高周波帯域通過フィルタ装置と同様の動作をして同様の効果を有する。
【０１３２】
＜第５の変形例＞
本発明に係る第５の変形例の高周波帯域通過フィルタ装置は、第３の実施例の高周波帯域通過フィルタ装置の出力キャパシタ２５に代えて出力ループインダクタ４３を備えたことを特徴とする。図２７（ｂ）は、第５の変形例の高周波帯域通過フィルタ装置の概略平面図である。図２７（ｂ）に示すように、当該高周波帯域通過フィルタ装置は、上記第３の実施例の高周波帯域通過フィルタ装置と同様の導体ケース８０を備え、上記ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３が互いに所定の間隔を隔てて並置されて構成される。そして、上記第５の変形例の高周波帯域通過フィルタ装置において、入力コネクタ４１は、上記導体ケース８０の上記ＴＭ２重モード誘電体共振器Ｒ１に隣接する一方の端面の中央部に設けられ、出力コネクタ４２は、上記導体ケース８０の上記ＴＭ２重モード誘電体共振器Ｒ３に隣接する端面の中央部に設けられる。
【０１３３】
図２４は、第５の変形例の高周波帯域通過フィルタ装置の等価回路を示す回路図である。図２４の等価回路において、上記ＴＭ２重モード誘電体共振器Ｒ１の分布定数線路ＬＮ１ａ乃至ＬＮ８ａの各接続点には、図１７の等価回路と同様の素子が接続され、上記ＴＭ２重モード誘電体共振器Ｒ２の分布定数線路ＬＮ１ｂ乃至ＬＮ８ｂの各接続点には、図１７の等価回路と同様の素子が接続される。また、上記ＴＭ２重モード誘電体共振器Ｒ３の分布定数線路ＬＮ１ｃ乃至ＬＮ８ｃの各接続点には、図２３の等価回路と同様の素子が接続される。
【０１３４】
以上のように、第５の変形例の高周波帯域通過フィルタ装置は、３つのＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３を備えている。そして、当該高周波帯域通過フィルタ装置において、上記入力コネクタ４１は、入力キャパシタ２２によって、上記ＴＭ２重モード誘電体共振器Ｒ１のｘ軸共振器と容量結合する。そして、上記ＴＭ２重モード誘電体共振器Ｒ１のｘ軸共振器は、上記誘電体１ａに設けられた切り欠き１１ａ，１２ａと上記誘電体２ａに設けられた切り欠き１３ａ，１４ａとによって、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸共振器と電磁的に結合する。また、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸共振器は、上記結合ループインダクタ２１によって、上記ＴＭ２重モード誘電体共振器Ｒ２のｙ軸共振器と誘導結合する。上記ＴＭ２重モード誘電体共振器Ｒ２のｙ軸共振器は、上記誘電体１ｂに設けられた切り欠き１１ｂ，１２ｂと上記誘電体２ｂに設けられた切り欠き１３ｂ，１４ｂとによって、上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器と電磁的に結合する。さらに、上記ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器は、結合キャパシタＣ８によって、上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸共振器と容量結合する。上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸共振器は、上記誘電体１ｃに設けられた切り欠き１１ｃ，１２ｃと上記誘電体２ｃに設けられた切り欠き１３ｃ，１４ｃとによって、上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸共振器と電磁的に結合する。またさらに、上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸共振器は、上記結合ループインダクタ４３によって、上記出力コネクタ４２と誘導結合する。以上のように構成することにより、入力コネクタ４１と出力コネクタ４２の間に、ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３を備えた６段の高周波帯域通過フィルタ装置が構成される。これによって、第５の変形例の高周波帯域通過フィルタ装置は、第３の実施例の高周波帯域通過フィルタ装置と同様の動作をして同様の効果を有する。
【０１３５】
以上詳述した第３の実施例の高周波帯域通過フィルタ装置と第２及至５の変形例の高周波帯域通過フィルタ装置とを含む、本発明に係る種々の高周波帯域通過フィルタ装置の動作について図２８乃至図３３を参照して説明する。図２８乃至図３３において、Ｒ１ｘ，Ｒ２ｘ，Ｒ３ｘはそれぞれＴＭ２重モード誘電体共振器Ｒ１のｘ軸共振器、ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器、ＴＭ２重モード誘電体共振器Ｒ３のｘ軸共振器を表し、Ｒ１ｙ，Ｒ２ｙ，Ｒ３ｙはそれぞれＴＭ２重モード誘電体共振器Ｒ１のｙ軸共振器、ＴＭ２重モード誘電体共振器Ｒ２のｙ軸共振器、ＴＭ２重モード誘電体共振器Ｒ３のｙ軸共振器を表す。また、図２８乃至図３３において、Ｃ結合は容量結合のことをいい、Ｌ結合は誘導結合のことをいう。以下の説明において、ＴＭ２重モード誘電体共振器Ｒ１のｘ軸共振器、ＴＭ２重モード誘電体共振器Ｒ２のｘ軸共振器、ＴＭ２重モード誘電体共振器Ｒ３のｘ軸共振器はそれぞれ、共振器Ｒ１ｘ，共振器Ｒ２ｘ，共振器Ｒ３ｘと称し、ＴＭ２重モード誘電体共振器Ｒ１のｙ軸共振器、ＴＭ２重モード誘電体共振器Ｒ２のｙ軸共振器、ＴＭ２重モード誘電体共振器Ｒ３のｙ軸共振器はそれぞれ、共振器Ｒ１ｙ，共振器Ｒ２ｙ，共振器Ｒ３ｙと称する。
【０１３６】
図２８は、第３の実施例の高周波帯域通過フィルタ装置の動作説明図である。図２８において、上記入力コネクタ４１は共振器Ｒ１ｘに容量結合し、上記共振器Ｒ１ｘは上記共振器Ｒ１ｙに電磁的に結合する。次に上記共振器Ｒ１ｙは上記共振器Ｒ２ｙに誘導結合する。上記共振器Ｒ２ｙは上記共振器Ｒ２ｘに電磁的に結合して、上記共振器Ｒ２ｘは上記共振器Ｒ３ｘに容量結合する。そして、上記共振器Ｒ３ｘは上記共振器Ｒ３ｙに電磁的に結合して、上記共振器Ｒ３ｙは上記出力コネクタ４２に容量結合する。以上のように入力コネクタ４１と出力コネクタ４２の間に上記各共振器を結合させたとき、上記ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３は、図１６にしめすように上記導体ケース８０の長手方向に並置されて設けられ、入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１に隣接する上記ケース８０の端面に設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸上の上記導体ケース８０の側面に設けられる。
【０１３７】
また、図２８において、第３の実施例とはとって代わり、上記入力コネクタ４１を共振器Ｒ１ｘに誘導結合させるとともに、上記出力コネクタ４２を上記共振器Ｒ３ｙに誘導結合させると、図２８の結合方法は、図２７（ａ）に示す第４の変形例の高周波帯域通過フィルタ装置の結合方法と同じになる。以上のように入力コネクタ４１と出力コネクタ４２の間に上記各共振器を結合させたとき、上記ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３は、上記導体ケース８０の長手方向に並置されて設けられ、入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸上の上記導体ケース８０の側面に設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３に隣接する上記導体ケース８０の端面に設けられる。
【０１３８】
さらに、図２８において、第３の実施例とはとって代わり、上記入力コネクタ４１を共振器Ｒ１ｘに容量結合させるとともに、上記出力コネクタ４２を上記共振器Ｒ３ｙに誘導結合させると、図２８の結合方法は、図２７（ｂ）に示す第５の変形例の高周波帯域通過フィルタ装置の結合方法と同じになる。以上のように入力コネクタ４１と出力コネクタ４２の間に上記各共振器を結合させたとき、上記ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３は、上記導体ケース８０の長手方向に並置され、入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１に隣接する上記導体ケース８０の端面に設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３に隣接する上記導体ケース８０の端面に設けられる。
【０１３９】
またさらに、図２８において、上記入力コネクタ４１を共振器Ｒ１ｘに誘導結合させるとともに、上記出力コネクタ４２を上記共振器Ｒ３ｙに容量結合させると、上記ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３は、上記導体ケース８０の長手方向に並置されて設けられ、入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸上の上記導体ケース８０の側面に設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸上の上記導体ケース８０の側面に設けられる。
【０１４０】
図２９は、図２８の結合方法とは異なる結合方法を用いて構成した動作説明図である。図２９において、上記入力コネクタ４１は共振器Ｒ１ｙに容量結合し、上記共振器Ｒ１ｙは上記共振器Ｒ１ｘと電磁的に結合する。次に上記共振器Ｒ１ｘは上記共振器Ｒ２ｘに容量結合する。上記共振器Ｒ２ｘは上記共振器Ｒ２ｙに電磁的に結合して、上記共振器Ｒ２ｙは上記共振器Ｒ３ｙに誘導結合する。そして、上記共振器Ｒ３ｙは上記共振器Ｒ３ｘに電磁的に結合して、上記共振器Ｒ３ｘは上記出力コネクタ４２と容量結合する。以上のように入力コネクタ４１と出力コネクタ４２の間に上記各共振器を結合させたとき、上記ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３は、上記導体ケース８０の長手方向に並置されて設けられ、入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸上の上記導体ケース８０の側面に設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３に隣接する上記導体ケース８０の端面に設けられる。
【０１４１】
また、図２９において、上記入力コネクタ４１と共振器Ｒ１ｙとを容量結合にに代えて誘導結合させるとともに、上記出力コネクタ４２と上記共振器Ｒ３ｘとを容量結合に代えて誘導結合させると、入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１に隣接する上記導体ケース８０の端面に設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸上の上記導体ケース８０の側面に設けられる。
【０１４２】
さらに、図２９において、上記出力コネクタ４２と上記共振器Ｒ３ｘとを容量結合に代えて誘導結合させると、入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸上の上記導体ケース８０の側面に設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３のｙ軸上の上記導体ケース８０の側面に設けられる。
【０１４３】
またさらに、図２９において、上記入力コネクタ４１と共振器Ｒ１ｙとを容量結合にに代えて誘導結合させると、入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１に隣接する上記導体ケース８０の端面に設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３に隣接する上記導体ケース８０の端面に設けられる。
【０１４４】
以上詳述したように、本発明に係るＴＭ２重モード誘電体共振器は、上記ＴＭ２重モード誘電体共振器Ｒ１と上記ＴＭ２重モード誘電体共振器Ｒ２との間の結合方法と、上記ＴＭ２重モード誘電体共振器Ｒ２と上記ＴＭ２重モード誘電体共振器Ｒ３との間の結合方法のうちの一方を容量結合として、他方を誘導結合にすることにより、上記３つのＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３を一列に並置して構成することができる。すなわち、各ＴＭ２重モード誘電体共振器を一列に並置して構成する高周波帯域通過フィルタ装置において、隣接する各ＴＭ２重モード誘電体共振器のｘ軸又はｙ軸共振器は、誘導結合と容量結合を用いて隣接する結合方法を互いに異ならせている。
【０１４５】
また、本発明に係るＴＭ２重モード誘電体共振器は、上記入力コネクタ４１と上記ＴＭ２重モード誘電体共振器Ｒ１との間の結合方法と、上記ＴＭ２重モード誘電体共振器Ｒ１と上記ＴＭ２重モード誘電体共振器Ｒ２との間の結合方法のうちの一方を容量結合として、他方を誘導結合にすることにより、上記入力コネクタ４１を上記導体ケース８０の端面に設けることができる。同様に、上記出力コネクタ４２と上記ＴＭ２重モード誘電体共振器Ｒ１との間の結合方法と、上記ＴＭ２重モード誘電体共振器Ｒ１と上記ＴＭ２重モード誘電体共振器Ｒ２との間の結合方法のうちの一方を容量結合として、他方を誘導結合にすることにより、上記出力コネクタ４２を上記導体ケース８０の端面に設けることができる。また、上記入力コネクタ４１と上記ＴＭ２重モード誘電体共振器Ｒ１との間の結合方法と、上記ＴＭ２重モード誘電体共振器Ｒ１と上記ＴＭ２重モード誘電体共振器Ｒ２との間の結合方法とをともに容量結合又は誘導結合とすることにより、上記入力コネクタ４１を上記導体ケース８０の側面に設けることができる。同様に、上記出力コネクタ４２と上記ＴＭ２重モード誘電体共振器Ｒ３との間の結合方法と、上記ＴＭ２重モード誘電体共振器Ｒ２と上記ＴＭ２重モード誘電体共振器Ｒ３との間の結合方法をともに容量結合又は誘導結合にすることにより、上記出力コネクタ４２を上記導体ケース８０の側面に設けることができる。
【０１４６】
以上の説明では、３つのＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３を備えた高周波帯域通過フィルタ装置について説明したが、さらに多くのＴＭ２重モード誘電体共振器を備えた高周波帯域通過フィルタ装置においても同様に説明できる。すなわち、上記入力コネクタ４１と初段のＴＭ２重モード誘電体共振器との間の結合方法と、隣接する各ＴＭ２重モード誘電体共振器間の結合方法と、終段のＴＭ２重モード誘電体共振器と上記出力コネクタ４２の間の結合方法とを、誘導結合と容量結合が交互になるように構成することにより、上記ＴＭ２重モード誘電体共振器ＴＭを一列に並置するように設けることができ、かつ上記入力コネクタ４１と上記出力コネクタ４２とを上記導体ケース８０の両端面に設けることができる。
【０１４７】
図３０は、第２の変形例の高周波帯域通過フィルタ装置の動作説明図である。図３０において、上記入力コネクタ４１は共振器Ｒ１ｘに容量結合し、上記共振器Ｒ１ｘは上記共振器Ｒ１ｙと電磁的に結合する。次に上記共振器Ｒ１ｙは上記共振器Ｒ２ｙに誘導結合する。上記共振器Ｒ２ｙは上記共振器Ｒ２ｘに電磁的に結合して、上記共振器Ｒ２ｘは上記共振器Ｒ３ｘに誘導結合する。そして、上記共振器Ｒ３ｘは上記共振器Ｒ３ｙに電磁的に結合して、上記共振器Ｒ３ｙは上記出力コネクタ４２と容量結合する。以上のように入力コネクタ４１と出力コネクタ４２の間に上記各共振器を結合させたとき、図２６（ａ）に示すように、上記ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３はＬ字型に設けられる。そして、入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１に隣接する上記導体ケース８１の端面８１ｔａに設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３に隣接する端面８１ｔｂに設けられる。
【０１４８】
図３０において、第２の変形例とはとって代わり、入力コネクタ４１と共振器Ｒ１ｘとを誘導結合させ、上記出力コネクタ４２と共振器Ｒ３ｙとを誘導結合させると、上記入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸上の上記導体ケース８１の側面８１ｓａに設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸上の上記導体ケース８１の側面８１ｓｂに設けられる。
【０１４９】
図３０において、第２の変形例とはとって代わり、入力コネクタ４１と共振器Ｒ１ｘとを誘導結合させると、上記入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸上の上記導体ケース８１の側面８１ｓａに設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３に隣接する端面８１ｔｂに設けられる。
【０１５０】
図３０において、第２の変形例とはとって代わり、上記出力コネクタ４２と共振器Ｒ３ｙとを誘導結合させると、上記入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１に隣接する上記導体ケース８１の端面８１ｔａに設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸上の側面８１ｓｂに設けられる。
【０１５１】
図３１は、図３０とは異なる結合方法の組み合わせを用いて構成した高周波帯域通過フィルタ装置の動作説明図である。図３１において、図３０と異なる部分は、上記共振器Ｒ１ｙを上記共振器Ｒ２ｙに容量結合させていることと、上記共振器Ｒ２ｘを上記共振器Ｒ３ｘに容量結合させている点である。以上のように入力コネクタ４１と出力コネクタ４２の間に上記各共振器を結合させたとき、上記ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３はＬ字型に設けられる。そして、上記入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸上の上記導体ケース８１の側面８１ｓａに設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸上の上記導体ケース８１の側面８１ｓｂに設けられる。
【０１５２】
図３１において、入力コネクタ４１と共振器Ｒ１ｘとを容量結合に代えて誘導結合させ、上記出力コネクタ４２と共振器Ｒ３ｙとを容量結合に代えて誘導結合させると、入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１に隣接する上記導体ケース８１の端面８１ｔａに設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３に隣接する端面８１ｔｂに設けられる。
【０１５３】
また、図３１において、入力コネクタ４１と共振器Ｒ１ｘとを容量結合に代えて誘導結合させると、入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１に隣接する導体ケース８１の端面８１ｔａに設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸上の上記導体ケース８１の側面８１ｓｂに設けられる。
【０１５４】
図３１において、上記出力コネクタ４２と共振器Ｒ３ｙとを容量結合に代えて誘導結合させると、入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸上の上記導体ケース８１の側面８１ｓａに設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３に隣接する端面８１ｔｂに設けられる。
【０１５５】
図３２は、さらに異なる結合方法の組み合わせを用いて構成した高周波帯域通過フィルタ装置の動作説明図である。図３２において、上記入力コネクタ４１は、共振器Ｒ１ｙに容量結合し、上記共振器Ｒ１ｙは上記共振器Ｒ１ｘと電磁的に結合する。次に上記共振器Ｒ１ｘは上記共振器Ｒ２ｘに誘導結合する。上記共振器Ｒ２ｘは上記共振器Ｒ２ｙに電磁的に結合して、上記共振器Ｒ２ｙは上記共振器Ｒ３ｙに誘導結合する。そして、上記共振器Ｒ３ｙは上記共振器Ｒ３ｘに電磁的に結合して、上記共振器Ｒ３ｘは上記出力コネクタ４２と容量結合する。以上のように入力コネクタ４１と出力コネクタ４２の間に上記各共振器を結合させたとき、上記ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３はＬ字型に設けられる。そして、入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１に隣接する上記導体ケース８１の端面８１ｔａに設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３に隣接する端面８１ｔｂに設けられる。
【０１５６】
図３２において、入力コネクタ４１と共振器Ｒ１ｙとを容量結合に代えて誘導結合させ、上記出力コネクタ４２と共振器Ｒ３ｘとを容量結合に代えて誘導結合させると、上記入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸上の上記導体ケース８１の側面８１ｓａに設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸上の上記ケース８１ｓｂの側面に設けられる。
【０１５７】
図３２において、入力コネクタ４１と共振器Ｒ１ｙとを容量結合に代えて誘導結合させると、上記入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸上の上記導体ケース８１の側面８１ｓａに設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３に隣接する端面８１ｔｂに設けられる。
【０１５８】
図３２において、上記出力コネクタ４２と共振器Ｒ３ｘとを容量結合に代えて誘導結合させると、上記入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１に隣接する上記ケース８１の端面８１ｔａに設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸上の上記導体ケース８１の側面８１ｓｂに設けられる。
【０１５９】
図３３は、第３の変形例の高周波帯域通過フィルタ装置の動作説明図である。図３３において、図３２と異なる部分は、上記共振器Ｒ１ｘを上記共振器Ｒ２ｘに容量結合させている点と、上記共振器Ｒ２ｙを上記共振器Ｒ３ｙに容量結合させている点である。以上のように入力コネクタ４１と出力コネクタ４２の間に上記各共振器を結合させたとき、上記ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３はＬ字型に設けられる。そして、入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸上の上記導体ケース８１の側面８１ｓａに設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸上の上記導体ケース８１の側面８１ｓｂに設けられる。
【０１６０】
図３３において、入力コネクタ４１と共振器Ｒ１ｙとを容量結合に代えて誘導結合させ、上記出力コネクタ４２と共振器Ｒ３ｘとを容量結合に代えて誘導結合させると、入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１に隣接する上記導体ケース８１の端面８１ｔａに設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３に隣接する端面８１ｔｂに設けられる。
【０１６１】
図３３において、入力コネクタ４１と共振器Ｒ１ｙとを容量結合に代えて誘導結合させると、入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１に隣接する上記導体ケース８１の端面８１ｔａに設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３のｘ軸上の上記導体ケース８１の側面８１ｓｂに設けられる。
【０１６２】
図３３において、上記出力コネクタ４２と共振器Ｒ３ｘとを容量結合に代えて誘導結合させると、入力コネクタ４１は、上記ＴＭ２重モード誘電体共振器Ｒ１のｙ軸上の上記導体ケース８１の側面８１ｓａに設けられ、上記出力コネクタ４２は、上記ＴＭ２重モード誘電体共振器Ｒ３に隣接する端面８１ｔｂに設けられる。
【０１６３】
以上詳述したように、本発明に係る高周波帯域通過フィルタ装置は、上記ＴＭ２重モード誘電体共振器Ｒ１と上記ＴＭ２重モード誘電体共振器Ｒ２との間の結合方法と、上記ＴＭ２重モード誘電体共振器Ｒ２と上記ＴＭ２重モード誘電体共振器Ｒ３との間の結合方法とをともに容量結合させることにより、又は上記ＴＭ２重モード誘電体共振器Ｒ１と上記ＴＭ２重モード誘電体共振器Ｒ２との間の結合方法と、上記ＴＭ２重モード誘電体共振器Ｒ２と上記ＴＭ２重モード誘電体共振器Ｒ３との間の結合方法とをともに誘導結合させることにより、上記３つのＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３をＬ字型にして構成することができる。すなわち、各ＴＭ２重モード誘電体共振器をＬ字型にして構成する高周波帯域通過フィルタ装置において、隣接する各ＴＭ２重モード誘電体共振器のｘ軸又はｙ軸共振器は、誘導結合と容量結合を用いて隣接する結合方法を互いに同一にしている。
【０１６４】
また、本発明に係る高周波帯域通過フィルタ装置は、上記入力コネクタ４１と上記ＴＭ２重モード誘電体共振器Ｒ１との間の結合方法と、上記ＴＭ２重モード誘電体共振器Ｒ１と上記ＴＭ２重モード誘電体共振器Ｒ２との間の結合方法のうちの一方の結合方法を容量結合として、他方の結合方法を誘導結合にすることにより、上記入力コネクタ４１を上記導体ケース８０の端面に設けることができる。同様に、上記出力コネクタ４２と上記ＴＭ２重モード誘電体共振器Ｒ３との間の結合方法と、上記ＴＭ２重モード誘電体共振器Ｒ２と上記ＴＭ２重モード誘電体共振器Ｒ３との間の結合方法のうちの一方を容量結合として、他方を誘導結合にすることにより、上記出力コネクタ４２を上記導体ケース８０の端面に設けることができる。また、上記入力コネクタ４１と上記ＴＭ２重モード誘電体共振器Ｒ１との間の結合方法と、上記ＴＭ２重モード誘電体共振器Ｒ１と上記ＴＭ２重モード誘電体共振器Ｒ２との間の結合方法とをともに容量結合又は誘導結合とすることにより、上記入力コネクタ４１を上記導体ケース８０の側面に設けることができる。同様に、上記出力コネクタ４２と上記ＴＭ２重モード誘電体共振器Ｒ３との間の結合方法と、上記ＴＭ２重モード誘電体共振器Ｒ２と上記ＴＭ２重モード誘電体共振器Ｒ３との間の結合方法をともに容量結合又は誘導結合にすることにより、上記出力コネクタ４２を上記導体ケース８０の側面に設けることができる。
【０１６５】
以上詳述したように、本発明に係るＴＭ２重モード誘電体共振器は、上記ＴＭ２重モード誘電体共振器Ｒ１と上記ＴＭ２重モード誘電体共振器Ｒ２との間の結合方法と、上記ＴＭ２重モード誘電体共振器Ｒ２と上記ＴＭ２重モード誘電体共振器Ｒ３との間の結合方法を、容量結合と誘導結合とを各種組み合わせて、上記高周波帯域通過フィルタ装置を構成することにより、導体ケース８０，８１の形状を直方体又はＬ字型等の各種形状で構成することができるとともに、上記形状において入力コネクタ４１と出力コネクタ４２を、端面に設けるようにもでき、側面に設けるようにもできる。
【０１６６】
＜他の変形例＞
以上の第１乃至第３の実施例及び第１乃至第５の変形例においては、誘電体１，２，１０１，１０２を円柱形状又は端面が正方形の柱形状に形成したが、本発明はこれに限らず、端面の断面形状が楕円又は辺の数が偶数の正多角形である柱形状に形成してもよい。以上のように構成しても第１乃至第３の実施例及び第１乃至第５の変形例と同様な動作をし同様な効果を有する。
【０１６７】
以上の第１乃至第３の実施例及び第１乃至第５の変形例は、導体ケース８，８０，１０８によって形成されるキャビティー８ａ，８０ａ，８０ｂ，８０ｃ，１０８ａを用いて構成したが、本発明はこれに限らず、上記誘電体１，１ａ，１ｂ，１ｃ，１０１の上端面に接する導体板と、上記誘電体２，２ａ，２ｂ，２ｃ，１０２の下端面に接する導体板とを用いて、上記各導体板の縁端部が上記誘電体１，２，１ａ，２ａ，１ｂ，２ｂ，１ｃ，２ｃ，１０１，１０２の外周から所定の距離だけ離れるように形成することにより構成してもよい。以上のように構成することにより、上記２つの導体板の間の自由空間は、電磁界エネルギーが減衰して電磁波が伝搬しない減衰領域として作用するので、第１乃至第３の実施例及び第１乃至第５の変形例と同様な動作をし同様な効果を有する。
【０１６８】
以上の第１乃至第３の実施例及び第１乃至第５の変形例においては、上記誘電体１，２の側面に上記切り欠きを形成することにより縮退を分離したが、本発明はこれに限らず、上記誘電体の側面に誘電率の異なる部分を設けても縮退を分離してもよいし、また上記切り欠きに代えて上記誘電体１，２の側面に凸部を設けてもよい。以上のように構成しても第１乃至第３の実施例及び第１乃至第５の変形例と同様な動作をし同様な効果を有する。
【０１６９】
以上の第１乃至第３の実施例及び第１乃至第５の変形例において、上記ＴＭ２重モード誘電体共振器の共振モードは、基本モードでありかつ２重に縮退したＴＭ_１１０モードを用いることが好ましいが、本発明はこれに限らず、ＴＭ_２１０モードやＴＭ_３１０モード等の他の２重縮退モードを用いもてもよい。以上のように構成しても第３の実施例及び第２乃至第５の変形例と同様な動作をし同様な効果を有する。
【０１７０】
以上の第３の実施例と第２乃至第５の変形例の高周波帯域通過フィルタ装置では、３つの上記ＴＭ２重モード誘電体共振器Ｒ１，Ｒ２，Ｒ３を用いて構成したが、本発明はこれに限らず、少なくとも１つの上記ＴＭ２重モード誘電体共振器を用いて構成してもよい。以上のように構成しても第３の実施例及び第２乃至第５の変形例と同様な動作をし同様な効果を有する。この場合においても、入力コネクタ４１とＴＭ２重モード誘電体共振器の結合及び出力コネクタ４２とＴＭ２重モード誘電体共振器の結合とを、容量結合と誘導結合とを各種組み合わせて構成することにより、上記入力コネクタ４１と上記出力コネクタ４２を設ける側面を自由に選ぶことができる。
【０１７１】
以上の第１乃至第３の実施例と第１乃至第５の変形例のＴＭ２重モード誘電体共振器及び高周波帯域通過フィルタ装置では、トーラス電極４，４ａ，４ｂ，４ｃ，１０４を、それぞれ平板電極３，３ａ，３ｂ，３ｃ，１０３に一体的に連結して形成したが、本発明はこれに限らず、トーラス電極４，４ａ，４ｂ，４ｃ，１０４が連結されていない平板電極３，３ａ，３ｂ，３ｃ，１０３を用いて構成してもよい。また、トーラス電極４，４ａ，４ｂ，４ｃ，１０４が連結された平板電極３，３ａ，３ｂ，３ｃ，１０３に代えて、図３４（ａ）に示すように、外周部の縦断面を半円形状に形成した平板電極３ｄを、平板電極３ｄのうちの半円形状に形成された部分３ｄａが、誘電体１，２の外周面から突出するように設けて構成してもよい。また、トーラス電極４，４ａ，４ｂ，４ｃ，１０４が連結された平板電極３，３ａ，３ｂ，３ｃ，１０３に代えて、図３４（ｂ）に示すように、外周部の縦断面を台形形状に形成した平板電極３ｅを、平板電極３ｅのうちの台形形状に形成された部分３ｅａが、誘電体１，２の外周面から突出するように設けて構成してもよい。以上のように構成しても、平板電極３ｄに形成された部分３ｄａ、又は平板電極３ｅに形成された部分３ｅａに電流を分散させることができるので、第１乃至第３の実施例及び第１乃至第５の変形例と同様な動作をし同様な効果を有する。
【０１７２】
以上の第１乃至第３の実施例と第１乃至第５の変形例のＴＭ２重モード誘電体共振器及び高周波帯域通過フィルタ装置では、縦断面が円形のトーラス電極４，４ａ，４ｂ，４ｃ，１０４を用いて構成したが、本発明はこれに限らず、図３４（ｃ）に示すように、縦断面が８角形状に形成された電極４ｄを用いて構成してもよいし、縦断面が８角形状以外の多角形に形成された電極を用いて構成してもよい。以上のように構成しても、当該電極４ｄの外周部に電流を分散させることができるので、第１乃至第３の実施例及び第１乃至第５の変形例と同様な動作をし同様な効果を有する。
【０１７３】
以上の第１乃至第３の実施例と第１乃至第５の変形例のＴＭ２重モード誘電体共振器及び高周波帯域通過フィルタ装置では、平板電極３，３ａ，３ｂ，３ｃ，１０３の直径を誘電体１，２の直径と同一になるように設定して、それぞれ平板電極３，３ａ，３ｂ，３ｃ，１０３に連結されたトーラス電極４，４ａ，４ｂ，４ｃ，１０４の内周が、誘電体１，２の外周面に接するように構成した。しかしながら、本発明はこれに限らず、図３５（ａ）に示すように、平板電極３の直径とトーラス電極４の縦断面の直径の２倍とを加えたトーラス電極４の外周の直径が、誘電体１，２の直径と同一になるように設定して、トーラス電極４の外周が誘電体１，２の外周面と一致するように構成してもよい。また、図３５（ｂ）に示すように、トーラス電極４の縦断面の外側の半円部分だけが誘電体１，２の外周面から突出するように構成してもよい。また、図３５（ｃ）に示すように、平板電極３の直径を誘電体１，２の直径より大きく設定して、トーラス電極４の内周が誘電体１，２の外周面から離れるように構成してもよい。以上のように構成しても、第１乃至第３の実施例及び第１乃至第５の変形例と同様な動作をし同様な効果を有する。
【０１７４】
【発明の効果】
本発明に係る請求項１記載のＴＭ２重モード誘電体共振器は、上記第１と第２の誘電体によって挟設された上記平板電極と、上記第１の誘電体の端面に形成された上記第１の導体板と、上記第２の誘電体の端面に形成された上記第２の導体板とを備えているので、比較的高い無負荷Ｑを有し、従来例に比較して小型でかつ薄型にできる。
【０１７５】
請求項２記載のＴＭ２重モード誘電体共振器は、第１と第２の誘電体は円柱形状を有するので、他の形状を有する誘電体を用いた場合に比較して、容易に構成できる。
【０１７６】
請求項３記載のＴＭ２重モード誘電体共振器は、上記平板電極の外周の全周に連結される上記トーラス電極をさらに備えているので、上記平板電極の縁端部で失われるエネルギーを減少させることができ、上記トーラス電極を備えない場合に比較して無負荷Ｑを高くすることができる。
【０１７７】
請求項４記載のＴＭ２重モード誘電体共振器は、縮退分離手段を備えているので、２重に縮退したモードを互いに異なる共振周波数を有する２つのモードに分離することができる。
【０１７８】
請求項５記載のＴＭ２重モード誘電体共振器は、上記第１と第２の誘電体の外周の一部に切り欠きを形成しているので、上記ＴＭ２重モード誘電体共振器における２重に縮退したモードを互いに異なる共振周波数を有する２つのモードに分離することができる。
【０１７９】
請求項６記載のＴＭ２重モード誘電体共振器は、上記キャビティーを備えているので、キャビティーを備えない場合に比較して、無負荷Ｑを高くすることができるとともに、共振周波数の変動を少なくできる。
【０１８０】
本発明に係る請求項７記載の高周波帯域通過フィルタ装置は、上記請求項１乃至６記載のＴＭ２重モード誘電体共振器を備えているので、小型でかつ薄型にできる。
【０１８１】
請求項８記載の高周波帯域通過フィルタ装置は、上記請求項１乃至６記載の少なくとも２つのＴＭ２重モード誘電体共振器と上記結合手段を備えているので、小型でかつ薄型にできるとともに、阻止帯域における減衰量を比較的大きくできる。
【０１８２】
請求項９記載の高周波帯域通過フィルタ装置は、誘導結合させる結合手段を備えているので、互いに隣接する上記各２つのＴＭ２重モード誘電体共振器を容易に結合させることができる。
【０１８３】
請求項１０記載の高周波帯域通過フィルタ装置は、容量結合させる上記結合手段を備えているので、互いに隣接する上記各２つのＴＭ２重モード誘電体共振器を容易に結合させることができる。
【図面の簡単な説明】
【図１】本発明に係る第１の実施例のＴＭ２重モード誘電体共振器の一部破断斜視図である。
【図２】図１のＴＭ２重モード誘電体共振器のＡ−Ａ’における断面図である。
【図３】図１のＴＭ２重モード誘電体共振器の平板電極３の上面における電流分布を示す平面図である。
【図４】図１のＴＭ２重モード誘電体共振器の図３に示すｘ軸に沿った縦断面における電界分布を示す断面図である。
【図５】図１のＴＭ２重モード誘電体共振器の図３に示すｙ軸に沿った縦断面における磁界分布を示す断面図である。
【図６】図１のＴＭ２重モード誘電体共振器におけるトーラス電極４の半径Ｒの値と無負荷Ｑ及び共振周波数の関係を示したグラフである。
【図７】図１のＴＭ２重モード誘電体共振器において、上記誘電体１，２の誘電率ε_ｒが２０，４０，６０のそれぞれの場合について、誘電体１，２の直径ｄと共振周波数ｆ_０の関係を示したグラフである。
【図８】図１のＴＭ２重モード誘電体共振器における上記誘電体１，２の厚さｈと無負荷Ｑ及び共振周波数ｆ_０の関係を示したグラフである。
【図９】図１のＴＭ２重モード誘電体共振器における上記誘電体１，２の比誘電率ε_ｒと無負荷Ｑ及び比誘電率ε_ｒとＴＭ２重モード誘電体共振器の体積の関係を示したグラフである。
【図１０】図１０は、本発明に係る第２の実施例のＴＭ２重モード誘電体共振器の横断面図である。
【図１１】（ａ）は、図１０のＴＭ２重モード誘電体共振器の平板電極３の表面における奇モードの電流分布を示す平面図であり、（ｂ）は、図１０のＴＭ２重モード誘電体共振器の平板電極３の表面における偶モードの電流分布を示す平面図である。
【図１２】図１０のＴＭ２重モード誘電体共振器の等価回路を示す回路図である。
【図１３】図１０のＴＭ２重モード誘電体共振器の負性容量の割合と結合係数を示すグラフである。
【図１４】図１の第１の実施例のＴＭ２重モード誘電体共振器の周波数０．５ＧＨｚから周波数２．５ＧＨｚにおける出力端伝送係数Ｓ_２１の測定値を示すグラフである。
【図１５】本発明に係る第３の実施例の高周波帯域通過フィルタ装置の縦断面図である。
【図１６】図１５の高周波帯域通過フィルタ装置のＣ−Ｃ’における横断面図である。
【図１７】図１５の高周波帯域通過フィルタ装置の等価回路を示す回路図である。
【図１８】図１７の等価回路に基づいて計算した図１５の高周波帯域通過フィルタ装置の出力端伝送係数Ｓ_２１と入力端反射係数Ｓ_１１のシミュレーション結果を示すグラフである。
【図１９】図１５の高周波帯域通過フィルタ装置の出力端伝送係数Ｓ_１２と入力端反射係数Ｓ_１１の測定値を示すグラフである。
【図２０】（ａ）は、本発明に係る第１の変形例のＴＭ２重モード誘電体共振器の縦断面図であり、（ｂ）は、（ａ）のＴＭ２重モード誘電体共振器の横断面図である。
【図２１】本発明に係る第２の変形例の高周波帯域通過フィルタ装置の等価回路を示す回路図である。
【図２２】本発明に係る第３の変形例の高周波帯域通過フィルタ装置の等価回路を示す回路図である。
【図２３】本発明に係る第４の変形例の高周波帯域通過フィルタ装置の等価回路を示す回路図である。
【図２４】本発明に係る第５の変形例の高周波帯域通過フィルタ装置の等価回路を示す回路図である。
【図２５】図１０のＢ−Ｂ’線についての第２の実施例のＴＭ２重モード誘電体共振器の縦断面図である。
【図２６】（ａ）は、第２の変形例の高周波帯域通過フィルタ装置の概略平面図であり、（ｂ）は、第３の変形例の高周波帯域通過フィルタ装置の概略平面図である。
【図２７】（ａ）は、第４の変形例の高周波帯域通過フィルタ装置の概略平面図であり、（ｂ）は、第５の変形例の高周波帯域通過フィルタ装置の概略平面図である。
【図２８】第３の実施例の高周波帯域通過フィルタ装置の動作説明図である。
【図２９】図２８に示した結合方法とは異なる結合方法を用いて構成した高周波帯域通過フィルタ装置の動作説明図である。
【図３０】第２の変形例の高周波帯域通過フィルタ装置の動作説明図である。
【図３１】図３０に示した結合方法とは異なる結合方法を用いて構成した高周波帯域通過フィルタ装置の動作説明図である。
【図３２】図３０，図３１に示した結合方法とは異なる結合方法を用いて構成した高周波帯域通過フィルタ装置の動作説明図である。
【図３３】第３の変形例の高周波帯域通過フィルタ装置の動作説明図である。
【図３４】（ａ）は、平板電極３，３ａ，３ｂ，３ｃ，１０３とは異なる平板電極３ｄを用いて構成したＴＭ２重モード誘電体共振器の部分縦断面図であり、（ｂ）は、（ａ）とは異なる平板電極３ｅを用いて構成したＴＭ２重モード誘電体共振器の部分縦断面図であり、（ｃ）は、トーラス電極４に代えて縦断面が８角形の電極４ｄを用いて構成したＴＭ２重モード誘電体共振器の部分縦断面図である。
【図３５】（ａ）は、トーラス電極４の設ける位置を変えたときのＴＭ２重モード誘電体共振器の部分縦断面図であり、（ｂ）は、さらにトーラス電極４の設ける位置を変えたときのＴＭ２重モード誘電体共振器の部分縦断面図であり、（ｃ）は、またさらにトーラス電極４の設ける位置を変えたときのＴＭ２重モード誘電体共振器の部分縦断面図である。
【符号の説明】
Ｒ１，Ｒ２，Ｒ３…ＴＭ２重モード誘電体共振器、
１，１ａ，１ｂ，１ｃ，２，２ａ，２ｂ，２ｃ，１０１，１０２…誘電体、
３，３ａ，３ｂ，３ｃ，３ｄ，３ｅ，１０３…平板電極、
４，４ａ，４ｂ，４ｃ，１０４…トーラス電極、
４ｄ…電極、
６，６ａ，６ｂ，６ｃ，７，７ａ，７ｂ，７ｃ，１０６，１０７…端面電極、
８，８０，８１，１０８…導体ケース、
８ａ，８０ａ，８０ｂ，８０ｃ，１０８ａ…キャビティー、
９ａ，９ｂ…導体板、
１１，１１ａ，１１ｂ，１１ｃ，１２，１２ａ，１２ｂ，１２ｃ，１３，１３ａ，１３ｂ，１３ｃ，１４，１４ａ，１４ｂ，１４ｃ…切り欠き、
２１，２１ａ…結合ループインダクタ、
２２…入力キャパシタ、
２３，２３ａ，２４，２４ａ…結合キャパシタ、
２５…出力キャパシタ、
２６ａ，２６ｂ，２６ｃ…結合調整ねじ、
３１…入力ループインダクタ、
４１…入力コネクタ、
４２…出力コネクタ、
４３…出力ループインダクタ、
５１，５２，５３，５４…周波数調整ねじ、
２１１，２１２，２１３，２１４，２１５，２３４…導体線。[0001]
[Industrial applications]
The present invention relates to a TM dual-mode dielectric resonator used in microwaves and a high-frequency band-pass filter device.
[0002]
[Prior art]
In recent years, devices used for mobile communication have been required to be smaller and have higher performance with the development of systems such as digitization of communication systems and microcells. Microwave filters and antenna duplexers used in wireless base stations are also required to be further reduced in size, thickness, and loss.
[0003]
The present inventors have previously disclosed a TM dual-mode dielectric in which a cross-shaped dielectric integrally formed with the above-described dielectric is provided at the center of a square cylindrical dielectric whose outer surface is metallized. A body resonator has been proposed in Japanese Patent Application No. 62-150021. The conventional TM dual-mode dielectric filter configured using the above-described conventional TM dual-mode dielectric resonator is one of the technologies capable of withstanding the high power of the radio base station and realizing a small and low-loss property. The present inventors are already at the stage of practical use.
[0004]
[Problems to be solved by the invention]
However, in the conventional TM dual-mode dielectric resonator, the dimensions of the resonator are uniquely determined according to the resonance frequency, and thus the unloaded Q is uniquely determined. There is a problem that it cannot be set higher than the determined no-load Q. Further, since the dimensions of the resonator are uniquely determined according to the resonance frequency, the thickness of the resonator cannot be set smaller than the thickness determined above, and it is difficult to reduce the thickness. There was a point.
[0005]
A first object of the present invention is to provide a TM dual-mode dielectric resonator that can be made smaller and thinner as compared with the conventional example and that can have a relatively high no-load Q.
[0006]
A second object of the present invention is to provide a high-frequency bandpass filter that can be made smaller and thinner than the conventional example, has relatively low loss in the passband, and has relatively large attenuation in the stopband. It is in.
[0007]
[Means for Solving the Problems]
The TM dual mode dielectric resonator according to claim 1 of the present invention comprises a first and a second dielectric having predetermined columnar shapes each having two end faces facing in parallel with each other;
Sandwiched between one end face of the first dielectric and one end face of the second dielectric;, Electromagnetically coupled to input or output terminalsA plate electrode,
The edge of the first conductor plate is separated from the outer periphery of the other surface of the first dielectric by a predetermined distance, and a part of the first conductor plate is formed on the other surface of the first dielectric. A first conductor plate formed so as to contact the entire surface of
The edge of the second conductor plate is separated from the outer periphery of the other surface of the second dielectric by a predetermined distance, and a part of the second conductor plate is placed on the other surface of the second dielectric. And a second conductor plate formed so as to be in contact with the entire surface of the substrate.
[0008]
A TM dual mode dielectric resonator according to a second aspect is characterized in that in the TM dual mode dielectric resonator according to the first aspect, the first and second dielectrics have a cylindrical shape.
[0009]
According to a third aspect of the present invention, there is provided a TM dual mode dielectric resonator according to the first or second aspect, further formed so as to be connected to the entire outer periphery of the flat plate electrode. An electrode for dispersing a current concentrated on the outer peripheral portion of the plate electrode when exciting the heavy mode dielectric resonator is provided.
[0010]
A TM dual mode dielectric resonator according to claim 4 is the TM dual mode dielectric resonator according to one of claims 1 to 3, further comprising a part of an outer periphery of the first and second dielectrics. Degenerate separation for separating a double degenerate mode in the TM dual mode dielectric resonator into two modes having different resonance frequencies from each other by making the permittivity of the above-described portion different from the permittivity of the other portion. It is characterized by further comprising means.
[0011]
A TM dual mode dielectric resonator according to a fifth aspect is the TM dual mode dielectric resonator according to the fourth aspect, wherein the degenerate separation means is formed on a part of the outer periphery of the first and second dielectrics. It is characterized by being a cutout.
[0012]
A TM dual-mode dielectric resonator according to claim 6 is the TM dual-mode dielectric resonator according to one of claims 1 to 5, further comprising the first conductor plate and the second conductor plate. And a cavity for confining the electromagnetic field of the TM dual mode dielectric resonator in the cavity.
[0013]
According to a seventh aspect of the present invention, there is provided a high-frequency band-pass filter device comprising: a TM dual-mode dielectric resonator according to the first to sixth aspects;
An input terminal for inputting a high-frequency signal to the TM dual-mode dielectric resonator,
An output terminal for outputting a high-frequency signal output from the TM dual-mode dielectric resonator., The input terminal and the output terminal are electromagnetically coupled to the plate electrodeIt is characterized by the following.
[0014]
The high-frequency band-pass filter device according to claim 8 includes at least two TM dual-mode dielectric resonators according to claim 1,
Electromagnetically coupled to the plate electrode,Coupling means for coupling each of the two TM dual-mode dielectric resonators adjacent to each other;
Electromagnetically coupled to the plate electrodeAn input terminal for inputting a high-frequency signal to the TM dual-mode dielectric resonator,
Electromagnetically coupled to the plate electrodeAn output terminal for outputting a high-frequency signal output from the TM dual-mode dielectric resonator.
[0015]
The high frequency band pass filter device according to the ninth aspect is the high frequency band pass filter device according to the eighth aspect, wherein at least one of the coupling means includes an inductive coupling.Therefore, it is connected to the above-mentioned plate electrode.It is characterized by the following.
The high frequency band pass filter device according to claim 10 is the high frequency band pass filter device according to claim 8, wherein at least one of the coupling means is capacitively coupled.Therefore, it is connected to the above-mentioned plate electrode.It is characterized by the following.
[0016]
[Action]
In the TM dual mode dielectric resonator according to claim 1 of the present invention, the plate electrode is sandwiched between one end face of the first dielectric and one end face of the second dielectric, and The other end face of each of the first and second dielectrics is in contact with the entire surface of the other end face and the edge of each of the first and second conductor plates from the outer periphery of the other end face. Are provided with a first conductor plate and a second conductor plate such that are separated by a predetermined distance. As a result, an attenuation region where electromagnetic field energy is attenuated and electromagnetic waves are not propagated is formed outside the side surfaces of the first and second dielectrics, so that the electromagnetic field of a high-frequency signal having a predetermined frequency is For example, as shown in FIGS. 4 and 5, the TM dual-mode dielectric resonator resonates in the first and second dielectrics and in the vicinity thereof.
[0017]
A TM dual mode dielectric resonator according to a second aspect is the TM dual mode dielectric resonator according to the first aspect, wherein the first and second dielectrics have a cylindrical shape.
[0018]
According to a third aspect of the present invention, there is provided a TM dual mode dielectric resonator according to the first or second aspect, further comprising a torus electrode connected to the entire outer periphery of the flat plate electrode. As a result, the current concentrated at the edge of the flat electrode is distributed and distributed around the periphery of the cross section of the torus electrode, reducing the energy lost at the edge of the flat electrode, The unloaded Q of the TM dual mode dielectric resonator is improved.
[0019]
A TM dual mode dielectric resonator according to a fourth aspect is the TM dual mode dielectric resonator according to any one of the first to third aspects, further comprising degenerate separation means. As a result, the double degenerated mode in the TM dual mode dielectric resonator is separated into two modes having different resonance frequencies.
[0020]
A TM dual mode dielectric resonator according to a fifth aspect is the TM dual mode dielectric resonator according to the fourth aspect, wherein a notch is formed in a part of the outer periphery of the first and second dielectrics. . Thereby, the dielectric constant of a part of the outer periphery of the first and second dielectrics is made different from the dielectric constant of the part other than the part, so that the double degenerate modes in the TM dual mode dielectric resonator are mutually Separation into two modes having different resonance frequencies.
[0021]
A TM dual-mode dielectric resonator according to claim 6 is the TM dual-mode dielectric resonator according to one of claims 1 to 5, further comprising the first conductor plate and the second conductor plate. And a cavity comprising: As a result, the electromagnetic field of the TM dual mode dielectric resonator is confined in the cavity, the unloaded Q of the TM dual mode dielectric resonator can be improved, and the fluctuation of the resonance frequency can be reduced.
[0022]
A high-frequency band-pass filter device according to a seventh aspect of the present invention includes the TM dual-mode dielectric resonator according to the first to sixth aspects, the input terminal, and the output terminal. Accordingly, the high-frequency signal having a predetermined frequency input from the input terminal passes through the TM dual-mode dielectric resonator and is output from the output terminal.
[0023]
A high-frequency band-pass filter device according to claim 8 includes at least two TM dual-mode dielectric resonators according to claim 1, the coupling unit, the input terminal, and the output terminal, and the coupling unit includes: , Two adjacent TM dual-mode dielectric resonators are coupled to each other. Accordingly, the high-frequency signal having a predetermined frequency input from the input terminal passes through each of the TM dual-mode dielectric resonators and is output from the output terminal.
[0024]
10. The high-frequency band-pass filter device according to claim 9, wherein said coupling means inductively couples two adjacent TM dual-mode dielectric resonators to each other.
11. The high-frequency band-pass filter device according to claim 10, wherein said coupling means capacitively couples two adjacent TM dual-mode dielectric resonators to each other.
[0025]
【Example】
Hereinafter, embodiments of the present invention will be described with reference to the drawings. In the accompanying drawings, the same components are denoted by the same reference numerals.
[0026]
<First embodiment>
FIG. 1 is a partially broken perspective view of a TM dual mode dielectric resonator according to a first embodiment of the present invention. FIG. 2 is a cross-sectional view of the TM dual mode dielectric resonator of FIG. 1 taken along the line A-A 'of FIG.
[0027]
The TM dual mode dielectric resonator of the first embodiment includes a dielectric 1, a dielectric 2, a plate electrode 3, a torus electrode 4, and a case 8, and the dielectric 1 and the dielectric 2 The flat electrode 3 is interposed therebetween, and the torus electrode 4 is connected to and integrated with the entire circumference of the flat electrode 3.
[0028]
In the TM dual mode dielectric resonator according to the first embodiment, as shown in FIGS. 1 and 2, a circular end face electrode 6 is first formed on the upper end face, and the diameter d and the axial length h are determined. And a cylindrical dielectric 2 having a circular end face electrode 7 formed on the lower end surface and having the same diameter d and axial length h as the dielectric 1. A circular plate electrode 3 having a diameter t and the same diameter d as the dielectrics 1 and 2 is coaxially sandwiched therebetween. Here, the plate electrode 3 is formed so that the upper surface thereof is in contact with the lower end surface of the dielectric 1 and the lower surface of the plate electrode 3 is in contact with the upper end surface of the dielectric 2. It is sandwiched between. A torus electrode 4 having a circular cross section having a diameter 2R larger than the thickness t of the flat plate electrode 3 is integrally formed and connected to the entire circumference of the flat plate electrode 3. You. Here, the torus electrode 4 is formed so that the inner periphery (inner surface) of the torus electrode 4 is in contact with the outer periphery of the flat plate electrode 3. Touch Further, the conductor case 8 which includes a conductor plate 9a and a conductor plate 9b opposed to each other and forms a columnar cavity 8a having a predetermined inner diameter D and a predetermined length K in the axial direction, The lower surface of the conductor plate 9a is in contact with the end surface electrode 6 formed on the dielectric 1 so as to be electrically conductive, and the upper surface of the conductor plate 9b is in contact with the end surface electrode 7 formed on the dielectric 2 It is provided so as to be electrically conductive. Here, the cavity 8a and the dielectrics 1 and 2 are provided so as to be coaxial, and the distance between the outer peripheral surface of the cavity 8a and the outer peripheral surfaces of the dielectrics 1 and 2 is a predetermined constant value. Is set to be The TM dual mode dielectric resonator according to the first embodiment is configured as described above.
[0029]
In the TM dual mode dielectric resonator configured as described above, the cavity 8a functions as an attenuation region in which electromagnetic field energy is attenuated and electromagnetic waves do not propagate, and the electromagnetic field of a high frequency signal having a predetermined frequency is It is distributed inside and near the dielectrics 1 and 2. Thus, when the TM dual-mode dielectric resonator is excited with a high-frequency signal, the TM dual-mode dielectric resonator resonates in various TM modes each having a unique resonance frequency. At this time, the side surfaces of the dielectrics 1 and 2 operate as magnetic walls that approximately satisfy the open condition.
[0030]
Next, the resonance mode of the TM dual mode dielectric resonator configured as described above will be described. The electromagnetic field distribution of the TM dual-mode dielectric resonator is basically the same as the electromagnetic field distribution of the balanced disk resonator. In the TM dual mode dielectric resonator, electromagnetic energy concentrates inside the dielectrics 1 and 2 because it is a free space having a dielectric constant lower than the dielectric constant. Here, the above-mentioned balanced disk resonator is formed by coaxially sandwiching a circular electrode having a diameter smaller than that of the dielectric substrate by two circular dielectric substrates having the same diameter and the same thickness. It is a disc resonator formed by forming a ground conductor on each surface of each dielectric substrate opposite to the surface in contact with the electrode.
[0031]
In the following description with reference to the drawings, the description will be made using orthogonal coordinates where the center point O on the axis of the plate electrode 3 is set as the origin and the z-axis in the axial direction of the plate electrode 3 as necessary. In this specification, a resonator having an electromagnetic field symmetric about the x-axis is called an x-axis resonator, and its eigenmode is called an x-mode. A resonator having a symmetric electromagnetic field distribution about the y-axis is called a y-axis resonator, and its eigenmode is called a y-mode. FIG. 3 shows that the TM dual mode dielectric resonator is a TM dual mode dielectric resonator.₁₁₀FIG. 9 is a plan view showing a current distribution of the x-axis resonator on the upper surface of the plate electrode 3 when resonating in a mode. As shown in FIG. 3, on the upper surface of the flat electrode 3, the current I is such that the outer circumference of the flat electrode 3 intersects the x-axis from an edge 31 which is one point where the outer circumference of the flat electrode 3 intersects the x-axis. It flows toward edge 32, which is the other point. Further, currents other than the current flowing through the center point O flow curving outward.
[0032]
FIG. 4 shows that the TM dual-mode dielectric resonator has the TM₁₁₀FIG. 4 is a cross-sectional view showing an electric field distribution of the x-axis resonator in a vertical cross section along the x-axis of the TM dual mode dielectric resonator when resonating in a mode. As shown in FIG. 4, in the cross section, in the region of the dielectric 1 where x is negative, an electric field E is distributed in the z-axis direction from the upper surface of the plate electrode 3 to the end surface electrode 6, and x is a negative dielectric. In the region of the body 2, an electric field E is distributed in the z-axis direction from the lower surface of the plate electrode 3 to the end surface electrode 7. In the region of the dielectric 1 where x is positive, an electric field E is distributed in the z-axis direction from the end face electrode 6 toward the upper surface of the flat plate electrode 3, and in the region of the dielectric 2 where x is positive, the end face electrode The electric field E is distributed in the z-axis direction from 7 to the lower surface of the plate electrode 3. Here, the length of the arrow indicating the electric field E in FIG. 4 represents the intensity of the electric field E, and the intensity of the electric field E becomes stronger as the electric field E is closer to the outer periphery of the dielectrics 1 and 2.
[0033]
FIG. 5 shows that the TM dual-mode dielectric resonator is TM₁₁₀It is sectional drawing which shows the magnetic field distribution of the said x-axis resonator in the longitudinal cross section along the y-axis of the TM dual mode dielectric resonator when it resonates in a mode. As shown in FIG. 5, the magnetic field H causes the torus electrode 4 to move from negative to positive in the dielectric 1 in the dielectric 1 and from positive to negative in the dielectric 2 in the dielectric 2. It is distributed so as to surround the connected plate electrodes 3. Here, the magnetic field H is distributed so as to be substantially parallel to the plate electrode 3.
[0034]
The y-axis resonator also has a similar electromagnetic field distribution, and the x mode and the y mode have the same resonance frequency and are degenerate.
[0035]
As shown in FIG. 5, the x-mode magnetic field makes a round around the x-axis, and the magnetic field intensity increases particularly at the edge of the plate electrode 3 on the y-axis. That is, since an edge effect occurs in which the current I is concentrated at the edge, the conductor loss increases at the edge. Similarly, the y-mode magnetic field makes a round around the y-axis, and the magnetic field intensity increases particularly at the edge of the plate electrode 3 on the x-axis. That is, since an edge effect occurs in which the current I is concentrated at the edge, the conductor loss increases at the edge. Therefore, in the TM dual mode resonator of the first embodiment, TM₁₁₀In order to reduce the loss due to the mode edge effect, a torus electrode 4 is provided on the outer periphery of the flat electrode 3, and current is dispersed on the outer periphery of the longitudinal section of the torus electrode 4 to reduce current concentration. Thereby, the conductor loss generated at the edge of the plate electrode 3 is reduced.
[0036]
FIG. 6 shows the radius R of the torus electrode 4, the no-load Q and the resonance frequency f of the TM dual mode dielectric resonator.₀5 is a graph showing the result of calculation using the two-dimensional finite element method for the relationship of.
Here, in the above calculation, each parameter was set as follows.
(1) dielectric constant ε of dielectrics 1 and 2_r= 40,
(2) the diameter d of the dielectrics 1 and 2 is 30 mm;
(3) The thickness h of the dielectrics 1 and 2 is 10 mm,
(4) The inner diameter D of the cavity 8a is 50 mm,
(5) The axial length K of the cavity 8a is K = 20 mm.
[0037]
As is clear from FIG. 6, as the radius R of the torus electrode 4 increases, the conductor loss at the edge of the plate electrode 3 decreases, and the no-load Q increases. However, when the radius R is larger than a certain value, the distance between the torus electrode 4 and the case 8 is reduced, and the conductor loss due to the case 8 is increased. Accordingly, it can be seen from FIG. 6 that there is an optimum radius R at which the no-load Q is maximum.
[0038]
FIG. 7 shows the TM dual mode dielectric resonator.₁₁₀Mode resonance frequency f₀And the relative dielectric constant ε_rAre graphs showing the cases of 20, 40, and 60, respectively. The resonance frequency f of FIG.₀And the diameter d of the dielectrics 1 and 2 were calculated using a two-dimensional finite element method.
Here, in the above calculation, each parameter was set as follows.
(1) The thickness h of the dielectrics 1 and 2 is 10 mm,
(2) The ratio d / D = 1.2 between the diameter d of the dielectrics 1 and 2 and the inner diameter D of the cavity 8a,
(3) The ratio R / D of the radius R of the torus electrode 4 to the inner diameter D of the cavity 8a is 0.03.
From the above results, the basic mode TM₁₁₀Mode resonance frequency f₀Is almost inversely proportional to the diameter d of the dielectrics 1 and 2, and the relative permittivity ε_rIt can be seen that it is almost inversely proportional to the square root of.
[0039]
FIG. 8 shows the thickness h, the unloaded Q, and the resonance frequency f of the dielectrics 1 and 2 of the TM dual mode dielectric resonator.₀3 is a graph showing the relationship of FIG.
Here, in the above calculation, each parameter was set as follows.
(1) dielectric constant ε of dielectrics 1 and 2_r= 40,
(2) the diameter d of the dielectrics 1 and 2 is 30 mm;
(3) The inner diameter D of the cavity 8a is 50 mm,
(4) The axial length K of the cavity 8a is K = 20 mm.
As is clear from FIG. 8, the no-load Q increases as the thickness h of the dielectrics 1 and 2 increases. On the other hand, the resonance frequency f₀Does not substantially depend on the thickness h of the dielectrics 1 and 2.
[0040]
FIG. 9 shows the relative permittivity ε of the dielectrics 1 and 2 of the TM dual mode dielectric resonator._r6 is a graph showing the relationship between the load Q, the no-load Q, and the volume of the cavity 8a of the dual mode dielectric resonator.
Here, in the above calculation, each parameter was set as follows.
(1) The thickness h of the dielectrics 1 and 2 is 10 mm,
(2) The ratio d / D = 1.2 between the diameter d of the dielectrics 1 and 2 and the inner diameter D of the cavity 8a,
(3) The axial length K of the cavity 8a is K = 20 mm,
(4) Resonance frequency f of TM dual mode dielectric resonator₀= 1000 MHz.
As is clear from FIG. 9, the relative permittivity ε_rBecomes larger, but the no-load Q hardly changes. That is, the no-load Q of the TM dual mode dielectric resonator is determined substantially by the thickness h of the dielectrics 1 and 2, and the dielectric constant ε_rIt does not depend.
[0041]
Next, the results of trial production and evaluation of the TM dual mode dielectric resonator of the first embodiment will be described.
[0042]
The parameters of the prototype TM dual mode dielectric resonator were set as shown in Table 1 below. Unloaded Q of the prototype TM dual mode dielectric resonator₀The measurement result of₀= 3100. On the other hand, the result calculated by the two-dimensional finite element method shows that the conductivity is σ = 4.97 × 10⁷The dielectric loss tangent of dielectrics 1 and 2 at S / m is tan δ = 2.2 × 10 at 1 GHz.^-5, The no-load Q of the resonator is Q₀= 3300. The measurement result was about 94% of the calculated value.
[0043]
Table 2 and FIG. 14 show the measured spurious mode resonance frequency and spurious characteristics, respectively. Table 2 also shows the calculation results of the resonance frequency of each mode by the two-dimensional finite element method. As shown in FIG. 14, the basic mode TM₁₁₀Following the mode, the higher order mode, TM₂₁₀, TM₀₁₀, TM₃₁₀The mode exists as a spurious mode. Basic mode TM₁₁₀TM adjacent to mode resonance frequency₂₁₀The resonance frequency of the mode is TM₁₁₀Mode resonance frequency f₀From the above TM₁₁₀Even if a bandpass filter is configured using the mode, the spurious resonance hardly affects the desired characteristics of the bandpass filter.
[0044]
[Table 1]

[0045]
Here, the dielectrics 1 and 2 are made of dielectric ceramics (Zr, Sn) TiO.₄The plate electrodes 3 and the end electrodes 6 and 7 were formed by baking silver paste on the upper and lower surfaces of the dielectrics 1 and 2, respectively.
[0046]
[Table 2]

[0047]
According to the TM dual mode dielectric resonator of the first embodiment described in detail above, when the unloaded Q of the TM dual mode dielectric resonator is set to a value between 3000 and 3500, the TM dual mode dielectric resonator The thickness of the dielectric resonator can be set to 23.5 mm, and the thickness of the dielectric resonator is set to a value between 7000 and 8000. Can be made thinner.
[0048]
According to the TM dual mode dielectric resonator of the first embodiment, the resonance frequency f₀Can be set to a desired no-load Q by setting the thickness h of the dielectrics 1 and 2 to a predetermined value without substantially changing.
[0049]
According to the TM dual-mode dielectric resonator of the first embodiment, the dielectrics 1 and 2 have a relative dielectric constant ε_rBy using a relatively large material having a relative dielectric constant of ε_rCan be reduced in the diameter direction without substantially deteriorating the no-load Q as compared with the case where a material using a relatively small material is used.
[0050]
<Second embodiment>
FIG. 10 is a transverse sectional view of the TM dual mode dielectric resonator of the second embodiment, and FIG. 25 is a longitudinal sectional view taken along line B-B 'in FIG. The TM dual-mode dielectric resonator according to the second embodiment differs from the TM dual-mode dielectric resonator according to the first embodiment in that notches 11, 12, 13, and 14 are provided on the outer circumferences of the dielectrics 1 and 2. It is characterized by having been provided.
[0051]
In the TM dual mode dielectric resonator according to the second embodiment, as shown in FIG. 10, opposing positions of the outer circumference of the dielectric 2 which are separated from the x axis by 45 degrees with respect to the axis of the dielectric 2 as a center. A notch 13 and a notch 14 are respectively formed by removing the dielectric in a semicircular groove shape from the upper end surface to the lower end surface of the dielectric 2. Further, as shown in FIG. 25, notches formed by removing the dielectric in a semicircular groove shape from the upper end surface to the lower end surface of the dielectric 1 at positions facing each other on the outer periphery of the dielectric 1. The notch 11 and the notch 12 are provided so as to be coaxial with the notches 13 and 14, respectively. Thus, the TM dual mode dielectric resonator of the second embodiment is configured to be asymmetric with respect to the x axis and the y axis.
[0052]
In the TM dual mode dielectric resonator according to the second embodiment configured as described above, the x-axis resonator and the y-axis resonator described in detail in the description of the first embodiment are provided on the dielectric 1. The cutouts 11 and 12 and the cutouts 13 and 14 provided in the dielectric 2 are coupled to each other to generate two independent modes having different resonance frequencies, an even mode and an odd mode. That is, the notches 11, 12, 13, and 14 separate the double degeneracy of the x mode and the y mode and generate two independent modes having different resonance frequencies, even mode and odd mode. It constitutes a degenerate separation means.
[0053]
FIG. 11A shows a current distribution on the upper surface of the plate electrode 3 in the odd mode. The electric current on the upper surface of the plate electrode 3 in the odd mode is changed from the edge 34 on the outer periphery of the plate electrode 3 facing the notch 12 to the edge on the outer periphery of the plate electrode 3 facing the notch 11. It is distributed in the direction toward the end 33. FIG. 11B shows a current distribution on the upper surface of the plate electrode 3 in the even mode. The current on the upper surface of the plate electrode 3 in the even mode is generated by the two opposite edges on the outer periphery separated from the edges 33 and 34 by 90 degrees about the center of the plate electrode 3 from each other. One of them is distributed in the direction from the edge 35 to the other edge 36.
[0054]
FIG. 12 is a circuit diagram showing an equivalent circuit of the TM dual mode dielectric resonator of the second embodiment. The equivalent circuit of FIG. 12 includes a rotationally symmetric ring distributed constant line formed by connecting distributed constant lines LN1 to LN8 in series in a ring shape. Here, each of the distributed constant lines LN1 to LN8 is set to have a length of ４ wavelength at the resonance frequency. Therefore, the ring distributed constant line has an electric length of 2π. A connection point between the distributed constant line LN1 and the distributed constant line LN2 is grounded via an internal coupling capacitor C3, and a connection point between the distributed constant line LN5 and the distributed constant line LN6 is grounded via an internal coupling capacitor C4. Be composed. Here, the internal coupling capacitors C3 and C4 are capacitors for coupling the x-axis resonator and the y-axis resonator, and correspond to the degenerate separation means.
[0055]
In the equivalent circuit of FIG. 12, input / output terminals T1 and T3 located on the x-axis respectively correspond to excitation points of the x-axis resonator, are input / output terminals of the x-axis resonator, respectively, and are located on the y-axis. Input / output terminals T2 and T4 correspond to the excitation point of the y-axis resonator, respectively, and are input / output terminals of the y-axis resonator, respectively. That is, the x-axis resonator is formed by connecting distributed constant lines LN1 to LN4 in series and connecting a distributed constant line having an electric length of π and distributed constant lines LN4 to LN8 in series and having an electric length of π. As a half-wave resonator connected in parallel with a distributed constant line. The y-axis resonator is formed by connecting distributed constant lines LN3, LN4, LN5 and LN6 in series, and connecting a distributed constant line having an electrical length of π and distributed constant lines LN7, LN8, LN1 and LN2 in series. And a half-wavelength resonator in which a distributed constant line having an electrical length of π is connected in parallel.
[0056]
Here, the capacitances of the internal coupling capacitors C3 and C4 in the equivalent circuit of FIG. 12 have negative capacitances equal to each other, and -ΔC Given by Where ω in Equation 1₀Is the resonance frequency f of the dual mode dielectric resonator.₀Which is given by Equation 2. The electrical length θ of each of the distributed constant lines LN1 to LN8 is given by the following equation (3). Further, k in Equations (1) and (3) is a coupling coefficient given when designing a high-frequency bandpass filter having a predetermined passband characteristic and a given stopband characteristic, and is given by the following Equation (4). F in Equations 2 and 4_oddAnd f_evenAre the resonance frequencies of the odd mode and the even mode, respectively.
[0057]
(Equation 1)
ΔC = (2Y_a/ Ω₀) · Tan {πk / (2-k)}
(Equation 2)
ω₀= 2πf₀= Π (f_even+ F_odd)
(Equation 3)
θ = (π / 4) · (1 + k / 2)
(Equation 4)
k = 2 (f_even−f_odd) / (F_even+ F_odd) = 1.418 (2ΔC / C₀)
[0058]
Where C in Equation 4₀Is the capacitance of the dielectric 1 sandwiched between the end face electrode 6 and the plate electrode 3 in the TM dual mode dielectric resonator. Further, since the dielectric 2 is formed in the same shape as the dielectric 1, the capacitance of the dielectric 2 sandwiched between the end face electrode 7 and the flat plate electrode 3 also becomes C.₀It is. Given that the TM dual mode dielectric resonator has the dielectrics 1 and 2, it is given by the following equation (5). Ε in Equation 5₀Is the dielectric constant in a vacuum, a is the radius of the dielectrics 1 and 2, and a = d / 2.
[0059]
(Equation 5)
C₀= (2 ε₀ε_r・ Πa²) / H
[0060]
The relationship between ΔC and k given by Equation 4 is shown by a solid line in the graph of FIG. In addition, the graph of FIG. 13 shows a result calculated by a two-dimensional finite element method using an ideal open model together with a measurement result. As is clear from FIG. 13, the relationship between ΔC and k given by Expression 4, the result calculated by the two-dimensional finite element method, and the measurement result are in good agreement. Here, in the TM dual mode dielectric resonator used for the above measurement, as described above, the low dielectric constant region on the outer periphery of the dielectrics 1 and 2 is formed by cutting a part of the dielectric to reduce the low dielectric constant bottom region. Formed.
[0061]
The electrical length θ given by Equation 3 is obtained by calculating the average value of the resonance frequencies of the even mode and the odd mode after coupling to the resonance frequency f before coupling.₀Is included in the correction coefficient (π / 4) × (k / 2). Also, the characteristic admittance Y of the line_aIs given by the following equation 6 using a susceptance slope b obtained by differentiating the susceptance of the TM dual mode dielectric resonator with respect to frequency.
[0062]
(Equation 6)
Y_a= B / π
[0063]
Here, the susceptance slope b of the TM dual-mode dielectric resonator is a perturbation using an analytical solution that satisfies the ideal open condition since the concentration of electric energy in the dielectrics 1 and 2 is as high as about 96%. By calculation, it is approximately given by the following equation 7. In addition, the perturbation calculation, the literature "Shuh-Han Chao," Measurements of microwave conductivity and dielectric constant by the cavity perturbation method and their errors ", IEEE Transaction on Microwave Theory Technology, vol.MTT-33, No.6, pp 519-526, 1985 ". Where J₀(•) is a 0th-order Bessel function of the first kind,₁(•) is a first-order Bessel function of the first kind. Also, k_cIs an eigenvalue determined from the boundary condition in the radial direction.
[0064]
(Equation 7)

[0065]
Next, the unloaded Q of the TM dual mode dielectric resonator is described.₀Is obtained by associating the unloaded Q of the TM dual-mode dielectric resonator with the transmission Q of the distributed constant lines LN1 to LN8, and using the attenuation constant α and the phase constant β of the distributed constant lines LN1 to LN8. Given by Equation 8.
[0066]
(Equation 8)
Q₀= Β / (2α)
[0067]
In the TM dual-mode dielectric resonator of the second embodiment configured as described above, since the notches 11, 12, 13, and 14 are formed on the side surfaces of the dielectrics 1 and 2, the x-axis resonance is performed. The device and the y-axis resonator can be coupled to generate two independent modes, an even mode and an odd mode.
[0068]
<Third embodiment>
FIG. 15 is a longitudinal sectional view of a high-frequency bandpass filter device according to a third embodiment of the present invention. FIG. 16 is a cross-sectional view of the high-frequency band-pass filter device according to the third embodiment, taken along line C-C 'in FIG.
[0069]
The high-frequency band-pass filter device according to the third embodiment includes three TM dual-mode dielectric resonators R1, R2, and R3 configured similarly to the TM dual-mode dielectric resonator according to the second embodiment. A coupling loop inductor 21 for inductively coupling the TM dual-mode dielectric resonator R1 and the TM dual-mode dielectric resonator R2, and the TM dual-mode dielectric resonator R2 and the TM double-mode dielectric resonator R3 And coupling capacitors 23 and 24 for capacitive coupling.
[0070]
Hereinafter, the configuration of the high-frequency bandpass filter device according to the third embodiment will be described in detail with reference to the drawings. In the high-frequency band-pass filter device of the third embodiment, as shown in FIGS. 15 and 16, the TM dual-mode dielectric resonator R1 includes a cylindrical dielectric 1a, a cylindrical dielectric 2a, It has a plate electrode 3a, a torus electrode 4a, and a cavity 80a formed in the conductor case 80, and is configured similarly to the TM dual mode dielectric resonator of the second embodiment. Here, the dielectric 1a includes an end face electrode 6a formed on an upper end face, and cutouts 11a and 12a formed at positions facing each other on the outer periphery, and the dielectric 2a is formed on a lower end face. An end face electrode 7a and cutouts 13a and 14a formed at opposing positions on the outer periphery are provided. In FIG. 16, reference numerals of the notches 13a and 14a are shown in parentheses above or below the reference numerals of the notches 11a and 12a, respectively.
[0071]
Similarly, the TM dual mode dielectric resonator R2 includes a dielectric 1b, a dielectric 2b, a plate electrode 3b, a torus electrode 4b, and a cavity 80b formed in the conductor case 80. Similarly, the TM dual-mode dielectric resonator R3 includes a dielectric 1c, a dielectric 2c, a plate electrode 3c, a torus electrode 4c, and a cavity 80c formed in the conductor case 80. Is done. The dielectric 1b includes an end face electrode 6b formed on the upper end face, and cutouts 11b and 12b formed at positions facing each other on the outer periphery, and the dielectric 2b is formed on the end face formed on the lower end face. It has an electrode 7b and notches 13b and 14b formed at opposing positions on the outer periphery. In FIG. 16, the symbols of the notches 13b and 14b are shown in parentheses above or below the symbols of the notches 11b and 12b, respectively. The dielectric 1c includes an end face electrode 6c formed on an upper end face, and cutouts 11c and 12c formed at positions facing each other on the outer periphery. The dielectric 2c includes an end face electrode 7c formed on a lower end face. And notches 13c and 14c formed at positions facing each other on the outer periphery. In FIG. 16, the symbols of the notches 13c and 14c are shown in parentheses above or below the symbols of the notches 11c and 12c, respectively. Here, the cavities 80a, 80b, 80c are formed in the conductor case 80 side by side at a predetermined interval. Therefore, the TM dual-mode dielectric resonators R1, R2, and R3 are provided side by side at a predetermined interval between both end surfaces of the conductor case 80.
[0072]
An input connector 41 for connecting the high-frequency band-pass filter device to an external circuit is provided at the center of one end face of the conductor case 80 in the longitudinal direction. A center conductor of the input connector 41 is electrically insulated from the end face and penetrates the end face to be connected to one electrode of the input capacitor 22. The other electrode of the input capacitor 22 is connected to the outer periphery (outer surface) on the x-axis of the torus electrode 4a, which is 45 degrees away from the notch 11a. As described above, the input connector 41 is capacitively coupled to the x-axis resonator of the TM dual-mode dielectric resonator R1 via the input capacitor 22. Here, the input capacitor 22 is formed by forming electrodes on both end surfaces of a cylindrical ceramic dielectric.
[0073]
The coupling loop inductor 21 includes five conductor wires 211, 212, 213, 214, and 215, and is connected in series so that the conductor wires 211, 212, 213, and 214 form a substantially rectangular loop. Both ends of the conductor wire 215 are connected to substantially the center points of the conductor wires 212 and 214, respectively. The coupling loop inductor 21 is positioned substantially on the same plane as the plate electrodes 3a and 3b, and the conductor wires 211, 213, and 215 are formed of the cavities 80a and 215. The conductor case 80 is provided so as to penetrate the center of the conductor wall of the conductor case 80 separating the conductor wall 80b in the longitudinal direction. Here, the conductor wires 211 and 213 penetrate the conductor wall while being electrically insulated from the conductor wall by through holes (not shown) provided to penetrate the conductor wall in the longitudinal direction of the conductor case 80. I do. Further, the conductor wire 215 penetrates the conductor wall so as to be electrically connected to the wall, whereby the conductor wire 215 is grounded at a central portion thereof. With the above configuration, a part of the magnetic field of the y-axis resonator of the TM dual-mode dielectric resonator R1 crosses the inside of the loop of the coupling loop inductor 21 around the conductor wire 212, and the TM2 Part of the magnetic field of the y-axis resonator of the heavy-mode dielectric resonator R2 crosses the inside of the loop of the coupling loop inductor 21 around the conductor line 214. As a result, the y-axis resonator of the TM dual-mode dielectric resonator R1 and the y-axis resonator of the TM dual-mode dielectric resonator R2 are inductively coupled via the coupling loop inductor 21.
[0074]
Further, a coupling capacitor 23 is connected to the outer peripheral surface of the torus electrode 4b on the x-axis located on the opposite side of the coupling loop inductor 21. Here, the coupling capacitor 23 is formed by forming electrodes on both end surfaces of a cylindrical dielectric, and the electrode formed on one end surface is connected to the torus electrode 4b. A coupling capacitor 24 is connected to the outer peripheral surface of the torus electrode 4c on the x-axis facing the coupling capacitor 23. Here, similarly to the coupling capacitor 23, the coupling capacitor 24 is formed by forming electrodes on both end surfaces of a cylindrical dielectric, and the electrodes formed on one end surface are connected to the torus electrode 4c. . Further, an electrode formed on the other end face of the coupling capacitor 23 and an electrode formed on the other end face of the coupling capacitor 24 are connected to each other by a conductor line 234. The conductor wire 234 is electrically insulated from a conductor wall separating the cavity 80b and the cavity 80c through a through hole (not shown) provided in the conductor wall, and connects the conductor wall with the conductor wall. Penetrate. The x-axis resonators of the TM dual-mode dielectric resonator R2 and the x-axis resonators of the TM dual-mode dielectric resonator R3 are formed by the coupling capacitors 23 and 24 and the conductor line 234 configured as described above. And is capacitively coupled.
[0075]
An output connector 42 having a center conductor and a ground conductor is provided on one side surface in the width direction of the conductor case 80 such that the center conductor coincides with the y-axis of the TM dual-mode dielectric resonator R3. . The center conductor of the output connector 42 is electrically insulated and penetrates the side surface of the conductor case 80 to be connected to one electrode of the output capacitor 25. The other electrode of the output capacitor 25 is connected to the outer periphery of the torus electrode 4c on the y-axis facing the output connector 42. As described above, the output connector 42 is capacitively coupled to the y-axis resonator of the TM dual-mode dielectric resonator R3 via the output capacitor 25.
[0076]
Amount of coupling between the x-axis resonator and the y-axis resonator of the TM dual-mode dielectric resonator R1 separated from the x-axis of the TM dual-mode dielectric resonator R1 by 45 degrees and close to the notch 13a. A coupling adjusting screw 26 a for adjusting the distance is provided on the lower bottom surface of the conductor case 80. The coupling adjustment screw 26a is provided so as to be parallel to the direction in which the notch 13a is formed, and when the protrusion length of the coupling adjustment screw 26a in the cavity 81a changes, x of the TM dual mode dielectric resonator R1 is changed. Since the coupling amount between the axis resonator and the y-axis resonator changes, the coupling amount can be adjusted by changing the protrusion length of the coupling adjusting screw 26a. Similarly, the x-axis resonator and the y-axis resonator of the TM dual-mode dielectric resonator R2 are separated from the x-axis of the TM dual-mode dielectric resonator R2 by 45 degrees and close to the notch 14b. A coupling adjusting screw 26b for adjusting the amount of coupling with the notch 14b is provided on the lower bottom surface so as to be parallel to the direction in which the notch 14b is formed. A coupling adjustment screw 26c for adjusting the coupling amount between the x-axis resonator and the y-axis resonator of the TM dual-mode dielectric resonator R3 is provided in the notch 14c. It is provided on the lower bottom surface so as to be parallel to the forming direction.
[0077]
A frequency adjusting screw 51 for adjusting the resonance frequency of each of the x-axis resonators of the TM dual-mode dielectric resonators R1, R2, and R3 is provided on the other side surface in the width direction of the conductor case 80, respectively. 52 and 53 are provided such that the axes of the respective screws coincide with the respective y-axes of the TM dual-mode dielectric resonators R1, R2 and R3. When the protrusion length of the frequency adjusting screws 51, 52, 53 in the cavities 81a, 81b, 81c changes, the resonance frequency of each x-axis resonator of the TM dual mode dielectric resonators R1, R2, R3 changes. The resonance frequency of each x-axis resonator can be adjusted by changing the protrusion length of the frequency adjustment screws 51, 52, 53. Further, on the other end face of the conductor case 80, a frequency adjusting screw 54 for adjusting the resonance frequency of the y-axis resonator of the TM dual mode dielectric resonator R3 is provided. It is provided so as to coincide with the x-axis of the TM dual-mode dielectric resonator R3. If the protrusion length of the frequency adjustment screw 54 in the cavity 81c changes, the resonance frequency of the y-axis resonator of the TM dual-mode dielectric resonator R3 changes, so that by changing the protrusion length of the frequency adjustment screw 54, The resonance frequency can be adjusted.
[0078]
As described above, the high-frequency band-pass filter device according to the third embodiment includes three TM dual-mode dielectric resonators R1, R2, and R3. In the high-frequency band-pass filter device, the input connector 41 is capacitively coupled to the x-axis resonator of the TM dual-mode dielectric resonator R1 by the input capacitor 22. The x-axis resonator of the TM dual-mode dielectric resonator R1 is formed by the notches 11a and 12a provided in the dielectric 1a and the notches 13a and 14a provided in the dielectric 2a. It is electromagnetically coupled with the y-axis resonator of the heavy mode dielectric resonator R1. The y-axis resonator of the TM dual-mode dielectric resonator R1 is inductively coupled to the y-axis resonator of the TM dual-mode dielectric resonator R2 by the coupling loop inductor 21. The y-axis resonator of the TM dual-mode dielectric resonator R2 is formed by the notches 11b and 12b provided in the dielectric 1b and the notches 13b and 14b provided in the dielectric 2b. It is electromagnetically coupled with the x-axis resonator of the dielectric resonator R2. Further, the x-axis resonator of the TM dual-mode dielectric resonator R2 is capacitively coupled to the x-axis resonator of the TM dual-mode dielectric resonator R3 by coupling capacitors 23 and 24. The x-axis resonator of the TM dual mode dielectric resonator R3 is formed by the notches 11c and 12c provided in the dielectric 1c and the notches 13c and 14c provided in the dielectric 2c. It is electromagnetically coupled with the y-axis resonator of the dielectric resonator R3. Further, the y-axis resonator of the TM dual-mode dielectric resonator R3 is capacitively coupled to the output connector 42 by the output capacitor 25. With the above configuration, a six-stage high-frequency band-pass filter device including the TM dual-mode dielectric resonators R1, R2, and R3 between the input connector 41 and the output connector 42 is configured.
[0079]
Next, the operation of the high-frequency band-pass filter device configured as described above will be described using an equivalent circuit.
[0080]
FIG. 17 is a circuit diagram showing an equivalent circuit of the high-frequency band-pass filter device. In the equivalent circuit, each of the TM dual-mode dielectric resonators R1, R2, and R3 can be represented in the same manner as the equivalent circuit of the TM dual-mode dielectric resonator of the second embodiment in FIG. That is, the equivalent circuit of each TM dual-mode dielectric resonator R1, R2, R3 is a double-mode equivalent circuit in which the x-axis resonator and the y-axis resonator are coupled, and each quarter-mode at each resonance frequency. Rotationally symmetric ring distributed constant lines having an electric length of 2π are connected to distributed constant lines LN1a to LN8a, distributed constant lines LN1b to LN8b, and distributed constant lines LN1c to LN8c each having a wavelength. In the equivalent circuit of the TM dual mode dielectric resonator R1, a connection point between the distributed constant line LN1a and the distributed constant line LN2a is grounded via an internal coupling capacitor C3a, and the distributed constant line LN5a and the distributed constant The connection point with the line LN6a is grounded via the internal coupling capacitor C4a.
[0081]
In the equivalent circuit of the TM dual-mode dielectric resonator R2, the connection point between the distributed constant line LN1b and the distributed constant line LN2b is grounded via the internal coupling capacitor C3b, and the distributed constant line LN5b and the distributed constant The connection point with the line LN6b is grounded via the internal coupling capacitor C4b. Further, in the equivalent circuit of the TM dual mode dielectric resonator R3, the connection point between the distributed constant line LN1c and the distributed constant line LN2c is grounded via the internal coupling capacitor C3c, and the distributed constant line LN5c and the distributed constant The connection point with the line LN6c is grounded via the internal coupling capacitor C4c. As described above, an equivalent circuit of each of the TM dual-mode dielectric resonators R1, R2, and R3 is configured.
[0082]
A connection point between the distributed constant line LN1a and the distributed constant line LN8a located on the x-axis of the TM dual mode dielectric resonator R1 is connected at one end to the other end of the input capacitor 22 having the input terminal T11. , Is grounded via a capacitor C2a. Here, the capacitor C2a is a capacitance for correcting the resonance frequency of the x-axis resonator, which changes when the input capacitor 22 is coupled to the x-axis resonator of the TM dual-mode dielectric resonator R1. And has a negative capacitance value. A connection point between the distributed constant lines LN2a and LN3a and a connection point between the distributed constant lines LN6a and LN7a are grounded via capacitors C5a and C6a, respectively. Here, the capacitors C5a and C6a correct the resonance frequency of the y-axis resonator that changes when the coupling loop inductor 21 is coupled to the y-axis resonator of the TM dual-mode dielectric resonator R1 as described later. And each has a negative capacitance value.
[0083]
Further, an inductor L1 is connected between the distributed constant line LN4a and the distributed constant line LN5a of the TM dual mode dielectric resonator R1, and the inductor L1 is inductively coupled to the inductor L11. The inductor L1 represents an equivalent inductor for generating a magnetic field inductively coupled to the coupling loop inductor 21 among magnetic fields generated by the y-axis resonator of the TM dual-mode dielectric resonator R1. The inductor L11, the inductor L12, and the inductor L13 are connected in parallel to form a coupled loop inductor 21, and one end of the coupled loop inductor 21 is grounded. Then, the inductor L13 is inductively coupled to the inductor L2 connected between the distributed constant line LN1b and the distributed constant line LN8b of the TM dual-mode dielectric resonator R2. Here, the inductor L2 represents an equivalent inductor for generating a magnetic field inductively coupled to the coupling loop inductor 21 among magnetic fields generated by the y-axis resonator of the TM dual-mode dielectric resonator R2. . Thus, the y-axis resonator of the TM dual-mode dielectric resonator R1 and the y-axis resonator of the TM dual-mode dielectric resonator R2 are inductively coupled.
[0084]
The connection point between the distributed constant lines LN2b and LN3b of the TM dual mode dielectric resonator R2 and the connection point between the distributed constant lines LN6b and LN7b are grounded via capacitors 5b and 6b, respectively. Here, the capacitors 5b and 6b serve as static capacitors for correcting the resonance frequency of the y-axis resonator that changes when the coupling loop inductor 21 is coupled to the y-axis resonator of the TM dual-mode dielectric resonator R2. Capacitance, each having a negative capacitance value.
[0085]
The connection point between the distributed coupling line LN4b and the distributed constant coupling line LN5b of the TM dual-mode dielectric resonator R2 is grounded via a capacitor C7b and is connected to one end of a coupling capacitor C8. Here, the coupling capacitor C8 corresponds to a capacitor configured by connecting the coupling capacitors 23 and 24 of FIG. 16 in series. The other end of the coupling capacitor C8 is grounded via the capacitor 7c, and is connected to a connection point between the distributed constant line LN1c and the distributed constant line LN8c of the TM dual mode dielectric resonator R3. As a result, the x-axis resonator of the TM dual-mode dielectric resonator R2 and the x-axis resonator of the TM dual-mode dielectric resonator R3 are capacitively coupled. Here, the capacitor C7b is a capacitance for correcting the resonance frequency of the x-axis resonator, which changes when the coupling capacitor C8 is coupled to the x-axis resonator of the TM dual-mode dielectric resonator R2. And has a negative capacitance value. Similarly, the capacitor C7c is provided with an electrostatic capacitor for correcting the resonance frequency of the x-axis resonator that changes when the coupling capacitor C8 is coupled to the x-axis resonator of the TM dual-mode dielectric resonator R3. A capacitance having a negative capacitance value.
[0086]
A connection point between the distributed coupling line LN6c and the distributed constant coupling line LN7c of the TM dual-mode dielectric resonator R3 is grounded via a capacitor C2c, and one end of the output capacitor 25 whose one end is an output terminal T12. Connected to. As a result, the y-axis resonator of the TM dual-mode dielectric resonator R3 is capacitively coupled to an external circuit.
[0087]
As described above, the equivalent circuit of the high-frequency band-pass filter device according to the third embodiment is configured. That is, in the high-frequency band-pass filter device of the third embodiment, the high-frequency signal having a predetermined frequency input to the input terminal T11 is applied to the x-axis of the dual mode dielectric resonator R1 via the input capacitor 22. The high frequency signal input to the resonator is transmitted to the y-axis resonator of the dual mode dielectric resonator R1 by the internal coupling capacitors C3a and C4a. The high-frequency signal transmitted to the y-axis resonator of the dual-mode dielectric resonator R1 is transmitted to the y-axis resonator of the dual-mode dielectric resonator R2 via the coupling loop inductor 21 and further transmitted to the inside. The light is transmitted to the x-axis resonator of the dual mode dielectric resonator R2 by the coupling capacitors C3b and C4b. The high-frequency signal transmitted to the x-axis resonator of the dual-mode dielectric resonator R2 is transmitted to the x-axis resonator of the dual-mode dielectric resonator R3 via the coupling capacitor C8, and further internally coupled. The signal is transmitted to the y-axis resonator of the dual mode dielectric resonator R3 by the capacitors C3c and C4c. Then, the high frequency signal transmitted to the y-axis resonator of the dual mode dielectric resonator R3 is output from the output terminal T12 via the output capacitor 25. As described above, the high-frequency band-pass filter device according to the third embodiment passes an input signal having a predetermined frequency and outputs the signal.
[0088]
Next, a method of setting each circuit constant in the equivalent circuit of FIG. 17 will be described.
[0089]
In the high-frequency band-pass filter device according to the third embodiment, the internal coupling between the x-axis resonator and the y-axis resonator in each of the TM dual-mode dielectric resonators R1, R2, R3, and the TM double-mode dielectric resonance The coupling between the devices R1, R2, and R3 can be represented by a circuit similar to a filter design method using a K inverter and a J inverter, respectively. Therefore, each circuit constant of the equivalent circuit shown in FIG._eAnd the coupling coefficient k. Here, in the equivalent circuit of FIG. 17, the K inverter and the J inverter are each used after modifying the lumped constant circuit expression.
[0090]
According to this, first, the capacitance C of the input capacitor 22 is calculated.₀₁And the capacitance C of the output capacitor 25₆₇Can be expressed by the following equation 9, and the capacitance C2 of the capacitor C2a is₁₁And the capacitance C of the capacitor C2c₆₆Can be expressed by the following equation (10). Here, J is an admittance inverter parameter of the input / output unit, and is expressed by the following equation (11)._LIs the input / output impedance, where both the input impedance and the output impedance are Z_L= 50Ω.
[0091]
(Equation 9)
C₀₁= C₆₇= (J / ω₀) ・ {1 /} (1-J²Z_L ²)｝
(Equation 10)
C₁₁= C₆₆= (J / ω₀) ・ √ (1-J²Z_L ²)
(Equation 11)
J = √ ｛(πY_a) / (Z_LQ_e)｝
[0092]
Also, the capacitance C of the internal coupling capacitors C3a and C4a₁₂And the capacitance C of the internal coupling capacitors C3b and C4b₃₄And the capacitance C of the internal coupling capacitors C3c and C4c.₅₆Is represented by the following equation 12. Here, in Equations 12 and 13, the subscripts (i, j) represent the capacitance C, respectively.₁₂, C₃₄, C₅₆(1, 2), (3, 4), and (5, 6). Also, k₁₂, K₃₄, K₅₆Is a coupling coefficient between the x-axis resonator and the y-axis resonator in the TM dual mode dielectric resonators R1, R2, and R3, respectively.
[0093]
(Equation 12)
C_ij= (-2Y_a/ Ω₀) · Tan ｛(πk_ij) / (2-k_ij)｝, (I, j) = (1,2), (3,4), (5,6)
[0094]
Further, the electrical length θ of each of the distributed constant lines LN1a to LN8a of the TM dual mode dielectric resonator R1₁₂And the electrical length θ of each of the distributed constant lines LN1b to LN8b of the TM dual mode dielectric resonator R2.₃₄And the electrical length θ of each distributed constant line LN1c to LN8c of the TM dual mode dielectric resonator R3.₅₆Is represented by the following Expression 13.
[0095]
(Equation 13)
θ_ij= (Π / 4) · (1 + k_ij/ 2), (i, j) = (1, 2), (3, 4), (5, 6)
[0096]
Furthermore, the inductance L of the coupling loop inductor 21₂₃And the capacitance C of the coupling capacitor C8₄₅Is represented by the following Expressions 14 and 15, respectively. Also, the capacitance C of the capacitors 5a, 5b, 6a, 6b₂₃₀Is the capacitance C expressed by the following equation (16).₂₃Is represented by Expression 17 using Here, k in Equations 14 to 16₂₃And k₄₅Are respectively the coupling coefficient between the y-axis resonator of the TM dual-mode dielectric resonator R1 and the y-axis resonator of the TM dual-mode dielectric resonator R2, and the x-axis of the TM dual-mode dielectric resonator R2. It is a coupling coefficient between the resonator and the x-axis resonator of the TM dual-mode dielectric resonator R3. Also, Z_aAre the characteristic impedances of the distributed constant transmission lines LN1a to LN8a, the distributed constant transmission lines LN1b to LN8b, and the distributed constant transmission lines LN1c to LN8c, all set equal.
[0097]
[Equation 14]
L₂₃= (K₂₃πZ_a) / Ω₀
(Equation 15)
C₄₅= (K₄₅πY_a) / Ω₀
(Equation 16)
C₂₃= (K₂₃πY_a) / Ω₀
[Equation 17]
C₂₃₀= -C₂₃/ 2
[0098]
As described above, each circuit constant of the equivalent circuit of FIG. 17 is set.
[0099]
Next, an electrical design example of the high-frequency band-pass filter device according to the third embodiment will be described. The Chebyshev type filter design theory can be applied to the filter design using the TM dual mode dielectric resonators R1, R2, and R3. According to the design theory of the Chebyshev type filter, the external Q and the coupling coefficient k₁₂, K₂₃, K₃₄, K₄₅, K₅₆Is obtained as shown in Table 4. Using the parameters of Table 4 and the above-described method of setting the circuit constants, the circuit constants of the equivalent circuit of FIG. 17 are calculated as shown in Table 5. Table 5 shows each circuit constant of the equivalent circuit of FIG. 17 calculated using the design theory of the Chebyshev-type filter and the output-end transmission coefficient S obtained by the design theory of the Chebyshev-type filter using these circuit constants as initial values.₂₁And input end reflection coefficient S₁₁Table 5 also shows the circuit constants optimized by a circuit simulator (MDS manufactured by Hewlett-Packard Company) so as to match the above. Output end transmission coefficient S at that time₂₁And input end reflection coefficient S₁₁And the transmission coefficient S at the output end obtained by the design theory of the Chebyshev type filter.₂₁And input end reflection coefficient S₁₁Is shown in FIG. Output end transmission coefficient S₂₁Are slightly different in the attenuation region, but are well matched in the pass band. Input end reflection coefficient S₁₁Are in good agreement over all frequency ranges.
[0100]
[Table 3]

[0101]
[Table 4]

[0102]
[Table 5]

[0103]
The inventor prototyped the high-frequency band-pass filter device according to the filter design specifications shown in Table 3, and evaluated the characteristics using a network analyzer (HP8753C). FIG. 19 shows characteristics evaluation results of the prototyped high-frequency bandpass filter device and simulation results performed using an equivalent circuit of the high-frequency bandpass filter device. The center frequency of the high-frequency band-pass filter device is 985 MHz, and the pass band width is 12 MHz. According to the above measurement results, the insertion loss at the center frequency is 1.3 dB, and the reflection loss is 22 dB. The bandwidth at the 3 dB attenuation point is 13.3 MHz, and the attenuation at a point 20 MHz away from the center frequency of 985 MHz is 62.8 dB. As is clear from FIG. 19, the measurement result and the simulation result of the equivalent circuit match well. Also, from the value of the insertion loss, it is inferred that the no-load Q of the resonator in each stage operates at about 2700 with the coupling mechanism and the adjustment mechanism mounted. The external dimensions of the prototyped high-frequency band-pass filter device were 158 × 54 × 23.5 mm, and a smaller and thinner device than the conventional high-frequency band-pass filter device could be realized.
[0104]
The high-frequency band-pass filter device of the third embodiment described above can be smaller and thinner than the conventional high-frequency band-pass filter device.
[0105]
In the high-frequency band-pass filter device according to the third embodiment, the coupling between the input terminal T11 and the x-axis resonator of the TM dual-mode dielectric resonator R1 and the y-axis resonator of the TM dual-mode dielectric resonator R3 are performed. The coupling between the input terminal 22 and the output terminal T12 uses capacitive coupling with small coupling loss by the input capacitor 22 and the output capacitor 25, respectively, so that the loss in the pass band of the high-frequency band-pass filter device can be reduced.
[0106]
<First Modification>
FIG. 20A is a longitudinal sectional view of a TM dual mode dielectric resonator according to a first modification of the present invention, and FIG. 20B is a diagram of the TM dual mode dielectric resonator shown in FIG. It is a transverse cross section about the EE 'line of a). The difference between the TM dual mode dielectric resonator and the TM dual mode dielectric resonator of the first embodiment is that each of the TM dual mode dielectric resonator traverses a cavity 108 a formed by the dielectrics 101 and 102, the plate electrode 103, and the conductor case 108. The point is that the surface shape is square. Here, the dielectrics 101 and 102 are provided with end face electrodes 106 and 107 on one end face, respectively. Further, a torus electrode 104 having a circular cross section having a diameter larger than the thickness of the plate electrode 103 is provided around the entire periphery of the plate electrode 103.
[0107]
The TM dual mode dielectric resonator of the first modification configured as described above operates similarly to the TM dual mode dielectric resonator of the first embodiment, and has the same effect.
[0108]
<Second Modification>
A high-frequency bandpass filter device according to a second modification of the present invention is characterized in that a coupling loop inductor 21a is provided instead of the coupling capacitors 23 and 24 of the high-frequency bandpass filter device according to the third embodiment. FIG. 26A is a schematic plan view of a high-frequency bandpass filter device according to a second modification. As shown in FIG. 26A, in the high-frequency band-pass filter device, the conductor case 81 includes three conductor portions 81a, 81b, and 81c having the same square pillar shape, and two conductor portions adjacent to each other. 81a and 81b are connected to each other via one side surface, and conductor portions 81b and 81c are connected to each other via one side surface to be integrally formed in an L-shape. The conductor 81a is provided with a TM dual-mode dielectric resonator R1, the conductor 81b is provided with a TM dual-mode dielectric resonator R2, and the conductor 81c is provided with a TM dual-mode dielectric resonator R2. A dielectric resonator R3 is provided. Here, the x-axes of the three TM dual-mode dielectric resonators R1, R2, and R3 are parallel to each other, and the y-axes of the TM dual-mode dielectric resonators R1, R2, and R3 are parallel to each other. , TM dual-mode dielectric resonators R1, R2, R3. The x-axis of the TM dual-mode dielectric resonator R1 and the x-axis of the TM dual-mode dielectric resonator R2 are on a straight line, while the y-axis of the TM dual-mode dielectric resonator R2 and the y-axis of the TM dual-mode dielectric resonator R2 The y-axis of the TM dual-mode dielectric resonator R3 exists in a straight line. The input connector 41 is provided at the center of the end face 81ta of the conductor case 81, and the output connector 42 is provided at the center of the end face 81tb of the conductor case 81. Further, the coupling loop inductor 21 is provided at the center of the conductor wall of the conductor case 81 that separates the TM dual mode dielectric resonator R1 and the TM dual mode dielectric resonator R2. It is provided at the center of the conductor wall of the conductor case 81 that separates the TM dual mode dielectric resonator R2 and the TM dual mode dielectric resonator R3.
[0109]
FIG. 21 is a circuit diagram showing an equivalent circuit of the high-frequency bandpass filter device according to the second modification. The equivalent circuit of FIG. 21 includes an equivalent circuit of the TM dual-mode dielectric resonators R1, R2, and R3, similarly to the equivalent circuit of FIG. 17, and is configured as follows.
[0110]
Elements similar to those in the equivalent circuit of FIG. 17 are connected to respective connection points of the distributed constant lines LN1a to L8a in the TM dual mode dielectric resonator R1. 17, the y-axis resonator of the TM dual-mode dielectric resonator R1 and the y-axis resonator of the TM dual-mode dielectric resonator R2 form a coupling loop inductor 21 with a configuration similar to that of the equivalent circuit of FIG. Inductively coupled via
[0111]
The junction of the distributed constant line LN1b of the TM dual mode dielectric resonator R2 and the inductor L2 is grounded via a capacitor C9b. A connection point between the distributed constant line LN2b and the distributed constant line LN3b and a connection point between the distributed constant line LN4b and the distributed constant line LN5b are grounded via capacitors 5b and 7b, respectively. An inductor L3 is connected in series between the distributed constant line LN6b and the distributed constant line LN7b, and a connection point between the distributed constant line LN7b and the inductor L3 is grounded via a capacitor C6b. Here, the inductor L3 represents an equivalent inductor for generating a magnetic field inductively coupled to the coupling loop inductor 21a among magnetic fields generated by the x-axis resonator of the TM dual-mode dielectric resonator R2. . Further, the capacitors C7b and C9b are provided with an electrostatic capacitance for correcting the resonance frequency of the x-axis resonator, which changes when the coupling loop inductor 21a is coupled to the x-axis resonator of the TM dual-mode dielectric resonator R2. Capacitance, each having a negative capacitance value.
[0112]
The inductor L3 is inductively coupled to the inductor L21. The inductor L21, the inductor L22, and the inductor L23 are connected in parallel to form a coupled loop inductor 21a, and one end of the coupled loop inductor 21a is grounded. Then, the inductor L23 is inductively coupled to an inductor L4 connected between the distributed constant line LN2c and the distributed constant line LN3c of the TM dual mode dielectric resonator R3. Here, the inductor L4 represents an equivalent inductor for generating a magnetic field inductively coupled to the coupling loop inductor 21a among magnetic fields generated by the x-axis resonator of the TM dual mode dielectric resonator R3. . As described above, the x-axis resonator of the TM dual-mode dielectric resonator R2 and the x-axis resonator of the TM dual-mode dielectric resonator R3 are inductively coupled.
[0113]
The connection point between the distributed constant lines LN1c and LN8c of the TM dual mode dielectric resonator R3 and the connection point between the distributed constant lines LN4c and LN5c are grounded via capacitors C9c and C10c, respectively. Here, the capacitors C9c and C10c serve as static capacitors for correcting the resonance frequency of the x-axis resonator, which changes when the coupling loop inductor 21a is coupled to the x-axis resonator of the TM dual-mode dielectric resonator R3. Capacitance, each having a negative capacitance value.
[0114]
A connection point between the distributed coupling line LN6c and the distributed constant coupling line LN7c of the TM dual-mode dielectric resonator R3 is grounded via a capacitor C2c, and one end of the output capacitor 25 whose one end is an output terminal T12. Connected to. This allows the y-axis resonator of the TM dual-mode dielectric resonator R3 to be capacitively coupled to an external circuit.
[0115]
As described above, the high-frequency bandpass filter device of the second modification includes three TM dual-mode dielectric resonators R1, R2, and R3. In the high-frequency band-pass filter device, the input connector 41 is capacitively coupled to the x-axis resonator of the TM dual-mode dielectric resonator R1 by the input capacitor 22. The x-axis resonator of the TM dual-mode dielectric resonator R1 is formed by the notches 11a and 12a provided in the dielectric 1a and the notches 13a and 14a provided in the dielectric 2a. It is electromagnetically coupled with the y-axis resonator of the heavy mode dielectric resonator R1. The y-axis resonator of the TM dual-mode dielectric resonator R1 is inductively coupled to the y-axis resonator of the TM dual-mode dielectric resonator R2 by the coupling loop inductor 21. The y-axis resonator of the TM dual-mode dielectric resonator R2 is formed by the notches 11b and 12b provided in the dielectric 1b and the notches 13b and 14b provided in the dielectric 2b. It is electromagnetically coupled with the x-axis resonator of the dielectric resonator R2. Further, the x-axis resonator of the TM dual-mode dielectric resonator R2 is inductively coupled to the x-axis resonator of the TM dual-mode dielectric resonator R3 by the coupling loop inductor 21a. The x-axis resonator of the TM dual mode dielectric resonator R3 is formed by the notches 11c and 12c provided in the dielectric 1c and the notches 13c and 14c provided in the dielectric 2c. It is electromagnetically coupled with the y-axis resonator of the dielectric resonator R3. Further, the y-axis resonator of the TM dual-mode dielectric resonator R3 is capacitively coupled to the output connector 42 by the output capacitor 25. With the above configuration, a six-stage high-frequency bandpass filter device including the TM dual-mode dielectric resonators R1, R2, and R3 between the input connector 41 and the output connector 42 is configured. Thus, the high-frequency band-pass filter device according to the second modified example operates in the same manner as the high-frequency band-pass filter device according to the third embodiment and has the same effect.
[0116]
<Third Modification>
A high-frequency band-pass filter device according to a third modification of the present invention includes a TM dual-mode dielectric resonator R1 and a TM double-mode dielectric instead of the coupling loop inductor 21 of the high-frequency band-pass filter device of the third embodiment. It is characterized in that coupling capacitors 23a and 24a for capacitively coupling with the resonator R2 are provided. FIG. 26B is a schematic plan view of a high-frequency bandpass filter device according to a third modification. As shown in FIG. 26B, the high-frequency band-pass filter device includes an L-shaped conductor case 81 and a TM dual-mode dielectric resonator R1, R2, as in the second modification. , R3 are configured in the same manner as in the second modification. The input connector 41 is provided on the side surface 81sa of the conductor case 81 such that the center conductor of the input connector 41 is located on the y-axis of the TM dual-mode dielectric resonator R1. The center conductor of 42 is provided on the side surface 81sb of the conductor case 81 so as to be located on the x-axis of the TM dual-mode dielectric resonator R3. Further, the coupling capacitors 23a and 24a are provided on the outer peripheral surface of the torus electrode 4a of the TM dual-mode dielectric resonator R1 and the outer peripheral surface of the torus electrode 4b of the TM dual-mode dielectric resonator R2, respectively. The central part of the conductor wall of the conductor case 81 separating the TM dual-mode dielectric resonator R1 and the TM dual-mode dielectric resonator R2 is connected by a conductor wire that is electrically insulated and penetrates. The coupling capacitors 23 and 24 are provided on the outer peripheral surface of the torus electrode 4b of the TM dual-mode dielectric resonator R2 and the outer peripheral surface of the torus electrode 4c of the TM dual-mode dielectric resonator R3, respectively. The central part of the conductor wall of the conductor case 81 separating the TM dual-mode dielectric resonator R2 and the TM dual-mode dielectric resonator R3 is connected by a conductor wire that is electrically insulated and penetrates.
[0117]
FIG. 22 is a circuit diagram showing an equivalent circuit of the high-frequency band-pass filter device according to the third modification. The equivalent circuit of FIG. 22 includes an equivalent circuit of the TM dual-mode dielectric resonators R1, R2, and R3, similarly to the equivalent circuit of FIG. 17, and is configured as follows.
[0118]
One end of the connection point between the distributed constant line LN2a and the distributed constant line LN3a of the TM dual mode dielectric resonator R1 is connected to the other end of the input capacitor 22 which is the input terminal T11, and is grounded via the capacitor C2a. You.
[0119]
Further, a connection point between the distributed constant line LN4a and the distributed constant line LN5a of the TM dual mode dielectric resonator R1 is grounded via a capacitor C11a, and one end of a coupling capacitor C12 is connected. Here, the coupling capacitor C12 is provided corresponding to a coupling capacitor configured by connecting the coupling capacitors 23a and 24a in series. A connection point between the distributed constant line LN1b and the distributed constant line LN8b of the TM dual-mode dielectric resonator R2 is connected to the other end of the coupling capacitor C12. Further, a connection point between the distributed constant line LN1b and the distributed constant line LN8b is grounded via the capacitor C11b. As a result, the x-axis resonator of the TM dual-mode dielectric resonator R1 and the x-axis resonator of the TM dual-mode dielectric resonator R2 are capacitively coupled. Here, the capacitor C11a is a capacitance for correcting the resonance frequency of the x-axis resonator that changes when the x-axis resonator of the TM dual-mode dielectric resonator R1 is coupled to the coupling capacitor C12. And has a negative capacitance value. The capacitor C11b is a capacitance for correcting the resonance frequency of the x-axis resonator that changes when the x-axis resonator of the TM dual-mode dielectric resonator R2 is coupled to the coupling capacitor C12. And has a negative capacitance value.
[0120]
Further, a connection point between the distributed constant line LN6b and the distributed constant line LN7b of the TM dual-mode dielectric resonator R2 is grounded via a capacitor C7b, and one end of a coupling capacitor C8 is connected. The connection point between the distributed constant line LN2c and the distributed constant line LN3c of the TM dual mode dielectric resonator R3 is grounded via a capacitor C7c, and the other end of the coupling capacitor C8 is connected. As a result, the y-axis resonator of the TM dual-mode dielectric resonator R2 and the y-axis resonator of the TM dual-mode dielectric resonator R3 are capacitively coupled.
[0121]
A connection point between the distributed coupling line LN4c and the distributed constant coupling line LN5c of the TM dual-mode dielectric resonator R3 is grounded via a capacitor C2c, and one end of the output capacitor 25 whose one end is an output terminal T12. Connected to. Thus, the x-axis resonator of the TM dual-mode dielectric resonator R3 can be capacitively coupled to an external circuit.
[0122]
As described above, the high-frequency bandpass filter device of the third modification includes three TM dual-mode dielectric resonators R1, R2, and R3. In the high-frequency band-pass filter device, the input connector 41 is capacitively coupled to the y-axis resonator of the TM dual-mode dielectric resonator R1 by the input capacitor 22. The y-axis resonator of the TM dual-mode dielectric resonator R1 is formed by the notches 11a and 12a provided in the dielectric 1a and the notches 13a and 14a provided in the dielectric 2a. It is electromagnetically coupled with the x-axis resonator of the heavy mode dielectric resonator R1. Further, the x-axis resonator of the TM dual-mode dielectric resonator R1 is capacitively coupled to the x-axis resonator of the TM dual-mode dielectric resonator R2 by the coupling capacitor C12. The x-axis resonator of the TM dual mode dielectric resonator R2 is formed by the notches 11b and 12b provided in the dielectric 1b and the notches 13b and 14b provided in the dielectric 2b. It is electromagnetically coupled with the y-axis resonator of the dielectric resonator R2. Further, the y-axis resonator of the TM dual-mode dielectric resonator R2 is capacitively coupled to the y-axis resonator of the TM dual-mode dielectric resonator R3 by the coupling capacitor C8. The y-axis resonator of the TM dual-mode dielectric resonator R3 is formed by the notches 11c and 12c provided in the dielectric 1c and the notches 13c and 14c provided in the dielectric 2c. It is electromagnetically coupled with the x-axis resonator of the dielectric resonator R3. Further, the x-axis resonator of the TM dual-mode dielectric resonator R3 is capacitively coupled to the output connector 42 by the output capacitor 25. With the above configuration, a six-stage high-frequency bandpass filter device including the TM dual-mode dielectric resonators R1, R2, and R3 between the input connector 41 and the output connector 42 is configured. Thus, the high-frequency band-pass filter device according to the third modified example operates in the same manner as the high-frequency band-pass filter device according to the third embodiment and has the same effect.
[0123]
<Fourth modification>
A high-frequency band-pass filter device according to a fourth modification of the present invention includes an input loop inductor 31 instead of the input capacitor 22 of the high-frequency band-pass filter device according to the third embodiment, and an output instead of the output capacitor 25. It is characterized by including a loop inductor 43. FIG. 27A is a schematic plan view of a high-frequency bandpass filter device according to a fourth modification. As shown in FIG. 27A, the high-frequency band-pass filter device includes a conductor case 80 similar to the high-frequency band-pass filter device of the third embodiment, and includes the TM dual-mode dielectric resonators R1 and R2. , R3 are juxtaposed at predetermined intervals. In the high-frequency band-pass filter device according to the fourth modified example, the input connector 41 has a center conductor on one side surface of the conductor case 80 and the y conductor of the TM dual-mode dielectric resonator R1. The output connector 42 is provided so as to coincide with the axis, and is provided at the center of the end face of the conductor case 80 adjacent to the TM dual-mode dielectric resonator R3.
[0124]
FIG. 23 is a circuit diagram showing an equivalent circuit of a high-frequency bandpass filter device according to a fourth modification. The equivalent circuit of FIG. 23 includes an equivalent circuit of the TM dual-mode dielectric resonators R1, R2, and R3, similarly to the equivalent circuit of FIG. 17, and is configured as follows.
[0125]
The other end of the inductor L5 whose one end is grounded is connected to the input terminal T11. Here, the inductor L5 is, for example, an inductor corresponding to a loop configured by grounding the end of the center conductor of the input connector. Then, the inductor L31 is inductively coupled to the inductor L5. The inductor L31, the inductor L32, and the inductor L33 are connected in parallel to form an input loop inductor 31, and one end of the input loop inductor 31 is grounded. Then, the inductor L33 is inductively coupled to the inductor L6 connected between the distributed constant line LN6a and the distributed constant line LN7a of the TM dual-mode dielectric resonator R1. Here, the inductor L6 represents an equivalent inductor for generating a magnetic field inductively coupled to the coupling loop inductor 31 among magnetic fields generated by the x-axis resonator of the TM dual mode dielectric resonator R1. . Thus, the x-axis resonator of the TM dual-mode dielectric resonator R1 and the input terminal T11 are inductively coupled.
[0126]
The connection point between the distributed constant line LN7a of the TM dual mode dielectric resonator R1 and the inductor L6 is grounded via a capacitor C6a.
[0127]
The connection point between the distributed constant line LN8a and the distributed constant line LN1a of the TM dual-mode dielectric resonator R1 is grounded via the capacitor C13a. The connection point between the distributed constant line LN2a and the distributed constant line LN3a is grounded via the capacitor C5a. An inductor L1 is connected in series between the distributed constant line LN4a and the distributed constant line LN5a, and a connection point between the distributed constant line LN4a and the inductor L1 is grounded via a capacitor C14a. Here, the capacitors C13a and C14a are used to correct the resonance frequency of the x-axis resonator, which changes when the input loop inductor L30 is coupled to the x-axis resonator of the TM dual-mode dielectric resonator R1. Capacitance, each having a negative capacitance value.
[0128]
The y-axis resonator of the TM dual-mode dielectric resonator R1 and the y-axis resonator of the TM dual-mode dielectric resonator R2 are configured in the same manner as the equivalent circuit of FIG. 17 and are inductively coupled. Elements similar to those in the equivalent circuit of FIG. 17 are connected to connection points of the distributed constant lines LN1b to LN8b of the TM dual mode dielectric resonator R2.
[0129]
A connection point between the distributed constant line LN1c and the distributed constant line LN8c of the dual mode dielectric resonator R3 is grounded via a capacitor C7c, and the other end of the coupling capacitor C8 is connected. As a result, the x-axis resonator of the TM dual-mode dielectric resonator R2 and the x-axis resonator of the TM dual-mode dielectric resonator R3 are capacitively coupled.
[0130]
Further, the connection point between the distributed constant line LN2c and the distributed constant line LN3c of the TM dual mode dielectric resonator R3 and the connection point between the distributed constant line LN6c and the distributed constant line LN7c are grounded via the capacitors C15c and 16c, respectively. Is done. An inductor L7 is connected in series between the distributed constant line LN4c and the distributed constant line LN5c. Here, the inductor L7 represents an equivalent inductor for generating a magnetic field inductively coupled to the coupling loop inductor 43 among magnetic fields generated by the y-axis resonator of the TM dual mode dielectric resonator R3. . Then, the inductor L41 is inductively coupled to the inductor L7. The inductor L41, the inductor L42, and the inductor L43 are connected in parallel to form an output loop inductor 43, and one end of the output loop inductor 43 is grounded. The inductor L43 is inductively coupled to an inductor L8 having one end connected to the output terminal T12 and the other end grounded. Here, the inductor L8 is, for example, an inductor corresponding to a loop formed by grounding the end of the center conductor of the output connector 42. Thus, the y-axis resonator of the TM dual-mode dielectric resonator R3 is inductively coupled to the output terminal T12 via the output loop inductor 43. Here, the capacitors C15c and 16c are used to correct the resonance frequency of the y-axis resonator, which changes when the output loop inductor 43 is coupled to the y-axis resonator of the TM dual-mode dielectric resonator R3. Capacitance, each having a negative capacitance value.
[0131]
As described above, the high-frequency bandpass filter device of the fourth modification includes three TM dual-mode dielectric resonators R1, R2, and R3. In the high-frequency band-pass filter device, the input connector 41 is inductively coupled to the x-axis resonator of the TM dual-mode dielectric resonator R1 by the input loop inductor 31. The x-axis resonator of the TM dual-mode dielectric resonator R1 is formed by the notches 11a and 12a provided in the dielectric 1a and the notches 13a and 14a provided in the dielectric 2a. It is electromagnetically coupled with the y-axis resonator of the heavy mode dielectric resonator R1. The y-axis resonator of the TM dual-mode dielectric resonator R1 is inductively coupled to the y-axis resonator of the TM dual-mode dielectric resonator R2 by the coupling loop inductor 21. The y-axis resonator of the TM dual-mode dielectric resonator R2 is formed by the notches 11b and 12b provided in the dielectric 1b and the notches 13b and 14b provided in the dielectric 2b. It is electromagnetically coupled with the x-axis resonator of the dielectric resonator R2. Further, the x-axis resonator of the TM dual-mode dielectric resonator R2 is capacitively coupled to the x-axis resonator of the TM dual-mode dielectric resonator R3 by the coupling capacitor C8. The x-axis resonator of the TM dual mode dielectric resonator R3 is formed by the notches 11c and 12c provided in the dielectric 1c and the notches 13c and 14c provided in the dielectric 2c. It is electromagnetically coupled with the y-axis resonator of the dielectric resonator R3. Further, the y-axis resonator of the TM dual-mode dielectric resonator R3 is inductively coupled to the output connector 42 by the output loop inductor 43. With the above configuration, a six-stage high-frequency bandpass filter device including the TM dual-mode dielectric resonators R1, R2, and R3 between the input connector 41 and the output connector 42 is configured. Thus, the high-frequency band-pass filter device according to the fourth modification operates in the same manner as the high-frequency band-pass filter device according to the third embodiment, and has the same effect.
[0132]
<Fifth Modification>
A high-frequency bandpass filter device according to a fifth modification of the present invention is characterized in that an output loop inductor 43 is provided instead of the output capacitor 25 of the high-frequency bandpass filter device according to the third embodiment. FIG. 27B is a schematic plan view of a high-frequency bandpass filter device according to a fifth modification. As shown in FIG. 27B, the high-frequency band-pass filter device includes a conductor case 80 similar to the high-frequency band-pass filter device of the third embodiment, and includes the TM dual-mode dielectric resonators R1 and R2. , R3 are juxtaposed at predetermined intervals. In the high-frequency band-pass filter device according to the fifth modified example, the input connector 41 is provided at the center of one end face of the conductor case 80 adjacent to the TM dual-mode dielectric resonator R1. Reference numeral 42 is provided at the center of the end face of the conductor case 80 adjacent to the TM dual-mode dielectric resonator R3.
[0133]
FIG. 24 is a circuit diagram showing an equivalent circuit of a high-frequency bandpass filter device according to a fifth modification. In the equivalent circuit of FIG. 24, the same element as that of the equivalent circuit of FIG. 17 is connected to each connection point of the distributed constant lines LN1a to LN8a of the TM dual mode dielectric resonator R1. Elements similar to those in the equivalent circuit of FIG. 17 are connected to each connection point of the distributed constant lines LN1b to LN8b of the circuit R2. Further, elements similar to those of the equivalent circuit of FIG. 23 are connected to connection points of the distributed constant lines LN1c to LN8c of the TM dual mode dielectric resonator R3.
[0134]
As described above, the high-frequency band-pass filter device according to the fifth modification includes three TM dual-mode dielectric resonators R1, R2, and R3. In the high-frequency band-pass filter device, the input connector 41 is capacitively coupled to the x-axis resonator of the TM dual-mode dielectric resonator R1 by the input capacitor 22. The x-axis resonator of the TM dual-mode dielectric resonator R1 is formed by the notches 11a and 12a provided in the dielectric 1a and the notches 13a and 14a provided in the dielectric 2a. It is electromagnetically coupled with the y-axis resonator of the heavy mode dielectric resonator R1. The y-axis resonator of the TM dual-mode dielectric resonator R1 is inductively coupled to the y-axis resonator of the TM dual-mode dielectric resonator R2 by the coupling loop inductor 21. The y-axis resonator of the TM dual-mode dielectric resonator R2 is formed by the notches 11b and 12b provided in the dielectric 1b and the notches 13b and 14b provided in the dielectric 2b. It is electromagnetically coupled with the x-axis resonator of the dielectric resonator R2. Further, the x-axis resonator of the TM dual-mode dielectric resonator R2 is capacitively coupled to the x-axis resonator of the TM dual-mode dielectric resonator R3 by the coupling capacitor C8. The x-axis resonator of the TM dual mode dielectric resonator R3 is formed by the notches 11c and 12c provided in the dielectric 1c and the notches 13c and 14c provided in the dielectric 2c. It is electromagnetically coupled with the y-axis resonator of the dielectric resonator R3. Further, the y-axis resonator of the TM dual-mode dielectric resonator R3 is inductively coupled to the output connector 42 by the coupling loop inductor 43. With the above configuration, a six-stage high-frequency bandpass filter device including the TM dual-mode dielectric resonators R1, R2, and R3 between the input connector 41 and the output connector 42 is configured. Thus, the high-frequency band-pass filter device according to the fifth modified example operates in the same manner as the high-frequency band-pass filter device according to the third embodiment, and has the same effect.
[0135]
The operation of various high-frequency band-pass filter devices according to the present invention, including the high-frequency band-pass filter device of the third embodiment described above and the high-frequency band-pass filter devices of the second to fifth modifications, will be described with reference to FIGS. This will be described with reference to FIG. 28 to 33, R1x, R2x, and R3x denote the x-axis resonator of the TM dual-mode dielectric resonator R1, the x-axis resonator of the TM dual-mode dielectric resonator R2, and the TM double-mode dielectric resonator R3, respectively. Where R1y, R2y, and R3y are the y-axis resonator of the TM dual-mode dielectric resonator R1, the y-axis resonator of the TM dual-mode dielectric resonator R2, and the TM double-mode dielectric resonator, respectively. R3 represents a y-axis resonator. 28 to 33, C-coupling refers to capacitive coupling, and L-coupling refers to inductive coupling. In the following description, the x-axis resonator of the TM dual-mode dielectric resonator R1, the x-axis resonator of the TM dual-mode dielectric resonator R2, and the x-axis resonator of the TM dual-mode dielectric resonator R3 each have a resonance. Are called a resonator R1x, a resonator R2x, and a resonator R3x, and are a y-axis resonator of the TM dual-mode dielectric resonator R1, a y-axis resonator of the TM double-mode dielectric resonator R2, and a TM-mode resonator R3. The y-axis resonators are referred to as a resonator R1y, a resonator R2y, and a resonator R3y, respectively.
[0136]
FIG. 28 is an explanatory diagram of the operation of the high-frequency band-pass filter device according to the third embodiment. In FIG. 28, the input connector 41 is capacitively coupled to a resonator R1x, and the resonator R1x is electromagnetically coupled to the resonator R1y. Next, the resonator R1y is inductively coupled to the resonator R2y. The resonator R2y is electromagnetically coupled to the resonator R2x, and the resonator R2x is capacitively coupled to the resonator R3x. The resonator R3x is electromagnetically coupled to the resonator R3y, and the resonator R3y is capacitively coupled to the output connector 42. When the resonators are coupled between the input connector 41 and the output connector 42 as described above, the TM dual-mode dielectric resonators R1, R2, and R3 are connected to the conductor case 80 as shown in FIG. The input connector 41 is provided side by side in the longitudinal direction, the input connector 41 is provided on the end face of the case 80 adjacent to the TM dual-mode dielectric resonator R1, and the output connector 42 is provided on the TM dual-mode dielectric resonator R3. On the side of the conductor case 80 on the y-axis.
[0137]
28, when the input connector 41 is inductively coupled to the resonator R1x and the output connector 42 is inductively coupled to the resonator R3y instead of the third embodiment, the coupling shown in FIG. The method is the same as the coupling method of the high frequency bandpass filter device of the fourth modification shown in FIG. When the resonators are coupled between the input connector 41 and the output connector 42 as described above, the TM dual-mode dielectric resonators R1, R2, and R3 are juxtaposed in the longitudinal direction of the conductor case 80. The input connector 41 is provided on a side surface of the conductor case 80 on the y-axis of the TM dual-mode dielectric resonator R1, and the output connector 42 is adjacent to the TM dual-mode dielectric resonator R3. It is provided on the end face of the conductor case 80.
[0138]
28, when the input connector 41 is capacitively coupled to the resonator R1x and the output connector 42 is inductively coupled to the resonator R3y instead of the third embodiment, the coupling shown in FIG. The method is the same as the coupling method of the high-frequency band-pass filter device of the fifth modification shown in FIG. When the resonators are coupled between the input connector 41 and the output connector 42 as described above, the TM dual-mode dielectric resonators R1, R2, and R3 are juxtaposed in the longitudinal direction of the conductor case 80, The input connector 41 is provided on an end face of the conductor case 80 adjacent to the TM dual-mode dielectric resonator R1, and the output connector 42 is connected to the conductor case 80 adjacent to the TM dual-mode dielectric resonator R3. It is provided on the end face.
[0139]
Further, in FIG. 28, when the input connector 41 is inductively coupled to the resonator R1x and the output connector 42 is capacitively coupled to the resonator R3y, the TM dual-mode dielectric resonators R1, R2, and R3 become The input connector 41 is provided on the side surface of the conductor case 80 on the y-axis of the TM dual-mode dielectric resonator R1, and the output connector 42 is It is provided on the side surface of the conductor case 80 on the y-axis of the TM dual mode dielectric resonator R3.
[0140]
FIG. 29 is an operation explanatory diagram configured using a coupling method different from the coupling method of FIG. In FIG. 29, the input connector 41 is capacitively coupled to a resonator R1y, and the resonator R1y is electromagnetically coupled to the resonator R1x. Next, the resonator R1x is capacitively coupled to the resonator R2x. The resonator R2x is electromagnetically coupled to the resonator R2y, and the resonator R2y is inductively coupled to the resonator R3y. The resonator R3y is electromagnetically coupled to the resonator R3x, and the resonator R3x is capacitively coupled to the output connector 42. When the resonators are coupled between the input connector 41 and the output connector 42 as described above, the TM dual-mode dielectric resonators R1, R2, and R3 are juxtaposed in the longitudinal direction of the conductor case 80. The input connector 41 is provided on a side surface of the conductor case 80 on the y-axis of the TM dual-mode dielectric resonator R1, and the output connector 42 is adjacent to the TM dual-mode dielectric resonator R3. It is provided on the end face of the conductor case 80.
[0141]
In FIG. 29, when the input connector 41 and the resonator R1y are inductively coupled instead of capacitive coupling, and the output connector 42 and the resonator R3x are inductively coupled instead of capacitive coupling, the input connector 41 is provided on the end face of the conductor case 80 adjacent to the TM dual-mode dielectric resonator R1, and the output connector 42 is connected to the conductor case 80 on the y-axis of the TM dual-mode dielectric resonator R3. Provided on the side.
[0142]
Further, in FIG. 29, when the output connector 42 and the resonator R3x are inductively coupled instead of capacitive coupling, the input connector 41 is connected to the conductor case 80 on the y-axis of the TM dual-mode dielectric resonator R1. The output connector 42 is provided on the side surface of the conductor case 80 on the y-axis of the TM dual mode dielectric resonator R3.
[0143]
Further, in FIG. 29, when the input connector 41 and the resonator R1y are inductively coupled instead of capacitive coupling, the input connector 41 is connected to the conductor case 80 adjacent to the TM dual-mode dielectric resonator R1. The output connector 42 is provided on an end face, and the output connector 42 is provided on an end face of the conductor case 80 adjacent to the TM dual-mode dielectric resonator R3.
[0144]
As described in detail above, the TM dual-mode dielectric resonator according to the present invention includes a coupling method between the TM dual-mode dielectric resonator R1 and the TM dual-mode dielectric resonator R2; One of the coupling methods between the mode dielectric resonator R2 and the TM dual-mode dielectric resonator R3 is capacitive coupling, and the other is inductive coupling, so that the three TM dual-mode dielectric resonators are used. R1, R2, and R3 can be arranged side by side in a line. That is, in the high-frequency band-pass filter device in which the TM dual-mode dielectric resonators are arranged side by side in a row, the x-axis or y-axis resonator of each adjacent TM dual-mode dielectric resonator is inductively coupled and capacitively coupled. Are used to make adjacent coupling methods different from each other.
[0145]
Further, the TM dual mode dielectric resonator according to the present invention includes a coupling method between the input connector 41 and the TM dual mode dielectric resonator R1, and a method for coupling the TM dual mode dielectric resonator R1 with the TM dual mode dielectric resonator. The input connector 41 can be provided on the end face of the conductor case 80 by using one of the coupling methods with the mode dielectric resonator R2 as capacitive coupling and the other as inductive coupling. Similarly, a coupling method between the output connector 42 and the TM dual-mode dielectric resonator R1, and a coupling method between the TM dual-mode dielectric resonator R1 and the TM dual-mode dielectric resonator R2 The output connector 42 can be provided on the end face of the conductor case 80 by making one of them a capacitive coupling and the other an inductive coupling. In addition, a coupling method between the input connector 41 and the TM dual-mode dielectric resonator R1, and a coupling method between the TM dual-mode dielectric resonator R1 and the TM dual-mode dielectric resonator R2. Are both capacitively coupled or inductively coupled, so that the input connector 41 can be provided on the side surface of the conductor case 80. Similarly, a coupling method between the output connector 42 and the TM dual-mode dielectric resonator R3 and a coupling method between the TM dual-mode dielectric resonator R2 and the TM dual-mode dielectric resonator R3 Are capacitively coupled or inductively coupled, so that the output connector 42 can be provided on the side surface of the conductor case 80.
[0146]
In the above description, the high-frequency band-pass filter device including three TM dual-mode dielectric resonators R1, R2, and R3 has been described. However, the high-frequency band-pass filter device including more TM dual-mode dielectric resonators. Can be similarly explained. That is, a coupling method between the input connector 41 and the first-stage TM dual-mode dielectric resonator, a coupling method between adjacent TM dual-mode dielectric resonators, and a final-stage TM double-mode dielectric resonator. And the coupling method between the output connector 42 and the output connector 42 are configured so that inductive coupling and capacitive coupling are alternated, so that the TM dual-mode dielectric resonators TM can be provided in a line. Further, the input connector 41 and the output connector 42 can be provided on both end surfaces of the conductor case 80.
[0147]
FIG. 30 is a diagram illustrating the operation of the high-frequency band-pass filter device according to the second modification. In FIG. 30, the input connector 41 is capacitively coupled to a resonator R1x, and the resonator R1x is electromagnetically coupled to the resonator R1y. Next, the resonator R1y is inductively coupled to the resonator R2y. The resonator R2y is electromagnetically coupled to the resonator R2x, and the resonator R2x is inductively coupled to the resonator R3x. The resonator R3x is electromagnetically coupled to the resonator R3y, and the resonator R3y is capacitively coupled to the output connector 42. When the respective resonators are coupled between the input connector 41 and the output connector 42 as described above, as shown in FIG. 26A, the TM dual-mode dielectric resonators R1, R2, and R3 are L-shaped. Provided in the mold. The input connector 41 is provided on an end face 81ta of the conductor case 81 adjacent to the TM dual-mode dielectric resonator R1, and the output connector 42 is provided on an end face 81tb adjacent to the TM dual-mode dielectric resonator R3. Is provided.
[0148]
In FIG. 30, instead of the second modification, the input connector 41 and the resonator R1x are inductively coupled, and the output connector 42 and the resonator R3y are inductively coupled. The output connector 42 is provided on the side surface 81sa of the conductor case 81 on the y-axis of the TM dual-mode dielectric resonator R1, and the side surface of the conductor case 81 on the x-axis of the TM dual-mode dielectric resonator R3. 81 sb.
[0149]
In FIG. 30, instead of the second modification, when the input connector 41 and the resonator R1x are inductively coupled, the input connector 41 is connected to the TM dual-mode dielectric resonator R1 on the y-axis. The output connector 42 is provided on a side surface 81sa of the conductor case 81, and the output connector 42 is provided on an end surface 81tb adjacent to the TM dual-mode dielectric resonator R3.
[0150]
In FIG. 30, instead of the second modification, when the output connector 42 and the resonator R3y are inductively coupled, the input connector 41 is connected to the conductor adjacent to the TM dual-mode dielectric resonator R1. The output connector 42 is provided on an end surface 81ta of the case 81, and the output connector 42 is provided on a side surface 81sb on the x-axis of the TM dual-mode dielectric resonator R3.
[0151]
FIG. 31 is a diagram illustrating the operation of a high-frequency band-pass filter device configured using a combination of coupling methods different from that in FIG. 31 differs from FIG. 30 in that the resonator R1y is capacitively coupled to the resonator R2y and that the resonator R2x is capacitively coupled to the resonator R3x. When the resonators are coupled between the input connector 41 and the output connector 42 as described above, the TM dual-mode dielectric resonators R1, R2, and R3 are provided in an L-shape. The input connector 41 is provided on a side surface 81sa of the conductor case 81 on the y-axis of the TM dual-mode dielectric resonator R1, and the output connector 42 is connected to the x of the TM dual-mode dielectric resonator R3. It is provided on the side surface 81sb of the conductor case 81 on the shaft.
[0152]
In FIG. 31, when the input connector 41 and the resonator R1x are inductively coupled instead of the capacitive coupling, and the output connector 42 and the resonator R3y are inductively coupled instead of the capacitive coupling, the input connector 41 becomes The output connector 42 is provided on an end face 81tb adjacent to the TM dual-mode dielectric resonator R3. The output connector 42 is provided on an end face 81ta of the conductor case 81 adjacent to the mode dielectric resonator R1.
[0153]
In FIG. 31, when the input connector 41 and the resonator R1x are inductively coupled instead of capacitive coupling, the input connector 41 is provided on the end face 81ta of the conductor case 81 adjacent to the TM dual-mode dielectric resonator R1. The output connector 42 is provided on a side surface 81sb of the conductor case 81 on the x-axis of the TM dual-mode dielectric resonator R3.
[0154]
In FIG. 31, when the output connector 42 and the resonator R3y are inductively coupled instead of capacitive coupling, the input connector 41 is connected to the side surface 81sa of the conductor case 81 on the y-axis of the TM dual-mode dielectric resonator R1. The output connector 42 is provided on an end face 81tb adjacent to the TM dual-mode dielectric resonator R3.
[0155]
FIG. 32 is a diagram illustrating the operation of a high-frequency bandpass filter device configured using a combination of different coupling methods. In FIG. 32, the input connector 41 is capacitively coupled to a resonator R1y, and the resonator R1y is electromagnetically coupled to the resonator R1x. Next, the resonator R1x is inductively coupled to the resonator R2x. The resonator R2x is electromagnetically coupled to the resonator R2y, and the resonator R2y is inductively coupled to the resonator R3y. The resonator R3y is electromagnetically coupled to the resonator R3x, and the resonator R3x is capacitively coupled to the output connector 42. When the resonators are coupled between the input connector 41 and the output connector 42 as described above, the TM dual-mode dielectric resonators R1, R2, and R3 are provided in an L-shape. The input connector 41 is provided on an end face 81ta of the conductor case 81 adjacent to the TM dual-mode dielectric resonator R1, and the output connector 42 is provided on an end face 81tb adjacent to the TM dual-mode dielectric resonator R3. Is provided.
[0156]
In FIG. 32, when the input connector 41 and the resonator R1y are inductively coupled instead of capacitive coupling, and the output connector 42 and the resonator R3x are inductively coupled instead of capacitive coupling, the input connector 41 The output connector 42 is provided on the side surface 81sa of the case 81sb on the x-axis of the TM dual-mode dielectric resonator R3 on the y-axis of the dual-mode dielectric resonator R1. Can be
[0157]
In FIG. 32, when the input connector 41 and the resonator R1y are inductively coupled instead of capacitive coupling, the input connector 41 becomes a side surface 81sa of the conductor case 81 on the y-axis of the TM dual-mode dielectric resonator R1. The output connector 42 is provided on an end face 81tb adjacent to the TM dual-mode dielectric resonator R3.
[0158]
In FIG. 32, when the output connector 42 and the resonator R3x are inductively coupled instead of capacitive coupling, the input connector 41 is provided on an end face 81ta of the case 81 adjacent to the TM dual-mode dielectric resonator R1. The output connector 42 is provided on a side surface 81sb of the conductor case 81 on the x-axis of the TM dual-mode dielectric resonator R3.
[0159]
FIG. 33 is an operation explanatory diagram of the high-frequency band-pass filter device of the third modification. FIG. 33 differs from FIG. 32 in that the resonator R1x is capacitively coupled to the resonator R2x and that the resonator R2y is capacitively coupled to the resonator R3y. When the resonators are coupled between the input connector 41 and the output connector 42 as described above, the TM dual-mode dielectric resonators R1, R2, and R3 are provided in an L-shape. The input connector 41 is provided on a side surface 81sa of the conductor case 81 on the y-axis of the TM dual-mode dielectric resonator R1, and the output connector 42 is provided on the x-axis of the TM dual-mode dielectric resonator R3. It is provided on the side surface 81sb of the conductor case 81 above.
[0160]
In FIG. 33, when the input connector 41 and the resonator R1y are inductively coupled instead of the capacitive coupling, and the output connector 42 and the resonator R3x are inductively coupled instead of the capacitive coupling, the input connector 41 becomes The output connector 42 is provided on an end face 81tb adjacent to the TM dual-mode dielectric resonator R3. The output connector 42 is provided on an end face 81ta of the conductor case 81 adjacent to the mode dielectric resonator R1.
[0161]
In FIG. 33, when the input connector 41 and the resonator R1y are inductively coupled instead of capacitive coupling, the input connector 41 is provided on the end face 81ta of the conductor case 81 adjacent to the TM dual-mode dielectric resonator R1. The output connector 42 is provided on a side surface 81sb of the conductor case 81 on the x-axis of the TM dual-mode dielectric resonator R3.
[0162]
In FIG. 33, when the output connector 42 and the resonator R3x are inductively coupled instead of capacitive coupling, the input connector 41 is connected to the side surface 81sa of the conductor case 81 on the y-axis of the TM dual-mode dielectric resonator R1. The output connector 42 is provided on an end face 81tb adjacent to the TM dual-mode dielectric resonator R3.
[0163]
As described in detail above, the high-frequency band-pass filter device according to the present invention includes a coupling method between the TM dual-mode dielectric resonator R1 and the TM dual-mode dielectric resonator R2; The capacitive coupling between the body resonator R2 and the coupling method between the TM dual-mode dielectric resonator R3 and the TM dual-mode dielectric resonator R1 and the TM dual-mode dielectric resonator R2 And the coupling method between the TM dual-mode dielectric resonator R2 and the TM dual-mode dielectric resonator R3 are both inductively coupled to form the three TM dual-mode dielectric resonators. The devices R1, R2, and R3 can be configured in an L-shape. That is, in the high-frequency band-pass filter device in which each TM dual-mode dielectric resonator is formed in an L-shape, the x-axis or y-axis resonator of each adjacent TM dual-mode dielectric resonator has inductive coupling and capacitive coupling. Are used to make adjacent coupling methods the same.
[0164]
Further, the high-frequency band-pass filter device according to the present invention includes a coupling method between the input connector 41 and the TM dual-mode dielectric resonator R1, a coupling method between the TM dual-mode dielectric resonator R1, and the TM dual-mode dielectric resonator. The input connector 41 can be provided on the end face of the conductor case 80 by setting one of the coupling methods to the body resonator R2 as capacitive coupling and the other coupling method as inductive coupling. . Similarly, a coupling method between the output connector 42 and the TM dual-mode dielectric resonator R3 and a coupling method between the TM dual-mode dielectric resonator R2 and the TM dual-mode dielectric resonator R3 The output connector 42 can be provided on the end face of the conductor case 80 by making one of them a capacitive coupling and the other an inductive coupling. In addition, a coupling method between the input connector 41 and the TM dual-mode dielectric resonator R1, and a coupling method between the TM dual-mode dielectric resonator R1 and the TM dual-mode dielectric resonator R2. Are both capacitively coupled or inductively coupled, so that the input connector 41 can be provided on the side surface of the conductor case 80. Similarly, a coupling method between the output connector 42 and the TM dual-mode dielectric resonator R3 and a coupling method between the TM dual-mode dielectric resonator R2 and the TM dual-mode dielectric resonator R3 Are capacitively coupled or inductively coupled, so that the output connector 42 can be provided on the side surface of the conductor case 80.
[0165]
As described in detail above, the TM dual-mode dielectric resonator according to the present invention includes a coupling method between the TM dual-mode dielectric resonator R1 and the TM dual-mode dielectric resonator R2; The coupling method between the mode dielectric resonator R2 and the TM dual-mode dielectric resonator R3 is configured by variously combining the capacitive coupling and the inductive coupling to constitute the high-frequency band-pass filter device. , 81 can be formed in various shapes such as a rectangular parallelepiped or an L-shape, and the input connector 41 and the output connector 42 can be provided on the end face or on the side face in the above-described shape.
[0166]
<Other modifications>
In the first to third embodiments and the first to fifth modifications, the dielectrics 1, 2, 101, and 102 are formed in a columnar shape or a columnar shape having a square end surface. Not limited to this, the cross-sectional shape of the end face may be formed in an elliptical shape or a columnar shape in which the number of sides is an even polygon. Even with the above-described configuration, the same operation as in the first to third embodiments and the first to fifth modifications is performed, and the same effect is obtained.
[0167]
The first to third embodiments and the first to fifth modifications described above are configured using the cavities 8a, 80a, 80b, 80c, and 108a formed by the conductor cases 8, 80, and 108. The present invention is not limited to this, and a conductor plate contacting the upper end surfaces of the dielectrics 1, 1a, 1b, 1c, 101 and a conductor plate contacting the lower end surfaces of the dielectrics 2, 2a, 2b, 2c, 102 may be used. Each of the conductor plates is formed such that the edge of the conductor plate is separated from the outer periphery of the dielectrics 1, 2, 1a, 2a, 1b, 2b, 1c, 2c, 101, 102 by a predetermined distance. You may. With the configuration described above, the free space between the two conductor plates acts as an attenuation region in which electromagnetic field energy is attenuated and electromagnetic waves are not propagated. Therefore, the first to third embodiments and the first to third embodiments have the following advantages. The same operation as that of the modified example 5 is performed and the same effect is obtained.
[0168]
In the above-described first to third embodiments and the first to fifth modifications, degeneration is separated by forming the cutouts on the side surfaces of the dielectrics 1 and 2, but the present invention is not limited to this. However, the present invention is not limited to this, and a portion having a different dielectric constant may be provided on the side surface of the dielectric to separate degeneracy, or a convex portion may be provided on the side surface of the dielectric 1, 2 in place of the cutout. . Even with the above-described configuration, the same operation as in the first to third embodiments and the first to fifth modifications is performed, and the same effect is obtained.
[0169]
In the first to third embodiments and the first to fifth modifications, the resonance mode of the TM dual-mode dielectric resonator is the fundamental mode and the TM degenerates twice.₁₁₀It is preferable to use the mode, but the present invention is not limited to this, and the TM₂₁₀Mode and TM₃₁₀Another double degenerate mode such as a mode may be used. Even with the above configuration, the same operation as in the third embodiment and the second to fifth modifications is performed, and the same effect is obtained.
[0170]
In the high-frequency band-pass filter devices of the third embodiment and the second to fifth modifications, the three TM dual-mode dielectric resonators R1, R2, and R3 are used. However, the present invention is not limited to this, and may be configured using at least one of the TM dual mode dielectric resonators. Even with the above configuration, the same operation as in the third embodiment and the second to fifth modifications is performed, and the same effect is obtained. Also in this case, the coupling between the input connector 41 and the TM dual-mode dielectric resonator and the coupling between the output connector 42 and the TM dual-mode dielectric resonator are configured by various combinations of capacitive coupling and inductive coupling. The side on which the input connector 41 and the output connector 42 are provided can be freely selected.
[0171]
In the TM dual-mode dielectric resonator and the high-frequency band-pass filter device according to the first to third embodiments and the first to fifth modifications, the torus electrodes 4, 4a, 4b, 4c, and 104 are flat plates, respectively. Although formed integrally with the electrodes 3, 3a, 3b, 3c, 103, the present invention is not limited to this, and the plate electrodes 3, 3a to which the torus electrodes 4, 4a, 4b, 4c, 104 are not connected. , 3b, 3c, and 103. Further, in place of the plate electrodes 3, 3a, 3b, 3c, 103 to which the torus electrodes 4, 4a, 4b, 4c, 104 are connected, as shown in FIG. The plate electrode 3d formed in a shape may be provided so that the semicircular portion 3da of the plate electrode 3d protrudes from the outer peripheral surfaces of the dielectrics 1 and 2. Further, in place of the plate electrodes 3, 3a, 3b, 3c, 103 to which the torus electrodes 4, 4a, 4b, 4c, 104 are connected, as shown in FIG. May be configured such that the trapezoidal portion 3ea of the plate electrode 3e protrudes from the outer peripheral surfaces of the dielectrics 1 and 2. Even with the above configuration, the current can be dispersed to the portion 3da formed on the plate electrode 3d or the portion 3ea formed on the plate electrode 3e, so that the first to third embodiments and the first embodiment can be used. The same operation as that of the fifth to fifth modifications is performed, and the same effect is obtained.
[0172]
In the TM dual-mode dielectric resonator and the high-frequency band-pass filter device according to the first to third embodiments and the first to fifth modifications, the torus electrodes 4, 4a, 4b, 4c, and 4c each have a circular longitudinal section. However, the present invention is not limited to this, and as shown in FIG. 34 (c), it may be configured using an electrode 4d having a vertical section formed in an octagonal shape. May be configured using electrodes formed in a polygon other than the octagon. Even with the above configuration, since the current can be dispersed to the outer peripheral portion of the electrode 4d, the same operation as the first to third embodiments and the first to fifth modifications is performed and the same operation is performed. Has an effect.
[0173]
In the TM dual mode dielectric resonator and the high-frequency bandpass filter device according to the first to third embodiments and the first to fifth modifications, the diameter of the plate electrodes 3, 3a, 3b, 3c and 103 is determined by the dielectric. The inner circumferences of the torus electrodes 4, 4a, 4b, 4c, 104 connected to the plate electrodes 3, 3a, 3b, 3c, 103, respectively, are set so as to be the same as the diameters of the bodies 1, 2, respectively. 1 and 2 were configured to be in contact with the outer peripheral surface. However, the present invention is not limited to this. As shown in FIG. 35A, the outer diameter of the torus electrode 4 obtained by adding the diameter of the plate electrode 3 and twice the diameter of the longitudinal section of the torus electrode 4 is: The diameter of the dielectrics 1 and 2 may be set to be the same, and the outer periphery of the torus electrode 4 may be configured to coincide with the outer peripheral surface of the dielectrics 1 and 2. Alternatively, as shown in FIG. 35B, only the outer semicircular portion of the longitudinal section of the torus electrode 4 may be configured to protrude from the outer peripheral surfaces of the dielectrics 1 and 2. Further, as shown in FIG. 35 (c), the diameter of the plate electrode 3 is set to be larger than the diameter of the dielectrics 1 and 2, so that the inner periphery of the torus electrode 4 is separated from the outer peripheral surfaces of the dielectrics 1 and 2. You may comprise. Even with the above-described configuration, the same operations and effects as those of the first to third embodiments and the first to fifth modifications are obtained.
[0174]
【The invention's effect】
The TM dual mode dielectric resonator according to claim 1 according to the present invention, wherein the flat electrode sandwiched between the first and second dielectrics and the end face of the first dielectric formed on the flat electrode. Since it has the first conductor plate and the second conductor plate formed on the end face of the second dielectric, it has a relatively high no-load Q, and is smaller and smaller than the conventional example. And it can be made thin.
[0175]
In the TM dual mode dielectric resonator according to the second aspect, since the first and second dielectrics have columnar shapes, they can be easily configured as compared with the case where a dielectric having another shape is used.
[0176]
The TM dual-mode dielectric resonator according to claim 3, further comprising the torus electrode connected to the entire periphery of the plate electrode, thereby reducing energy lost at the edge of the plate electrode. The no-load Q can be increased as compared with the case where the torus electrode is not provided.
[0177]
The TM dual mode dielectric resonator according to the fourth aspect includes the degenerate separation means, so that the double degenerate mode can be separated into two modes having different resonance frequencies.
[0178]
In the TM dual-mode dielectric resonator according to the fifth aspect, the notch is formed in a part of the outer circumference of the first and second dielectrics, so that the TM dual-mode dielectric resonator is provided with a double. The degenerated mode can be separated into two modes having different resonance frequencies.
[0179]
Since the TM dual mode dielectric resonator according to claim 6 includes the cavity, it is possible to increase the no-load Q as compared with a case where the cavity is not provided, and to reduce the variation of the resonance frequency. Can be reduced.
[0180]
A high-frequency band-pass filter device according to a seventh aspect of the present invention includes the TM dual-mode dielectric resonator according to the first to sixth aspects, and thus can be small and thin.
[0181]
The high-frequency band-pass filter device according to the eighth aspect includes at least two TM dual-mode dielectric resonators according to the first to sixth aspects and the coupling means. Can be relatively large.
[0182]
Since the high-frequency band-pass filter device according to the ninth aspect includes the coupling means for inductive coupling, the two TM dual-mode dielectric resonators adjacent to each other can be easily coupled.
[0183]
Since the high-frequency band-pass filter device according to the tenth aspect includes the coupling means for capacitively coupling, the two adjacent TM dual-mode dielectric resonators adjacent to each other can be easily coupled.
[Brief description of the drawings]
FIG. 1 is a partially broken perspective view of a TM dual mode dielectric resonator according to a first embodiment of the present invention.
FIG. 2 is a cross-sectional view taken along line A-A 'of the TM dual-mode dielectric resonator of FIG.
FIG. 3 is a plan view showing a current distribution on the upper surface of a plate electrode 3 of the TM dual mode dielectric resonator of FIG.
4 is a cross-sectional view showing an electric field distribution in a vertical cross section along the x-axis shown in FIG. 3 of the TM dual mode dielectric resonator of FIG.
5 is a cross-sectional view showing a magnetic field distribution in a vertical cross section along the y-axis shown in FIG. 3 of the TM dual-mode dielectric resonator of FIG.
6 is a graph showing the relationship between the value of the radius R of the torus electrode 4 and the no-load Q and the resonance frequency in the TM dual mode dielectric resonator of FIG.
FIG. 7 shows the dielectric constant ε of the dielectrics 1 and 2 in the TM dual mode dielectric resonator of FIG._rAre 20, 40, and 60, respectively, the diameter d of the dielectrics 1 and 2 and the resonance frequency f₀3 is a graph showing the relationship of FIG.
8 is a diagram showing the thickness h, the unloaded Q, and the resonance frequency f of the dielectrics 1 and 2 in the TM dual mode dielectric resonator shown in FIG. 1;₀3 is a graph showing the relationship of FIG.
FIG. 9 shows a relative dielectric constant ε of the dielectrics 1 and 2 in the TM dual mode dielectric resonator shown in FIG._rAnd no-load Q and relative permittivity ε_r6 is a graph showing the relationship between the volume of a TM dual mode dielectric resonator and that of a TM dual mode dielectric resonator.
FIG. 10 is a cross-sectional view of a TM dual mode dielectric resonator according to a second embodiment of the present invention.
11A is a plan view showing an odd mode current distribution on the surface of the plate electrode 3 of the TM dual mode dielectric resonator shown in FIG. 10, and FIG. 11B is a plan view showing the TM dual mode dielectric shown in FIG. FIG. 4 is a plan view showing an even mode current distribution on the surface of a plate electrode 3 of the body resonator.
FIG. 12 is a circuit diagram showing an equivalent circuit of the TM dual mode dielectric resonator of FIG.
FIG. 13 is a graph showing a ratio of a negative capacitance and a coupling coefficient of the TM dual mode dielectric resonator of FIG. 10;
14 is an output-end transmission coefficient S of the TM dual-mode dielectric resonator of the first embodiment of FIG. 1 from a frequency of 0.5 GHz to a frequency of 2.5 GHz.₂₁5 is a graph showing measured values of the above.
FIG. 15 is a longitudinal sectional view of a high-frequency bandpass filter device according to a third embodiment of the present invention.
16 is a cross-sectional view taken along line C-C 'of the high-frequency band-pass filter device of FIG.
17 is a circuit diagram showing an equivalent circuit of the high-frequency band-pass filter device of FIG.
18 is a graph showing an output-side transmission coefficient S of the high-frequency band-pass filter device of FIG. 15 calculated based on the equivalent circuit of FIG.₂₁And input end reflection coefficient S₁₁6 is a graph showing simulation results of FIG.
19 is an output-end transmission coefficient S of the high-frequency band-pass filter device of FIG.₁₂And input end reflection coefficient S₁₁5 is a graph showing measured values of the above.
20A is a longitudinal sectional view of a TM dual-mode dielectric resonator according to a first modification of the present invention, and FIG. 20B is a cross-sectional view of the TM dual-mode dielectric resonator of FIG. FIG.
FIG. 21 is a circuit diagram showing an equivalent circuit of a high-frequency bandpass filter device according to a second modification of the present invention.
FIG. 22 is a circuit diagram showing an equivalent circuit of a high-frequency bandpass filter device according to a third modification of the present invention.
FIG. 23 is a circuit diagram showing an equivalent circuit of a high-frequency bandpass filter device according to a fourth modification of the present invention.
FIG. 24 is a circuit diagram showing an equivalent circuit of a high-frequency bandpass filter device according to a fifth modification of the present invention.
FIG. 25 is a longitudinal sectional view of the TM dual-mode dielectric resonator of the second embodiment taken along line B-B ′ of FIG. 10;
FIG. 26A is a schematic plan view of a high-frequency bandpass filter device according to a second modification, and FIG. 26B is a schematic plan view of a high-frequency bandpass filter device according to a third modification.
FIG. 27A is a schematic plan view of a high-frequency band-pass filter device according to a fourth modification, and FIG. 27B is a schematic plan view of a high-frequency band-pass filter device according to a fifth modification.
FIG. 28 is a diagram illustrating the operation of the high-frequency band-pass filter device according to the third embodiment.
FIG. 29 is an operation explanatory diagram of a high-frequency bandpass filter device configured using a coupling method different from the coupling method shown in FIG. 28;
FIG. 30 is a diagram illustrating the operation of the high-frequency band-pass filter device according to the second modification.
FIG. 31 is a diagram illustrating the operation of a high-frequency band-pass filter device configured using a coupling method different from the coupling method shown in FIG. 30;
FIG. 32 is an operation explanatory diagram of a high-frequency band-pass filter device configured using a coupling method different from the coupling methods shown in FIGS. 30 and 31;
FIG. 33 is a diagram illustrating the operation of the high-frequency band-pass filter device according to the third modification.
FIG. 34 (a) is a partial longitudinal sectional view of a TM dual mode dielectric resonator constituted by using a plate electrode 3d different from the plate electrodes 3, 3a, 3b, 3c, 103, and FIG. (A) is a partial longitudinal sectional view of a TM dual-mode dielectric resonator constituted by using a plate electrode 3e different from that of (a), and (c) shows an electrode 4d having an octagonal longitudinal section instead of the torus electrode 4. FIG. 3 is a partial vertical cross-sectional view of a TM dual mode dielectric resonator configured using the same.
35 (a) is a partial longitudinal sectional view of a TM dual-mode dielectric resonator when a position where a torus electrode 4 is provided is changed, and FIG. 35 (b) is a position where a torus electrode 4 is further provided; FIG. 4 is a partial longitudinal sectional view of the TM dual mode dielectric resonator at the time, and FIG. 4C is a partial longitudinal sectional view of the TM dual mode dielectric resonator when the position where the torus electrode 4 is provided is further changed.
[Explanation of symbols]
R1, R2, R3 ... TM dual mode dielectric resonator,
1, 1a, 1b, 1c, 2, 2a, 2b, 2c, 101, 102 ... dielectric,
3,3a, 3b, 3c, 3d, 3e, 103 ... plate electrode,
4, 4a, 4b, 4c, 104 ... a torus electrode,
4d ... electrode,
6, 6a, 6b, 6c, 7, 7a, 7b, 7c, 106, 107 ... end face electrodes,
8, 80, 81, 108 ... conductor case,
8a, 80a, 80b, 80c, 108a ... cavities,
9a, 9b ... conductor plate,
11, 11a, 11b, 11c, 12, 12a, 12b, 12c, 13, 13a, 13b, 13c, 14, 14a, 14b, 14c...
21, 21a ... coupled loop inductor,
22 input capacitor,
23, 23a, 24, 24a ... coupling capacitors,
25 ... output capacitor,
26a, 26b, 26c ... coupling adjusting screw,
31 ... input loop inductor,
41 ... input connector,
42 ... output connector,
43 ... output loop inductor,
51, 52, 53, 54 ... frequency adjustment screw,
211, 212, 213, 214, 215, 234 ... conductor wires.

Claims (10)

First and second dielectrics each having a predetermined column shape having two end surfaces facing each other in parallel,
A plate electrode sandwiched between one end surface of the first dielectric and one end surface of the second dielectric and electromagnetically coupled to an input terminal or an output terminal ;
The edge of the first conductor plate is separated from the outer periphery of the other surface of the first dielectric by a predetermined distance, and a part of the first conductor plate is formed on the other surface of the first dielectric. A first conductor plate formed so as to contact the entire surface of
The edge of the second conductor plate is separated from the outer periphery of the other surface of the second dielectric by a predetermined distance, and a part of the second conductor plate is formed on the other surface of the second dielectric. And a second conductor plate formed so as to be in contact with the entire surface of the TM dual mode dielectric resonator. 2. The TM dual mode dielectric resonator according to claim 1, wherein said first and second dielectrics have a cylindrical shape. An electrode formed to be connected to the entire outer periphery of the plate electrode and dispersing a current concentrated on the outer periphery of the plate electrode when the TM dual-mode dielectric resonator is excited; 3. The TM dual mode dielectric resonator according to claim 1, wherein: By making the permittivity of a part of the outer periphery of the first and second dielectrics different from the permittivity of the part other than the part, the double degenerate modes in the TM dual mode dielectric resonator are different from each other. 4. The TM dual mode dielectric resonator according to claim 1, further comprising degenerate separation means for separating into two modes having a frequency. 5. The TM dual mode dielectric resonator according to claim 4, wherein said degenerate separation means is a notch formed in a part of an outer periphery of said first and second dielectrics. 10. A structure comprising the first conductor plate and the second conductor plate, further comprising a cavity for confining the electromagnetic field of the TM dual mode dielectric resonator in the cavity. Item 6. A TM dual-mode dielectric resonator according to one of Items 1 to 5. A TM dual-mode dielectric resonator according to any one of claims 1 to 6,
An input terminal for inputting a high-frequency signal to the TM dual-mode dielectric resonator,
An output terminal for outputting a high-frequency signal output from the TM dual-mode dielectric resonator , wherein the input terminal and the output terminal are electromagnetically coupled to the plate electrode. apparatus. At least two TM dual-mode dielectric resonators according to claim 1;
Coupling means for electromagnetically coupling to the plate electrode and coupling the two adjacent TM dual-mode dielectric resonators to each other;
An input terminal that is electromagnetically coupled to the plate electrode and that inputs a high-frequency signal to the TM dual-mode dielectric resonator;
An output terminal that is electromagnetically coupled to the plate electrode and outputs a high-frequency signal output from the TM dual-mode dielectric resonator. At least one high-frequency band-pass filter apparatus according to claim 8, wherein the inductive coupling thus be coupled to the plate electrode of said coupling means. At least one high-frequency band-pass filter apparatus according to claim 8, wherein the binding to the thus the plate electrode in the capacitive coupling of said coupling means.

Images