JP3287540B2 - Digital detection method and its detector - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a digital signal detection method and a detector capable of receiving various modulated signals and transmission signals of various code transmission rates.

[0002]

2. Description of the Related Art In order to realize multimedia communication in the future, there is a demand for transmitting data, voice, and images over the same transmission path. As a means for efficiently transmitting data, voice, and images by digital wireless communication, a method using an optimal code transmission rate and an optimal modulation / demodulation method can be considered. For example, in mobile communication, a base station can provide a still image such as a television or a data bank, and the mobile station can perform normal voice communication and can extract a still image from the base station with simple operation and simple equipment. desired. In this case, QPSK modulation is usually used in voice communication. However, at the time of still image transmission, it is necessary to transmit a larger amount of information compared to voice, so it is necessary to use multi-level modulation such as QAM. In this case, as shown in FIG. 1A, for example, a QAM modulation transmitter 11, a PSK modulation transmitter 12, and an FSK modulation transmitter 13 are provided as a transmitter group 10 on the transmission side for each modulation and demodulation method, and the receiver is QAM receiver 14, PSK receiver 1 as group 20
5, a method of preparing an independent transmitter and a receiver for each modulation scheme so as to provide an FSK receiver 16, and a transmitter as shown in FIG. 1B in which a QAM modulation transmitter 11, a PSK modulation transmitter 12, The transmitter group 10 of the FSK modulation transmitter 13 is provided, but the QAM detector 2 is provided in one receiver 21 on the receiving side.
2. It is conceivable to prepare a device provided with a PSK detector 23 and an FSK detector 24. To have a plurality of detectors in the same wireless device as shown in FIG.
A method of incorporating a detector dedicated to the system is conceivable.

There are two types of mobile communication services, one using the 800 MHz band and the other using the 1500 MHz band. At present, only one of them can be used. However, if both bands can be easily switched for both configuration and operation, 1500 MHz
Utilizing the property of higher linearity of radio wave than 800MHz band, 1500MHz band is used indoors and closed space, 800MHz band outdoors.
By using the z band, co-channel interference can be reduced.

However, the device configuration shown in FIG.
Since a plurality of detectors are independently built-in, the device configuration becomes large and complicated. In addition, in the case of digital wireless communication for transmitting data, voice, and images by dynamically changing the demodulation method and carrier frequency, since each detector is independent,
It is difficult to switch the detector instantaneously. The receiver 21 performs quadrature demodulation of the received signal. For this purpose, it is necessary to generate a local oscillation signal synchronized with the carrier of the input received signal.
When _{f 1, f 2, f 3} , f 4 and temporally sequentially changes as shown in, there is a need to change the frequency of the local oscillation signal in response thereto. For this reason, conventionally, as shown in FIG. 2B, the oscillation frequency of the local oscillator 25 of the phase locked loop type is represented by f ₁ , f ₂ , f ₃ , f ₃ as shown by the local oscillators 25 ₁ , 25 ₂ , 25 ₃ , 25 _{4. 4} and the switching means 17 are sequentially switched, the output of the switched local oscillator and the input modulated wave signal are multiplied by the multiplier 18 and passed through the filter 19 to obtain the baseband signal. In the phase-locked loop local oscillator 25, the frequency switching speed was several milliseconds even when using a digital loop preset type frequency synthesizer. For this reason, it was not possible to sufficiently respond to frequency switching during a call.

[0005] Also, the code transmission rate of the received signal is, for example, as shown in FIG.
As shown in FIG. 2C, when the signal temporally changes to B ₁ , B ₂ , B ₃ , and B ₄ , a filter for filtering the output of the quadrature demodulator is conventionally replaced with a switch 2 as shown in FIG.
Filters 26 ₁ , 2 adapted to the transmission rate by 7, 28
Are sequentially switched to the 6 _2, 26 _3, 26 _4. As described above, since the filter is configured by hardware, the switching speed of the filter cannot be increased due to the transient characteristics of the filter element.

[0006]

The object of the present invention is to
Provided is a digital signal detection method and a detector that realizes compatibility with a plurality of modulation / demodulation and a plurality of local oscillation frequencies and a plurality of code transmission speeds by sharing various functions used for a plurality of detection systems. Is to do. It is another object of the present invention to provide a digital signal detection method and a detector capable of responding to a high speed change of a modulation format and a change of a code transmission speed.

[0007]

A feature of the present invention resides in that a baseband signal is obtained by digital signal processing implemented by software from an analog-to-digital converted reception signal.

According to the present invention, the received modulated wave signal that has been converted from analog to digital is subjected to quadrature demodulation in a quadrature demodulation step, and the result of the demodulation is filtered in a filter step to obtain a baseband signal. , At least one processing variable in at least one of the quadrature demodulation step and the filter step is changed in the control step.

In the quadrature demodulation step, the digital signal of the input modulated wave is subjected to n-point interpolation processing in the interpolation step (n
Is a real number greater than or equal to 1). The interpolation processing result is complex-multiplied by a local oscillation signal in a multiplication step, and the multiplication result is subjected to n-point thinning processing in a thinning step. The processing variables that can be changed in this quadrature demodulation step are the local oscillation frequency, its amplitude and phase, and the number of n.

In the filter step, the result of the orthogonal demodulation operation is smoothed by the smoothing processing step to reduce the number of sample points, and the result of the smoothing operation is subjected to the band limitation operation by the digital filter step. . The processing variables in this filter step are the number of smoothing points and the characteristics of the digital filter. Further, the gain of the input modulated wave signal is controlled by an automatic gain adjuster and supplied to an analog / digital converter as a signal within a predetermined level range.

The various processes described above are performed by the microprocessor decoding and executing the program.

[0012]

FIG. 3 shows an embodiment of a digital signal detector according to the present invention. The analog signal received at the input terminal 30 is controlled so as to suppress the amplitude fluctuation of the received signal within a certain range by the amplification gain of the automatic gain adjuster 31 after passing through a band filter (not shown). An analog signal output from the automatic gain adjuster 31 is converted into a digital signal by an analog / digital converter 32. The digitally converted received signal is demodulated by quadrature demodulation means 33 and further spectrally shaped by filter processing by digital filter means 34, thereby obtaining a demodulated digitized baseband signal at output terminal 40. Can be. The baseband signal is determined by a determination unit 39 for each of the in-phase component and the quadrature component.
The determination is made in the symbol period, and further, which signal point is determined based on the determination results. That is, for example, in the case of a QPSK signal, it is determined whether each of the in-phase component and the quadrature component is +1 or −1, and both determination results are any of the four signal points on the IQ coordinate. Is determined.

In FIG. 3, the quadrature demodulation means 33 and the digital filter means 34 are realized by software using digital frequency as an argument (variable) by using a sampling frequency, a code transmission speed, a modulation format, and a local oscillation frequency as arithmetic processing for a digital signal. . The control means 35 includes software for controlling the automatic gain adjuster 31, the quadrature demodulation means 33, and the digital filter means 34. Automatic gain adjuster 31
The operation of the control means 35 changes the amplification gain so that the amplitude fluctuation of the baseband signal is suppressed within a certain range. The control of the control means 35 for the quadrature demodulation means 33 and the digital filter means 34 is performed in accordance with changes in the sampling frequency, code transmission speed, modulation format, and local oscillation frequency of the digitized modulated wave signal.
3 and the argument set in the digital filter means 34 are changed. That is, the setting input means 36 such as a keyboard
Is connected to the control means 35, and the setting input means 36 is provided with a plurality of keys respectively indicating a predetermined number of sampling frequencies, a number of code transmission rates, and a number of local oscillation frequencies. After operating any key for each parameter, or operating any key indicating the sampling frequency, code transmission speed, local oscillation frequency, it is possible to set and input a numerical value by using the ten keys, Further, a plurality of keys each indicating a modulation format are provided so that the modulation format can be input.

In this way, the quadrature demodulation means 33 and the digital filter means 35 can realize digital signal processing using the sampling frequency, code transmission speed, modulation format, and local oscillation frequency as software. By controlling the gain of the gain adjuster 31 and the above-mentioned variables, a digital signal for performing an operation corresponding to a selectively specified one of various modulation types and a plurality of local oscillation frequencies and a plurality of code transmission rates. A detector can be configured.

FIG. 4 shows the orthogonal demodulation means 33 in FIG.
The configuration shown is preferable. In other words, analog
An output digital signal from the digital converter 32 has n points.
Interpolation means 41I and 41Q perform n-point interpolation calculation on the time axis.
It is. These interpolated signals are output from the local oscillator 45.
Local oscillation signals f that are 90 ° out of phase with each other_LI(k), f
_LQMultiplication means 42 using (k)_I, 42_QMultiply by
Is done. Multiplication means 42 _I, 42_QOutput is n-point thinning
Means 43_I, 43_QIs used to thin out n points on the time axis
You. This thinning processing is performed by the n-point interpolation unit 41._I, 41_Qso
The interpolated number of sample points is thinned out and the orthogonal demodulation means 33
In-phase component and quadrature component of the demodulated signal are obtained.
This n-point interpolation means 41_I, 41_QImproves time resolution
Multiplication can be performed. Increase time resolution
By doing so, the digitized modulated wave signal and local oscillation signal
With high accuracy and multiplication output
Is the n-point thinning means 43_I, 43_QTime resolution
Drop, and digital signal processing after the quadrature demodulation means 33.
The load can be reduced. The multiplication means 42_I, 4
2_QConstitutes the complex multiplication means 42.

The arithmetic processing in the orthogonal demodulation means 33 will be described below. The input analog signal (IF signal) y (t) of the analog / digital converter 32 can be expressed by the following equation. y (t) = A (t) cos {ωt + ψ (t)} (1) where t is time, ω is 2πf, f is carrier frequency, and A (t)
Indicates the envelope, and ψ (t) indicates the phase. This analog signal y
(t) is the sample values are sampled every sampling time T _s by an analog-to-digital converter 32 is a digital signal. The relationship between time t and T _s when the m is an integer becomes the following equation.

T = mT _s (2) The digitally converted digital signal y _s (mT _s ) of y (t)
Can be expressed by the following equation. _{y s (mT s) = A} s (mT s) cos {ωmT s + ψ s (mT s)} ... (3) where, A _s (mT _s) specimens of the envelope A (t) at time mT _s reduction value, ψ _s (mT _s) is a sampled phase value of [psi (t) at time mT _s.

[0018] (3) a to normalize (4) the expression of the time in the sampling time T _s. y _s (m) = A _s (m) cos {ωm + ψ _s (m)} (4) Next, the time series represented by the equation (4) is subjected to an n-point interpolation operation.
The digitized received signal is interpolated by inserting n-point samples between adjacent points in the time series. The n-point interpolation operation result y _u is expressed as follows.

[0019] _{y u (k) = y s} (k / n) ··· (5) = A s (k / n) cos {ω (k / n) + ψ s (k / n)} ... (6) This interpolation value is obtained by an interpolation algorithm described later.
This y _u (k) is obtained by multiplying y _s (m) by n times the number of data on the time axis. From the local oscillation means 45, the angular velocity ω
, A local oscillation frequency signal f _L (k) is output.

F _L (k) = B _L exp {jω (k / n)} (7) where B _L is the amplitude of the local oscillation signal f _L (k), and exp (·) is an exponential function , (7) are complex numbers. The digital multiplication means 42 multiplies the n-point interpolation calculation result _yu (k) by the local oscillation signal f _L (k). z _u (k) = _yu (k) · f _L (k) (8) The multiplication result z _u (k) is the output of the multiplication means 42 at time k and is a complex number. This z _u (k) is subjected to an n-point thinning operation for extracting every n points in the time series by the n-point thinning means 43. The calculation result z _d is _{z d (p) = z u} (pn) ··· (9) = y u (pn) · f L (pn) ··· (10) = y s (pn / n) · f _L (pn) ··· (11) = y _s (p) · f _L (pn) ··· (12) In z _d (p), the time axis of z _u (k) becomes 1 / n . As for the sampling interval of z _d (p), the sampling time T _s in the analog / digital converter 32 is maintained as long as the n-point interpolation means and the n-point thinning means are used.

At the start of the demodulation process, the quadrature demodulation means 33
Is processed by the filter means 34 as a demodulation result, the local oscillation signal f _L (m'T _s ) of the local oscillation means 45 is used.
Is the received signal, that is, the input signal y _s (mT
Synchronize with _s ) so that synchronous detection processing can be performed correctly. This synchronization processing is performed, for example, according to the procedure shown in FIG. 4B. 4A, n of the n-point interpolation processing means 41 _I and 41 _Q is set to 0, that is, a signal y _s (mT _s ) for which no interpolation processing is performed and a local oscillation signal f _L (m′T _s ) of a frequency f ′. (S1).

Z _s (m) = y _s (mT _s ) f _L (m′T _s ) (13) z _s (m) = A _s (m) cos (2πfmT _s + θ) B _L cos (2πf'm'T _s ) (14) z _s (m) = (1/2) A _s (m) B _L [cos (2π (fm-f'm ') T _s + θ) + cos (2π (fm + f'm') T _s + θ)] (15) This z _s (m) is subjected to low-pass filtering to extract the difference frequency component z ＾ _s (m) given by equation (16) (S2).

Z ＾ _s (m) = (1/2) A _s (m) B _L cos [2π (fm-f′m ′) T _s + θ] (16) Amplitude of equation (16) , And the initial phase, the following equation is obtained. z ′ _s (m) = cos (fm−f′m ′) (17) 2πf = ω and 2πf ′ = ω ′. This expression is called an evaluation function, and the local oscillation frequency f 'is controlled so that z _s ' (m) is maximized. That is, it is checked whether the evaluation function has become maximum (S3), and if not, the frequency f 'of the local oscillation signal f _L (m'T _s ) is adjusted so that z _s ' (m) becomes maximum. It returns to step S1 (S4). When z _s ′ (m) becomes maximum, n-point interpolation processing is performed on the input signal y _s (mT _s ) and the local oscillation signal f _L (m′T _s ) (S5), and these interpolated signals are processed. y
_{u (kΔ), f L (} k'Δ) ( where, Enuderuta = a T _s is) multiplied by (S6), to give (S7) by the multiplication result low-pass filtering the difference frequency component by, step S3 Check whether the evaluation function z _s '(k) = cos (fk-fk') similar to
(S8) If it is not the maximum, it is checked whether the interpolation number n needs to be changed (S9). If no change is needed, the local oscillation signal f
K ′ of L (k′Δ) is adjusted so that z _s ′ (k) is maximized, and the process returns to step S6 (S10). Whether the interpolation number n needs to be changed, that is, the evaluation function z _s ′ (k) does not become larger than the threshold value, or is larger than the threshold value, and z _s ′ ( When it is determined whether or not k) has hardly changed, the number of interpolations n is increased by 1 or a number close to this, and the process returns to step S5 (S11). When z _s ′ (k) becomes the maximum, it is determined that synchronization has been achieved, and the synchronization control ends.

If the evaluation function z _s ′ (k) hardly increases even if n is increased, the smaller n is used for the subsequent processing, that is, z _s ′ (k) is maximized, and the number of interpolations is increased. Synchronization is performed with high accuracy by minimizing n, and the amount of calculation is reduced as much as possible. From the above description, since the minimum adjustment value of the local oscillation signal z _s (k′Δ) in step S10 is k ′ = 1, that is, the sample interval Δ in an interpolated state, the interpolation number n is increased. As the input signal y _s (mT _s ) increases, the synchronization accuracy of the local oscillation signal f _L (m'T _s ) increases. It should be noted that the low-pass filter means 101 in FIG.
Means for executing each process of S7 will be described.

Next, a specific interpolation processing method in the interpolation means 41 _I and 41 _Q will be described. As shown in FIG. 5A, a time-series signal y _s (m) having a sampling period T _s is converted into a frequency-domain signal y (f) by a fast Fourier transform (FFT) every time Ta, and this signal y (f) ), A zero point is inserted on the frequency axis to obtain a signal y ′ (f), and the signal y ′ (f) is subjected to inverse fast Fourier transform (IFFT) to increase the number of samples at time Ta, thereby increasing the number of samples. get y _u (k). In the interpolation method using the FFT, the reliability of the interpolated signal is high when the number of interpolation points is large. This method is described in, for example, Toshinori Yoshikawa et al., "Numerical Calculation Method in Engineering," p.

Alternatively, as shown in FIG. 5B, for each time Ta, a sample value determined by a linear function at + b substantially continuous with the peak value of the time-series signal y _s (m) is represented by a dotted line between the actual sample values. To obtain a time-series signal _yu (k) in which the number of samples has increased at time Ta. Alternatively, interpolation is performed by interpolating a sample value determined by a quadratic function at ² + bt + c, which is substantially continuous with a peak value, between actual sample values as indicated by a dotted line and estimating a time-series signal y in which the number of samples has increased at time Ta. The method is a simple method, and when the number of interpolation points is small, the amount of calculation is small and relatively high reliability can be obtained.

Further, as shown in FIG. 5C, the time-series signal y _s (m)
A sample to be interpolated next is estimated by an adaptive algorithm from the past Q samples of the time-series signal _yu (k) obtained by interpolation, and the time-series signal _yu (k) is interpolated as indicated by a dotted line. obtain. This adaptive algorithm only needs to be optimized for Gaussian noise, such as the Kalman Filter algorithm, the least mean square algorithm, the recursive least squares algorithm (Recursive LeastSquare Algorithm), the Newton method,
There is the steepest descent method. This method can perform interpolation on a time-series signal while compensating for some degree of signal deterioration due to a transmission line.

Next, a specific example of the thinning processing method by the thinning means 43 _I and 43 _Q in FIG. 4 will be described. For example, as shown in FIG. 6A, the time series in which the number of samples in the thinning time Ta is simply reduced by the number of points interpolated by the interpolation means 41 _I and 41 _Q from the time series signal z _u (k) for each time Ta. Let it be a signal z _d (p). This method is very simple to process. As another method of the decimation process, as shown in FIG. 6B, an evaluation function is calculated using a plurality of (Q, three in the figure) samples in the time-series signal z _u (k) to obtain one sample. Using this in place of the plurality of samples to perform thinning,
The time series signal z _d (p) is obtained. As an evaluation function, a function for obtaining an average value, a function for obtaining a center of gravity, or the like can be used. According to this method, information on the thinned sample can be reflected on the remaining sample by an evaluation function.

The digital filter means 34 in FIG. 3 performs a digital filter operation and a smoothing operation operation. That is, as shown in FIG. 7, the in-phase component output and the quadrature component output on the quadrature demodulation means 33, that is, the n-point thinning means 43 in FIG.
_The outputs of _I and 43 _Q are respectively supplied to smoothing processing means 5.
The number of sampling points on the time axis is reduced within the range that satisfies Nyquist's sampling theorem by being smoothed by 1 _I and 51 _Q. That is, a plurality of sample points are reduced to one point by the averaging process. This makes it possible to perform the thinning process on a substantial time axis. These smoothing processing means 51 _I ,
The respective processing results at 51 _Q, it is possible to reduce the filter orders, respectively subsequent digital filter means 52 _I, 52 _Q. Smoothing a plurality of signal points is represented by the following equation.

Z _ds (p _s ) = g (..., Z _d ((k−1) p), z _d (kp),...) (18) where g (·) is a smoothing process. , P _s is a normalized variable of the smoothing processing output, and z _ds is a complex number signal output from the smoothing processing means 51. In the smoothing process, for example, an average of signals at m points is obtained, and the number of sample points is reduced to 1 / m. Signal z
_ds ( _ps ) is band-limited by the filter means 52.
This is represented by the following equation.

Z∧ (p _s ) = h (z _ds (p _s )) (19) where h (·) is a signal processing function of the filter means 52, and z
∧ indicates an output signal of the filter processing. z∧ (·) is a complex number. As shown in FIGS. 4 and 7, the quadrature demodulation operation, the smoothing operation, and the filter operation may be separated into an in-phase component and an orthogonal component to perform a complex operation. You may.

A specific processing method of the smoothing processing means 51 in FIG. 7 will be described. In the simplest method, as shown in FIG. 8A, the time series signal z _d (p) from the quadrature demodulation means is thinned out by N samples every N + 1 samples and smoothed time series signal z _ds
(p _s ). The smoothing process by the linear weighting method is shown in the figure.
As shown in FIG. 8C , the time series signal z _d (p) is directly supplied to the multiplication means M ₀ and also supplied to the delay means D ₁ ,..., D _L, and the sampling period T _{d of the} time series signal z _d (p), respectively. _s , 2
T _s, ..., multiplier means M ₁ is LT _s delay, ..., it is supplied to the M _L. Multiplying means M _0, M _1, ..., weights f _0, f ₁ respectively M _L, ..., f _L is multiplied by these multiplication results are added by the adding means S _u, smooth and the addition result for each L + 1 samples It is output as the converted time series signal z _ds (p _s ). That is, the signal z _d (p) is given one weight f ₀ ,..., F _L for each L sample, added, and smoothed to form one sample of the signal z _ds (p _s ). That is, the following arithmetic processing is performed.

In the case of _{z ds (p s) = f} 0 · z d (p) + f 1 · z d (p-1) + ... + f L · z d (pL) L = 3, as shown in FIG. 8B Four samples z _d (p-3) to z _d (p) are linearly weighted and one sample z
Output as _ds (p _s ). In order to perform the smoothing processing by the adaptive filtering method, as shown by a dotted line in FIG. 8C , the adaptive algorithm processing means 50 uses an adaptive algorithm so that the judgment result from the received data judgment means becomes the best, and the weighting coefficients f ₀ ,. , F _L. The first determination of the weight coefficient is performed, for example, in a pilot signal (training signal) section in which received data is known. As the adaptive algorithm, a Kalman Filter algorithm and an algorithm derived therefrom can be used.

Next, a specific example of detection according to the present invention for a plurality of modulation formats will be described with reference to FIG. FIG.
The received analog signal of A is sent to the automatic gain adjuster 31 (FIG. 3).
After suppressing the amplitude component of the received signal within a certain range, the analog-to-digital converter 32 converts the signal into the digital signal shown in FIG. 9B. Local oscillation means 35 controlled by software generates local oscillation signals of the in-phase component and the quadrature component in FIGS. 9C and 9D. Multiplying means 42 _I , 42 _Q
In FIG. 9B, the digital signal of FIG. 9B is multiplied by the respective local oscillation signals of FIGS. 9C and 9D. As a result, the in-phase component of the output of the quadrature demodulation means 33 becomes as shown in FIG. 9E, and the quadrature component becomes as shown in FIG. 9F. The multiplied in-phase and quadrature components are spectrally shaped by independent digital filter means 34 controlled by software. As a result, as shown in FIG. 9G, the in-phase component in the spectrum-shaped baseband signal has the quadrature component shown in FIG.
Are obtained as shown in FIG. As described above, even if the modulation format is sequentially changed to QPSK, 16QAM, or BPSK in a short time of 4 milliseconds during transmission, the signal can be correctly reproduced by the digital signal detector of the present invention. The received modulated wave signal is transmitted and received per symbol at 2 bits by QPSK, 4 bits by 16QAM, and 1 bit by BPSK. As described above, reception detection in variable bit transmission is realized by the digital signal detector of the present invention. In this case, these QP
It is known in advance that SK, 16QAM, and BPSK modulated signals are sequentially received every 4 milliseconds.
Reference numeral 5 switches a variable for each means in synchronization with the received signal.

Next, switching to a plurality of local oscillation frequencies is performed in the present invention, as shown in FIG.
Perform for 5 The local oscillating means 45 is basically realized by software.
Calculation of _LI B _L exp _{(j2πfmT s)} is made. To perform the switching shown in FIG. 2A, the frequency is set to f ₁ ,
An instruction to sequentially switch to f ₂ , f ₃ , and f ₄ is given.
This instruction is performed in an instruction execution cycle of a microprocessor that executes software in principle. Therefore, the frequency switching speed can be realized in a few nanoseconds. As described above, according to the present invention, the frequency switching of the local oscillation means 45 can be performed at a higher speed than in the conventional configuration, and it is possible to sufficiently cope with the frequency switching during a call. The present invention can also be applied to high-speed frequency hopping that dynamically switches

In the present invention, similarly, the change of the filter means 34 is realized by software programmability with respect to the change of the code transmission rate, and the control means 35 gives an instruction to change the transmission rate. That is, the filter means 34 and the control means 35 realize digital signal processing by software. For example, when the digital signal processing is made to correspond to the change in the transmission speed shown in FIG. 2C, the control means 35 sends the digital signal processing to the filter means 34 as shown in FIG. 10B. The code transmission rates B ₁ , B ₂ , B ₃ , B ₄ are sequentially instructed. Accordingly, the transmission speed change is performed in the instruction execution cycle of the microprocessor that executes the software in principle. As described above, according to the present invention, the characteristics of the filter unit 34 can be changed by the control unit 35 at high speed. The characteristic control for the filter means 34 is also performed by the modulation format of the received signal. For example, when the modulation format changes as shown in FIG. 9G, the roll-off of the filter characteristic is accordingly changed to QP.
0.5 for SK, 0.3 for 16QAM, 0.5 for BPSK
Respectively. As described above, the various processes of the present invention are performed by software calculations. That is, FIG.
Microprocessor 37 in control means 35 as shown in FIG.
Controls the automatic gain adjuster 31, the quadrature demodulator 33, and the digital filter 34 by decoding and executing the program 38 in the controller 35. The control of the automatic gain adjuster 31 is performed by detecting the output level of the automatic gain adjuster 31 with a level detector (not shown), converting the detected level into a digital value and taking it into the microprocessor 37, By controlling the amplification gain in a stepwise manner, the output of the automatic gain adjustment period 31 falls within a predetermined amplitude fluctuation range. Control of the quadrature demodulation unit 33 is performed on the local oscillation unit 45, the n-point interpolation unit 41, and the n-point thinning unit 43. For the local oscillation means 45, the local oscillation frequency f, the phase ψ, and the amplitude B _L are specified by the microprocessor 37. For the n-point interpolation means 41 and the n-point thinning means 43, the sampling frequency 1 / T _s and the number n of interpolation and thinning points are specified. The control on the digital filter means 34 is performed on the smoothing processing means 51 and the digital filter means 52. For the smoothing processing means 51, the sampling frequency 1 / T _s , the number of smoothing points, and the smoothing method are specified. For the digital filter means 52, a sampling frequency 1 / T _s , a filter coefficient, and the number of symbols required for filtering are specified. That is, when the carrier frequency of the input modulated wave signal of the analog-digital converter 32 is set and input, the minimum value of the sampling frequency 1 / T _s of the analog-digital converter 32 in accordance with the frequency determined automatically, i.e. The carrier frequency of the input modulated wave signal is 13
If it is 0 MHz, set the sampling frequency to 260 MHz,
When the minimum sampling frequency is used, a digital signal having a sampling frequency at least four times the carrier frequency of the input modulated wave signal is required to accurately synchronize the local oscillation signal with the carrier of the input modulated wave signal. Yes,
From this point, the number of interpolation points and the number of thinning points n are automatically determined.
If the conversion speed of the analog / digital converter 32 is high, the input modulated wave signal can be sampled at the minimum sampling frequency or higher, and the number of interpolation and decimation points n can be reduced accordingly.

Further, in the state where the signal is demodulated into a baseband signal, if the frequency is, for example, 20 KHz, the sampling frequency required for this signal processing may be 40 KHz.
Since the output digital signal of the quadrature demodulation means 33 has an unnecessarily large number of signals, these signals are effectively used, and the baseband signal is as faithful as possible and smoothed in order to reduce the sampling frequency. Processing means
The processing at 51 is performed, and the number of smoothing points is automatically determined from the set carrier frequency of the input modulated wave signal and the set code transmission rate. As described with reference to FIGS. 2D and 10B, it is necessary to set appropriate filter characteristics in accordance with the code transmission rate. Therefore, the filter coefficient and the number of symbols required for filtering are automatically set in accordance with the set code rate. Is decided. The filter coefficient is automatically determined by the modulation format set as described above. The filter means 34, the local oscillation means 45,
A program is created in advance so that various parameters (variables) for the interpolation means 41, the thinning means 43, and the like are automatically specified.

Microprocessors are provided for the quadrature demodulation means 33 and the digital filter means 34, respectively. These microprocessors are specified with the above-mentioned various parameters (variables) from the microprocessor 37, and use these as arguments to perform interpolation processing, A so-called distributed processing method may be used in which programs such as orthogonal demodulation processing and thinning processing, and programs such as smoothing processing and filter processing are executed, respectively. A centralized processing method may be used in which programs such as interpolation, orthogonal demodulation, decimation, smoothing, and filtering are executed by time sharing to perform substantially simultaneous processing.

The processing procedure of this digital detector is as follows. That is, it is checked whether there is a variable change request as shown in FIG. 12 (S1). That is, as described above, a change in the carrier frequency, a change in the code transmission rate, or the like is newly set and input from the setting input unit 36, or the time at which the modulation format or the like is changed as described with reference to FIG. When it is known, or when it is changed, or as described in the Background of the Invention, switching between use in the 800 MHz band and use in the 1500 MHz band, and switching between audio information reception and still image information reception Is input by the setting input means when there is a variable change request. Further, for example, as shown in FIG. 14A, the received signal is a time-division multiplexed signal of three channels, and each channel is composed of a preamble 71 and data 72 as shown in FIG. 14B. As shown in FIG.
And the base station number 74, and the modulation format information 75 such as the data modulation format and code transmission rate used for the next data frame 72. In this manner, the received signal modulates the data in the data frame 72, etc. Is indicated when the data indicating the modulation format or the like is demodulated when a variable change request is issued.

If there is such a variable change request, a variable to be changed and its value are determined in accordance with the variable change request (S2), and the determined variable is converted into a processor for executing a corresponding program. That is, for example, in order to execute a program for performing quadrature modulation processing, the number of interpolations n, the local carrier frequency, its amplitude and phase, etc. are set as arguments to the processor that executes this program (S3). afterwards,
The digital detection processing program is executed (S4).

As shown in FIG. 13, the digital detection processing procedure performs automatic gain control on the automatic gain adjuster 31 in FIG. 11 so that the level of the received signal falls within a predetermined range.
(S1), the signal subjected to the automatic gain control is converted into a digital signal by the analog / digital converter 32 shown in FIG.
2) Perform n-point interpolation on the digital signal
(S3) A quadrature demodulation operation is performed on the interpolated digital signal (S4). An n-point thinning process is performed on the demodulation operation result (S5), and a smoothing process is further performed (S6).
A low-pass filter process is performed on the smoothing processing result (S7), and a signal point determination is performed on the processing result (S7).
8).

As shown in FIG. 15, the microprocessor 37 may be controlled by time sharing of the digital detector 53, the radio line controller 54, and the voice encoding / decoding processor 55 according to the present invention. Dedicated control programs 38, 56, and 57 are also prepared for the digital detector 53, the radio line controller 54, and the audio encoding / decoding processor 55, respectively. The microprocessor 37 performs processing by switching the control target and the control program for the digital detector 53, the radio line controller 54, and the audio encoding / decoding processor 55 at appropriate time intervals by a time sharing method. . As a result, basically 1
A plurality of control targets can be processed by one microprocessor 37. This wireless device is used, for example, for a mobile station of mobile communication, and the wireless channel controller 54 controls a control channel,
Switch the communication channel.

FIG. 16 shows a transceiver for digital radio communication to which the digital detector of the present invention is applied, and portions corresponding to those in FIG. 3 are denoted by the same reference numerals. The modulated modulated wave signal from the antenna 60 is amplified by the low noise amplifier 61, and then converted into an intermediate frequency signal by the frequency converting means 62.
The intermediate frequency signal is band-limited by the filter 63 and supplied to the automatic gain adjuster 31. That is, it is supplied to the digital detector 53 according to the present invention shown in FIG.
4 and further to, for example, an exchange of a communication network not shown. The signal from this exchange is supplied to the filter means 64 through the radio line controller 54, and after being subjected to band limitation, the intermediate frequency carrier is quadrature-modulated by the quadrature modulation means 65. The signal is converted to an analog signal by an analog converter 66, and further converted to a high-frequency signal by a frequency conversion means 67. The high-frequency modulated wave signal is power-amplified by a transmission power amplifier 68 and transmitted by an antenna 69.

Next, a specific example to which the present invention is applied will be described. (i) In the case of mobile multimedia Here, the control operation of the present invention is based on (1) manual setting by the user of the mobile device, (2) automatic setting by base station command, and (3) autonomous judgment of the mobile device. An example of automatic setting will be described. Here, the voice service is a conventional PDC (Personal Digital Cellular), and the media service (a service for broadcasting and receiving a still image or the like) is a multi-level modulation (for example, 16QAM). The transmission band is constant regardless of the service. Further, the service is an example in which the service is changed from a voice to a still image. Media Services
It is assumed that the broadcast is performed in a predetermined frequency band.

(1) When a user of a mobile device receives a media service such as an image from a state of receiving a voice service,
Change the service mode of the mobile device. This change is made by a dial key or a dedicated mode switch provided on the setting input means 36 (FIG. 11). When the mode change request signal is input, the control means 35 of the mobile device receives a change request for changing the mode from the voice service mode to the media service mode in this example. (Not shown), and the processing variables of the gain adjuster 31, the quadrature demodulator 33, and the filter 34 are changed. For the synthesizer, the set frequency is changed to a predetermined frequency band. For the gain adjuster 31, the setting of the maximum amplitude value of the input signal is changed. That is, since the modulation format is changed from QPSK to QAM, the maximum setting value of the input signal is made larger than QPSK. For the orthogonal modulating means 33, the setting of the number of interpolation points n and the number of thinning points n is changed. n is larger than QPSK. For the filter means 34, the smoothing method, the filter coefficient and the filter order are changed.

As for the smoothing method, the optimum smoothing method determined in advance by changing the modulation method is obtained, and the algorithms of the various optimum smoothing methods are stored in the ROM. The optimum smoothing method for this mode is read out from the ROM based on the above, and the smoothing process is performed. For the filter means 52 (FIG. 7), the roll-off rate is made smaller than that of QPSK, and in accordance with this, The coefficient and order of the filter means 52 are also changed.

In order to change variables in various processes according to the mode thus set, for example, as shown in FIG. 11, a processing variable storage unit 102 and a smoothing procedure storage unit 103 are provided in the control means 35. , Processing variable storage unit 10
2 includes, for example, as shown in FIG. 17A, the maximum amplitude value of the automatic gain adjuster for each of the audio mode and the image mode, the number of interpolation points n, the number of the ROM storing the algorithm used for the smoothing process, and the roll-off rate. , Filter coefficients are stored, and the control means 35 reads out these variables and uses them according to the mode set and inputted. In the smoothing procedure storage unit 103, algorithms or procedures necessary to execute various smoothing processes are stored by being distinguished by ROM numbers. Note that a processing variable corresponding to the service mode can be stored in the processing variable storage unit 102 so that other services can be received in addition to the audio service and the image service.

(2) The user of the mobile device changes the service mode setting of the mobile device using the setting input means 36 as in (1). Upon receiving the service mode change request, the control means 35 of the mobile station sets a flag indicating the mode change on the transmission channel communicating with the base station and transmits the flag. To set this flag means to set 0 of a predetermined bit on the transmission frame to 1 or 1 to 0. When the base station receives the service change request flag of the mobile station, the base station notifies the mobile station of transmission information on the media service through the downlink since the request is a media service request in this example. The mobile station detects transmission information from the base station by the radio channel controller 54 (FIG. 16), and includes the modulation method, transmission speed, and TDMA frame used in the data frame of the radio channel in the detected transmission information. Time slots and the like. If necessary, the mobile station and the base station exchange signals several times over the radio line regarding the service mode to be changed. The detected transmission information is input to the control unit 35 as a variable change request.

The control means 35 changes each variable for the synthesizer, gain adjuster, quadrature demodulation means and filter means. The changes are the same as in (1). In this case, as shown in FIG. 17B, the maximum amplitude value, the number of interpolation points n, and the ROM number of the smoothing method are read out from the processing variable storage unit 102 according to the modulation format designated by the base station.
As shown in (1), the roll-off rate and the filter coefficient can be read out according to the code transmission rate specified by the base station.

(3) The user of the mobile station changes the setting of the service mode of the mobile station as in (1). In this case, even in the voice service mode, some combinations of the modulation format and the code transmission rate are used. Similarly, in the media service, some combinations of the modulation format and the code transmission rate are determined, and different mode numbers are assigned to the respective combinations, and each processing number is processed for each mode number as shown in FIG. 17A. It is stored in the variable storage unit 102. The same relationship among the mode number, the modulation format, and the code transmission speed is held in the base station. When the service mode change is set and input, the control means 35 transmits a code indicating the set mode number to the base station via a transmission channel for communicating with the base station. The control unit 35 causes the storage unit 102 to read the processing variables corresponding to the mode input and set, and performs a detection process using the read processing variables.

The base station receives the service mode change request mode number from the mobile station, and in response to this, the modulation scheme, transmission rate, and TDMA frame used in the radio line for information transmission to the mobile station. The information such as the number of time slots is changed and the information requested by the mobile device is transmitted.
If necessary, the mobile station and the base station exchange signals several times over the radio line for the service mode to be changed. (ii) In the case of a compatible mobile station Here, the control operation of the present invention is based on (1) manual setting by the user of the mobile station, (2) automatic setting by base station command, and (3) autonomous judgment of the mobile station. An example of automatic setting will be described. It is assumed that the user moves from outdoors to indoors (closed space such as an underground shopping mall). PDC outdoors and 16 QA indoors
M is assumed. It is assumed that the transmission band is constant.

(1) The user of the mobile device changes the call mode setting of the mobile device when moving from outdoors to indoors. This change is made using the dial keys of the setting input means 36 or a dedicated mode change switch. When receiving the call mode change request, the control unit 35 changes the setting of the oscillation frequency of the synthesizer. For example, when the 800 MHz band is used outdoors,
When using the 1500MHz band indoors, set the local oscillation frequency to be supplied to the frequency converter for converting to an intermediate frequency signal.
Change to 1500MHz band. Since the modulation method is changed from QPSK to 16QAM as in the case of the mobile multimedia of (i), the control means 35 increases the maximum amplitude value of the gain adjuster and sets the number n of interpolation and decimation in the quadrature demodulation processing. Is larger than QPSK, and a ROM storing the optimum smoothing method is used, and the transmitting side filter means 64 (FIG. 1) is used.
6) The roll-off rate of the filter means 34 (FIG. 16) on the receiving side is reduced, and the filter coefficient is changed.

(2) When the user of the mobile device moves from outdoor to indoor, the user sets the communication mode of the mobile device by setting input means 3
6 to change. The mobile station sets a flag on the transmission frame of the radio channel with the communication mode change signal for the base station. After receiving the service change request flag of the mobile station, the base station notifies the mobile station of transmission information on the communication mode via the downlink. The mobile station is connected to the radio line controller 54.
By detecting the transmission information from the base station, the detected transmission information includes the carrier frequency used in the radio line, the modulation method,
It includes the transmission speed, the time slot of the TDMA frame to be used, and the like. The mobile device changes variables used in a synthesizer, a quadrature modulation unit, a gain adjuster, a quadrature demodulation unit, and a filter unit among the units of the mobile device based on the transmission information. The changes are the same as in (1).

(3) As in (1) and (2), when moving from outdoors to indoors, the user of the mobile device changes the call mode setting of the mobile device. In this case, similar to (3) of (i), a plurality of modes corresponding to combinations of several modulation formats and code transmission rates are determined in advance, and the mobile station changes the communication mode setting to the base station. Is notified through a wireless line. The base station sets a predetermined call mode according to the notification signal from the mobile station. At this time,
The settings such as the carrier frequency, modulation scheme, transmission speed, TDMA frame, and number of time slots used in the wireless line are changed. The base station and the mobile device perform communication several times over a wireless line, and then perform communication in the changed call mode. The mobile device reads the processing variable storage unit 102 based on the changed call mode, and changes variables used in the synthesizer, the quadrature modulator, the gain adjuster, the quadrature demodulator, and the filter. The changes are described in (1) and
Same as (2).

In the above, all the variables used in the gain adjuster 31, the quadrature demodulator 33, and the digital filter 34 have all been changed according to the mode change. This is a desirable example. For example, when changing to the QAM modulation format, at least only the maximum amplitude value of the gain adjuster 31 is changed so that the amplitude value of the received signal is correctly detected, and other variables are not changed. Conversely, when changing to the QPSK modulation format, amplitude information is not used, so that it is not necessary to give priority to changing the maximum amplitude value of the gain adjuster 31. It is relatively important to change the roll-off rate in the filter means 34 and the accompanying change in the filter coefficient and its order. Next, the number of interpolation points n, and then the selection of the smoothing method, the mode is changed in the latter order. Even if there is, the change may be omitted.

In the above description, wireless communication is assumed,
The present invention can also be applied to reception of communication in which a modulation format and a code transmission rate suitable for various types of information are used in wired communication. In that case, the automatic gain adjuster 31 can be omitted.

[0057]

As described above, the present invention has the following effects in comparison with the conventional method. (I)
It can support multiple modulation formats, (ii) can support multiple local oscillation frequencies, (iii) can support multiple code transmission rates,
(Iv) variable bit transmission and variable code rate transmission can be realized by one receiver, (v) perfect orthogonal demodulated waves can be generated,
(Vi) Applicable to fast frequency hopping.

[Brief description of the drawings]

1A is a block diagram showing an example of digital mobile radio communication in which a plurality of modulation methods are mixed, and FIG. 1B is an example of digital mobile radio communication using a receiver having a plurality of detectors in the mobile communication of FIG. 1A. FIG.

FIG. 2A is a diagram showing how a carrier frequency of a received signal changes with time, and FIG. 2B is a diagram showing a conventional configuration in which a local oscillation frequency of a detector is changed in response to a change in the carrier frequency of FIG. 2A. C is a diagram showing how the code transmission rate of the received signal changes with time, and D is a block diagram showing a conventional configuration for switching the band limiting filter of the detector according to the change in the code transmission rate in FIG. 2C. .

FIG. 3 is a block diagram showing a functional configuration of a detector according to the present invention.

4A is a block diagram showing a specific example of a functional configuration of a quadrature demodulation unit 33 in FIG. 3, and FIG. 4B is a flowchart showing an example of a processing procedure for automatically synchronizing a local oscillation signal.

5A is a diagram illustrating a method of performing n-point interpolation using FFT, FIG. 5B is a diagram illustrating a method of performing linear interpolation of the n-point interpolation using an m-order function, and FIG. FIG. 3 is an explanatory diagram of a method of performing nonlinear interpolation.

6A is a diagram illustrating an n-point thinning process using a simple thinning method, and FIG. 6B is a diagram illustrating an n-point thinning process using a weight replacement method.

FIG. 7 is a block diagram showing a specific functional configuration example of a filter means 43 in FIG. 3;

8A is a diagram illustrating a smoothing process by a simple extraction method, FIG. 8B is a block diagram illustrating a functional configuration example of another smoothing process, and FIG. 8C is a diagram illustrating the operation of FIG. 8B.

FIGS. 9A to 9H are diagrams showing states of signals in respective sections of the digital detector according to the present invention.

10A is a block diagram illustrating a functional configuration example of a process of switching a local oscillation frequency in the quadrature demodulation unit 33 in FIG. 3; and FIG. 10B is a functional configuration of a process of switching a filter characteristic in the filter unit 34 in FIG. It is a block diagram showing an example.

FIG. 11 is a block diagram showing an example of a functional configuration for implementing the method of the present invention.

FIG. 12 is a flowchart illustrating an example of a processing procedure of the detection method.

FIG. 13 is a flowchart illustrating an example of a procedure of digital detection processing.

FIG. 14 is a diagram illustrating a frame configuration example of a received signal.

FIG. 15 is a functional configuration example showing an example in which a microprocessor used in the present invention is also used for other processing.

FIG. 16 is a block diagram showing a functional configuration of a transceiver to which the present invention is applied.

17A is a diagram illustrating a storage example of a processing variable storage unit, FIG. 17B is a diagram illustrating a part of another storage example of a processing variable storage unit, and FIG.
It is a figure showing the example of storage of other parts in the memory part of 7B.

Continuation of front page (56) References JP-A-3-66244 (JP, A) JP-A-3-283744 (JP, A) JP-A-5-291859 (JP, A) JP-A-5-199267 (JP) , A) JP-A-57-155856 (JP, A) (58) Fields investigated (Int. Cl. ⁷ , DB name) H04L 27/00-27/38

Claims (25)

(57) [Claims] An A / D conversion step of converting a received modulated wave signal into a digital signal; an orthogonal demodulation step of performing an orthogonal demodulation operation on the converted digital signal; A filter step for obtaining a baseband signal by performing a filter operation; a sampling frequency, a code transmission rate, a local oscillation frequency,
Setting at least one variable of the modulation type
And the quadrature demodulation step is performed at n points on the digital signal.
An interpolation step for performing an interpolation process, where n is one or more real numbers.
And the result of the above n-point interpolation processing and the local oscillation signal
Multiplication step for complex multiplication and the result of the complex multiplication
and a thinning step for performing n-point thinning processing. The filter step includes:
Reduce the number of samples on the time axis by smoothing
Smoothing step and the result of the smoothing process
Digital filter step to perform band limiting operation and more
It becomes, and the quadrature demodulation step at the request of change the variable,
A control step of changing at least one processing variable in at least one of the filter steps by software . 2. The detection method according to claim 1 , wherein the interpolating step performs a discrete Fourier transform on the digital signal, increases a coefficient of zero with respect to a result of the Fourier transform, and performs an inverse discrete Fourier transform to perform the interpolation. Obtain the processing result. 3. The detection method according to claim 1 , wherein the step of interpolating comprises:
According to a linear interpolation method that approximates an m-th order function and interpolates a sample that matches the m-th order function, m is a real number of 1 or more. 4. The detection method according to claim 1 , wherein the interpolation step estimates and interpolates a next sample by an adaptive algorithm using a predetermined number of the digital signal sequences. 5. The detection method according to claim 1 , wherein the thinning-out step is a process of thinning out the sample of the sample point interpolated in the interpolation step from the result of the multiplication. 6. The detection method according to claim 1 , wherein the decimation step processes each of a plurality of samples of the series of the multiplication result by an evaluation function, and
This is a process for making a sample. 7. The detection method according to claim 1 , further comprising a synchronization step of synchronizing the local oscillation signal with the digital signal. 8. The detection method according to claim 7 , wherein the synchronizing step synchronizes the frequency of the local oscillation signal with the digital signal, and interpolates the local oscillation signal whose frequency is synchronized with the n-point interpolation signal. Time synchronizing the processed digital signal. 9. The detection method according to claim 1 , wherein the number n of interpolation points is used as the processing variable in the quadrature demodulation step. 10. The detection method according to claim 1 , wherein the local oscillation signal frequency is used as the processing variable in the quadrature demodulation step. 11. The detection method according to claim 1 , wherein the processing variable of the filter step is a number of smoothing points for reducing the number of samples. 12. The detection method according to claim 1 , wherein the processing variable in the filter step is a filter characteristic in the digital filter step. 13. The detection method according to claim 1 , further comprising a gain adjustment step of controlling a maximum amplitude of the received modulated wave signal to a set value, and processing the set value of the maximum amplitude. Variable. 14. The detection method according to claim 13 , wherein a change request step for detecting a variable change request, and a processing variable determination for determining which processing variable is to be changed and when the variable change request is detected. And a step of changing a corresponding processing variable to the determined processing variable. 15. The detection method according to claim 14 , wherein each of the steps is performed by decoding and executing a program. 16. The detection method according to claim 15 , wherein the processing variable determining step reads out a processing variable stored in advance in response to the change request. 17. The detection method according to claim 16 , wherein the change request step is input by operating setting input means. 18. The detection method according to claim 16 , wherein the change requesting step detects the change request from a received signal. 19. An analog-to-digital converter for converting a received modulated wave signal into a digital signal, quadrature demodulation means for performing quadrature demodulation on an output digital signal of the analog-digital converter, and the quadrature demodulated digital signal Filter means for obtaining a baseband signal by performing a filter operation on the sampling frequency, code transmission speed, local oscillation frequency,
Input means for inputting a request for changing at least one processing variable of a modulation type, and the quadrature demodulating means in response to the input request for change;
At least one of at least one of the above filter means
Control means for changing the processing parameters of the analog-to-digital converter.
Interpolation means for performing n-point interpolation processing on output digital signals
And n is a real number greater than or equal to 1 and computes and generates a local oscillation signal.
And a complex multiplication of the interpolation processing result and the local oscillation signal.
Multiplying means, and performing n-point thinning processing on the digital signal resulting from the multiplication.
More becomes thinning means for performing, in said filtering means to smooth the output of the orthogonal demodulating means
Smoothing processing means for performing calculations to reduce the number of sample points
And performs band limiting calculation on the calculation result of the smoothing processing means.
Digital detection with digital filter means
vessel. 20. The detector according to claim 19 , further comprising an automatic gain adjuster provided before the analog-to-digital converter and performing gain control so that a maximum amplitude of the received modulated wave signal becomes a set value. The set value can be changed by the control means as the process variable. 21. The detector according to claim 20 , further comprising a storage unit storing a correspondence table between each change request and each processing variable, wherein the control unit reads out the storage unit in response to the input change request. Means for changing the processing variable corresponding to the read processing variable. 22. The detector according to claim 21 , wherein said input means is a setting input means for inputting said change request by manual operation. 23. The detector according to claim 21 , wherein said input means is means for detecting said change request from a received signal. 24. A detector according to any one of claims 1 to 23, wherein the control means is further a program and a microprocessor to decrypt do this. 25. The detector according to claim 24 , wherein said control means also serves as control means for a digital radio transceiver.