JP2022080081A - Switching power source apparatus, and control device and control method therefor - Google Patents

Abstract

To reduce conduction loss by reducing an in-circuit current when load is light in a state where a voltage difference between an input and an output is large.SOLUTION: A switching power source apparatus includes a primary-side inverter 10, a transformer 20, a secondary-side inverter 30, and a control device 40 that generates drive pulses S11-S16, S31-S36 for turning on/off switches 11-16, 31-36 of the inverters 10, 30. The control device 40 has a function of controlling output power Po of the secondary-side inverter 30 by changing a phase difference φ between an output value of the primary-side inverter 10 and an input value of the secondary-side inverter 30, and a function of comparing a triangular carrier TC and a comparison value CV having large and small modulation factors MR to obtain a comparison result, and modulating the drive pulses S11-S16, S31-S36 on the basis of the comparison result.SELECTED DRAWING: Figure 1

Description

The present invention relates to a switching power supply device such as a dual active bridge (hereinafter referred to as "DAB") type DC / DC converter, a control device thereof, and a control method.

Conventionally, a DAB type DC / DC converter, which is one of switching power supply devices, has a phase shift of a full bridge inverter on the primary side and a secondary side of a transformer, for example, as described in Patent Documents 1 and 2. It is a DC / DC converter that can transmit power in both directions by controlling it.

FIG. 8 is a block diagram of a conventional three-phase DAB type DC / DC converter (hereinafter referred to as “conventional DAB”) described in Patent Document 1.
In this conventional DAB, the primary side inverter 10 is connected in parallel with the primary side smoothing capacitor 1 that smoothes the DC primary side voltage E1 and the primary side current I1. The primary side inverter 10 is a circuit that switches between the smoothed primary side voltage E1 and the primary side current I1 to convert them into a three-phase AC voltage and a three-phase AC current, and is a high level U-phase (hereinafter referred to as “H”). () Side switch 11, U-phase low-level (hereinafter referred to as "L") side switch 12, V-phase H-side switch 13, V-phase L-side switch 14, W-phase H-side switch 15, and It is composed of a full bridge circuit of the W-phase L-side switch 16. The connection point between the switches 11 and 12, the connection point between the switches 13 and 14, and the connection point between the switches 15 and 16 are connected to the three-phase transformer 20 via the three-phase reactors 17, 18 and 19. The primary winding is connected.

A secondary side inverter 30 is connected to the secondary winding of the transformer 20. The black circles attached to the vicinity of the upper ends of the primary winding and the secondary winding of the transformer 20 indicate the beginning of winding of the winding. The secondary side inverter 30 is a circuit that rectifies the three-phase AC voltage and the three-phase AC current output from the secondary winding of the transformer 20, and is a U-phase H-side switch 31 and a U-phase L-side switch 32. , V-phase H-side switch 33, V-phase L-side switch 34, W-phase H-side switch 35, and W-phase L-side switch 36.
The DC voltage and DC current rectified by the secondary side inverter 30 are smoothed by the secondary side smoothing capacitor 37, and the smoothed DC secondary side voltage E2 and the secondary side current I2 are output. ing.

The switches 11 to 16, 31 to 36 constituting the primary side inverter 10 and the secondary side inverter 30 are elements that are turned on / off by drive pulses S11 to S16 and S31 to S36 supplied from a control device (not shown), respectively. It is composed of power semiconductor elements such as a metal oxide semiconductor conductor type field effect transistor (hereinafter referred to as "PWM") and an insulated gate bipolar transistor (hereinafter referred to as "IGBT"). For example, when the drive pulses S11 to S16 and S31 to S36 are H, the switches 11 to 16, 31 to 36 are turned on, and when the drive pulses S11 to S16 and S31 to S36 are L, the switches 11 to 16, 31 are turned on. ~ 36 turns off. Regenerative diodes are connected in antiparallel to each of the switches 11 to 16 and 31 to 36.

The drive pulses S11 to S16 and S31 to S36 for driving the power conversion unit of the conventional DAB in FIG. 8 have a constant frequency ω, a fixed duty ratio D of 0.5, and 120 U, V, and W phases, respectively. It has a phase difference β of each °.
In the primary side inverter 10, the switches 11 to 16 are turned on / off by the primary side drive pulses S11 to S16 supplied from a control device (not shown), and the DC primary side voltage E1 and the primary side current I1 are generated. It is converted into a three-phase AC voltage vp (hereinafter referred to as "output voltage vp") and a three-phase AC current. In the secondary side inverter 30, the switches 31 to 36 are turned on / off by the secondary side drive pulses S31 to S36 supplied from a control device (not shown), and the three phases are induced in the secondary winding of the transformer 20. The AC voltage vs. (hereinafter referred to as "input voltage vs.") and the three-phase AC current are converted into a DC secondary voltage E2 and a secondary current I2.

The input voltage (or input current) and output voltage (or) are due to the phase difference φ between the output voltage vp (or output current) of the primary side inverter 10 and the input voltage vs (or input current) of the secondary side inverter 30. Output current) and power flow can be controlled. When the voltage bl (= vp-vs) between the primary winding and the secondary winding of the transformer 20 passes through the reactors 17 to 19, the transformer current (U-phase reactor current IL_U, V-phase, which is the reactor current IL). (Same as reactor current IL_V and W phase reactor current IL_W) flows. At this time, a U-phase reactor voltage VL_U, a V-phase reactor voltage VL_V, and a W-phase reactor voltage VL_W, which are reactor voltages VL, are generated. The output power Po can be calculated from the effective value ILT of the reactor current IL.
As described above, in the conventional DAB, the buck-boost operation and the bidirectional power conversion can be easily performed by changing the phase difference φ.

US Pat. No. 5,027,264 Japanese Unexamined Patent Publication No. 2020-102933

FIG. 9 is a diagram showing on / off patterns of the drive pulses S11 to S16 and S31 to S36 of FIG. 8 and operation waveforms of the power conversion unit. As the operation waveform of FIG. 9, for example, the U-phase reactor voltage VL_U, the U-phase reactor current IL_U, and the primary side current I1 when the secondary side voltage E2 is in a drooping state of 0 V (phase difference φ = 0 °) are shown. It is shown. The dead time when switching on / off of switches 11 to 16 and 31 to 36 is not included. T is one control cycle.
FIG. 10 is a pattern diagram of the drive pulses S11 to S16 described in the conventional patent document 2.

The conventional DAB of FIG. 8 can be insulated and can easily perform buck-boost operation and bidirectional power conversion. However, especially when the voltage difference between input and output is large, the current circulating in the circuit increases. In that case, since the conduction loss mainly increases, there is a problem that the power conversion efficiency tends to decrease. Further, even when one of the input / output voltage differences having a substantially maximum value becomes a short circuit or a state close to the short circuit, the current circulating in the circuit cannot be suppressed.

That is, as shown in FIG. 9, since power conversion cannot be performed when the secondary voltage E2 is 0 V, the phase difference φ of the control parameter, which is the output command value, is controlled to 0 ° on the primary side. The phases of the drive pulses S11 to S16 and the secondary drive pulses S31 to S36 are in the same phase. However, it can be confirmed that the voltage brought about by the primary side voltage E1 is applied to the U-phase reactor voltage VL_U waveform at any time (similarly, the V-phase reactor voltage VL_V waveform and the W-phase reactor voltage VL_W). The waveform is also the same as the U-phase reactor voltage VL_U waveform, except that there is a phase difference of 120 °). Therefore, in principle, the reactor current IL_U is generated in the reactor 17, and the reactor currents IL_V and IL_W are also generated in the other reactors 18 and 19 as well). Due to the current path, the primary side current I1 is also generated in the primary side input section, but this becomes a triangular wave waveform with an average value of 0A, and is smoothed to DC 0A by an external filter circuit (not shown). Therefore, no effective power is generated, and all the flowing current becomes an invalid current.
As described above, in the conventional DAB of FIG. 8, since there is no section in which the switch states of the U, V, and W phases are all the same (on or off) at any time, the current can be reduced at the time of a short circuit or the like. Can not.

Therefore, in the three-phase DAB type DC / DC converter described in Patent Document 2, as shown in FIG. 10, a duty D addition pattern for forcibly inverting the drive pulses S11 to S16 (and / or S31 to S36) is added. By inserting PA, the current component is reduced.
However, in recent years, in order to improve the degree of freedom in circuit design, the drive pulse modulation method is changed without changing the configuration of the basic circuit in the conventional DAB, and the DAB type having the same effect as that of Patent Document 2 is obtained. There has been a demand for the realization of a switching power supply device such as a DC / DC converter.

In the switching power supply device of the present invention, a plurality of switches that operate on / off by a plurality of primary side drive pulses are bridge-connected, and the DC primary side voltage and the primary side current are switched to the AC voltage and the AC current. It has a primary side inverter that converts and outputs, a primary winding and a secondary winding, and the output voltage and output current of the primary side inverter are input to the primary winding to induce an AC voltage. A transformer that outputs AC current from the secondary winding and a plurality of switches that operate on / off by a plurality of secondary drive pulses are bridge-connected to rectify the output voltage and output current of the secondary winding. Then, the secondary side inverter that outputs the DC secondary voltage and the secondary side current, the plurality of primary side drive pulses supplied to the primary side inverter, and the plurality of units supplied to the secondary side inverter. A control device that outputs a secondary side drive pulse and controls the output power of the secondary side inverter by changing the phase difference between the output value of the primary side inverter and the input value of the secondary side inverter. , Is equipped.

Then, the control device includes a comparative value having a large and small modulation factor controlled by an increase from an intermediate value of carriers generated in a plurality of cycles during one control cycle to the + side and the-side, and the carrier. The above-mentioned primary side drive pulse and the secondary side drive pulse are modulated based on the comparison result. Here, the comparison value is a value that has the modulation factor, which is an output command value, and changes to a plurality of levels during one control cycle.

In the control device of the switching power supply device of the present invention, a plurality of switches that operate on / off by a plurality of primary side drive pulses are bridge-connected, and AC voltage and AC voltage and primary side current are switched by switching the DC primary side voltage and the primary side current. It has a primary side inverter that converts to alternating current and outputs, a primary winding and a secondary winding, and is induced by inputting the output voltage and output current of the primary side inverter to the primary winding. A transformer that outputs the AC voltage and AC current from the secondary winding and a plurality of switches that operate on / off by a plurality of secondary drive pulses are bridge-connected, and the output voltage and output of the secondary winding are connected. It is a control device of a switching power supply device including a secondary side inverter that rectifies a current and outputs a DC secondary side voltage and a secondary side current.

Then, the control device outputs the plurality of primary side drive pulses supplied to the primary side inverter and the plurality of secondary side drive pulses supplied to the secondary side inverter, and the primary side inverter. Between the function of controlling the output power of the secondary side inverter by changing the phase difference between the output value of and the input value of the secondary side inverter and the carrier that generates multiple cycles during one control cycle. A comparison value having a large and small modulation factor controlled by an increase from the value to the + side and the-side is compared with the carrier to obtain a comparison result, and based on the comparison result, the primary side drive pulse and the primary side drive pulse and It has a function of modulating the secondary drive pulse. Here, the comparison value is a value that has the modulation factor, which is an output command value, and changes to a plurality of levels during one control cycle.

In the control method of the switching power supply device of the present invention, a plurality of switches that operate on / off by a plurality of primary side drive pulses are bridge-connected, and an AC voltage and an AC voltage and a primary side current are switched by switching the DC primary side voltage and the primary side current. It has a primary side inverter that converts to alternating current and outputs, a primary winding and a secondary winding, and is induced by inputting the output voltage and output current of the primary side inverter to the primary winding. A transformer that outputs the AC voltage and AC current from the secondary winding and a plurality of switches that operate on / off by a plurality of secondary drive pulses are bridge-connected, and the output voltage and output of the secondary winding are connected. It is a control method of a switching power supply device including a secondary side inverter that rectifies a current and outputs a DC secondary side voltage and a secondary side current.

Then, the control method outputs the plurality of primary side drive pulses supplied to the primary side inverter and the plurality of secondary side drive pulses supplied to the secondary side inverter, and outputs the plurality of secondary side drive pulses to the primary side inverter. The output power of the secondary side inverter is controlled by changing the phase difference between the output value of the above and the input value of the secondary side inverter, and the intermediate value of the carrier generated in a plurality of cycles during one control cycle. A comparison value having a large and small modulation factor controlled by an increase from the + side to the-side and the carrier are compared to obtain a comparison result, and based on the comparison result, the primary side drive pulse and the said. Modulates the secondary drive pulse. Here, the comparison value is a value that has the modulation factor, which is an output command value, and changes to a plurality of levels during one control cycle.

According to the present invention, when the voltage ratio between the primary side voltage and the secondary side voltage is large, it circulates in the circuit by reducing the modulation factor without changing the configuration of the basic circuit from the conventional DAB. Reactive current can be suppressed. Further, like the conventional DAB, bidirectional power conversion by phase difference is also possible. Moreover, as in Patent Document 2, the conduction loss can be reduced by reducing the current in the circuit at the time of a light load when the voltage difference between the primary side voltage and the secondary side voltage is large. Further, since the drive pulse pattern of each phase is completely different, the options for improving the problem of the conventional DAB can be increased.

Configuration diagram of the three-phase DAB type DC / DC converter according to the first embodiment of the present invention. An operation waveform diagram showing a drive pulse modulation method in the control device 40 of FIG. Typical pattern diagram of the drive pulse of FIG. Pattern diagram showing an example of the drive pulse of FIG. 1 (E2 = 0V, φ = 0 °, primary side modulation factor MR1 = secondary side modulation factor MR2 = 0) Comparison diagram of output characteristics between the conventional DAB and the DAB type DC / DC converter of the first embodiment. The figure which shows the drooping characteristic of a transformer current at the time of output short circuit of FIG. Operational waveform diagram showing the drive pulse modulation method of the second embodiment of the present invention. Configuration diagram of a conventional 3-phase DAB type DC / DC converter (conventional type DAB) The figure which shows the on / off pattern of the drive pulse of FIG. 8 and the operation waveform of a power conversion part. Conventional pattern diagram of drive pulse described in Patent Document 2.

The embodiments for carrying out the present invention will become clear when the following description of preferred embodiments is read in light of the accompanying drawings. However, the drawings are for illustration purposes only and do not limit the scope of the present invention.

(Structure of Example 1)
FIG. 1 is a block diagram of a three-phase DAB type DC / DC converter according to the first embodiment of the present invention.
The three-phase DAB type DC / DC converter of the first embodiment has the same primary side smoothing capacitor 1, primary side inverter 10, three-phase reactors 17, 18, 19, and three-phase transformer 20 as in the conventional DAB. It is composed of a power conversion unit having a secondary side inverter 30 and a secondary side smoothing capacitor 37, and a control device 40 different from the conventional one.

Similar to the conventional DAB, the switches 11 to 16, 31 to 36 constituting the primary side inverter 10 and the secondary side inverter 30 are the primary side drive pulses S11 to S16 and the secondary side supplied from the control device 40. It is an element that is turned on / off by each of the drive pulses S31 to S36, and is composed of a power semiconductor element such as a MOSFET or an IGBT. For example, when the drive pulses S11 to S16 and S31 to S36 are H, the switches 11 to 16, 31 to 36 are turned on, and when the drive pulses S11 to S16 and S31 to S36 are L, the switches 11 to 16, 31 are turned on. ~ 36 turns off. Regenerative diodes are connected in antiparallel to each of the switches 11 to 16 and 31 to 36. When each switch 11 to 16, 31 to 36 is composed of a MOSFET, for example, a parasitic diode of the MOSFET may be used.

Further, a three-phase reactor is connected in series to each of the primary winding and the secondary winding of the transformer 20. Those reactors may be replaced by the leakage inductance of the transformer 20. In FIG. 1, in order to simplify the illustration, the reactors 17, 18, and 19 are connected in series to the primary winding side of the transformer 20, respectively. For example, when the primary side voltage E1 is input, the U-phase reactor voltage VL_U, the V-phase reactor voltage VL_V, and the W-phase reactor voltage VL_W, which are the reactor voltages VL, are generated in the reactors 17, 18 and 19, respectively. A U-phase reactor current IL_U, a V-phase reactor current IL_V, and a W-phase reactor current IL_W, which are reactor currents IL, flow, respectively.

The control device 40 inputs, for example, the measured secondary side current I2 and the secondary side target current value Is, and reduces the error between the secondary side current I2 and the secondary side target current value Is. The phase difference φ of the control parameter, which is the output command value, between the output voltage vp (or output current) of the primary side inverter 10 and the input voltage vs (or input current) of the secondary side inverter 30 is obtained, and a plurality of them are obtained. Primary side drive pulses S11 to S16 and a plurality of secondary side drive pulses S31 to S36 are generated, and the primary side inverter 10 and the secondary side inverter 30 are switched and controlled to generate the output power Po of the secondary side inverter. It has a function to control.

Further, the control device 40 inputs, for example, the measured primary side voltage E1 and the secondary side voltage E2 to obtain the primary side / secondary side voltage ratio E1 / E2, and obtains the voltage ratio E1 / E2. When is large (in the worst case, the secondary side is short-circuited), the comparative value CV having a large and small modulation factor MR controlled by the increase from the intermediate value of the carrier (for example, the triangular wave carrier TC) to the + side and the-side, and its It has a function of obtaining a comparison result by comparing with the triangular wave carrier TC and modulating the primary side drive pulses S11 to S16 and the secondary side drive pulses S31 to S36 based on the comparison result. Here, the triangular wave carrier TC generates a plurality of cycles (for example, 3 cycles) during one control cycle T. The large and small modulation factor MRs are control parameters that are output command values, and have a primary side modulation factor MR1 and a secondary side modulation factor MR2. The comparison value CV having the modulation factor MR has a U-phase comparison value CV_U, a V-phase comparison value CV_V, and a W-phase comparison value CV_W, and has a plurality of levels (for example, 0, + side) during one control cycle T. It is a value that changes to (and three levels on the − side) and is generated by 120 ° in one control cycle T.

The relationship between the phase difference φ, which is a control parameter, and the modulation factor MR is as follows.
The phase difference φ and the modulation factor MR are independent control parameters. Therefore, in the control device 40, for example, the drive pulses S11 to S16 and S31 to S36 in which the phase difference φ is determined and the primary side modulation factor MR1 and the secondary side modulation factor MR2 are determined with respect to the phase difference φ are generated. Either a control method for adjusting the phase difference φ by generating drive pulses S11 to S16 and S31 to S36 for determining the primary side modulation factor MR1 and the secondary side modulation factor MR2 is adopted. can.
Such a control device 40 is composed of, for example, a central processing unit (CPU) or an individual circuit such as a semiconductor element.

(Control method of Example 1)
FIG. 2 is an operation waveform diagram showing an example of a modulation method of the primary side drive pulses S11 to S16 and the secondary side drive pulses S31 to S36 in the control device 40 of FIG.
The control device 40 obtains the primary side / secondary side voltage ratio E1 / E2 based on the measured primary side voltage E1 and the secondary side voltage E2, and when the voltage ratio E1 / E2 is large (at worst). (The secondary side is short-circuited), and the U, V, W phase comparison values CV_U, CV_V, CV_W are based on the U, V, W phase comparison values CV_U, CV_V, CV_W having the magnitude and small modulation factor MRs, which are control parameters. The comparison result is obtained by comparing with the triangular wave carrier TC, and the primary side drive pulses S11a to S16a and the secondary side drive pulses 31a to 36a are generated (modulated) and output based on the comparison result.

Here, the triangular wave carrier TC is generated in three cycles during one controlled cycle T. As shown in the upper waveform diagram of FIG. 2, the U, V, and W phase comparison values CV_U, CV_V, and CV_W are generated in three ways, 0, + side, and-side, respectively, within one control cycle T, respectively. It occurs by 120 °. As shown in the waveform diagrams of (a), (b), and (c) at the bottom of FIG. 2, the magnitude (= minimum 0, medium, large) of the modulation factor MR, which is the amplitude, is on the + side and-side. It is controlled by the increase from the intermediate value of the triangular wave carrier TC. The U, V, and W phase comparison values CV_U, CV_V, and CV_W have a phase difference of 120 ° from each other. The modulation factor MR is the same for each U, V, and W phase.

Next, a control operation (A1) when the primary side voltage E1 and the secondary side voltage E2 are close to each other and a control operation (A2) when the secondary side short circuit is performed when no load is applied will be described.

(A1) Control operation when the primary side voltage E1 and the secondary side voltage E2 are close to each other When the primary side voltage E1 and the secondary side voltage E2 are close to each other and the output is high such as near the rated output, FIG. ), The modulation factor MR becomes large, and the phase difference between the output voltage vp (or output current) of the primary side inverter 10 and the input voltage vs (or input current) of the secondary side inverter 30. φ also becomes large, and the corresponding drive pulses S11 to S16 and S31 to S36 are output from the control device 40, and the primary side inverter 10 and the secondary side inverter 30 are switched.

For example, the U-phase H-side switch 11 in the primary-side inverter 10 is off, the L-side switch 12 is on, the V-phase H-side switch 13 is on, the L-side switch 14 is off, and the W-phase H-side switch 15. Is off, and the L-side switch 16 is on. Further, the phase difference φ is deviated, the U-phase H-side switch 31 in the secondary-side inverter 30 is off, the L-side switch 32 is on, the V-phase H-side switch 33 is on, and the L-side switch 34 is off. The H-side switch 35 of the W phase is turned off, and the L-side switch 36 is turned on.

Then, in FIG. 1, the + side of the primary side voltage E1 source → the H side switch 13 → the reactor 18 → the primary winding of the transformer 20 → the reactor 17 → the L side switch 12 → the negative side of the primary side voltage E1 source. And the path on the + side of the primary voltage E1 source → H side switch 13 → reactor 18 → primary winding of the transformer 20 → reactor 19 → L side switch 16 → negative side path of the primary voltage E1 source And, the primary side current I1 flows through. Correspondingly, an induced electromotive current is generated in the secondary winding of the transformer 20, and the secondary winding of the transformer 20 → the diode of the H side switch 33 → the load → the diode of the L side switch 32 → the secondary winding. The secondary side current I2 flows in the wire path and the secondary winding of the transformer 20 → the diode of the H side switch 33 → the load → the diode of the L side switch 36 → the secondary winding path. The switching operation of one cycle T ends.

(A2) Control operation in the case of a secondary side short circuit when there is no load For example, when the primary side / secondary side voltage ratio E1 / E2 is large due to the secondary side short circuit when there is no load and the output is throttled, the modulation factor Both MR and the phase difference φ become small, and the corresponding drive pulses S11 to S16 and S31 to S36 are output from the control device 40, and the primary side inverter 10 and the secondary side inverter 30 are switched.

FIG. 3 is a typical pattern diagram of the drive pulse of FIG. 1 (for example, the primary side drive pulse S11 to S16).
U-phase complementary drive pulses S11, S12, V-phase complementary drive pulses S13, S14, and W-phase complementary drive pulses S15, S16 have large and medium pulse widths (duty ratio D = 0. 5) By exchanging the small ones in order, all the switches 11 to 16 are in the same state (for example, all the L side switches 12, 14 and 16 are in the on state, or all the H side switches 11, 13 and 15 are in the on state). Is generated.
The typical pattern diagram of the secondary side drive pulses S31 to S36 is the same as that in FIG.

For example, when the secondary side voltage E2 becomes 0V (short circuit state) due to load fluctuation, all switches 11 to 16 (and 31 to 36) are in the same state (for example, all L side switches) as shown in FIG. 12, 14, 16 are on, or all H-side switches 11, 13, 15 are on. Similarly, all L-side switches 32, 34, 36 are on, or all H-side switches 31, 33, A section in which 35 is on) is generated.
When the secondary side voltage E2 becomes 0V, the electric charge accumulated in the smoothing capacitor 37 is discharged, and a large pulse-shaped current is instantaneously generated, but after that, the secondary side current I2 is maintained at a constant value. At this time, in the control device 40, since the primary side / secondary side voltage ratio E1 / E2 is the maximum value, as shown in FIG. 2, the desired primary side modulation factor MR1 and the secondary side modulation factor MR2 ( For example, MR1 = MR2 = minimum 0, medium or large) is calculated.

For example, as shown in (a) to (c) at the bottom of FIG. 2, the modulation factor MR (MR1, MR2) of the U, V, W phase comparison values CV_U, CV_V, CV_W is the minimum 0, medium or large. , U-phase drive pulses S11, S12 (S31, S32), V-phase drive pulses S13, S14 (S33, S34), and W-phase drive pulses S15, S16 (S35, S36) change pulse widths. However, as shown in the waveform diagram of (a) at the bottom of FIG. 2, when the modulation factor MR is 0, which is the minimum, the U, V, and W phase comparison values CV_U, CV_V, and CV_W are intermediate values of the triangular wave carrier TC. Since it becomes flat, the pulse widths of the primary side drive pulses S11 to S16 and the secondary side drive pulses S31 to S36 are all the same.
Then, the control device 40 generates the primary side drive pulses S11 to S16 and the secondary side drive pulses S31 to S36 based on the comparison result between the U, V, W phase comparison values CV_U, CV_V, CV_W and the triangular wave carrier TC. Then, the switches 11 to 16 in the primary side inverter 10 and the switches 31 to 36 in the secondary side inverter 30 are turned on / off.

Then, as shown in FIG. 3, there is a section in which all L-side switches 12, 14, 16 (and 32, 34, 36) and all H-side switches 11, 13, 15 (and 31, 33, 35) are turned on. The generated current that circulates in the circuit is theoretically 0A. Therefore, it is possible to suppress the reactive current circulating in the circuit, which becomes remarkable especially when the voltage difference between the input and output is large.

FIG. 4 is a pattern diagram (E2 = 0V, φ = 0 °, primary side modulation factor MR1 = secondary side modulation factor MR2 = 0) showing an example of the drive pulses S11 to S16 and S31 to S36 of FIG. ..
FIG. 4 shows an operating waveform in a drooping state in which the secondary voltage E2 in the three-phase DAB type DC / DC converter of FIG. 1 is 0 V. The primary side drive pulses S11 to S16 and the secondary side drive pulses S31 to S36 do not include a dead time. Since power conversion cannot be performed when the secondary voltage E2 is 0 V, the control parameters phase difference φ and the modulation factor MR (= primary side modulation factor MR1 and secondary side modulation factor MR2) are all used. It is set to 0.

Similar to the waveform diagram of (a) in the lower part of FIG. 2, when the primary side modulation factor MR1 and the secondary side modulation factor MR2 are the minimum 0, the primary side drive pulses S11 to S16 and the secondary side drive pulse S31 In ~ S36, the U, V, and W phases have the same waveform state (that is, the pulse widths are all the same). Then, all H side switches 11, 13, 15, 31, 33, 35 are on, all L side switches 12, 14, 16, 32, 34, 36 are off, or all H side switches 11, 13, 15, 31, 33, 35 is off, and all L-side switches 12, 14, 16, 32, 34, 36 are on. In this operating state, the circuit connection state in which the voltage is applied to the reactor voltage VL (for example, the U-phase reactor voltage VL_U) does not occur (VL_U = 0V). Therefore, the reactor current IL (for example, the U-phase reactor current IL_U) does not generate a current and is always 0 A. The V phase and the W phase are also the same as the U phase reactor voltage VL_U and the U phase reactor current IL_U. Therefore, the primary side current I1 is not generated (0A), and the reactive current component is completely suppressed. As described above, it can be seen that the reactive current circulating in the circuit, which becomes remarkable especially when the voltage difference between the input and output is large, is suppressed.

(Effect of Example 1)
According to the first embodiment, the following effects (1) to (4) are obtained.
(1) As shown in the pattern diagram of the primary side drive pulses S11 to S16 (and the same applies to the secondary side drive pulses S31 to S36) in FIG. 3, the pulse widths of the U, V, and W phases are large and medium (duty). By exchanging the ratio D = 0.5) and the small in order, a section in which all the switches 11 to 16 (and / or 31 to 36) are in the same state is created. This drive pulse modulation method is shown in FIG. As shown in FIG. 2, under the condition that the modulation factor MR is the minimum (= 0) (the state of (a) in the lower part of FIG. 2 and the state of FIG. 4), the U, V, and W all-phase switches 11 The operating states of ~ 16 (and / or 31 ~ 36) are the same. At this time, there are two circuit states (U-phase H-side switch 11, V-phase H-side switch 13, W-phase H-side switch 15 are on, U-phase L-side switch 12, V-phase L-side) at any timing. Switch 14, W-phase L-side switch 16 is off, or U-phase L-side switch 12, V-phase L-side switch 14, W-phase L-side switch 16 is on, U-phase H-side switch 11, V-phase H-side. Only the switch 13 or the W-phase H-side switch 15 is in the off state) can be generated (the same applies to the switches S31 to S36). In this state, as shown in FIG. 4, no voltage can be generated between the lines of the winding portion (the part where the reactors 17 to 19 and the transformer 20 are located) in either of the two patterns (for example, U). Phase reactor voltage VL_U = 0V), the mode in which the voltage is applied to the winding portion disappears. Thereby, the current circulating in the circuit can be reduced (for example, U-phase reactor current IL_U = 0A, primary side current I1 = 0A).
As described above, according to the first embodiment, when the primary side / secondary side voltage ratio E1 / E2 is large, the modulation factor MR (MR1, MR2) does not change the configuration of the basic circuit from the conventional DAB. By reducing the value of, the reactive current circulating in the circuit can be suppressed. However, if the modulation factor MR (MR1, MR2) is reduced, the active power component is also suppressed, so if the primary side / secondary side voltage ratio E1 / E2 is small, the modulation factor MR (MR1, MR2) is too small. MR2) should not be reduced.

(2) FIG. 5 is a comparison diagram of output characteristics between the conventional DAB and the DAB type DC / DC converter of the first embodiment. The horizontal axis of FIG. 5 is the phase difference φ (°), and the vertical axis is the conversion power (normalized) which is the output power Po.
As shown in FIG. 5, in the first embodiment, bidirectional power conversion by the phase difference φ is also possible as in the conventional DAB.

(3) FIG. 6 is a diagram showing a drooping characteristic of the transformer current at the time of output short circuit of FIG. 1 (= secondary side voltage E2 is 0V). The horizontal axis of FIG. 6 is the modulation factor MR (primary side modulation factor MR1 = secondary side modulation factor MR2), and the vertical axis is the transformer current (Arms) (same as the reactor current IL). The modulation factor MR has a minimum of 0.0 and a maximum of 1.0.
In the first embodiment, as shown by the solid transformer current effective value in FIG. 6, the modulation factor MR (MR1, MR2), which is a control parameter, is changed when the output is short-circuited (= the secondary voltage E2 is 0V). Thereby, the effective value of the transformer current (same as the reactor current IL) can be changed. For example, when the modulation factor MR (MR1, MR2) is 0.0, which is the minimum, the effective value of the transformer current is 0A.
In the control of the phase difference φ of the conventional DAB, as shown by the current value (image) shown by the broken line in FIG. 6, the maximum current always flows, so that the transformer current cannot be reduced. On the other hand, in the first embodiment, as in Patent Document 2, the conduction loss can be reduced by reducing the in-circuit current at the time of a light load when the voltage difference between the input and output is large. ..

(4) In the first embodiment, as shown in FIG. 3, since the patterns of the U, V, and W phase drive pulses S11 to S16 (and S31 to S36) are completely different, there is an option for improving the problem of the conventional DAB. Can also be increased.

The switching power supply device of the present invention can be applied to a single-phase or four-phase or more DAB-type DC / DC converter other than the three-phase DAB-type DC / DC converter of the first embodiment.
Hereinafter, as Example 2, for example, a single-phase DAB type DC / DC converter will be described.

(Structure of Example 2)
In the single-phase DAB type DC / DC converter of the second embodiment, for example, in FIG. 1, the W-phase switches 15, 16, 35, 36 and the reactor 19 are omitted, and the three-phase transformer 20 is a single-phase transformer. It is replaced with a transformer (hereinafter, designated by reference numeral "20A"), and further, the drive pulses S15, S16, S35, and S36 output from the control device 40 are omitted.
The control device of the second embodiment (hereinafter, referred to as “40A”) has the measured secondary side current I2 and the secondary side target current value Is in substantially the same manner as the control device 40 of the first embodiment. The output voltage vp (or output current) of the primary side inverter (hereinafter referred to as "10A") that is input and reduces the error between the secondary side current I2 and the secondary side target current value Is. And the phase difference φ of the control parameter, which is the output command value, between the input voltage vs (or the input current) of the secondary side inverter (hereinafter referred to as “30A”) is obtained, and a plurality of primary side drives are obtained. A function to generate pulses S11 to S14 and a plurality of secondary side drive pulses S31 to S34, switch and control the primary side inverter 10A and the secondary side inverter 30A, and control the output power Po of the secondary side inverter 30A. Have.

Further, the control device 40A inputs, for example, the measured primary side voltage E1 and the secondary side voltage E2 to obtain the primary side / secondary side voltage ratio E1 / E2, and obtains the voltage ratio E1 / E2. When is large (in the worst case, the secondary side is short-circuited), the comparative value CV having a large and small modulation factor MR controlled by the increase from the intermediate value of the carrier (for example, the triangular wave carrier TC) to the + side and the-side, and its It has a function of obtaining a comparison result by comparing with the triangular wave carrier TC and modulating the primary side drive pulses S11 to S14 and the secondary side drive pulses S31 to S34 based on the comparison result. Here, the triangular wave carrier TC generates a plurality of cycles (for example, two cycles) during one controlled cycle T. The large and small modulation factor MRs are control parameters that are output command values, and have a primary side modulation factor MR1 and a secondary side modulation factor MR2. The comparison value CV having the modulation factor MR has a U-phase comparison value CV_U, a V-phase comparison value CV_V, and a W-phase comparison value CV_W, and has a plurality of levels (for example, + side and-) during one control cycle T. It is a value that changes to two levels on the side) and is generated by 180 ° in one control cycle T.

The relationship between the phase difference φ, which is a control parameter, and the modulation factor MR is as follows.
Similar to the first embodiment, the phase difference φ and the modulation factor MR are independent control parameters. Therefore, in the control device 40A, for example, the drive pulses S11 to S14 and S31 to S34 in which the phase difference φ is determined and the primary side modulation factor MR1 and the secondary side modulation factor MR2 are determined with respect to the phase difference φ are generated. Either a control method for adjusting the phase difference φ by generating drive pulses S11 to S14 and S31 to S34 for determining the primary side modulation factor MR1 and the secondary side modulation factor MR2 is adopted. can.

(Control method and effect of Example 2)
FIG. 7 is an operation waveform diagram showing an example of a modulation method of the primary side drive pulses S11 to S14 and the secondary side drive pulses S31 to S34 in the control device 40A of the second embodiment of the present invention.
As shown in FIG. 7, the triangular wave carrier TC is generated in two cycles during one controlled cycle T. As shown in the upper waveform diagram of FIG. 7, the U and V phase comparison values CV_U and CV_V are generated in two ways, the + side and the-side, respectively, and are generated by 180 ° in one control cycle T, respectively. As shown in the waveform diagrams of (a), (b), and (c) at the bottom of FIG. 7, the magnitude (= minimum 0, medium, large) of the modulation factor MR, which is the amplitude, is on the + side and-side. It is controlled by the increase from the intermediate value of the triangular wave carrier TC. The U and V phase comparison values CV_U and CV_V have a phase difference of 180 ° from each other. The modulation factor MR is the same for each U and V phase.

According to the control device 40A of the second embodiment, the operation is substantially the same as that of the first embodiment, and the same effect as that of the first embodiment is obtained.

(Modification example)
The present invention is not limited to the above-mentioned Examples 1 and 2, and various usage forms and modifications are possible. For example, the configuration of the power conversion unit in the DAB type DC / DC converter shown in FIG. 1 may be changed to a configuration other than that shown in the figure.

1 Primary side smoothing capacitor 10 Primary side inverter 11 to 16, 31 to 36 Switch 17 to 19 Reactor 20 Transformer 30 Secondary side inverter 37 Secondary side smoothing capacitor 40 Control device

Claims (5)

Multiple switches that operate on / off by multiple primary side drive pulses are bridge-connected, and the primary side that switches the DC primary side voltage and primary side current to convert them into AC voltage and AC current and outputs them. With an inverter
It has a primary winding and a secondary winding, the output voltage and output current of the primary side transformer are input to the primary winding, and the induced AC voltage and AC current are output from the secondary winding. Transformers and
A plurality of switches that operate on / off by a plurality of secondary drive pulses are bridge-connected, rectify the output voltage and output current of the secondary winding, and output the DC secondary voltage and secondary current. Secondary side inverter and
The plurality of primary side drive pulses supplied to the primary side inverter and the plurality of secondary side drive pulses supplied to the secondary side inverter are output, and the output value of the primary side inverter and the secondary side inverter are output. A control device that controls the output power of the secondary inverter by changing the phase difference with the input value of the side inverter, and
Equipped with
The control device is
It has a large and small modulation factor, which is an output command value, controlled by an increase from the intermediate value of carriers generated in multiple cycles during one control cycle to the + side and-side, and during the control cycle. The comparison value that changes to a plurality of levels is compared with the carrier to obtain a comparison result, and the primary side drive pulse and the secondary side drive pulse are modulated based on the comparison result.
A switching power supply that features. The carrier is a triangular wave carrier generated in three cycles during one controlled cycle.
The comparison value changes to three levels of 0, + side and-side during one control cycle, and is generated by 120 ° each within one control cycle.
The switching power supply device according to claim 1. The carrier is a triangular wave carrier generated in two cycles during one controlled cycle.
The comparison value changes to two levels, + side and-side, during one control cycle, and occurs by 180 ° each within one control cycle.
The switching power supply device according to claim 1. Multiple switches that operate on / off by multiple primary side drive pulses are bridge-connected, and the primary side that switches the DC primary side voltage and primary side current to convert them into AC voltage and AC current and outputs them. With an inverter
It has a primary winding and a secondary winding, the output voltage and output current of the primary side transformer are input to the primary winding, and the induced AC voltage and AC current are output from the secondary winding. Transformers and
A plurality of switches that operate on / off by a plurality of secondary drive pulses are bridge-connected, rectify the output voltage and output current of the secondary winding, and output the DC secondary voltage and secondary current. Secondary side inverter and
It is a control device of a switching power supply device provided with
The plurality of primary side drive pulses supplied to the primary side inverter and the plurality of secondary side drive pulses supplied to the secondary side inverter are output, and the output value of the primary side inverter and the secondary side inverter are output. The function to control the output power of the secondary side inverter by changing the phase difference with the input value of the side inverter, and
It has a large and small modulation factor, which is an output command value, controlled by an increase from the intermediate value of carriers generated in multiple cycles during one control cycle to the + side and-side, and during the control cycle. A function to obtain a comparison result by comparing a comparison value that changes to a plurality of levels with the carrier, and to modulate the primary side drive pulse and the secondary side drive pulse based on the comparison result.
A control device for a switching power supply device, characterized by having. Multiple switches that operate on / off by multiple primary side drive pulses are bridge-connected, and the primary side that switches the DC primary side voltage and primary side current to convert them into AC voltage and AC current and outputs them. With an inverter
It has a primary winding and a secondary winding, the output voltage and output current of the primary side transformer are input to the primary winding, and the induced AC voltage and AC current are output from the secondary winding. Transformers and
A plurality of switches that operate on / off by a plurality of secondary drive pulses are bridge-connected, rectify the output voltage and output current of the secondary winding, and output the DC secondary voltage and secondary current. Secondary side inverter and
It is a control method of a switching power supply device provided with
The plurality of primary side drive pulses supplied to the primary side inverter and the plurality of secondary side drive pulses supplied to the secondary side inverter are output, and the output value of the primary side inverter and the secondary side inverter are output. While controlling the output power of the secondary side inverter by changing the phase difference with the input value of the side inverter,
It has a large and small modulation factor, which is an output command value, controlled by an increase from the intermediate value of carriers generated in multiple cycles during one control cycle to the + side and-side, and during the control cycle. The comparison value that changes to a plurality of levels is compared with the carrier to obtain a comparison result, and the primary side drive pulse and the secondary side drive pulse are modulated based on the comparison result.
A method of controlling a switching power supply, which is characterized by the fact that.

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