JP2022036558A - Drive circuit of switching element - Google Patents

Abstract

To provide a drive circuit of a switching element which can reduce an unnecessary loss in a normal operation while appropriately suppressing overcurrent in a short circuit.SOLUTION: A drive circuit Dr of a switch SW comprises a Zener diode ZD, a clamp control switch T3 and a diode D1. When voltage Vds between a drain and a source of the switch SW is larger than a threshold, power supply voltage is applied to a gate of the clamp control switch T3, the clamp control switch T3 switches between a gate of the switch SW and the Zener diode ZD to an energization state, and limits gate voltage of the switch SW to clamp voltage or lower by the Zener diode ZD. On the other hand, when the voltage Vds is equal to or less than the threshold, the gate voltage of the clamp control switch T3 is discharged via the diode D1 to cut off between the gate of the switch SW and the Zener diode ZD.SELECTED DRAWING: Figure 2

Description

The present invention relates to a drive circuit for a switching element.

Conventionally, a drive circuit for driving a switching element on and off at high speed by controlling the gate voltage of the switching element has been known. Some drive circuits of this type have a configuration for preventing short-circuit accidents (see, for example, Patent Document 1 and Patent Document 2).

The drive circuit described in Patent Document 1 determines a short-circuit accident based on the voltage between the collector and the emitter by the short-circuit detection circuit, and when the short-circuit accident is determined, the gate voltage clamp circuit determines the gate voltage. Overcurrent is suppressed by limiting it to below the clamp voltage.

On the other hand, in the drive circuit described in Patent Document 2, the gate voltage is limited to a predetermined clamp voltage or less until a predetermined predetermined time elapses, and overcurrent is suppressed.

Japanese Patent No. 4901083 Japanese Patent No. 5716711

By the way, in the drive circuit described in Patent Document 1, after the short circuit accident is determined by the short circuit detection circuit, the gate voltage clamp circuit limits the gate voltage to a predetermined clamp voltage or less, so that the gate voltage is reached after the short circuit accident occurs. There was a problem that there was a delay before limiting the voltage, and during that time, the overcurrent could not be sufficiently suppressed.

On the other hand, in the drive circuit described in Patent Document 2, although there is no such problem, the gate voltage is excessively suppressed in order to limit the gate voltage until the specified time elapses regardless of the presence or absence of a short circuit accident. Unnecessary loss could occur.

The present invention has been made in view of the above circumstances, and provides a drive circuit for a switching element capable of appropriately suppressing an overcurrent at the time of a short circuit and reducing unnecessary loss during normal operation. The main purpose.

A means for solving the above problem is a clamp that limits the gate voltage of the switching element to a predetermined clamp voltage or less in the drive circuit of the switching element that opens and closes the switching element by controlling the gate voltage of the switching element. An anode is connected to a clamp control switch that switches between an energized state and an energized cutoff state between the unit, the gate of the switching element, and the gate of the clamp control switch, and the cathode is on the high potential side of the switching element. A first diode connected to the terminal is provided, and the clamp control switch is predetermined from the power supply to the gate of the clamp control switch when the voltage at the high potential side terminal of the switching element is larger than the threshold value. The power supply voltage of the above is applied to switch between the gate of the switching element and the clamp portion into an energized state, and the clamp portion limits the gate voltage of the switching element to the clamp voltage or less, while the switching element. When the voltage at the high potential side terminal is equal to or lower than the threshold voltage, the gate voltage of the clamp control switch is discharged via the first diode, and the energization is cut off between the gate of the switching element and the clamp portion. Switch to the state.

It is known that whether or not a short circuit has occurred can be determined by the voltage at the high potential side terminal of the switching element. Therefore, in the above means, when the voltage at the high potential side terminal of the switching element is larger than the threshold value, a predetermined power supply voltage is applied from the power supply to the gate of the clamp control switch, and the gate of the switching element is set by the clamp control switch. The connection with the clamp part was switched to the energized state, and the gate voltage of the switching element was limited to the clamp voltage or less by the clamp part. On the other hand, when the voltage at the high potential side terminal of the switching element is equal to or less than the threshold value, the gate voltage of the clamp control switch is discharged via the diode, and the clamp control switch between the gate of the switching element and the clamp portion. Was switched to the power cutoff state.

When a short circuit occurs, even if the switching element is turned on, the voltage on the high potential side terminal side of the switching element does not decrease and becomes a value larger than the threshold value. Therefore, the gate and clamp of the switching element. The element remains energized, and the clamp portion keeps limiting the gate voltage of the switching element to the clamp voltage or less. That is, in the case of a short circuit, the gate voltage is continuously limited to the clamp voltage or less by the clamp portion. Does not occur. Therefore, even if a short circuit occurs, the occurrence of overcurrent can be appropriately suppressed.

On the other hand, when a short circuit does not occur, the gate voltage of the clamp control switch is discharged via the first diode as the voltage of the high potential side terminal of the switching element drops, and the clamp control switch is turned off. .. Therefore, the gate of the switching element and the clamp portion are cut off, and the limitation of the gate voltage by the clamp portion is released. Therefore, it is possible to prevent the gate voltage of the switching element from being unnecessarily limited and reduce the loss.

The figure which shows the whole structure of a control system. The figure which shows the drive circuit. Time chart of gate voltage Vgs (SW) and voltage Vds between drain and source. Time chart of gate voltage Vgs (SW) and voltage Vds between drain and source. The figure which shows the drive circuit of 2nd Embodiment. The figure which shows the drive circuit of 3rd Embodiment. The figure which shows the drive circuit in another example. The figure which shows the drive circuit of 4th Embodiment. The figure which shows the drive circuit of 5th Embodiment. The figure which shows the drive circuit of another example. The figure which shows another example of the clamp part. The figure which shows the drive circuit in the modification. (A) is a time chart of the gate voltage Vgs (SW) in the modified example, (b) is a time chart of the voltage of the connection point M52 in the modified example, and (c) is the time of the output signal of the comparator CMP1 in the modified example. chart.

<First Embodiment>
Hereinafter, the first embodiment in which the drive circuit according to the present invention is embodied will be described with reference to the drawings. The control system and drive circuit according to the present invention are applied to a vehicle (for example, an electric vehicle or a hybrid vehicle) in this embodiment. In each of the following embodiments, the parts that are the same or equal to each other are designated by the same reference numerals in the drawings.

As shown in FIG. 1, the control system includes a rotary electric machine 10 and an inverter. The inverter includes a switching device unit 20 and a control unit 30 for controlling the rotary electric machine 10. In this embodiment, the rotary electric machine 10 includes a three-phase winding 11 connected in a star shape. The control system of this embodiment is mounted on a vehicle. The rotor of the rotary electric machine 10 is connected to the drive wheel of the vehicle so as to be able to transmit power. The rotary electric machine 10 is, for example, a synchronous machine.

The rotary electric machine 10 is connected to the DC power supply 21 via the switching device unit 20. In the present embodiment, the DC power supply 21 is a secondary battery. The switching device unit 20 includes a smoothing capacitor 22.

The switching device unit 20 includes a series connection body of the upper and lower arm switches SW for each of the U, V, and W phases. In the present embodiment, each switch SW is an N-channel MOSFET, but may be an IGBT. The switch SW corresponds to a switching element. A freewheel diode is connected in antiparallel to each switch SW. In each switch SW of the present embodiment, the high potential side terminal is a drain (collector in the case of an IGBT), and the low potential side terminal is a source (emitter in the case of an IGBT).

In each phase, the first end of the winding 11 is connected to the connection point between the low potential side terminal of the upper arm switch SW and the high potential side terminal of the lower arm switch SW. The second end of the winding 11 of each phase is connected at the neutral point.

The control unit 30 drives each switch SW of the switching device unit 20 in order to control the control amount of the rotary electric machine 10 to a command value. The control amount is, for example, torque. The control unit 30 individually sets the drive signal IN corresponding to the upper and lower arm switch SW to the upper and lower arm switch SW in order to alternately turn on the upper and lower arm switch SW while sandwiching the dead time. Output to the provided drive circuit Dr. The drive signal IN takes either an on command instructing the switch to be switched to the on state or an off command instructing the switch to be switched to the off state.

Subsequently, the drive circuit Dr will be described with reference to FIG. Each drive circuit Dr of the upper and lower arms of the present embodiment has basically the same configuration.

The drive circuit Dr includes a constant voltage power supply 40, a charging switch T1, and a charging resistor R1. The charging switch T1 of this embodiment is a P-channel MOSFET. The gate (open / close control terminal) of the switch SW is connected to the constant voltage power supply 40 via the charging switch T1 and the charging resistor R1. The output voltage Vcc (for example, 15V) of the constant voltage power supply 40 is the power supply voltage supplied to the gate of the switch SW, and corresponds to the upper limit value of the gate voltage Vgs (SW) of the switch SW. The constant voltage power supply 40 does not have to be built in the drive circuit Dr, and may be input from the outside.

The drive circuit Dr includes a discharge resistor R2 and a discharge switch T2. The discharge switch T2 of this embodiment is an N-channel MOSFET. The source of the switch SW as a ground portion is connected to the gate of the switch SW via the discharge resistor R2 and the discharge switch T2.

The drive circuit Dr includes a drive controller 50. The drive controller 50 acquires the drive signal IN output from the control unit 30. When the acquired drive signal IN is an ON command, the drive controller 50 performs a charging process. The charging process is a process in which the charging switch T1 is turned on and the discharge switch T2 is turned off. According to the charging process, the gate voltage Vgs (SW) of the switch SW becomes equal to or higher than the threshold voltage Vth (SW), and the switch SW is switched to the ON state.

When the acquired drive signal IN is an off command, the drive controller 50 performs a discharge process. The discharge process is a process in which the charge switch T1 is turned off and the discharge switch T2 is turned on. According to the discharge process, the gate voltage Vgs (SW) of the switch SW becomes less than the threshold voltage Vth (SW), and the switch SW is switched to the off state.

The function provided by the drive controller 50 can be provided by, for example, software recorded in a substantive memory device, a computer for executing the software, hardware, or a combination thereof.

The drive circuit Dr includes a short circuit detection circuit 60 and a clamp circuit 70. The short circuit detection circuit 60 employs a DESAT detection method for detecting the occurrence of a short circuit. The DESAT detection method will be described in detail. Normally, during the ON operation of the switch SW, the voltage Vds between the drain and the source of the switch SW drops to the saturation voltage (drain saturation voltage, in the case of the IGBT, the collector saturation voltage). However, when a large current is generated due to a short circuit, the voltage Vds between the drain and the source once dropped increases, and the voltage Vds between the drain and the source becomes a voltage (unsaturated voltage) that is not a saturated voltage. Therefore, in the DESAT detection method, the occurrence of a short circuit is detected by detecting the occurrence of such an unsaturated voltage while the switch SW is on.

A specific configuration of the short circuit detection circuit 60 will be described. The short circuit detection circuit 60 includes a diode D2, a comparator CMP, a capacitor C1, a reset switch T4, and a resistor R4. The diode D2 is a high withstand voltage diode, the cathode side is connected to the drain side of the switch SW, and the anode side is connected to one end of the capacitor C1. The other end of the capacitor C1 is connected to the source side of the switch SW. The constant voltage power supply 40 is connected to the connection point M1 between the diode D2 and the capacitor C1 via the resistor R4.

As a result, the capacitor C1 is charged by the constant voltage power supply 40. Further, the diode D2 is configured to limit the voltage between the terminals of the capacitor C1 and clamp (fix) to the magnitude of the saturation voltage when the voltage Vds between the drain and the source of the switch SW is the saturation voltage. ing. On the other hand, when the voltage is unsaturated, the voltage between the terminals of the capacitor C1 is not clamped (fixed) to the saturated voltage. That is, the potential on the cathode side of the diode D2 becomes high, and the voltage between the terminals of the capacitor C1 rises.

The reset switch T4 is an N-channel MOSFET and is connected in parallel to the capacitor C1. The gate of the reset switch T4 is connected to the drive controller 50, and the reset switch T4 is on / off controlled by an instruction from the drive controller 50. Specifically, while the switch SW is off, the reset switch T4 is turned on to discharge the capacitor C1, and while the switch SW is on, the reset switch T4 is turned off to charge the capacitor C1. ..

As a result, during the on operation of the switch SW, the voltage between the terminals of the capacitor C1 corresponds to the voltage Vds between the drain and the source of the switch SW. That is, when the voltage Vds between the drain and the source is a saturated voltage, the voltage between the terminals of the capacitor C1 becomes a value corresponding to the saturated voltage, and when it is an unsaturated voltage, it becomes a value corresponding to the unsaturated voltage. Become. Therefore, by monitoring the voltage at the connection point M1 between the diode D2 and the capacitor C1, it is possible to detect the unsaturated voltage in the voltage Vds between the drain and the source of the switch SW.

The inverting input terminal (-input terminal) of the comparator CMP is connected to the connection point M1, and the voltage at the connection point M1 is input. Further, a voltage source (not shown) of the reference voltage is connected to the non-inverting input terminal (+ input terminal) of the comparator CMP, and a determination value having the same magnitude as the saturation voltage is input. The comparator CMP is configured to compare the voltages input to the inverting input terminal and the non-inverting input terminal and output the result to the drive controller 50. That is, when the voltage of the connection point M1 (that is, the voltage between the terminals of the capacitor C1) reaches the determination value, the comparator CMP outputs a detection signal notifying that a short circuit has been detected to the drive controller 50.

The clamp circuit 70 includes a Zener diode ZD (TVS for surge absorption, power Zener) as a clamp portion cl that limits the gate voltage (open / close control voltage) of the switch SW to a predetermined clamp voltage or less, and a gate (open control) of the switch SW. It is provided with a clamp control switch T3 for switching between the energization and the energization cutoff state between the terminal) and the Zener diode ZD. The clamp control switch T3 is an N-channel MOSFET and is connected in series with the Zener diode ZD. Specifically, one end (high potential side terminal, drain) of the clamp control switch T3 is connected to the gate of the switch SW, and the other end (low potential side terminal, source) is on the cathode side of the Zener diode ZD. It is connected. The anode side of the Zener diode ZD is connected to the source of the switch SW.

The gate of the clamp control switch T3 is connected to the constant voltage power supply 40 via the resistor R3, and is configured so that a voltage is applied from the constant voltage power supply 40. Further, the connection point M2 between the gate of the clamp control switch T3 and the resistance R3 is connected to the drain side of the switch SW via the diode D1. More specifically, the diode D1 is a high withstand voltage diode, the cathode side is connected to the drain side of the switch SW, and the anode side is connected to the connection point M2.

Therefore, similarly to the diode D2, the diode D1 limits the gate voltage Vgs (T3) of the clamp control switch T3 when the voltage Vds between the drain and the source of the switch SW is the saturation voltage, and the saturation voltage is increased. It is configured to be clamped (fixed) to the size. On the other hand, when the voltage is unsaturated, the potential on the cathode side of the diode D1 becomes high, and the gate voltage Vgs (T3) of the clamp control switch T3 rises.

Next, the operation in the clamp circuit 70 configured in this way will be described. The explanation is based on the premise that the switch SW is designed so that when the gate voltage Vgs (SW) of the switch SW reaches the voltage Vgmax, the maximum current specified in the control system is sufficiently passed and the switch SW reaches a saturated state. do.

First, the operation in a normal time (that is, in a non-short circuit) will be described with reference to FIG. As shown in FIG. 3A, when the charging process is started and the charging switch T1 is turned on, the gate voltage Vgs (SW) of the switch SW rises. Then, the gate voltage Vgs (SW) of the switch SW rises, and in the process of reaching the voltage Vgmax, the voltage Vds between the drain and the source of the switch SW drops to the saturation voltage.

When the voltage Vds between the drain and the source of the switch SW becomes equal to or less than a predetermined threshold in the process of decreasing (time point t11), specifically, Vds = {V (ZD) + Vth (T3) -Vf (D1). )} When it becomes less than or equal to, the gate voltage Vgs (T3) of the clamp control switch T3 is discharged at high speed via the diode D1, and the clamp control switch T3 is turned off. Note that V (ZD) is the breakdown voltage of the Zener diode ZD and corresponds to the clamp voltage. Vth (T3) is the threshold voltage of the gate voltage Vgs (T3) when the clamp control switch T3 is turned on. Further, Vf (D1) is a forward voltage of the diode D1. Further, these values satisfy V (ZD) + Vth (T3) -Vf (D1) <Vgmax.

When the clamp control switch T3 is turned off, the Zener diode ZD is disconnected from the gate of the switch SW (the energization is cut off). After that, the gate voltage Vgs (SW) of the switch SW rises to the power supply voltage Vcc (> Vgmax) without being affected by the Zener diode ZD.

On the other hand, the operation at the time of a short circuit will be described with reference to FIG. At the time of a short circuit, as described above, even if the gate voltage Vgs (SW) of the switch SW reaches the voltage Vgmax, the voltage Vds does not decrease until the saturation voltage is reached. That is, the voltage Vds between the drain and the source of the switch SW does not fall below a predetermined threshold value. Therefore, the clamp control switch T3 is not turned off, and the gate voltage Vgs (SW) of the switch SW is limited by the Zener diode ZD and is clamped to the clamp voltage V (ZD). As a result, it is possible to prevent the gate voltage Vgs (SW) of the switch SW from reaching its maximum value, the power supply voltage Vcc, and suppress the short-circuit current.

The configuration of the first embodiment described above has the following effects.

It is known that whether or not a short circuit has occurred can be detected by the voltage on the drain side (high potential side terminal side) of the switch SW, that is, the voltage Vds between the drain and the source. Therefore, when the voltage Vds is larger than the threshold value, that is, when Vds> V (ZD) + Vth (T3) -Vf (D1), the power supply voltage (predetermined) is sent from the constant voltage power supply 40 to the gate of the clamp control switch T3. The voltage) is applied to turn on the clamp control switch T3, and the gate of the switch SW and the Zener diode ZD are switched to the energized state. Then, the gate voltage Vgs (SW) of the switch SW is limited to the clamp voltage V (ZD) or less by the Zener diode ZD.

On the other hand, when the voltage Vds between the drain and the source is equal to or less than the threshold value, and when Vds ≦ V (ZD) + Vth (T3) −Vf (D1), the gate of the clamp control switch T3 via the diode D1. The voltage Vgs (T3) was discharged, the clamp control switch T3 was turned off, and the gate of the switch SW and the Zener diode ZD were switched to the energization cutoff state.

When a short circuit occurs, even if the switch SW is turned on, the voltage on the drain side of the switch SW does not decrease and the voltage Vds becomes a value larger than the threshold value. , The gate of the switch SW and the Zener diode ZD remain energized (connected). As a result, the gate voltage Vgs (SW) of the switch SW is continuously limited to the clamp voltage V (ZD) or less by the Zener diode ZD. That is, since the gate voltage Vgs (SW) is limited to the clamp voltage V (ZD) or less from the beginning by the Zener diode ZD, unlike the case where the limit of the gate voltage Vgs (SW) is instructed after determining the presence or absence of a short circuit. There is no delay in processing to limit the gate voltage Vgs (SW). Therefore, even if a short circuit occurs, the occurrence of overcurrent can be appropriately suppressed.

On the other hand, when a short circuit does not occur, the voltage Vds becomes equal to or less than the threshold value as the voltage on the drain side of the switch SW decreases. As a result, the clamp control switch T3 is turned off, the Zener diode ZD is cut off, and the limitation of the gate voltage Vgs (SW) by the Zener diode ZD is released. Therefore, it is possible to prevent the gate voltage Vgs (SW) of the switch SW from being unnecessarily limited and reduce the loss. That is, the gate voltage Vgs (SW) can be quickly raised to Vcc, and a sufficient current can flow.

Further, a Zener diode ZD, which is a TVS for absorbing surge, is used as the clamp portion cl. Since the current rating is high, the clamp operation can be performed independently, and the number of parts can be suppressed.

(Second Embodiment)
A part of the configuration of the first embodiment may be modified as described below. The same configuration as that of the first embodiment is designated by the same reference numerals and the description thereof is omitted.

As shown in FIG. 5, in the drive circuit Dr of the second embodiment, the gate of the switch SW is connected to the negative power supply VEE (<0V) via the discharge resistor R2 and the discharge switch T2.

Further, a diode D3 is provided between the gate of the switch SW and the clamp control switch T3. The anode of the diode D3 is connected to the gate side of the switch SW, and the cathode is connected to the drain side (high potential side terminal side) of the clamp control switch T3. That is, the diode D3 corresponds to a second diode that regulates the flow of current (indicated by a two-dot chain line in FIG. 5) from the source side of the switch SW to the gate side of the switch SW.

The configuration of the second embodiment described above has the following effects.

According to the second embodiment, when the discharge switch T2 is turned on by connecting the gate of the switch SW to the negative power supply Vee via the discharge resistor R2 and the discharge switch T2, the gate voltage Vgs of the switch SW ( SW) can be lowered quickly.

Further, by providing the diode D3 between the gate of the switch SW and the clamp control switch T3, even if the potential on the source side of the switch SW is higher than the potential of the negative power supply Vee, the switch is switched from the source side of the switch SW. It is possible to prevent the current (indicated by a two-point chain line) from wrapping around the gate side of the SW.

(Third Embodiment)
A part of the configuration of the second embodiment may be modified as described below. The same configuration as that of the second embodiment is designated by the same reference numerals and the description thereof is omitted.

As shown in FIG. 6, in the drive circuit Dr of the third embodiment, a resistance R5 as a resistance element is provided on an electric path connecting the gate side and the source side of the clamp control switch T3. More specifically, one end is connected to a connection point M31 on the electrical path between the source of the clamp control switch T3 and the cathode of the Zener diode ZD, on the electrical path between the gate of the clamp control switch T3 and the resistor R3. A resistor R5 having the other end connected to the connection point M32 is provided.

The configuration of the second embodiment described above has the following effects.

According to the configuration of the third embodiment, the resistance R5 from the constant voltage power supply 40 before the clamp control switch T3 is switched from off to on, that is, before the gate of the switch SW and the Zener diode ZD are energized. The current will be supplied via. Therefore, the parasitic capacitance (not shown) existing in parallel with the Zener diode ZD can be precharged in advance before the clamp control switch T3 is turned on. Therefore, it is possible to prevent the waveform of the gate voltage Vgs (SW) of the switch SW from being disturbed by the parasitic capacitance when the gate voltage Vgs (SW) of the switch SW is charged.

Further, the diode D3 can prevent the current due to precharging from wrapping around the gate of the switch SW.

In the third embodiment, the Zener diode ZD as the clamp portion cl may have a circuit configuration using the transistor TR8, the Zener diode ZD8, and the resistor R10, as shown in FIG. 7. Explaining this circuit configuration in detail, the transistor TR8 is a PNP type bipolar transistor, the emitter of the transistor TR8 is connected to the connection point M31, that is, the source of the clamp control switch T3, and the collector is the source of the switch SW. It is connected to the. Further, the base of the transistor TR8 is connected to the cathode of the Zener diode ZD8 and is also connected to the emitter (that is, the connection point M31) via the resistor R10. Further, the anode of the Zener diode ZD8 is connected to the source of the switch SW.

In the circuit configuration shown in FIG. 7, when the current I flowing into the clamp circuit 70 exceeds VF / R10, the transistor TR8 conducts (turns on). “VF” is the forward voltage between the emitter and the base, and “R10” is the resistance value of the resistor R10. When the transistor TR8 conducts, the clamp portion cl clamps so that the gate voltage of the switch SW becomes the voltage of Vz + VF. "Vz" is the yield voltage of the Zener diode ZD8.

In the third embodiment, when the clamp portion cl has the circuit configuration shown in FIG. 7, the parasitic capacitance C2 existing in parallel with the Zener diode ZD8 is connected to the parasitic capacitance C2 in parallel with the Zener diode ZD8 via the resistor R5 before the clamp control switch T3 is turned on. A current (indicated by the dashed line) is supplied. As a result, it is possible to prevent the transistor TR8 from being erroneously turned on due to the current flowing into the parasitic capacitance C2 and being clamped at a voltage lower than a predetermined clamping voltage.

(Fourth Embodiment)
A part of the configuration of the third embodiment may be modified as described below. The same configuration as that of the third embodiment is designated by the same reference numerals and the description thereof is omitted.

As shown in FIG. 8, in the drive circuit Dr of the fourth embodiment, the connection point M1 between the resistor R4 and the capacitor C1 is connected to the anode side of the diode D1 via the diode D4. More specifically, the anode of the diode D4 is connected to the connection point M1 and the cathode is connected to the connection point M41 on the electrical path between the connection point M2 and the anode of the diode D1. That is, the diode D2 is omitted, and the short-circuit detection circuit 60 and the clamp control switch T3 are configured to share a high withstand voltage diode. Although the diode D2 is omitted in FIG. 8, the diode D1 may be omitted and the diode D2 may be shared.

Further, in the electric path on the anode side of the diode D1, the diode D4 is provided between the connection point M1 and the connection point M41. The cathode side of the diode D4 is connected to the connection point M41, and the anode side is connected to the connection point M1. As a result, as shown by the two-dot chain line in FIG. 8, it is possible to prevent the current from wrapping around from the constant voltage power supply 40 via the resistance R3 and the connection point M41.

The effect of the fourth embodiment will be described.

Since a high voltage is generally applied to the drain side (high potential side terminal side) of the switch SW, a high withstand voltage diode is required when connecting a circuit to the electric path on the drain side. Therefore, with the above configuration, it is not necessary to provide a high withstand voltage diode for each of the shunt detection circuit 60 and the clamp control switch T3, and the number of high withstand voltage diodes can be reduced.

Although a part of the configuration of the third embodiment is changed in the fourth embodiment, the same change may be made in the first embodiment and the second embodiment.

(Fifth Embodiment)
A part of the configuration of the third embodiment may be modified as described below. The same configuration as that of the third embodiment is designated by the same reference numerals and the description thereof is omitted.

As shown in FIG. 9, in the drive circuit Dr of the fifth embodiment, the transistor T6 in which the collector is connected to the constant voltage power supply 40 and the emitter is connected to the charging resistor R1 is provided. The base of the transistor T6 is connected to the source side of the charging switch T1 via the resistor R8. Therefore, when the charging switch T1 is turned on, the transistor T6 is also turned on, and the power supply voltage of the constant voltage power supply 40 is applied to the gate of the switch SW via the charging resistor R1.

Further, the anode side of the diode D5 is connected to the connection point M51 between the resistor R8 and the base of the transistor T6. The cathode side of the diode D5 is connected to the collector of the transistor T7. The emitter of the transistor T7 is connected to the source side of the switch SW.

Further, a series connection body of the detection resistor R7 and the detection resistor R6 is provided on the electric path connecting the anode side of the Zener diode ZD as the clamp portion cl and the source side of the switch SW. The detection resistor R7 and the detection resistor R6 are connected in series to the Zener diode ZD in this order, and the connection point M52 between the detection resistor R7 and the Zener diode ZD is connected to the inverting input terminal (-input terminal) of the comparator CMP. It is connected.

Further, the connection point M53 between the detection resistor R6 and the detection resistor R7 is connected to the base of the transistor T7. When the base voltage of the transistor T7 exceeds the threshold voltage Vth (T7), the transistor T7 is configured to be turned on.

The operation of the drive circuit Dr configured in this way will be described.

Since the clamp control switch T3 is not turned off at the time of a short circuit, the gate voltage Vgs (SW) of the switch SW is clamped by the Zener diode ZD. At that time, the current I supplied from the constant voltage power supply 40 via the charging resistor R1 flows into the detection resistors R7 and R6 via the clamp circuit 70. When the current I exceeds the current value Is determined by Vth (T7) / R6, the transistor T7 is turned on. In addition, "R6" is the resistance value of the detection resistor R6.

When the transistor T7 is turned on, the base voltage of the transistor T6 is suppressed so that the current I flowing through the charge resistor R1 (current I flowing into the clamp circuit 70) is maintained at the current value Is.

When the current value of the current I flowing through the charging resistor R1 is maintained at the current value Is at the time of a short circuit, the voltage of the connection point M52 between the detection resistor R7 and the Zener diode ZD becomes Vth (T7) + Is.・ It will be maintained at R7. In addition, "R7" is the resistance value of the detection resistor R7. The comparator CMP detects a short circuit by monitoring whether or not the voltage input from the connection point M52 reaches Vth (T7) + Is · R7.

It should be noted that, in the fifth embodiment, it will be described that the threshold value when the gate voltage Vgs (T3) of the clamp control switch T3 is discharged via the diode D1 differs depending on the provision of the detection resistors R6 and R7. Not to mention. However, each value is set so that the threshold value is smaller than Vgmax.

Further, in the fifth embodiment, when the gate voltage of the switch SW is clamped by the clamp portion cl, the current I flowing through the clamp portion cl corresponds to the clamp current, and the detection resistors R6 and R7 correspond to the clamp current detection portion. do. Further, the comparator CMP corresponds to the short circuit detection unit. Further, since the operation in the normal state is almost the same except that the threshold value is different, the description thereof will be omitted.

The effect of the configuration of the fifth embodiment described above will be described.

In the above configuration, a detection resistor R6 and a detection resistor R7 connected to the Zener diode ZD and through which the current I flows when the gate voltage Vgs (SW) of the switch SW is limited by the Zener diode ZD are provided. Further, the comparator CMP detects a short circuit based on the current I flowing through the detection resistor R6 and the detection resistor R7. Specifically, the comparator CMP detects a short circuit by monitoring whether or not the voltage input from the connection point M52 reaches Vth (T7) + Is · R7. Thereby, even if the short circuit detection circuit 60 is not provided, the short circuit can be detected by using the clamp circuit 70.

Further, since the current I does not flow when the clamp is released, it does not affect the gate voltage Vgs (SW) of the switch SW in the normal state (when not short-circuited). That is, it is possible to prevent the switch SW from being affected by opening / closing drive.

In the fifth embodiment, the Zener diode ZD serving as the clamp portion cl may be changed to a circuit composed of the transistor TR8, the Zener diode ZD8, and the resistor R10 as shown in FIG. The configuration of the clamp portion cl in FIG. 10 is the same as the configuration of the clamp portion cl described with reference to FIG. 7.

When the clamp portion cl is changed to the circuit shown in FIG. 10, the collector of the transistor TR8 is connected to the source of the switch SW via the detection resistor R7 and the detection resistor R6. In this case, it is possible to detect a short circuit as in the fifth embodiment. Further, unlike the circuit configuration shown in FIG. 9, the Zener diode ZD8 that determines the clamp voltage and the detection resistors R7 and R6 are not connected in series. Therefore, it is possible to suppress the fluctuation of the clamp voltage due to the current I flowing through the detection resistors R7 and R6.

(Modification example)
A modification of the above embodiment is shown below.

-In each of the above embodiments, a configuration other than the Zener diode ZD (surge absorption TVS, power Zener) may be adopted as the clamp portion cl. For example, as shown in FIG. 11A, low power (small current) Zener diodes ZD1 to ZD4 connected in parallel may be adopted as the clamp portion cl. At that time, it is desirable to share the clamp current so as not to exceed the rating of each Zener diode ZD1 to ZD4. This makes it possible to improve the clamping accuracy.

Further, for example, as shown in FIG. 11B, the clamp portion cl may have a circuit configuration using a transistor TR8, a Zener diode ZD8, and a resistor R10. Since this circuit configuration is the same as that in FIG. 7, the description thereof will be omitted. With such a circuit configuration, it is possible to obtain good clamp voltage accuracy while suppressing the number of parts as compared with the circuit configuration shown in FIG. 11A.

-In the fifth embodiment, the detection resistor R7 may not be provided. Further, if it is not necessary to maintain (clamp) the current at the current value Is, the detection resistor R6 or the like may not be provided. That is, the clamp current detection unit may be only one of the detection resistor R6 and the detection resistor R7.

-In the fifth embodiment, as shown in FIG. 12, the circuit configuration may be changed. As shown in FIG. 12, a detection resistor R17 is provided on the electric path connecting the anode side of the Zener diode ZD as the clamp portion cl and the source side of the switch SW. The connection point M52 between the detection resistor R17 and the Zener diode ZD is connected to the inverting input terminal (−input terminal) of the comparator CMP1 and the comparator CMP2.

When the voltage of the connection point M52 reaches the determination value Vt1, the comparator CMP1 outputs a detection signal notifying that a short circuit has been detected to the drive controller 50. The output terminal of the comparator CMP2 is connected to the base of the transistor T7, and is configured to turn on the transistor T7 when the voltage of the connection point M52 exceeds the determination value Vt2 (> Vt1).

The operation of the drive circuit Dr configured in this way will be described with reference to FIG. Since the clamp control switch T3 is not turned off at the time of a short circuit, the gate voltage Vgs (SW) of the switch SW is clamped by the Zener diode ZD (time point t100). At that time, the current I supplied from the constant voltage power supply 40 via the charging resistor R1 flows into the detection resistor R17 via the clamp circuit 70. When the current I exceeds the current value determined by Vt1 / R17 (time point t101), the comparator CMP1 outputs a detection signal notifying that a short circuit has been detected to the drive controller 50. In addition, "R17" is the resistance value of the detection resistor R17. That is, when the voltage of the connection point M52 exceeds the determination value Vt1, the comparator CMP1 outputs a detection signal notifying that a short circuit has been detected.

After that, when the current I exceeds the current value determined by Vt2 / R17 (time point t102), the comparator CMP2 turns on the transistor T7. That is, the comparator CMP2 turns on the transistor T7 when the voltage of the connection point M52 exceeds the determination value Vt2. When the transistor T7 is turned on, the base voltage of the transistor T6 is suppressed, and the current I flowing through the charging resistor R1 (current I flowing into the clamp circuit 70) is maintained at a predetermined value. That is, the voltage at the connection point M52 is maintained constant (specifically, the determination value Vt2).
In this modification, the detection resistor R17 corresponds to the clamp current detection unit. Further, the comparator CMP1 corresponds to the short circuit detection unit. Thereby, the same effect as that of the fifth embodiment can be obtained.

40 ... constant voltage power supply, D1 ... diode, Dr ... drive circuit, SW ... switch, T3 ... clamp control switch, cl ... clamp part, ZD ... Zener diode.

Claims (6)

In the drive circuit (Dr) of the switching element that opens and closes the switching element by controlling the gate voltage of the switching element (SW).
A clamp portion (cl) that limits the gate voltage of the switching element to a predetermined clamp voltage or less,
A clamp control switch (T3) that switches between an energized state and an energized cutoff state between the gate of the switching element and the clamp portion.
A first diode (D1), in which the anode is connected to the gate of the clamp control switch and the cathode is connected to the high potential side terminal of the switching element, is provided.
The clamp control switch is
When the voltage at the high potential side terminal of the switching element is larger than the threshold value, a predetermined power supply voltage is applied from the power supply (40) to the gate of the clamp control switch, and the gate of the switching element and the clamp portion While switching to an energized state, the gate voltage of the switching element is limited to the clamp voltage or less by the clamp portion.
When the voltage at the high potential side terminal of the switching element is equal to or lower than the threshold value, the gate voltage of the clamp control switch is discharged via the first diode, and the gate voltage of the switching element is between the gate of the switching element and the clamp portion. The drive circuit of the switching element that switches to the energization cutoff state. A second diode (D3) connected in series to the clamp portion is provided.
One end of the clamp portion is connected to the low potential side terminal of the switching element.
The drive circuit for a switching element according to claim 1, wherein the second diode regulates the flow of a current from the low potential side terminal side of the switching element to the gate side of the switching element. The high potential side terminal of the clamp control switch is connected to the gate of the switching element via the second diode.
The low-potential side terminal of the clamp control switch is connected to the low-potential side terminal of the switching element via the clamp portion.
The second aspect of claim 2, wherein one end of the clamp portion is connected to the low potential side terminal of the clamp control switch and is connected to the gate or the power supply of the clamp control switch via a resistance element (R5). Driving circuit of the switching element. A short circuit detection circuit (60) for detecting a short circuit based on the voltage at the high potential side terminal of the switching element is provided.
The drive circuit for a switching element according to any one of claims 1 to 3, wherein the short-circuit detection circuit is connected to the high potential side terminal of the switching element via the first diode. A clamp current detection unit connected to the clamp unit and through which a clamp current flows when the gate voltage of the switching element is limited by the clamp unit.
The drive circuit for a switching element according to any one of claims 1 to 4, further comprising a short circuit detection unit that detects a short circuit based on a clamp current flowing through the clamp current detection unit. The clamp portion has a transistor (TR8) in which an emitter is connected to the gate side of the switching element via the clamp control switch, a cathode is connected to the base of the transistor, and an anode is a low potential side terminal of the switching element. With a Zener diode (ZD8) connected to,
The drive circuit for a switching element according to claim 5, wherein the clamp current detection unit is a detection resistor having one end connected to the collector of the transistor and the other end connected to the low potential side terminal of the switching.

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