JP2019122173A - Power converter - Google Patents

Abstract

To suppress the switching loss.SOLUTION: The power converter 10 according to the present invention includes: a switching element 101 and a switching element 103 serially connected to each other; a switching element 102 and a switching element 104 serially connected to each other; a controller 120 for outputting the conduction ratio for each PWM cycle; and a PWM signal generator 130 for generating a PWM signal controlling the switch elements 101 to 104 on the basis of the output conduction ratio. The controller 120 calculates the conduction ratio D(n-1) in the first PWM cycle and the conduction ratio D(n) in the second PWM cycle on the basis of the input voltage and the output voltage instruction, and outputs the conduction ratio D'(n-1) at which the switching elements 101, 102 are constantly in the on-state and the switching elements 103 and 104 are constantly in the off-state in the frist PWM cycle and outputs the conduction ratio D'(n) according to the conduction ratios D(n-1) and D(n) in the second PWM cycle.SELECTED DRAWING: Figure 1

Description

  The present invention relates to a power converter which converts direct current voltage into equal or smaller direct current voltage.

  In a railway vehicle (electric vehicle) that operates by taking in DC power from an overhead wire, a DC-DC power converter (hereinafter referred to as a DC-DC power converter) that stabilizes input power to an auxiliary power supply device as a countermeasure against sudden changes in overhead wire voltage , Referred to as a power converter). Non-Patent Document 1 describes a serial-parallel continuously regulated chopper (SPCH) as such a power converter.

  FIG. 6 is a circuit diagram showing a configuration example of the power converter 10a which is the series-parallel continuous switching chopper described above.

  Power converter 10a shown in FIG. 6 includes switching elements 101 to 104, capacitors 105 and 107, loads 106 and 108, inductors 109 and 110, control unit 120a, and PWM (Pulse Width Modulation) signal generation unit 130. And

  Each of the switching elements 101 to 104 is configured by reverse parallel connection of a switching element such as an IGBT (Insulated Gate Bipolar Transistor) that can be turned on and off, and a diode.

  The switching element 101 (first switching element) and the switching element 103 (third switching element) are connected in series to constitute a leg 113 (first leg). The switching element 102 (second switching element) and the switching element 104 (fourth switching element) are connected in series to constitute a leg 114 (second leg).

  A capacitor 105 (first capacitor) and a load 106 (first load) are connected in parallel to the leg 113. Also, a capacitor 107 (second capacitor) and a load 108 (second load) are connected in parallel to the leg 114. Each of the loads 106 and 108 is, for example, a resistive load in which a capacitor and a resistor are connected in parallel, an inverter or the like, but is not limited thereto.

  An inductor 109 (first inductor) is connected between the end of the switching element 103 not connected to the switching element 101 and the connection point between the switching element 102 and the switching element 104. In addition, an inductor 110 (second inductor) is connected between a connection point of the switching element 101 and the switching element 103 and an end of the switching element 104 not connected to the switching element 102.

  A DC power supply 20 is connected between the end of the switching element 101 not connected to the switching element 103 and the end of the switching element 102 not connected to the switching element 104. In addition, DC power supply 20 is corresponded to an overhead wire, when power converter 10a is mounted in an electric vehicle.

In the following, the input voltage (power supply voltage) from the DC power supply 20 is V _in , the current flowing _in the inductor 109 is I _L1 , the current flowing in the inductor 110 is I _L2 , the voltage of the capacitor 105 is V _C1 , and the voltage of the capacitor 107 is V _C2 , output load power to load 106 P _out1 , output voltage to load 106 V _out1 , output voltage command indicating output voltage V _out1 V _{out1_ref} , output load current flowing to load 106 I _out1 , load 108 The output load power to the load 108 is defined as P _out2 , the output voltage to the load 108 as V _out2 , the output voltage command designating the output voltage V _out2 as V _{out2 — ref} , and the output load current flowing through the load 108 as I _out2 . Further, hereinafter, the capacitance of the capacitor 105 is defined as C ₁ , the capacitance of the capacitor 107 as C ₂ , the inductance of the inductor 109 as L ₁ , and the inductance of the inductor 110 as L ₂ .

L ₁ = L ₂ = L, C ₁ = C ₂ = C, I _L1 = I _L2 = I _L , V _C1 = V _C2 = V _C , I _out1 = I _out2 = I _out , V _out1 = V _out2 = V _{out, V out1_ref = V out2_ref =} V out_ref, by setting the _{P out1 = P out2 = P out} , the control unit 120a, the following values _D 1, which is determined by equation (1), _{D 2,} respectively, switching elements 101 , 102 as a duty command value for controlling the duty ratio (conduction ratio D = D ₁ = D ₂ ), and outputs it to the PWM signal generation unit 130. PWM signal generating unit 130 outputs a triangular wave, a PWM signal _S 1 to S ₄ of the pulse width corresponding to the magnitude relation between the amplitude and the duty ratio of the PWM carrier, such as a sawtooth wave to the switching element 101 to 104.

JP, 2017-127142, A

  In order to improve the characteristics and economy of the power converter, it is effective to improve the carrier frequency of the PWM carrier. On the other hand, in a power device such as an IGBT, switching loss occurs in a transition state of switch operation. Switching losses increase in proportion to the operating frequency. Therefore, suppression of switching loss is an important issue for power conversion.

  An object of the present invention made in view of the above problems is to provide a power converter capable of suppressing switching loss.

  In order to solve the above-mentioned subject, the power converter concerning the present invention is the 1st leg which connected the 1st switching element and the 3rd switching element in series, and the 1st connected in parallel with the 1st leg. And a second leg in which a second switching element and a fourth switching element are connected in series, a second capacitor connected in parallel to the second leg, and a second load A first switching element connected between the end of the third switching element not connected to the first switching element and the connection point between the second switching element and the fourth switching element A contact is made between an inductor, a connection point between the first switching element and the third switching element, and an end of the fourth switching element not connected to the second switching element. An end of the first switching element not connected to the third switching element, and an end not connected to the fourth switching element of the second switching element And a control unit for outputting a conduction ratio at each PWM cycle, and the first to fourth power conversion devices based on the conduction ratio output from the control unit. A PWM signal generation unit for generating a PWM signal for controlling the switching elements of the first and second control units, the control unit further comprising: an input voltage from the DC power supply; and an output voltage to the first load and the second load. Calculating a first conduction rate in a first PWM cycle and a second conduction rate in a second PWM cycle subsequent to the first PWM cycle, based on the output voltage command instructing the In the PWM cycle Outputting a third conduction ratio to the PWM signal generator, wherein the first and second switching elements are always on and the third and fourth switching elements are always off, and the second PWM In the cycle, a fourth conduction rate corresponding to the first conduction rate and the second conduction rate is output to the PWM signal generation unit.

  Further, in the power converter according to the present invention, the control unit is configured to calculate the time integral value of the output voltage according to the fourth current conduction rate according to the time of the output voltage according to the first current conduction rate Preferably, the fourth conduction ratio is determined to be equal to the sum of the integral value and the time integration value of the output voltage according to the second conduction ratio.

  Further, in the power converter according to the present invention, the first conduction ratio is output to the PWM signal generation unit in the first PWM cycle, and the second conduction ratio is output in the second PWM cycle. To perform the first control, or to the PWM signal generation unit to output the third conduction ratio in the first PWM cycle, and to output the fourth conduction in the second PWM cycle. It is preferable to further include a switching control unit that switches whether to output the ratio or to perform the second control.

  Further, in the power converter according to the present invention, the switching control unit may be configured such that current ripples generated in the inductor when the fourth conduction rate performs the second control perform the first control. Iron core loss generated in the inductor in the case of performing the second control, and the inductor in the case of performing the first control in a range smaller than the current ripple generated in the inductor in the case of performing From the difference between the switching loss generated in the switching element when the first control is performed and the switching loss generated in the switching element when the second control is performed. Is also smaller, it is preferable to determine to perform the second control, and otherwise to determine to perform the first control

  Further, in the power converter according to the present invention, preferably, the iron core loss is calculated based on a formula (6) described later.

  According to the power converter of the present invention, the switching loss can be suppressed.

It is a figure showing an example of composition of a power converter concerning a 1st embodiment of the present invention. It is a figure for demonstrating the production | generation of the PWM signal in the conventional power converter. It is a figure for demonstrating the production | generation of the PWM signal in the power converter shown in FIG. It is a figure which shows the structural example of the power converter which concerns on the 2nd Embodiment of this invention. It is a flowchart which shows operation | movement of the power converter shown in FIG. It is a figure which shows the structural example of the conventional power converter.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings.

First Embodiment
FIG. 1 is a view showing a configuration example of a power converter 10 according to a first embodiment of the present invention. The power converter 10 according to the present embodiment is different from the power converter shown in FIG. 6 in that the control unit 120a is changed to a control unit 120. That is, the power converter 10 according to the present embodiment includes the switching elements 101 to 104, the capacitors 105 and 107, the loads 106 and 108, the inductors 109 and 110, the control unit 120, and the PWM signal generation unit 130. Prepare. The connection relationship between switching elements 101 to 104, capacitors 105 and 107, loads 106 and 108, and inductors 109 and 110 is the same as that of power converter 10a shown in FIG.

  The control unit 120 outputs the conduction ratio to the PWM signal generation unit 130 for each PWM cycle in PWM control according to the magnitude relationship between the amplitude of the PWM carrier and the conduction ratio.

The PWM signal generation unit 130 generates PWM signals S _{1 to} S ₄ based on the conduction ratio output from the control unit 120, and outputs the PWM signals S _{1 to} S ₄ to the switching elements 101 to 104, respectively.

  Next, the conduction ratio output by the control unit 120 will be described.

The control unit 120 calculates the conduction rate D based on the above equation (1) for each PWM cycle. More specifically, the control unit 120, the input voltage _{V in} from the DC power source 20, based on the output voltage command _{V Out_ref} instructing output voltage _{V out,} the first n-1 primary PWM cycle (n an even number ) (The first PWM cycle) (hereinafter referred to as “the conduction rate D (n−1)”), and the nth PWM cycle (the second cycle following the n−1th PWM cycle) The conduction rate D (hereinafter referred to as the conduction rate D (n)) in the PWM cycle of 1) is calculated.

  Next, in the nth-1st PWM cycle (odd-order PWM cycle), the control unit 120 continuously changes the switching ratio D (the switching elements 101 and 102 are always on and the switching elements 103 and 104 are always off). Hereinafter, the conduction ratio D ′ (n−1) is output to the PWM signal generation unit 130. That is, the control unit 120 outputs, to the PWM signal generation unit 130, a conduction ratio D '(n-1) represented by the following equation (2) in the (n-1) th PWM cycle.

  Further, in the nth PWM cycle (even-order PWM cycle), control unit 120 determines a conduction rate D (hereinafter referred to as “D”) corresponding to conduction rate D (n−1) and conduction rate D (n). The conduction ratio D ′ (n) is output to the PWM signal generator 130. Specifically, the control unit 120 outputs, to the PWM signal generation unit 130, a conduction ratio D '(n) represented by the following equation (3) in the nth PWM cycle.

That is, control unit 120 outputs input voltage V _in from DC power supply 20, and output voltage command V out — _ref instructing output voltage V _out to load 106 (first load) and load 108 (second load). , The conduction rate D (n-1) (first conduction rate) in the n-1st PWM cycle (first PWM cycle), and the 1st PWM cycle following the n-1th PWM cycle The conduction rate D (n) (second conduction rate) in the n-th PWM cycle (second PWM cycle) is calculated. Then, in the control unit 120, in the (n−1) th PWM cycle, the switching elements 101 and 102 are always on, and the switching elements 103 and 104 are always off. A conduction ratio D ′ (n−1) (third Current ratio) to the PWM signal generation unit 130, and in the nth PWM cycle, the current ratio D ′ (n) according to the current ratio D (n−1) and the current ratio D (n). n) (the fourth conduction rate) is output to the PWM signal generator 130.

FIG. 2 is a diagram for explaining generation of PWM signals S _{1 to} S _{4 in} the conventional power converter 10 a shown in FIG. 6. FIG. 2 shows an example in which the PWM carrier is a triangular wave whose amplitude changes in the range of 0 to 1.

The PWM signal generation unit 130 compares the conduction ratio D (n−1) output from the control unit 120 a with the amplitude of the PWM carrier in the n−1th period (n−1 th order PWM period), duty ratio D (n-1) is the case where more than PWM carrier the PWM signal _S 1, _{S 2} and High level, the PWM signal _S 3, _{S 4} Low level. Further, when the conduction ratio D (n−1) is smaller than the PWM carrier, the PWM signal generation unit 130 sets the PWM signals S ₁ and S ₂ to Low level, and sets the PWM signals S ₃ and S ₄ to High level. Do. Similarly, also in the nth period (nth PWM period) following the n-1th period, the PWM signal generation unit 130 outputs the conduction ratio D (n) output from the control unit 120a and the PWM carrier. depending on the magnitude relationship, it changes the logic level of the PWM signal _S 1 to S _4. Thus, in the conventional power converter 10a, switching of the switching elements 101 to 104 is performed for each PWM cycle.

FIG. 3 is a diagram for describing generation of PWM signals S _{1 to} S ₄ in the power converter 10 according to the present embodiment. FIG. 3 shows an example in which the PWM carrier is a triangular wave whose amplitude changes in the range of 0 to 1.

In the present embodiment, the control unit 120 sets the conduction ratio D ′ (n−1) = 1 in the n−1 th cycle (n−1 th order PWM cycle). Since the conduction rate D '(n-1) = 1, as shown in FIG. 3, the conduction rate D' (n-1) is always equal to or higher than the PWM carrier in the (n-1) th cycle. Therefore, the PWM signal generation unit 130 keeps the PWM signals S ₁ and S ₂ at the High level and keeps the PWM signals S ₃ and S ₄ at the Low level in the n−1th cycle. Therefore, in the (n−1) th cycle (the n−1 th order PWM cycle), the switching elements 101 and 102 are always on, and the switching elements 103 and 104 are always off.

In the nth period (nth PWM period) subsequent to the n-1th period, the PWM signal generation unit 130 compares the magnitude of the conduction ratio D ′ (n) output from the control unit 120 with the PWM carrier. Correspondingly, changes the logic level of the PWM signal _S 1 to S _4.

Therefore, in the power converter 10 according to the present embodiment, switching of the switching elements 101 to 104 is not performed in the n-1st period (the n-1th PWM period). Therefore, the power converter 10 according to the present embodiment can be compared with the power converter 10a shown in FIG. 6 in terms of a period in which the n-1st PWM period and the nth PWM period are combined. The switching of the switching elements 101 to 104 is halved. Therefore, the switching loss due to the switching of the switching elements 101 to 104 can be suppressed. It is noted that control unit 120 sets the time integral value of output voltage V _out according to conduction ratio D ′ (n) in the n th PWM period to the conduction ratio D (n th a time integration value of the output voltage V _out corresponding to -1), to be equal to the sum of the time integral value of the output voltage V _out corresponding to the duty ratio in the n-th PWM cycle D (n), Determine the conduction rate D '(n). Therefore, the switching loss can be suppressed while following the output voltage command V out — _ref .

As described above, in the present embodiment, the control unit 120 determines _whether or not the (n−1) th PWM cycle (the first PWM cycle) is based on the input voltage V _in from the DC power supply 20 and the output voltage command V out — _ref . The conduction ratio D (n-1) (first conduction ratio) and the conduction ratio D (n) in the nth PWM cycle (second PWM cycle) following the n-1st PWM cycle The second PWM cycle is calculated, and in the n-1st PWM cycle, the switching elements 101 and 102 are always on, and the switching elements 103 and 104 are always off. A conduction ratio D '(n-1) (The third conduction rate) is output to the PWM signal generation unit 130, and in the nth PWM cycle, the conduction rate according to the conduction rate D (n-1) and the conduction rate D (n) D ′ (n) (fourth conduction ratio) is output to the PWM signal generator 130.

  Therefore, switching of the switching elements 101 to 104 is not performed in the (n−1) th PWM cycle, so that switching loss can be suppressed.

Second Embodiment
As described above, in the conventional power converter 10a, the conduction rate D (n-1) is output to the PWM signal generation unit 130 in the n-1st PWM cycle, and in the nth PWM cycle. Control (hereinafter, referred to as first control) for outputting the current conduction rate D (n) is performed. On the other hand, in the first embodiment, the conduction ratio D '(n-1) is output to the PWM signal generation unit 130 in the n-1st PWM cycle, and the conduction ratio is output in the nth PWM cycle. Control for outputting D ′ (n) (hereinafter, referred to as second control) is performed.

  Here, in the case of performing the second control, the current ripple of the inductors 109 and 110 in the case of performing the second control does not exceed the upper limit of the current ripple of the inductors 109 and 110 in the case of performing the first control. It is desirable to satisfy the condition (first condition) that the current ripple in the case of performing control 2 is smaller than the current ripple in the case of performing the first control. In addition, when performing the second control, the condition that the increase in iron core loss of the inductors 109 and 110 in the case of performing the second control is smaller than the reduction in switching loss in the case of performing the second control It is desirable to satisfy the condition 2).

  Therefore, in the second embodiment of the present invention, whether to perform the first control or to perform the second control is switched depending on whether the above-described condition is satisfied.

  FIG. 4 is a view showing a configuration example of a power converter 10A according to a second embodiment of the present invention. In addition, although description is abbreviate | omitted about the direct-current power supply 20, the switching elements 101-104, the capacitors 105 and 107, the loads 106 and 108, and the inductors 109 and 110 in FIG. Are similar to the power converter 10 shown in FIG.

  Power converter 10A shown in FIG. 4 is different from power converter 10 shown in FIG. 1 in that control unit 120 is changed to control unit 120A, switch 121, switching control unit 122, iron core loss calculation unit 123. And a switching loss calculation unit 124 is added.

  The control unit 120A switches the conduction rate D (n-1) in the (n-1) th PWM cycle and the conduction rate D (n) in the nth PWM cycle, which are calculated according to equation (1). Output to 121. The control unit 120A further calculates the conduction rate D ′ (n−1) of the (n−1) th PWM cycle and the conduction rate of the nth PWM cycle, which are calculated based on the equations (2) and (3). D ′ (n) is output to the switch 121. Further, the control unit 120A outputs the calculated conduction ratio D '(n) to the switching control unit 122.

  The switching device 121 outputs the conduction rate according to the first control (the conduction rate D (n-1) and the conduction rate D (n)) output from the control unit 120A according to the control of the switching control unit 122. And one of the conduction rates (the conduction rate D ′ (n−1) and the conduction rate D ′ (n)) corresponding to the second control are output to the PWM signal generation unit 130.

  The switching control unit 122 controls the conduction rate D ′ (n) output from the control unit 120A, the iron core loss calculated by the iron core loss calculating unit 123 described later, and the switching calculated by the switching loss calculating unit 124 described later. Based on the loss, it is determined whether the first and second conditions described above are satisfied. The switching control unit 122 determines that the second control is to be performed when the first condition and the second condition are satisfied. Then, the switching control unit 122 controls the switching device 121 such that the conduction ratio according to the second control is input to the PWM signal generation unit 130. Further, in other cases, the switching control unit 122 determines to perform the first control. Then, the switching control unit 122 controls the switching device 121 such that the conduction ratio according to the first control is input to the PWM signal generation unit 130.

  Specifically, the switching control unit 122 determines whether or not the current conduction rate D '(n) output from the control unit 120A satisfies the first condition described above.

Current ripple [Delta] I _L in the inductor 109 and 110 of the SPCH can be calculated based on the following equation (4). In equation (4), T is the carrier period of the PWM carrier.

  From the equation (4), the range of the conduction ratio D in which the first condition (the current ripple in the case of performing the second control is smaller than the current ripple in the case of performing the first control) is satisfied The flow rate operating range is expressed by the following equation (5).

In the equation (5), L ′ is an inductance in which the inductor current I _L increases at a constant power load when the conduction ratio D approaches 1 and decreases due to magnetic saturation.

  The switching control unit 122 determines that the first condition is satisfied when the conduction rate D ′ (n) calculated by the control unit 120 is within the conduction rate operation range, and in other cases, , It is determined that the first condition is not satisfied. The determination as to whether or not the second condition is satisfied by the switching control unit 122 will be described later.

The iron core loss calculation unit 123 calculates the iron core loss P _fe in the inductors 109 and 110 based on the following equation (6).

In equation (6), V _fe is the volume of the inductor core of inductors 109 and 110, K _c , α and β are constant parameters according to the core loss characteristics in the core material data sheet, and f is the carrier frequency , N _P is the number of turns of the inductors 109 and 110, and A _C is a cross-sectional area of the inductor core of the inductors 109 and 110.

In the following, assuming that the output voltage _Vout is constant in the conduction ratio operation range shown in the equation (5), the iron core loss of one inductor in the case of performing the first control is defined as P _fe (D, f) The iron core loss of one inductor in the case of performing the second control is defined as P _fe (D, f / 2). The iron core loss calculation unit 123 calculates the iron core loss P _fe (D, f) and the iron core loss P _fe (D, f / 2), and outputs the iron core loss P _fe (D, f / 2) to the switching control unit 122.

Switching loss calculation unit 124 uses the switching loss characteristics in the data sheet of the power device in consideration of the switching loss of switching elements 101 to 104 generated by inductor current I _L according to the current ripple corresponding to conduction ratio D. calculate. In the following, the switching loss per leg at the carrier frequency f in the case of performing the first control is defined as P _SW (I _L , D, f), and in the case of performing the second control, per switching. The switching loss is defined as P _SW (I _L , D, f / 2). The switching loss calculation unit 124 calculates the switching loss P _SW (I _L , D, f) and the switching loss P _SW (I _L , D, f / 2), and outputs this to the switching control unit 122. The calculation of the switching loss as described above is well known to those skilled in the art, and thus the detailed description is omitted.

The switching control unit 122 outputs the iron core loss P _fe (D, f) and the iron core loss P _fe (D, f / 2) output from the iron core loss calculation unit 123 and the switching loss P _SW ( _IL , D, f). And, based on the switching loss P _SW (I _L , D, f / 2), it is determined whether or not the second condition described above is satisfied.

The increase ΔP _fe (D, f / 2) of the core loss due to one inductor in the case of performing the second control is expressed by the following equation (7).

Further, the decrease ΔP _SW (I _L , D, f / 2) of the switching loss per one leg when the second control is performed is expressed by the following equation (8).

The switching control unit 122 is configured to calculate an increase ΔP _fe (D, f / 2) of iron core loss due to one inductor calculated based on Expression (7) and switching loss per one leg calculated based on Expression (8). The decrease amount ΔP _SW (I _L , D, f / 2) is compared to determine whether or not the following equation (9) is established.

  The switching control unit 122 determines that the second condition is satisfied when Expression (9) is satisfied. Then, the switching control unit 122 determines that the second control is to be performed when the first condition and the second condition are satisfied, and determines that the first control is to be performed in other cases. That is, in the switching control unit 122, the current ripple generated in the inductor when the conduction ratio D ′ (n) performs the second control is higher than the current ripple generated in the inductor when the first control is performed. The difference between the iron core loss generated in the inductor when the second control is performed and the iron core loss generated in the inductor when the first control is performed performs the first control. If it is smaller than the difference between the switching loss in the case and the switching loss in the case of performing the second control, it is determined to perform the second control, otherwise it is determined to perform the first control.

  Next, the operation of the power converter 10A according to the present embodiment will be described with reference to the flowchart shown in FIG.

  First, the switching control unit 122 determines whether or not the current conduction rate D '(n) calculated by the control unit 120A is within the current conduction rate operation range (step S11).

If the current conduction rate D '(n) is within the current conduction rate operating range (step S11: Yes), the iron core loss calculation unit 123 calculates the iron core loss P _fe (D, f) and the iron core based on equation (6). The loss P _fe (D, f / 2) is calculated (step S12). Further, the switching loss calculation unit 124 calculates the switching loss P _SW (I _L , D, f) and the switching loss P _SW (I _L , D, f / 2) (step S 13).

The switching control unit 122 includes iron core loss P _fe (D, f), iron core loss P _fe (D, f / 2), switching loss P _SW (I _L , D, f), and switching loss P _SW (I _L , D). , F / 2), the decrease in switching loss per leg ΔP _SW (I _L , D, f / 2) by the second control is the increase in iron core loss due to one inductor. It is determined whether it is larger than ΔP _fe (D, f / 2) (step S14).

If the conduction rate D '(n) is not within the conduction rate operating range (step S11: No) and the decrease in switching loss ΔP _SW (I _L , D, f / 2) increases the iron core loss When it is not larger than the minute ΔP _fe (D, f / 2) (step S14: No), the switching control unit 122 determines to perform the first control (step S15). Then, the switching control unit 122 controls the switching device 121, and the conduction rate (the conduction rate D (n-1) and the conduction rate D (n)) corresponding to the first control is a PWM signal generation unit. Make it output to 130.

When the decrease ΔP _SW (I _L , D, f / 2) of the switching loss is larger than the increase ΔP _fe (D, f / 2) of the core loss (step S 14: Yes), the switching control unit 122 It is determined that the second control is to be performed (step S16). Then, the switching control unit 122 controls the switching device 121, and the conduction rate (the conduction rate D '(n-1) and the conduction rate D' (n)) corresponding to the second control is a PWM signal. It is output to the generation unit 130.

  The PWM signal generation unit 130 outputs a conduction rate (a conduction rate according to the first control or a conduction rate according to the second control) output from the control unit 120 via the switch 121; A PWM signal is generated according to the magnitude relationship with the PWM carrier (step S17).

  As described above, in the present embodiment, the power converter 10A further includes the switching control unit 122 that switches whether to perform the first control or the second control. Therefore, the switching loss can be suppressed more reliably.

  Although the embodiments described above have been described as representative examples, it will be obvious to those skilled in the art that many modifications and substitutions are possible within the spirit and scope of the present invention. Accordingly, the present invention should not be construed as being limited by the above-described embodiments, and various modifications and changes are possible without departing from the scope of the claims. For example, it is possible to combine a plurality of configuration blocks described in the configuration diagram of the embodiment into one, or to divide one configuration block.

DESCRIPTION OF SYMBOLS 10 Power converter 20 DC power supply 101, 102, 103, 104 Switching element 105, 107 Capacitor 106, 108 Load 109, 110 Inductor 113, 114 Leg 120, 120A Control part 121 Switching device 122 Switching controller 123 Iron core loss calculation part 124 Switching loss calculation unit 130 PWM signal generation unit

Claims (5)

A first leg in which a first switching element and a third switching element are connected in series;
A first capacitor and a first load connected in parallel with the first leg;
A second leg in which a second switching element and a fourth switching element are connected in series;
A second capacitor and a second load connected in parallel with the second leg;
A first inductor connected between an end of the third switching device not connected to the first switching device, and a connection point between the second switching device and the fourth switching device; ,
A second inductor connected between a connection point between the first switching element and the third switching element and an end of the fourth switching element not connected to the second switching element; Equipped with
A DC power supply is connected between an end of the first switching element not connected to the third switching element and an end of the second switching element not connected to the fourth switching element A power converter,
A control unit that outputs a conduction rate at each PWM period;
And a PWM signal generation unit that generates a PWM signal for controlling the first to fourth switching elements based on the conduction ratio output from the control unit.
The control unit is configured to generate a first current in a first PWM cycle based on an input voltage from the DC power supply and an output voltage command instructing an output voltage to the first load and the second load. Rate and a second conduction rate in a second PWM cycle subsequent to the first PWM cycle, and in the first PWM cycle, the first and second switching elements are always on, and A third conduction ratio at which the third and fourth switching elements are always off is output to the PWM signal generation unit, and in the second PWM cycle, the first conduction ratio and the second conduction ratio are output. Outputting a fourth conduction ratio corresponding to the ratio to the PWM signal generation unit. In the power converter according to claim 1,
The control unit is configured such that a time integral value of the output voltage according to the fourth conduction rate is a time integral value of the output voltage according to the first conduction rate, and the second conduction rate And determining the fourth conduction ratio to be equal to the sum of the output voltage and the time integral value of the output voltage. The power converter according to claim 1 or 2
Is the first control of outputting the first conduction rate in the first PWM cycle and outputting the second conduction rate in the second PWM cycle to the PWM signal generation unit? Performing second control of outputting the third conduction ratio in the first PWM cycle and outputting the fourth conduction ratio in the second PWM cycle to the PWM signal generation unit; Power converter characterized by further comprising a switching control unit for switching between In the power converter according to claim 3,
The switching control unit
The fourth current conduction rate is within a range in which the current ripple generated in the inductor when performing the second control is smaller than the current ripple generated in the inductor when performing the first control. Yes, and
The switching in the case where the first control is performed when the difference between the iron core loss generated in the inductor in the case of performing the second control and the core loss generated in the inductor in the case of performing the first control If it is smaller than the difference between the switching loss generated in the element and the switching loss generated in the switching element when performing the second control, it is determined that the second control is performed; A power converter characterized in that it is determined to perform control of 1. In the power converter according to claim 4,
Let V _{fe be} the volume of the inductor core of the inductor, K _c , α, and β be constant parameters according to the core loss characteristics, f be the carrier frequency, V _out be the output voltage, and N _{p be} the number of turns of the inductor _Assuming that A _c is a cross-sectional area of the inductor core of the inductor and D is a flow rate, the core loss is calculated based on the following equation.

Images