JP2019115194A - Power converter control device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a power converter control device capable of suppressing a three-phase current imbalance generated by a DC offset current during a transient response to a high response. An inverter control device 501 controls the operation of an inverter 60 that converts DC power into three-phase AC power and supplies it to a motor 80. The basic voltage command value calculation unit 201 calculates the dq-axis voltage command values Vq * and Vd * based on the dq-axis current command values iq * and id *, and further performs three-phase conversion of the basic voltage command values Vub * and Vvb *. Is calculated. Compensated voltage command value calculation unit 401 is based on the three-phase current deviation Δiu, Δiv, which is the deviation between the three-phase current command values iu *, iv * and the three-phase current values iu, iv, and the compensated voltage command value Vuoffset *, Calculate Vvoffset *. The PWM signal generation unit 56 outputs a gate signal commanded to the inverter 60 based on the voltage command values Vu * and Vv * obtained by adding the basic voltage command values Vub * and Vvb * and the compensation voltage command values Vuoffset * and Vvoffset *. Generate. [Selection diagram] FIG. 7

Description

  The present invention relates to a power converter controller.

  BACKGROUND ART Conventionally, there are known techniques for reducing a direct current component included in a three-phase alternating current supplied from a power converter to a load. For example, according to Patent Document 1, when the three-phase AC voltage is unbalanced, a DC current flows through the load, causing the transformer or motor to generate a biased magnetism. In this case, a DC component is superimposed on the three-phase AC. It becomes a waveform. " Therefore, in the embodiment shown in FIG. 4 of Patent Document 1, each phase current is input to the integrator, and if the output of the integrator includes a DC component, each phase is removed so as to remove the DC component. The voltage command is biased and corrected.

  Further, the controller of the multiphase rotating machine disclosed in FIG. 1 and FIG. 2 of Patent Document 2 multiplies the current of each phase by the gain K corresponding to the virtual resistance of the stator to generate negative feedback for the voltage command of each phase. And controllably reduce the DC current component by increasing the apparent stator resistance. Furthermore, the controller for multi-phase rotation disclosed in FIG. 3 and FIG. 4 of Patent Document 2 detects a direct current component using a direct current component detection unit such as a low pass filter, and the direct current component output by the direct current component detection unit Only by the virtual resistance gain K.

JP-A-9-117154 JP 2003-88192 A

  In the prior art of Patent Document 1, the generation factor of the direct current component is due to unbalance of the power converter or the control unit. In the prior art of patent document 1, in order to extract a direct-current component, in order to integrate the electric current for 1 electrical-period period using an integrator, there exists a problem that the responsiveness at the time of a transient falls.

  In the prior art of Patent Document 2, the generation factor of the DC component is the variation of the on resistance of the switching element or the fluctuation of the inverter DC intermediate voltage. In the prior art of Patent Document 2, since a virtual resistor is used, the voltage applied to the load is lowered, and the voltage utilization rate is lowered. Further, in the configuration of FIG. 4 of Patent Document 2, in order to extract a direct current component, a filter is provided to process a current for one cycle of an electrical angle, and there is a problem that the responsiveness during transition is degraded.

  Furthermore, when an initial current flows in the inductance of the circuit in a transient response in which the output of the power converter changes, a DC offset current is generated, resulting in an unbalance of three-phase currents. However, neither of the patent documents 1 and 2 mentions the DC offset current.

  This invention is created in view of such a point, The objective is to provide the power converter control device which can suppress the three-phase current imbalance generated by DC offset current at the time of transient response to high response. It is.

  The present invention is a power converter control device that converts DC power into three-phase AC power and controls the operation of a power converter (60) that supplies a load (80) having resistance and inductance. The power converter control device includes a basic voltage command value calculation unit (20), a compensation voltage command value calculation unit (40), and a gate signal generation unit (56).

The basic voltage command value calculation unit calculates the dq axis voltage command values (Vq ^* , Vd ^* ) based on the dq axis current command values (iq ^* , id ^* ), and further converts the dq axis voltage command values into three phases. basic voltage command values _{^{(Vu b *, Vv b *}} ) for calculating a. The compensation voltage command value calculation unit is a three-phase current that is a deviation between a three-phase current command value (iu ^* , iv ^* ) obtained by three-phase conversion of the dq axis current command value and the three-phase current value (iu, iv) Based on the deviations (Δiu, Δiv), the compensation voltage command values (Vu _offset ^* , Vv _offset ^* ) are calculated. The gate signal generation unit generates a gate signal for instructing the power converter based on a voltage command value (Vu ^* , Vv ^* ) obtained by adding the basic voltage command value and the compensation voltage command value.

  In the present invention, by adding the compensation voltage command value to the basic voltage command value, it is possible to suppress the unbalance of the three-phase current generated by the DC offset current at the time of transient response of the power converter. At this time, the compensation voltage command value calculation unit can control the DC offset component directly and immediately based on the three-phase current deviation. Therefore, the responsiveness for reducing the DC component can be improved as compared with the prior art of Patent Documents 1 and 2 in which it is necessary to process the current for one cycle of the electrical angle.

  Preferably, the load according to the present invention is a motor, and the compensation voltage command value computing unit computes a basic voltage command value based on the three-phase current deviation and the resistance or inductance which is an equipment constant of the motor. For example, the compensation voltage command value computing unit computes a compensation voltage command value using a resistance value as a proportional gain of PD control that receives a three-phase current deviation as input and using an inductance value as a differential gain. Thus, it is possible to calculate the voltage command value for following the command value fastest.

  Preferably, the compensation voltage command value computing unit further computes a compensation voltage command value based on the three-phase current value. Specifically, if the three-phase current deviation is the same, the compensation voltage command value calculation unit increases the compensation voltage command value as the three-phase current value increases. That is, when the load is low and the three-phase current value is small, the influence of the current imbalance is small, so the compensation voltage command value is set small with emphasis on control stability. On the other hand, when the load is high and the three-phase current value is large, in order to suppress the current imbalance, the compensation voltage command value is set large with emphasis on high response.

BRIEF DESCRIPTION OF THE DRAWINGS The whole block diagram of the motor drive system to which the inverter control apparatus by each embodiment is applied. The figure explaining the three-phase current imbalance which generate | occur | produces at the time of torque instruction | command fluctuation | variation. The circuit model figure which extracted the single phase of a three-phase inverter. The current waveform figure at the time of the transient response (at the time of inverter output change) in the inverter control apparatus of a comparative example. The circuit model figure isolate | separated into alternating current (AC) component and direct current (DC) component. The current waveform figure at the time of a transient response when dead beat control is applied. FIG. 2 is a control block diagram of the inverter control device according to the first embodiment. FIG. 7 is a control block diagram of an inverter control device according to a second embodiment. FIG. 10 is a control block diagram of an inverter control device according to a third embodiment. The figure which shows the relationship between (a), (b) 3-phase current and the gain of a compensation voltage computing device, (c) The relationship between a 3-phase current and a 3-phase current deviation, and a compensation voltage command value. FIG. 10 is a control block diagram of an inverter control device according to a fourth embodiment. FIG. 15 is a control block diagram of a basic voltage command value calculation unit according to a fifth embodiment. The figure which shows switching of the direct-current component extraction structure by motor rotation speed. The control block diagram of the inverter control apparatus of a comparative example.

  Hereinafter, a plurality of embodiments of a power converter control device are described based on a drawing. In the plurality of embodiments and the comparative example, substantially the same configurations are denoted by the same reference numerals and descriptions thereof will be omitted. The power converter control device of the present embodiment converts DC power into three-phase AC power, and controls the operation of the power converter that supplies the load having resistance and inductance. In the present embodiment, the motor corresponds to a load, and the inverter corresponds to a “power converter”. Further, the inverter control device corresponds to the "power converter control device".

  First, referring to FIG. 1, the overall configuration of the motor drive system 90 will be described. In the inverter 60, six switching elements 61 to 66 of the upper and lower arms are bridge-connected. Specifically, switching elements 61, 62 and 63 are switching elements of upper arms of U phase, V phase and W phase, respectively, and switching elements 64, 65 and 66 are respectively below U phase, V phase and W phase It is a switching element of an arm. The switching elements 61 to 66 are, for example, IGBTs, and parallel connection is made of reflux diodes that allow current from the low potential side to the high potential side.

  The smoothing capacitor 15 is provided at the input of the inverter 60 to smooth the voltage of the battery 10. A boost converter may be provided between battery 10 and inverter 60. Inverter 60 converts DC power of battery 10 into three-phase AC power by switching elements 61-66 operating according to gate signals UH, UL, VH, VL, WH, and WL instructed from inverter control device 50. . Then, the inverter 60 applies the three-phase voltages Vu, Vv, Vw to the phase windings 81, 82, 83 of the motor 80.

  The motor 80 is, for example, a permanent magnet synchronous three-phase AC motor, and is typically a motor generator capable of powering and regenerating operation. The current path connected to the two-phase winding of the three-phase windings 81, 82, 83 of the motor 80 is provided with a current sensor for detecting a phase current value. In the example of FIG. 1, current sensors 71 and 72 for detecting a U-phase current value iu and a V-phase current value iv are provided in current paths connected to the U-phase winding 81 and the V-phase winding 82, respectively. . The rotation angle sensor 85 is a rotation angle sensor such as a resolver, and detects an electrical angle θ of the motor 80. Further, the angular velocity obtained by time-differentiating the electrical angle θ by the differentiator 86 is converted to calculate the number of rotations ω.

  The inverter control device 50 calculates a voltage command value to be applied to the motor 80 based on a torque command from a host ECU (not shown) and feedback information from the current sensors 71 and 72 and the rotation angle sensor 85. Then, gate signals UH, UL, VH, VL, WH, and WL are generated based on the calculated voltage command values, and the switching elements 61 to 66 of the inverter 60 are commanded.

  Here, a configuration of a known inverter control device 509 that performs dq axis current feedback control is shown in FIG. 14 as a comparative example. The configuration relating to the dq-axis current feedback control includes the other-phase current calculation unit 31, the three-phase / dq conversion unit 32, the dq-axis current deviation calculation unit 36, the controllers 371, 372, and the like. The other-phase current calculation unit 31 calculates the current value (for example, iw) of another one phase from the current value (for example, iu, iv) of the two phases of the three phases using Kirchhoff's law.

The three-phase / dq conversion unit 32 converts the three-phase current values iu, iv, iw into dq axis current values iq, id based on the electrical angle θ. Hereinafter, according to the order of the upper and lower sides in the figure, the q-axis current and the like will be described first in the description order of symbols such as the dq-axis current value and the like. The dq axis current deviation calculation unit 36 calculates dq axis current deviations Δiq, Δid that are deviations between the dq axis current command value iq ^* , id ^* and the dq axis current value iq, id. The controllers 371 and 372 respectively calculate the q-axis voltage command value Vq ^* and the d-axis voltage command value Vd ^* by PI control or the like.

Further, in the comparative example of FIG. 14, a non-interference control unit 38 that calculates a non-interference term is provided so as to compensate the interference voltage between the dq axes based on the dq axis current values iq, id. The non-interference terms calculated by the non-interference control unit 38 are added to the outputs of the controllers 371 and 372 by the non-interference term addition unit 39. The distinction between the symbols of the dq axis voltage command values Vq ^* and Vd ^* before and after the addition of the non-interference term is omitted.

The dq / 3 phase conversion unit 24 converts the dq axis voltage command values Vq ^* and Vd ^* into three phases based on the electrical angle θ to calculate U phase and V phase voltage command values Vu ^* and Vv ^* . The other-phase voltage calculation unit 55 calculates a voltage command value flow value Vw ^* of another W phase, which is one phase, from the U-phase and V-phase voltage command values Vu ^* and Vv ^* . The PWM signal generation unit 56 generates a gate signal by PWM control based on the three-phase voltage command values Vu ^* , Vv ^* and Vw ^* , and outputs the gate signal to the inverter 60.

  In the inverter control device 509 of the comparative example, when the voltage command value fluctuates with the fluctuation of the torque command, unbalance of three-phase current occurs. The three-phase current imbalance will be described with reference to FIG. In the three-phase current, a solid line, a broken line, and an alternate long and short dash line indicate the current of each phase. Before time tx, the current command is constant at i1, and the three-phase currents are balanced. When the torque command changes at time tx and the current command changes from i1 to i2, an imbalance occurs in the three-phase current, and an overcurrent occurs in one phase indicated by a solid line. As a result, damage to the inverter 60 and torque ripple may occur. In addition, a reactive current is also generated in the battery current on the DC side, and there is a possibility that the life may be reduced due to the heat generation of the battery 10 and the smoothing capacitor 15.

  The cause of occurrence of the three-phase unbalance and the solution thereof will be described with reference to FIGS. FIG. 3 shows a circuit model in which a single-phase component of a three-phase inverter is extracted taking the U-phase as an example. Vu shown by an alternating current symbol is an inverter output, and eu is an induced voltage. The U-phase current iu flows to the resistance R and the inductance L of the circuit. When an initial current flows in the inductance L, a voltage VL reverse to the U-phase current iu is generated. At this time, the inverter output Vu is expressed by equation (1). The symbol P in the equation is a differential operator. In addition, with regard to the notation of subscript letters of symbols in the following mathematical formulas, portions corresponding to the subscript letters may be written in normal letters in the description or in the drawings depending on the legibility.

  Here, as shown in FIG. 4, it is assumed that the inverter output Vu changes. Va is the amplitude of Vu. When the U-phase current iu is zero at tx when the inverter output Vu changes, the center of the current amplitude remains zero even after time tx. On the other hand, when the U-phase current iu is not zero at tx when the inverter output Vu changes, an offset current ΔIdc is generated in the negative direction in the example of FIG. 4 and the center of the current amplitude is offset from zero.

  Next, FIG. 5 shows a circuit model in which the U-phase current iu is separated into an alternating current (AC) component iu_ac and a direct current (DC) component iu_dc. The equation (1) of the inverter output Vu is separated into the equation (2.1) of the AC voltage Vu and the equation (2.2) of the DC offset voltage Vdc.

  The alternating current component iu_ac of the U-phase current changes in a sinusoidal manner. Further, in the configuration of the comparative example, which is a well-known technology, the direct current component iu_dc is attenuated according to the LR time constant as the equation (3) with the offset current −ΔIdc as an initial value as shown by the solid line.

  Thus, when the initial current flows in the inductance L at the time of transition, a DC offset current is generated. That is, in the three-phase inverter, since the DC offset current is always generated at the time of transition, the three-phase current unbalance is generated. Therefore, attention is directed to the point of controlling the DC offset current by the DC offset voltage from the equation (2.2) of the DC component. Then, as indicated by a broken line in the circuit model of the direct current component in FIG. 5, the direct current offset voltage Vdc is determined so that the direct current offset current becomes zero at the next sample timing after tx. Control for suppressing the DC offset current in this manner is called "dead beat control". Deadbeat control is expressed by equation (4).

  FIG. 6 shows a current waveform when a voltage obtained by adding the AC voltage Vu and the DC offset voltage Vdc determined by the dead beat control is applied to the inverter. The two-dot chain line is a current waveform when the dead beat control shown in FIG. 4 is not performed. Even when the U-phase current iu is not zero at tx when the inverter output Vu changes, dead-beat control is performed at tx when the u-phase current iu coincides with zero at the next sample timing.

  As described above, by applying the dead beat control to the DC offset component ΔIdc, the U-phase current iu can be controlled to reach the target value instantaneously. Therefore, the three-phase current unbalance at the time of transient response caused by the DC offset current is suppressed. As a result, damage to the inverter due to an overcurrent, occurrence of torque ripple, heat generation of a battery or a smoothing capacitor, and the like can be appropriately prevented.

  The inverter control device 50 of the present embodiment aims to suppress three-phase current unbalance at the time of a transient response in which a torque command fluctuates, using the above principle. Subsequently, the configuration and operation of the inverter control device 50 will be described for each embodiment. In the inverter control device of each embodiment, the third digit following “50” is assigned the embodiment number.

First Embodiment
The first embodiment will be described with reference to FIG. The inverter control device 501 of the first embodiment includes a basic voltage command value calculation unit 201, a compensation voltage command value calculation unit 401, a PWM signal generation unit 56 as a "gate signal generation unit", and the like. Here, in the reference numerals of the basic voltage command value calculation unit and the compensation voltage command value calculation unit of the first embodiment, “1” is added to the third digit following “20” and “40”, respectively. Regarding the reference numerals of the basic voltage command value calculation unit and the compensation voltage command value calculation unit of the following embodiments, when the configurations are substantially the same as those of the above embodiments, the reference characters of the above embodiments are used. On the other hand, when the configuration is different from that of the above-described embodiment, the third digit following “20” or “40” is newly assigned the number of that embodiment.

  Further, in the present embodiment, in response to the current sensors 71 and 72 detecting the current values iu and iv of the two phases of the U phase and the V phase, of the three phase current values, the two phases of the U phase and the V phase are detected. The current value is controlled, and the current value of the other one phase W phase iw is calculated using Kirchhoff's law. In other embodiments, any two phases may be used, and three phase current values may be detected and controlled. In the specification, the terms “three-phase current value” and “three-phase voltage value” are also used for the current value or voltage value of the two phases of the U phase and the V phase.

Basic voltage command value calculating section 201 includes a FF control calculation unit 21 and dq / 3-phase conversion unit 24, ^* the basic voltage command values Vu _b by the feed forward control, calculates the Vv _b ^*. The FF control calculation unit 21 calculates the dq axis voltage command values Vq ^* and Vd ^{* according} to the voltage equation shown in Expression (5). dq / 3-phase conversion unit 24, dq-axis voltage command value based on the electric angle theta Vq ^*, to convert the Vd ^* 3-phase, calculated U-phase, the basic voltage command values V phase Vu _b ^*, the Vv _b ^* Do.

The compensation voltage command value calculation unit 401 has a dq / three-phase conversion unit 41, a three-phase current deviation calculation unit 42, and three-phase current controllers 431, 432. The dq / three-phase conversion unit 41 performs three-phase conversion on the dq-axis current command values iq ^* and id ^* based on the electrical angle θ to calculate three-phase current command values iu ^* and iv ^* . Three-phase current deviation calculation unit 42 calculates three-phase current deviations Δiu and Δiv, which are deviations between three-phase current command values iu ^* and iv ^* and three-phase current values iu and iv detected by current sensors 71 and 72, respectively. Do.

U-phase current controller 431 and V-phase current controller 432 calculate U-phase and V-phase compensation voltage command values Vu _offset ^* and Vv _offset ^* based on current deviations Δiu and Δiv of U-phase and V-phase, respectively. . The current deviations Δiu and Δiv correspond to DC offset components during transient response. That is, the technical significance of the compensation voltage command value calculation unit 401 is that the three-phase current controllers 431 and 432 directly control the DC offset component.

And encompasses each phase current deviation .DELTA.i, represents a compensation voltage to encompass command value V _offset ^*, described compensating voltage command value V _offset ^* computation methods. The calculation of the compensation voltage command value V _offset ^* is generally performed by PD control according to equation (6.1) or P control according to equation (6.2). Here, Kp is a proportional gain, Kd is a differential gain, and (Δi / Δt) is a time differential value of the three-phase current deviation Δi. Formula (6.2) of P control is equivalent to what considered the differential term Kd ((DELTA) i / (DELTA) t) of Formula (6.1) of PD control as zero.

Further, in the above-described dead beat control, the DC offset voltage Vdc is determined by the equation (4) including the resistance R and the inductance L which are the machine constants of the motor 80 so as to suppress the DC offset current Δidc. The three-phase current controllers 431 and 432 may calculate the compensation voltage command value V _offset ^* by the equation (7) obtained by multiplying the right side of the equation (4) by the gain K (K ≦ 1). By using the machine constant of the motor 80, it is possible to calculate the voltage command value for following the command value most quickly.

  The equation (7) can be regarded as an equation in which the differential gain Kd in equation (6.1) of PD control is (K × L) and the proportional gain Kp is (K × R). Further, ignoring the differential term of the equation (7), the equation for P control is obtained with the proportional gain Kp as (K × R).

Compensation voltage adding unit 54, U-phase, the basic voltage command value for each V-phase Vu _b ^*, Vv _b ^* and the compensation voltage command value Vu _offset ^*, adds the Vv _offset ^*, the voltage command value after addition Vu ^* , Vv ^* is output. The other-phase voltage calculation unit 55 calculates a voltage command value flow value Vw ^* of another W phase, which is one phase, from the U-phase and V-phase voltage command values Vu ^* and Vv ^* .

The PWM signal generation unit 56 generates a gate signal by PWM control based on the three-phase voltage command values Vu ^* , Vv ^* and Vw ^* , and outputs the gate signal to the inverter 60. In another embodiment, the gate signal generation unit may generate a gate signal by a pulse pattern method or the like which selects an appropriate pattern from a plurality of voltage output patterns set in advance, not limited to PWM control.

  As described above, in the first embodiment, the AC component is controlled by the feedforward control by the basic voltage command value calculation unit 201, and the DC offset is controlled by the three-phase current feedback control by the compensation voltage command value calculation unit 401. High response can be achieved.

Second Embodiment
The second embodiment will be described with reference to FIG. In the inverter control device 502 of the second embodiment, the basic voltage command value computing section 202, the basic voltage command values Vu _b ^* by conjugation with a feed-forward control and the dq-axis current feedback control, and calculates the Vv _b ^*. The configuration of the compensation voltage command value calculation unit 401 is the same as that of the first embodiment.

The configuration related to feedforward control in the basic voltage command value calculation unit 202 is the same as that of the basic voltage command value calculation unit 201 in the first embodiment. The FF control calculation unit 21 calculates feed forward terms Vq ^* _ff and Vd ^* _ff of the dq axis voltage command value by the voltage equation.

  In the configuration related to dq axis current feedback control, other phase current calculation unit 31, three phase / dq conversion unit 32, low pass filter ("LPF" in the drawings and the following text) 34, dq axis current deviation calculation unit 36, control Vessels 371 and 372 are included. Except for the LPF 34, the configuration is substantially the same as the configuration of the comparative example of FIG.

The LPF 34 generates high frequency components of dq axis current values iq, id at a predetermined frequency or higher, and outputs dq axis current values iq _LPF , id _LPF after the LPF processing. In other words, the LPF 34 feeds back only the AC component to the dq axis current command values iq ^* and id ^* except for the DC offset of the dq axis current value iq and id. That is, the LPF 34 corresponds to a “DC component extraction unit” that extracts DC components of the dq-axis current values iq and id.

The dq-axis current deviation calculation unit 36 calculates dq-axis current deviations Δiq and Δid that are deviations between the dq-axis current command values iq ^* and id ^* and the dq-axis current values iq _LPF and id _LPF after LPF processing. The controllers 371 and 372 calculate the feedback term Vq ^* _fb of the q-axis voltage command value and the feedback term Vd ^* _fb of the d-axis voltage command value by PI control or the like. The voltage command value adding unit 22 adds the feedforward terms Vq ^* _ff and Vd ^* _ff of the dq-axis voltage command value and the feedback terms Vq ^* _fb and Vd ^* _fb .

  As described above, the basic voltage command value calculation unit 202 according to the second embodiment can improve the controllability in the steady state by feeding back only the AC component excluding the DC offset by the LPF 34. That is, since the dq axis current iq, id vibrates when the DC offset current is generated, only a DC component is extracted using a filter, and the dq axis current iq, id is controlled to realize a desired torque. Therefore, in the second embodiment, high response can be achieved by combining feedback control of the current excluding the DC offset by the compensation voltage command value calculation unit 401 and feedback control of the DC offset current by the basic voltage command value calculation unit 20. Control without steady-state deviation can be realized.

Third Embodiment
A third embodiment will be described with reference to FIG. 9 and FIG. The compensation voltage command value calculation unit 403 of the inverter control device 503 of the third embodiment differs from the compensation voltage command value calculation unit 401 of the first and second embodiments in the U-phase current controller 431 and the V-phase current controller 432. Except that the U-phase current value iu and the V-phase current value iv are input respectively, and the other configuration is the same. U-phase current controller 431 calculates U-phase compensation voltage command value Vu _offset ^* based on U-phase current value iu in addition to U-phase current deviation Δiu. V-phase current controller 432 calculates V-phase compensation voltage command value Vv _offset ^* based on V-phase current deviation iv in addition to V-phase current deviation Δiv.

  For example, in the configuration for performing dead beat control according to equation (7), the compensation voltage command value calculation unit 403 sets the gain K of the three-phase current controllers 431 and 432 according to the U-phase current value iu and the V-phase current value iv. Change from 0 to 1. Preferably, the gain K is set larger as the three-phase current values iu and iv are larger.

That is, when the load is low and the three-phase current values iu and iv are small, the influence of the current imbalance is small, and the gain K decreases so that the compensation voltage command value V _offset ^* becomes smaller with emphasis on control stability. Be done. On the other hand, when the load is high and the three-phase current values iu and iv are large, the gain K is increased so that the compensation voltage command value V _offset ^* becomes large with emphasis on high responsiveness in order to suppress current imbalance. Ru.

  FIGS. 10A and 10B show change patterns of the gain K according to the three-phase current values iu and iv. In the example of FIG. 10A, the gain K monotonously increases to 1 as the three-phase current values iu and iv increase in a region where the three-phase current values iu and iv are less than the saturation current value Isat. The gain K is 1 in the region where the three-phase current values iu and iv are greater than or equal to the saturation current value Isat. Further, in the example of FIG. 10B, the gain K is 0 in the region where the three-phase current values iu and iv are less than or equal to the critical value Icrt. That K is 0 is equivalent to not performing DC offset current suppression control.

  Further, in the configuration in which the PD control according to equation (6.1) is performed, the proportional gain Kp and the differential gain Kd are set to be larger as the three-phase current values iu and iv are larger. In the configuration that implements P control according to equation (6.2), the proportional gain Kp is set larger as the three-phase current values iu and iv are larger.

Further, the compensation voltage command value V _offset ^* is not limited to the configuration calculated by the equation using the gain K, but three-phase current values iu and iv and three-phase current deviations Δiu and Δiv as shown in FIG. It may be calculated directly by the map which takes as input. In this map, as the three-phase current values iu and iv are larger and as the three-phase current deviations Δiu and Δiv are larger, the compensation voltage command value V _offset ^* becomes larger. The characteristic lines of the map are not limited to straight lines as shown in FIG. 10C, and may be broken lines or curves.

Fourth Embodiment
The fourth embodiment will be described with reference to FIG. Inverter controller 504 in the basic voltage command value calculating portion 204 of the fourth embodiment does not perform the feedforward control, the basic voltage command values Vu _b ^* only by dq-axis current feedback control, and calculates the Vv _b ^*. The configuration of the compensation voltage command value calculation unit 401 is the same as that of the first embodiment.

  The configuration relating to dq axis current feedback control is the same as in the second and third embodiments. Further, in the example of FIG. 11, as in the comparative example of 14, the non-interference control unit 38 which calculates the non-interference term to compensate the interference voltage between the dq axes based on the dq axis current values iq and id is provided. ing. The portion Ex enclosed by a frame including the LPF 34 and the dq-axis current deviation calculation unit 36 in FIG. 11 is a portion cited in the description of the fifth embodiment shown in FIG. In the fourth embodiment, the controllability at the time of steady state can be improved by feedback controlling the AC component excluding the DC offset by the LPF 34 as the “DC component extraction unit”.

Fifth Embodiment
The fifth embodiment will be described with reference to FIGS. 12 and 13. The basic voltage command value calculation unit 205 of the fifth embodiment differs from the basic voltage command value calculation unit 204 of the fourth embodiment only in the configuration corresponding to the Ex unit of FIG. The other configuration is the same as that of FIG. The basic voltage command value calculation unit 205 is provided with a DC component extraction unit 330 including the switching determination unit 33, the LPF 34, and the moving average calculation unit 35.

The LPF 34 performs an LPF process for removing high frequency components of dq axis current values iq and id not lower than a predetermined frequency as in the fourth embodiment, and outputs dq axis current values iq _LPF and id _LPF after the LPF process. The moving average calculation unit 35 performs moving average processing for calculating a moving average of dq axis current values iq and id in a predetermined electric angle range or predetermined time, and outputs dq axis current values id _MVAVR and iq _MVRVR after moving average processing. Do. The switching determination unit 33 switches between the LPF processing and the moving average processing as the extraction processing of the DC component according to the number of rotations of the motor.

  In the fourth embodiment having only the LPF 34 as the DC component extraction configuration, the cut-off frequency is fixed. Therefore, in the frequency domain where the frequency of the AC component of the dq-axis current values iq and id, which is the sixth-order component of the rotational frequency ω, is equal to or lower than the cutoff frequency, the DC component can not be extracted. It becomes unstable. In addition, if the cut-off frequency is lowered too much to allow the LPF 34 to function in a wide rotation speed region, the convergence after the transient response is deteriorated. Therefore, as shown in FIG. 13, in the low rotation speed region where the frequency of the AC component of the dq-axis current values iq and id is equal to or lower than the cutoff frequency of the LPF 34, the switching determination unit 33 switches from the LPF processing to the moving average processing To determine. The section of the moving average process is set to a predetermined electrical angle range or a predetermined time corresponding to, for example, one cycle of the sixth order component of the rotational frequency.

  As described above, in the fifth embodiment, in the dq-axis current feedback control of the basic voltage command value calculation unit 205, the LPF processing is performed in the high rotation speed region larger than the cutoff frequency according to the rotation speed of the motor. The processing is switched so that the DC component is extracted by the moving average processing. Therefore, it is possible to properly balance the control stability in the steady state and the convergence after the transient response.

(Other embodiments)
The load of the inverter (power converter) in the present invention is not limited to the motor 80 shown in the above embodiment. However, the present invention is effective for a system for supplying power to a load having resistance and inductance, on the premise that unbalance of three-phase current becomes a problem.
As mentioned above, the present invention is not limited to the above-mentioned embodiment at all, and can be implemented in various forms in the range which does not deviate from the meaning of an invention.

201, 202, 204, 205 ... Basic voltage command value calculation unit,
401, 403 ... compensation voltage command value calculation unit,
50 (501-504) ... inverter control device (power converter control device),
56: PWM signal generator (gate signal generator),
60 · · · inverter (power converter),
80: Motor (load).

Claims (6)

A power converter control device that controls the operation of a power converter (60) which converts DC power into three-phase AC power and supplies the load (80) with resistance and inductance,
dq-axis current command value (iq ^*, ib) dq-axis voltage command value based on (Vq ^*, Vd ^*) is calculated and further the dq-axis voltage command value 3-phase converted basic voltage command values (Vu _b ^* , Vv _b ^* ), and a basic voltage command value calculation unit (201, 202, 204, 205);
The three-phase current deviation (Δiu, Δiv), which is the deviation between the three-phase current command value (iu ^* , iv ^* ) obtained by three-phase conversion of the dq-axis current command value, and the three-phase current value (iu, iv) A compensation voltage command value computing unit (401, 403) for computing a compensation voltage command value (Vu _offset ^* , Vv _offset ^* ) based on
A gate signal generation unit (56) for generating a gate signal for instructing the power converter based on a voltage command value (Vu ^* , Vv ^* ) obtained by adding the basic voltage command value and the compensation voltage command value;
Power converter controller comprising: The load is a motor,
The power converter control device according to claim 1, wherein the compensation voltage command value calculation unit calculates the basic voltage command value based on the three-phase current deviation and a resistance or an inductance which is an equipment constant of the motor. The compensation voltage command value calculation unit (403)
The power converter control device according to claim 1, further calculating the compensation voltage command value based on the three-phase current value. The basic voltage command value calculation unit (201, 202)
The power converter control device according to any one of claims 1 to 3, wherein the basic voltage command value is calculated by feedforward control using a voltage equation. The basic voltage command value calculation unit (202, 204, 205)
A DC component extraction unit (34, 330) for extracting a DC component of the dq-axis current value (iq, id) subjected to dq conversion from the three-phase current value;
The basic voltage command value is calculated by feedback control based on a dq-axis current deviation (Δiq, Δid) which is a deviation between the dq-axis current command value and the dq-axis current value excluding the DC component by the DC component extraction unit. The power converter control device according to any one of claims 1 to 4 which calculates. The load is a motor,
The DC component extraction unit (330) of the basic voltage command value calculation unit (205) performs low-pass filter processing for removing high frequency components of a predetermined frequency or more of the dq axis current value according to the rotation speed of the motor. The power converter control device according to claim 5, switching between moving average processing for calculating a moving average at a predetermined electric angle range or a predetermined time of the dq axis current value.

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