JP2019068601A - Control method of motor, and control arrangement of motor - Google Patents

Abstract

To improve accuracy of rotation control of a motor.SOLUTION: A control method of a motor is a control method of a variable flux type motor where the magnetic flux from a permanent magnet toward a stator can be changed according to the application current to the motor, by a magnetic barrier layer provided in a rotor including a permanent magnet. This control method calculates a gain for use in the feedback control of the application current at multiple calculation timings in the route from the current working point indicated by the application current to a target working point indicated by the target current, and controls the application current by feedback control using the gain.SELECTED DRAWING: Figure 3

Description

  The present invention relates to a control method of a motor and a control device of the motor.

  In a motor, when the rotational speed is controlled, the inductance of the motor may fluctuate due to magnetic non-linearity and magnetic saturation. Therefore, as in the technology disclosed in Patent Document 1, the flux linkage number of each winding with respect to the current value is obtained in advance by analysis of the nonlinear finite element method of the motor, etc., and these current values and the flux linkage number There is known a technique for controlling motor torque and the like using a data table corresponding to the above. By this technique, control parameters can be optimized and motor control with higher accuracy can be realized.

JP, 2008-141835, A

  Here, among the motors, there is a motor called a variable magnetic flux type (or variable leakage magnetic flux type) in which a rotor is provided with a magnetic barrier such as a physical air gap. In this variable flux motor, a part of the magnetic flux from the permanent magnet of the rotor to the stator is blocked by the magnetic barrier, and the amount of flux blocked by the magnetic barrier is the motor It changes according to the current applied to it. Therefore, since the amount of magnetic flux from the rotor to the stator changes according to the magnitude of the applied current, the amount of change in inductance is relatively large.

  In this variable magnetic flux motor, the provision of the magnetic barrier causes a large change in the inductance due to the applied current. Therefore, even if the analysis method as disclosed in Patent Document 1 is used, the calculation accuracy of the inductance does not become sufficiently high. Therefore, even if torque, speed, position and the like are controlled using the calculated inductance, the accuracy is high. There is a risk that rotation control can not be performed. From such a thing, the technique which improves the precision of rotation control further is calculated | required.

  The present invention has been made in view of such problems, and it is an object of the present invention to provide a motor control method and motor control device capable of improving the accuracy of motor rotation control.

  According to one aspect of the motor control method of the present invention, a magnetic barrier provided on a rotor provided with a permanent magnet can change the magnetic flux from the permanent magnet to the stator according to the current applied to the motor It is a control method of a variable magnetic flux type motor. This control method calculates gains used for feedback control of the applied current at a plurality of calculation timings in a path from the current operating point indicated by the applied current to the target operating point indicated by the target current, and uses the gains. The applied current is controlled by feedback control.

  According to one aspect of the present invention, the accuracy of rotation control of the motor can be improved.

FIG. 1 is a cross-sectional view of a variable magnetic flux type motor according to a first embodiment. FIG. 2 is a block diagram showing a part of the motor. FIG. 3 is a block diagram of a motor and its control device. FIG. 4 is a diagram showing the timing of calculating the gain. FIG. 5A is a diagram showing timing of calculating gains in the first modified example. FIG. 5B is a diagram showing an example of a change in inductance. FIG. 5C is a diagram showing an example of a change in inductance. FIG. 6A is a diagram showing an example of calculation timing of gain in the second modified example. FIG. 6B is a diagram showing another example of gain calculation timing. FIG. 7 is a diagram showing the relationship between the torque of the motor and time in the third modification. FIG. 8 is a block diagram of a motor and a controller of the second embodiment. FIG. 9 is a block diagram of a motor and a control device according to a third embodiment. FIG. 10 is a diagram showing an operating point when the target current vibrates. FIG. 11 is a diagram showing an operating point when the target current vibrates. FIG. 12 is a diagram showing an operating point when the target current vibrates in the fourth modification. FIG. 13 is a diagram showing an operating point when the target current vibrates.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings.

First Embodiment
In general, a motor referred to as variable flux type or variable leakage flux type is configured to be capable of changing the amount of magnetic flux from the rotor to the stator due to the physical configuration of the rotor or the like. It is known that the amount of change of the magnetic flux changes according to the current applied to the motor, and the inductance can be changed by controlling the applied current, so that an optimum operating point according to the required load is realized. be able to.

  FIG. 1 is a cross-sectional view perpendicular to the axial direction of the variable magnetic flux type electric motor (rotating electric machine) 100 according to the first embodiment, showing a quarter of the entire configuration. In the remaining three quarters of the overall configuration, the partial configuration shown in FIG. 1 is continuously repeated. The motor 100 according to the present embodiment includes a stator 1 having an annular shape, and a rotor 2 disposed concentrically with the stator 1 and having an air gap 13 between the stator 1 and the stator 1. And a plurality of permanent magnets 3 fitted to the rotor 2 to constitute an electric motor or a generator.

  The stator 1 includes a ring-shaped stator core 11, a plurality of teeth 8 protruding from the stator core 11 toward the inner circumferential side, and a slot 9 which is a space between the adjacent teeth 8. The stator winding 10 is wound around the teeth 8. The stator core 11 is formed of, for example, an electromagnetic steel plate which is a soft magnetic material.

  The rotor 2 has a rotor core 12. The rotor core 12 is formed in a cylindrical shape by a so-called laminated steel plate structure configured by axially laminating a large number of electromagnetic steel plates formed by punching a steel plate having a high magnetic permeability in an annular shape. ing. Further, in the vicinity of the peripheral portion of rotor core 12 facing stator 1, along the circumferential direction, the plurality of permanent magnets 3 are equally spaced from each other, and the polarities of the permanent magnets 3 adjacent to each other are different. It is provided to be polar. The rotor core 12 according to the variable magnetic flux type motor 100 of the present embodiment has an eight-pole structure in which eight permanent magnets 3 are provided along the circumferential direction, as can be inferred from the partial configuration shown in FIG. Have.

  Further, the rotor 2 is a magnetic barrier (also referred to as a flux barrier) which is a space portion formed by punching an electromagnetic steel plate forming the rotor core 12 between magnetic poles formed by the adjacent permanent magnets 3. ) 4,5. The magnetic barriers 4 and 5 have larger magnetic resistance than the magnetic steel sheet. Thus, the magnetic barriers 4, 5 act as a flux barrier for the magnet flux in the magnetic circuit that the permanent magnets 3 constitute on the rotor 2. As shown in the figure, the magnetic barrier 4 is formed in a substantially triangular shape whose width on the rotation center side is narrower than the outer peripheral side of the rotor 2. The magnetic barrier 5 is formed in a substantially U shape so as to cover the magnetic barrier 4 closer to the center of rotation than the magnetic barrier 4. However, the shapes of the magnetic barriers 4 and 5 are not limited to the shapes shown in the drawings as long as the technical effects described later can be obtained. Further, in the rotor 2, the magnetic barriers 4 and 5 are formed as shown in the figure, so that the magnetic flux emitted from the permanent magnet 3 constituting one magnetic pole constitutes another adjacent permanent magnet 3. A magnetic flux bypass path 6 is formed which is a path for leakage to the magnetic pole side. Further, the rotor 2 is provided on the outer peripheral side of the magnetic barrier 4 with a narrow bridge-shaped portion 7 connected to the magnetic flux bypass path 6.

  The permanent magnet 3 is fixed to the rotor core 12 by being fitted into an air gap formed in a corresponding portion of the rotor core 12. Further, in the permanent magnet 3, the radial direction of the rotor 2 is the magnetization direction. Here, in the present embodiment, a geometric center of the permanent magnet 3 is defined as a d-axis, and a position electrically orthogonal to the d-axis is defined as a q-axis. In addition, since the variable magnetic flux type motor 100 of this embodiment has an eight-pole structure, a position of 22.5 degrees in mechanical angle from the d axis is defined as the q axis. The above-mentioned magnetic barriers 4 and 5 are formed on the q axis.

  The above is the basic configuration of the variable magnetic flux type motor 100 according to the first embodiment. Here, before the detailed description of the present embodiment, the basic principle of the variable magnetic flux type motor 100 which is an object of the present invention will be described.

  The motor 100 of the variable magnetic flux type targeted by the present invention is a part of the magnetic flux emitted from the permanent magnet 3 provided in the rotor 2 by the action of the load current (stator current) applied to the stator winding 10 (leakage flux The magnetic path of the above can be changed. With this feature, the motor 100 can passively change the stator interlinkage magnetic flux by controlling the stator current, so that the magnetic force of the permanent magnet 3 included in the rotor 2 can be apparently varied by current control. Have the ability to Due to such characteristics, as described above, the variable flux motor 100, which is the subject of the present invention, is also referred to as a variable leakage flux type.

  FIG. 2 is a configuration diagram showing a part of the motor 100, for explaining an outline of operation in the variable magnetic flux type. The shapes and arrangements of the components of the variable magnetic flux type motor 100 shown in FIG. 2 are general ones for explaining the outline of the operation.

  The motor 100 includes a stator 1 and a rotor 2 disposed to form an air gap 13 between the stator 1 and the rotor 1. The rotor 2 includes a permanent magnet 3, magnetic barriers 4 and 5, and a magnetic flux bypass 6 connected via magnetic barriers 4 and 5 provided between two adjacent permanent magnets. Have. The direction indicated by the arrow shown on the left side of the drawing indicates the rotation direction of the rotor 2.

  The magnetic path direction 15 shown in FIG. 2 indicates the flow direction of the leakage flux when no current is applied to the stator winding 10 of the stator 1. As shown in FIG. 2, in the rotor, a part of the magnetic flux emitted from the permanent magnet 3 leaks to the adjacent different pole side through the magnetic flux bypass path 6.

  For this reason, the amount of magnetic flux of the main magnetic flux component from the permanent magnet 3 to the stator 1 side, that is, the stator interlinkage flux is relatively reduced, and the magnetic force of the permanent magnet 3 is apparently weakened. Thereby, the iron loss generated by the stator interlinkage magnetic flux can be reduced. This effect is particularly effective in improving the efficiency of the motor 100 in a low load, high speed region.

  On the other hand, the magnetic path direction 20 indicates the flow direction of the magnetic flux when current is supplied to the stator winding 10. The flow of the stator current causes the magnetic flux emitted from the permanent magnet 3 to be drawn to the stator side in the rotational direction of the rotor 2 as the magnetic path direction 20 points. Therefore, when the current is not supplied to the stator winding 10, the magnetic flux leaking in the rotor 2 as a leakage magnetic flux can be efficiently converted into the stator interlinkage magnetic flux. Thereby, the motor 100 can output the high torque equivalent to the state which the magnetic flux which comes out of the permanent magnet 3 does not leak.

  Thus, by providing the magnetic flux bypass path 6 between the magnetic poles formed in the rotor 2 and changing the leakage flux in the rotor by controlling the stator current, a characteristic in which the stator interlinkage magnetic flux changes passively The motor having the above is called a variable magnetic flux type (variable leakage magnetic flux type).

  The above is the basic principle of the variable magnetic flux type motor 100 to which the present invention is applied. Hereinafter, details of a control device that controls such a variable magnetic flux type motor 100 will be described.

  FIG. 3 is a block diagram of the motor 100 and the control device 200. The control device 200 is configured by a microcomputer including a central processing unit (CPU), a read only memory (ROM), a random access memory (RAM), and an input / output interface (I / O interface). The control device 200 stores a program, and executes the stored program to execute control described later.

Control device 200 controls three-phase AC currents I _u , I _v and I _w used for rotation control based on d-axis target current I _d ^* and q-axis target current I _q ^* calculated by target current calculation unit 230. Is output to the motor 100. In target current calculation unit 230, d-axis target current I _d ^* and q-axis target current I _q ^* for achieving a desired rotational speed in motor 100 based on the driver's operation of an accelerator, a brake, etc. It is calculated. Further, the rotational speed ω of the motor 100 is feedback input to the control device 200.

A subtraction value obtained by subtracting the d-axis current I _d output from the inverter 211 from the d-axis target current I _d ^* is input to the d-axis PI control unit 201 from the subtractor 202 provided in the previous stage. Further, in the d-axis PI control section 201, the d-axis gain G _d obtained by gain map 212 is inputted. The d-axis gain G _d includes a proportional control gain and an integral control gain, and the d-axis PI control unit 201 performs d-axis by proportional and integral control using the input d-axis gain G _d. The current I _d is feedback controlled.

A subtractor 203 is provided at the subsequent stage of the d-axis PI control unit 201, and the d-axis non-interference control unit 204 is obtained from the feedback-controlled d-axis target current I _d ^* output from the d-axis PI control unit 201. The subtraction value obtained by subtracting the d-axis non-interference control current output from V.sub.2 is output to the inverter 211.

Also regarding the q axis, the q axis target current I _q ^* input to the control device 200 is subjected to feedback control by the q axis PI control unit 221 and the subtracter 222, and then the subtractor 223 and the q axis noninterference The interference component is canceled and decoupled by the control unit 224 and output to the inverter 211. Note that, in the process of this calculation, the q-axis gain G _q is input from the gain map 212.

In inverter 211, the switching elements are controlled according to d-axis target current I _d ^* and q-axis target current I _q ^* input, whereby three-phase AC currents Iu, Iv and Iw are output to motor 100. Ru. Thus, rotation control of the motor 100 is performed. An ammeter 101 is provided between the inverter 211 and the motor 100, and the uvw phase is converted to the dq axis with respect to the three-phase AC currents Iu, Iv and Iw measured by the ammeter 101. Is performed, and the d-axis current I _d and the q-axis current I _q are feedback input to the controller 200.

  Here, processing in the gain map 212 will be described. In the variable magnetic flux motor 100, the inductance largely changes according to the applied current. In general, in motor 100, the responsiveness in feedback control of applied current changes according to the inductance. That is, in the variable magnetic flux type motor 100, the responsiveness of feedback control is largely different according to the applied current.

  If the gains (proportional control gain and integral control gain) used for feedback control in the d-axis PI control unit 201 and the q-axis PI control unit 221 are large, the time for the current to reach a steady state is short, but the vibration is And there is a risk of On the other hand, if the gain is small, the risk of divergence is small, but the time to reach a steady state is long. Therefore, according to the current applied to the motor 100, it is assumed that the gain map 212 stores in advance a gain that is in the shortest steady state without diverging.

The gain map 212 is. The d-axis gain G _d and the q-axis gain G _q are calculated according to the d-axis current I _d and the q-axis current I _q input to the motor 100 using such a prestored map. calculate. The calculation of the gain by the gain map 212 is performed according to the calculation timing TM generated by the calculation timing generation unit 214. Then, the calculated gain is used until the next calculation timing.

The inductance estimation unit 213 obtains the d-axis inductance L _d and the q-axis inductance L _{q of} the motor 100 from the input of the d-axis gain G _d and the q-axis gain G _q . This is because the current applied to the motor 100 corresponds to each of the gain and the inductance, so the inductance can be estimated according to the gain to be obtained. The inductance can also be estimated from the current applied to the motor 100.

In the calculation timing generation unit 214, the current operating point determined by the input d-axis current I _d and q-axis current I _q , the d-axis target current I _d ^* , and the q-axis target current I _q ^* The calculation timing of the gain is calculated from the target operation point to be determined, and the calculation timing TM indicating this timing is output to the gain map 212. The calculation timing generation unit 214 may receive an operation state such as a control cycle of the motor 100, a load condition, and a rotational speed, and the calculation timing may be calculated using these operation states.

The d-axis non-interference control unit 204 and the q-axis non-interference control unit 224 are a d-axis non-interference control current and a q-axis non-interference control current for realizing mutual non-interference in each of the d-axis and q-axis Are respectively output to the subtractor 203 and the subtractor 223. The d-axis non-interference control unit 204 performs d-axis non-interference control from the rotational speed ω of the motor, the d-axis current I _d output from the inverter 211, and the d-axis inductance L _d estimated by the inductance estimation unit 213. Determine the current. Similarly, the q-axis non-interference control unit 224 determines the q-axis from the motor rotational speed ω, the q-axis current I _q output from the inverter 211, and the q-axis inductance L _q estimated by the inductance estimation unit 213. Determine non-interference control current.

  Here, in general motor control, the gain often uses a fixed value. On the other hand, in the variable magnetic flux type motor 100, since the inductance largely changes according to the applied current, the use of a fixed value as the gain for feedback control causes poor followability, which leads to overshoot and stabilization. Sometimes it takes time. Therefore, by sequentially calculating the gains in the gain map 212 at the calculation timing instructed by the calculation timing generation unit 214, it is possible to shorten the time to stabilization without diverging by feedback control.

  FIG. 4 is a diagram showing calculation timing by the calculation timing generation unit 214. As shown in FIG.

In this figure, the d-axis current I _d is shown on the horizontal axis, and the q-axis current I _q is shown on the vertical axis, and the operating point of the motor 100 is shown. From the initial operating point P0 where the d-axis current I _d and the q-axis current I _q are both zero, to the target operating point P ^* indicated by the d-axis target current I _d ^* and the q-axis target current I _q ^* And the path when changing the applied current is shown. Then, in this example, in the path from the initial operating point P0 to the target operating point P ^* , the gain is calculated by the gain map 212 at a plurality of calculation timings from P1 to P7. Note that the calculation timings of P1 to P7 do not have to be determined at one time, and for example, the calculation timings may be determined sequentially so that P2 is determined after P2 is determined in P1.

  Specifically, for example, at the initial operation point P0, the gain is calculated from the current value, and feedback control from the initial operation point P0 to the operation point P1 which is the next calculation timing is performed using the calculated gain. It will be. At the operating point P1, the gain is calculated from the current value at the operating point P1, and feedback control is performed from the operating point P1 to the operating point P2, which is the timing for calculating the next gain. Thus, the control accuracy of feedback control can be improved by calculating the gain sequentially.

  In the present embodiment, the gains are calculated at seven operating points on the path, but the present invention is not limited to this. The gains may be calculated sequentially at a plurality of operating points. In addition, the calculation timing generation unit 214 can appropriately determine the calculation timing of the gain, the calculation timing of the gain, and the calculation period according to the control period of the motor 100, the load condition, the operating condition such as the rotation speed.

  According to the first embodiment, the following effects can be obtained.

The motor 100 controlled by the control method of the first embodiment is of the variable magnetic flux type, and has the rotor 2 provided with the magnetic barriers 4 and 5, and the rotor 2 according to the applied current. The magnetic flux from the permanent magnet 3 to the stator 1 can be changed. In order to control this motor 100, the gain is a gain map used for feedback control in the d-axis PI control section 201, the q-axis PI control section 221 using the d-axis current I _d and the q-axis current I _q is applied to the motor 100 It is determined at 212.

The calculation timing generation unit 214 sets a plurality of operation points P1 to P7 on the path as calculation timings in the path from the initial operation point P0 to the target operation point P ^* indicated by the target current. Then, the gain map 212 sequentially calculates the gain based on the current at the operating point at the calculation timing. The feedback control is performed in the d-axis PI control unit 201 and the q-axis PI control unit 221 using the gains sequentially calculated in this manner, and the currents I _d and I _q (I _u , I) applied to the motor 100 _v , I _w ) are obtained.

The variable magnetic flux motor 100 is characterized in that the inductance changes significantly in response to the applied current. In general, in motor 100, responsiveness in feedback control of applied current differs depending on the inductance. That is, in the variable magnetic flux type motor 100, the response of the feedback control in accordance with the current input is greatly different, calculates the d-axis gain G _d and q-axis gain G _q is used in the feedback control in accordance with the applied current There is a need to.

Therefore, when controlling the change from the initial operating point P0 to the target operating point P ^* in the motor 100, a plurality of calculation timings are set in the change path, and at these calculation timings, the gain corresponding to the applied current is set. Calculate. In this way, even if the inductance changes significantly, the gain can be calculated appropriately, so feedback control can be appropriately performed, and divergence can be suppressed and stabilization time can be shortened.

  According to the control method of the motor 100 of the first embodiment, the calculation timing generation unit 214 determines the calculation timing of the gain in accordance with the operation condition such as the control cycle, load condition, and rotational speed of the motor 100. By doing this, for example, when the amount of change in the inductance of the motor 100 is large, optimal feedback control can be realized by shortening the calculation interval of the gain. On the other hand, when the change amount of the inductance of the motor 100 is small, an unnecessary change in the gain can be suppressed by lengthening the calculation interval.

  According to the control method of the motor 100 of the first embodiment, the gain map 212 calculates the gain based on the current value indicating the current calculation timing operation point at the gain calculation timing. By doing this, the gain can be sequentially calculated with relatively simple control, so that the whole process can be simplified.

(First modification)
In the example of the first embodiment described above, an example has been described in which the gain is corrected for each predetermined calculation cycle based on the current of the current operating point at the calculation timing of the gain. In this embodiment, a modification of the method of setting the calculation timing of the gain will be described.

  FIG. 5A is a diagram showing a first modification of the calculation timing of the gain. In this figure, only the path of change of the operating point is shown in one dimension.

According to this example, the calculation timing generation unit 214 sets the operation point from P1 to P6 as the calculation timing of the gain in the path from the initial operation point P0 to the target operation point P ^* . Then, at each gain calculation timing, an intermediate operation point between the current calculation timing operation point and the operation point at the next calculation timing is determined, and the gain is calculated using the current indicating this intermediate operation point.

  Specifically, for example, at the initial operation point P0, the calculation timing generation unit 214 sets an intermediate operation point between the initial operation point P0 of the current calculation timing and the operation point P1 at the next calculation timing on the route. It is generated halfway between the initial operating point P0 and the operating point P1. Then, the intermediate operation point is transmitted to the gain map 212 using the calculation timing TM. Thus, since the gain at the intermediate operating point is calculated in the gain map 212, feedback control from the initial operating point P0 to the operating point P1 can be performed using the gain.

  Here, at the timing of calculating the gain, the case where the inductance at the next calculation timing is smaller than the inductance at the current operating point will be examined.

  FIG. 5B is a diagram showing the relationship between the current and the inductance when the inductance is reduced. This figure shows the operating point at the current calculation timing and the operating point at the next calculation timing. Then, it is shown that the inductance L becomes smaller from the current operating point to the operating point of the next calculation timing.

  Considering the section from the current operation point to the operation point of the next calculation timing, the inductance is relatively large at the current operation point which is the start timing of this section, but based on the intermediate operation point Since the relatively small gain calculated in the above manner is used for feedback control, the responsiveness becomes low. On the other hand, at the operation point of the next calculation timing which is the end timing of this section, although the inductance is relatively small, a relatively large gain calculated based on the intermediate operation point is used. , Become more responsive. As described above, in the section that changes from the initial operating point P0 to the operating point P1, an improvement in stability due to the gradually higher response is expected.

  Next, the case where the inductance at the next calculation timing is larger than the inductance at the current operating point at the timing of calculating the gain will be examined.

  FIG. 5C is a diagram showing the relationship between the current and the inductance when the inductance is increased. In this figure, it is shown that the inductance L increases from the current operating point to the operating point of the next calculation timing.

  In the section from the current operation point to the operation point of the next calculation timing, at the current operation point which is the start timing, although the inductance is relatively small, the comparison calculated based on the intermediate operation point Response is increased because a large gain is used. On the other hand, at the operation point of the next calculation timing which is the end timing of this section, although the inductance is relatively large, a relatively small gain calculated based on the intermediate operation point is used. , Responsiveness is low. As described above, in the section in which the operating point changes from P0 to P1, the responsiveness gradually decreases. Therefore, the response is high at the beginning of the section and is likely to diverge, but the suppression of the divergence is expected at the end.

  According to the first modification, the following effects can be obtained.

  According to the control method of motor 100 of the first modification, calculation timing generation unit 214 sets an intermediate operation point between the current operation point and the operation point at the next calculation timing at the gain calculation timing as a path between the two. Generate at Then, the gain map 212 calculates the gain based on the current indicating the intermediate operation point. By doing this, even when the inductance of the motor 100 changes significantly until the next gain calculation timing, feedback is obtained by using the gain calculated using the current at the intermediate operating point. It is possible to suppress the divergence due to control and the increase in convergence time.

  According to the control method of the motor 100 of the first modification, as shown in FIG. 5B, the inductance of the motor 100 decreases from the current operating point toward the operating point at the next calculation timing. In such a case, in the section from the current operation point to the operation point of the next calculation timing, stability can be improved by gradually increasing the responsiveness.

  According to the control method of the motor 100 of the first modification, as shown in FIG. 5C, the inductance of the motor 100 increases from the current operating point to the operating point at the next calculation timing. In such a case, in the section from the current operating point to the operating point of the next calculation timing, the responsiveness can be gradually reduced, so that the divergence can be suppressed.

(2nd modification)
In the previous examples in the first embodiment, the example in which the calculation period of the gain is constant has been described. In the second modification, an example in which the calculation cycle is changed will be described.

  FIG. 6A and FIG. 6B are diagrams showing changes in the operating point in the second modification.

In the case shown in FIG. 6A, the difference between the q-axis current I _q between the initial operating point P0 and the target operating point P ^* is large as compared with the case shown in FIG. 6B. Here, since the q-axis current I _q affects the torque of the motor 100, the case shown in FIG. 6A is different from the case shown in FIG. 6B from the initial operating point P0 to the target operating point P ^* . The change of the torque of the motor 100 between is small.

Therefore, when the change in torque is relatively large as shown in FIG. 6A, a large change in gain is expected, so the gain calculation interval is short between the initial operating point P0 and the target operating point P ^*. Set On the other hand, when the change in torque is relatively small as shown in FIG. 6B, the calculation interval is set long.

  As a result, when the torque fluctuation is large and it is necessary to increase the calculation frequency of the gain, by shortening the calculation interval of the gain, the calculation frequency of the gain becomes high, and the followability in feedback control can be improved. . On the other hand, when the fluctuation of torque is small and it is not necessary to calculate a large amount of gain, the frequency of calculating the gain is lowered by lengthening the interval of calculating the gain, and unnecessary processing can be omitted.

  According to the second modification, the following effects can be obtained.

According to the control method of the motor 100 of the second modification, when the change in q-axis current is large in the path from the initial operating point P0 to the target operating point P ^* , the calculation timing generation unit 214 calculates the gain calculation timing. Increase the interval between By doing this, since the gain can be calculated frequently when the fluctuation of the torque is large, the followability can be improved. On the other hand, when the change of the d-axis current is small and the change of the torque is small, frequent change of the gain is unnecessary. Therefore, unnecessary processing can be omitted by increasing the calculation interval of the gain.

(Third modification)
In the second modification, an example in which the calculation period of the gain is changed has been described. In the third modified example, an example in which calculation periods of a plurality of gains are variably set will be described.

FIG. 7 is a diagram showing the relationship between the torque of the motor 100 and the elapsed time when controlled by the control device 200. As shown in FIG. On the vertical axis, torque is shown, and torques at the initial operating point P0 and the target operating point P ^* are shown as T0 and T ^* . The times at which the torque T0 and T ^* become are shown as t0 and te.

According to this figure, the torque has a large change rate when the value is small, and the change rate is small when the value is large. Therefore, for torque, T1 to T8 are set such that the intervals from torque T0 to T ^* are equal. Then, times corresponding to the torques T1 to T8 are set as t1 to t8.

  For example, in a section where the torque changes from T0 to T1, since the rate of change in torque is relatively large, the calculation interval of gain from time t0 to t1 becomes relatively short. On the other hand, since the rate of change of the torque is relatively small in the section from when the torque changes from T8 to the target torque Tf, the gain calculation interval from the time t8 to the target time tf becomes relatively long.

  As described above, the calculation timing generation unit 214 sets the calculation timings of the gains t1 to t8 so that the amounts of change in torque become equal intervals using the relationship between torque and time as shown in FIG. 7. Do. By doing this, when the rate of change of torque is large, the number of times of gain calculation increases, so that the followability can be enhanced, and when the rate of change of torque is small, the number of times of gain calculation decreases. Unnecessary processing can be omitted.

  According to the third modification, the following effects can be obtained.

According to the control method of motor 100 of the third modification, calculation timing generation unit 214 determines that the gain change rate is relatively large in the path from initial operation point P0 to target operation point P ^* . Reduce the calculation interval. By doing this, the gain is frequently updated, so that the followability can be improved. On the other hand, when the rate of change of the torque is relatively small, it is less necessary to frequently change the gain, so unnecessary processing can be omitted by increasing the calculation interval.

Second Embodiment
In the first embodiment, an example in which the inductance estimation unit 213 is provided at the subsequent stage of the gain map 212 has been described. In the present embodiment, an example in which a gain calculator is provided downstream of the inductance estimation unit 213 will be described.

  FIG. 8 is a diagram showing a control device 200 of the present embodiment.

  According to this figure, the d-axis voltmeter 231 is added between the subtractor 203 and the inverter 211 as compared with the control device 200 of the first embodiment in the control device 200 of the present embodiment, and the subtractor A q-axis voltmeter 232 is added between the H.223 and the inverter 211.

Furthermore, in addition to the d-axis current I _d and the q-axis current I _q , the inductance estimation unit 213 further includes the d-axis voltage V _d measured by the d-axis voltmeter 231 and the q-axis voltmeter 232. The q-axis voltage V _q to be measured is input. Inductance estimation section 213, together with obtaining the d-axis inductance L _d and a d-axis current I _d and the d-axis voltage V _d, obtains the q-axis inductance Lq of the q-axis current I _q and the q-axis voltage V _q. Then, the obtained d-axis inductance L _d is output to the d-axis gain calculator 233 and the d-axis non-interference control unit 204, and the q-axis inductance L q is obtained by the q-axis gain calculator 234 and the q-axis non-interference control unit 224. Output to

d-axis gain calculator 233 calculates the d-axis gain G _d used in the d-axis PI control section 201 based on the d-axis inductance L _d. The q-axis gain calculator 234 obtains the q-axis gain G _q used by the q-axis PI control unit 221 based on the q-axis inductance L _q . This is because, in the variable magnetic flux motor 100, the response changes in accordance with the inductance, so that the gain used for feedback control can be determined based on the inductance.

In the d-axis gain calculator 233 and the q-axis gain calculator 234, the calculation timing generated by the calculation timing generation unit 214, that is, the initial timing, as in the case shown in FIG. 4 in the first embodiment. The gains are calculated at a plurality of timings in the path from the operating point P0 to the target operating point P ^* . By calculating the gain successively in this manner, the control accuracy of feedback control can be improved.

  Also in this embodiment, as shown in the first to third modifications of the first embodiment, feedback control can be performed more accurately by determining the timing of calculating the gain.

  According to the second embodiment, the following effects can be obtained.

According to the control method of the second embodiment, the d-axis gain calculator 233 and the q-axis gain calculator 234 calculate gains based on the inductance calculated by the inductance estimation unit 213. Here, since the response of the motor 100 changes according to the inductance, it is possible to obtain the gain even based on the inductance. Therefore, besides the embodiment in which the gain is calculated based on the d-axis current I _d and the q-axis current I _{q as} in the first embodiment, the gain is set in different embodiments using an inductance to stabilize feedback control. Since it is possible to improve the degree of freedom in design.

Third Embodiment
In the first and second embodiments, examples of measuring the inductance using the feedback input from the motor 100 have been described. In the third embodiment, an example will be described in which the inductance is measured by slightly changing the operating point to measure the inductance.

FIG. 9 is a diagram showing a control device 200 in the present embodiment. As compared with the control device 200 of the second embodiment, the control device 200 of the present embodiment is notified that the calculation timing TM of the gain is notified from the calculation timing generation unit 214 of the control device 200 to the target current calculation unit 230. It is different. When receiving the calculation timing TM from the calculation timing generation unit 214, the target current calculation unit 230 vibrates the d-axis target current I _d ^* with an amplitude of ΔI _d for a predetermined time. By doing this, the applied current to the motor 100 also vibrates.

FIG. 10 shows the operating point when the d-axis target current I _d ^* oscillates.

According to this figure, it is assumed that the lower limit of the d-axis current is _id1 and the upper limit is _id2 for an operating point that vibrates at a certain gain calculation timing. Here, when the d-axis current is _id1 , the q-axis voltage is V _q1 and the q-axis current is i _q1 . d-axis current to the V _q2, q-axis current q-axis voltage i _q2 Then when it is i _d2, the following expression holds.

However, the resistance of the motor 100 is denoted by Ra, and the differential operator is denoted by ρ. Further, the magnetic flux of the motor 100 is denoted by _aa, and the flux linkage is denoted by λ.

By using these equations, without using a map as in the first embodiment and the second embodiment, it is possible to estimate the d-axis inductance L _d and the q-axis inductance Lq.

  According to the third embodiment, the following effects can be obtained.

  According to the control method of the motor 100 of the third embodiment, the current value indicating the operating point at the calculation timing is oscillated with a predetermined amplitude. In such a case, for example, with respect to two operating points of the upper limit current of the vibration and the corresponding operating point, and the lower limit current and the corresponding operating point, current values and voltages corresponding to the respective operating points and The inductance of the motor 100 can be determined using Therefore, as in the first and second embodiments, there is no need to have the gain map 212, so the capacity of the storage memory can be reduced.

According to the control method of the motor 100 of the third embodiment, the d-axis target current I _d ^{* of the} target current is vibrated. By doing this, since the q-axis target current I _q ^* that affects the torque of the motor 100 does not change, it is possible to estimate the impedance without affecting the rotation control of the motor 100.

In the present embodiment, although the influence of the torque of the motor 100 to vibrate the less the d-axis current I _d, is not limited thereto. The q-axis current I _q may be oscillated, or the d-axis current indicating the operation point of the calculation timing in order to change both the d-axis current I _d and the q-axis current I _q as shown in FIG. And, the current may be changed to a circular locus whose radius is r around the q-axis current.

In such a case, it is possible to estimate the following equation instead of equation (1), (2) and (3) and the d-axis inductance L _d and the q-axis inductance Lq by is used. In this example, since the q-axis current also changes, the d-axis voltages V _d1 and V _d2 are further used.

(4th modification)
In the example of the third embodiment, the example in which the current indicating the operating point at the calculation timing is oscillated with a predetermined amplitude has been described. In the fourth modified example, an example in which this amplitude is changed for each operating point will be described.

FIG. 12 shows a plurality of operating points at which the d-axis target current I _d ^* oscillates.

According to this figure, the amplitude is set to increase in order of Δ _d1 , Δ _d2 and Δ _d3 as the operating points P 1, P 2 and P 3 which are the calculation timing of the gain change. The amplitude is determined at a constant rate according to the magnitude of the d-axis current at the operating point. That is, when the absolute value of the d-axis target current I _d ^* is relatively small as in the operating point P ₁ , oscillation is performed with a relatively small amplitude Δ _{d 1} . On the other hand, the absolute value of the d axis target current I _d ^* as the operating point P3 is when relatively large vibrates at a relatively large amplitude delta _d3. By doing this, vibration with an unnecessary size can be suppressed because vibration is performed with an amplitude according to the current for each operating point.

  According to the fourth modification, the following effects can be obtained.

According to the control method of motor 100 of the fourth modification, when the d-axis current increases as operating points P1, P2 and P3 change, the amplitude of vibration at each operating point is Δ _d1 and Δ _d2 , Δ _d3 and so on. In this way, since the amplitude according to each operating point can be set, the vibration with an unnecessarily large amplitude is suppressed, so the vibration for measuring the inductance controls the rotation of the motor 100. Can reduce the impact on

In this modification, although the influence of the torque of the motor 100 to vibrate the less the d-axis current I _d, is not limited thereto. The q-axis current I _q may be oscillated, or the d-axis current indicating the operation point of the calculation timing in order to change both the d-axis current I _d and the q-axis current I _q as shown in FIG. And, the current may be changed to a circular locus whose radius is r around the q-axis current. In such a case, the radius r changes according to the magnitude of the d-axis current.

  As mentioned above, although the embodiment of the present invention was described, the above-mentioned embodiment showed only a part of application example of the present invention, and in the meaning of limiting the technical scope of the present invention to the concrete composition of the above-mentioned embodiment. Absent.

100 motor 200 controller 201 d axis PI control unit 212 gain map 213 inductance estimator 214 calculation timing generation unit 221 q axis PI control unit 230 target current calculation unit

Claims (11)

A control method of a variable magnetic flux type motor capable of changing a magnetic flux from the permanent magnet to the stator according to an applied current to the motor by a magnetic barrier provided on a rotor provided with a permanent magnet. ,
The gains used for feedback control of the applied current are calculated at a plurality of calculation timings in a path from the current operating point indicated by the applied current to the target operating point indicated by the target current,
A control method of a motor, wherein the applied current is controlled by the feedback control using the gain. In the motor control method according to claim 1,
A motor control method, wherein an inductance of the motor is obtained using a current of the operating point in the path, and the gain is calculated based on the inductance of the motor. In the motor control method according to claim 2,
A motor control method, wherein the applied current is oscillated by oscillating the target current, and the inductance of the motor is calculated using a plurality of values of the oscillated applied current. In the motor control method according to claim 3,
The control method of a motor, wherein the target current to be oscillated is a d-axis current. In the motor control method according to claim 3 or 4,
A control method of a motor, wherein an amplitude of the vibration is determined according to the operating point. The motor control method according to any one of claims 1 to 5,
A control method of a motor, wherein a plurality of calculation timings are determined in accordance with an operation state of the motor. In the motor control method according to claim 6,
The motor control method, wherein an interval of the calculation timing is set to be longer as a torque change amount in the path is larger. In the motor control method according to claim 6,
A control method of a motor, wherein an interval of the calculation timing is set so that torque variation amounts become equal in the interval. The motor control method according to any one of claims 1 to 8.
The control method of the motor, wherein a plurality of the calculation timings are determined based on the current at the current operating point. The motor control method according to any one of claims 1 to 9.
At the calculation timing, an intermediate operating point between the current operating point and the operating point at the next calculation timing is generated, and the motor control method calculates the gain based on the current at the intermediate operating point. . A control device of a variable magnetic flux type motor capable of changing a magnetic flux from permanent magnets of the rotor to the stator according to an applied current to the motor by a magnetic barrier provided on the rotor. ,
A calculation timing generation unit that determines a plurality of calculation timings of gain used for feedback control of the applied current in a path from an operating point indicated by the current applied current to a target operating point indicated by the target current;
A gain calculation unit that calculates the gain at the calculation timing;
An inverter for applying a current obtained by the feedback control using the gain to the motor.

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