JP2016131414A - Switching power source - Google Patents

Abstract

PROBLEM TO BE SOLVED: To miniaturize a resonance converter while suppressing AM radio noise.SOLUTION: A switching power source has means 4 for setting a target value, means 3 for measuring a control target 2, means 5 for performing erroneous amplification on the deviation between the control target and the target value, means 6 for determining an operation amount from the amplified error, frequency variation determining means 7 for determining any one of frequency increasing, decreasing and maintaining operations according to the operation amount, frequency determining means 8 for determining the frequency upon the determination so that the frequency is a multiple of a predetermined frequency, and means 9 for generating a driving signal from the operation amount and the determined frequency, frequency modulation and other modulation methods are combined, and the frequency modulation is discretely executed.SELECTED DRAWING: Figure 1

Description

  The present invention relates to a modulation method for a switching power supply, and more particularly to a method for discretely performing frequency modulation.

  Conventionally, a method using frequency modulation is well known as a modulation method for controlling a series resonant converter. An example of a series resonant converter is shown in FIG. MOSFET 12, MOSFET 13, MOSFET 14, and MOSFET 15 are bridge-connected, a DC power source 11 is connected to the DC input, and a resonance circuit 19 including a choke 17, a choke 18, and a capacitor 16 is connected to the AC output. A primary winding of the transformer 20 is connected in parallel to the choke 18, and an AC input of a rectifier circuit in which a diode 21, a diode 22, a diode 23, and a diode 24 are bridge-connected is connected to the secondary winding of the transformer 20. . A capacitor 25 is connected to the DC output of the rectifier circuit, and a load 26 is connected to both ends thereof.

  The impedance of the resonance circuit 19 included in this converter varies depending on the frequency. Therefore, when the switching frequency of the series resonant converter is changed, the energy passing through the resonant circuit 19 can be changed. This is the reason why the series resonant converter can be controlled using frequency modulation.

  Non-Patent Document 1 clarifies how the ratio between the input voltage and the output voltage changes when the frequency of the converter is changed. The input frequency is Vin, the output voltage is Vo, the load resistance is RL, the inductance of the choke 17 is Lr, the inductance of the choke 18 is Lm, the capacitance of the capacitor 16 is Cr, the resonance frequency of the inductance Lr of the choke 17 and the capacitance Cr of the capacitor 16 Is fr, the switching frequency is f, and the transformer turns ratio is n: 1, the input / output voltage ratio nVo / Vin is expressed by the following equation.

  Using this equation, you can see how the input / output voltage ratio changes when the switching frequency is changed. FIG. 4 shows the switching frequency when the load resistance RL is 30Ω, the inductance Lr of the choke 17 is 10 μH, the inductance Lm of the choke 18 is 100 μH, the capacitance Cr of the capacitor 16 is 0.33 μF, and the transformer turns ratio n is 1. It shows the relationship of the output voltage ratio.

  In the case of a series resonant converter, a switching frequency higher than the resonant frequency is generally used to realize zero voltage switching. In the above constant, the resonance frequency of the inductance Lr of the choke 17 and the capacitance Cr of the capacitor 16 is about 87.6 kHz, and the input / output voltage ratio at that time is 1. As is apparent from FIG. 4, the input / output voltage ratio decreases as the frequency is increased in a frequency region higher than the resonance frequency. This means that if the input voltage is constant, the output voltage decreases. That is, in order to maintain a constant output voltage, the frequency is lowered when the output voltage is too low, and the frequency is raised when the output voltage is too high.

  The switching power supply has a problem of generating noise of harmonic components of the switching frequency on the principle of switching high voltage and large current at the switching frequency. In order to prevent this noise from affecting the AM radio, Patent Document 1 discloses a method of setting the switching frequency to a multiple of the frequency interval of the carrier wave.

  However, when the series resonant converter is controlled by frequency modulation, the switching frequency is continuously changed in accordance with fluctuations in the input voltage and load, so that the multiple of the frequency interval of the carrier wave is deviated. Therefore, there is a problem that the above-mentioned noise countermeasure cannot be used.

  On the other hand, there is also a method of controlling the series resonant converter using a modulation method other than frequency modulation. If the effective value of the voltage applied to the resonance circuit 19 is increased, the energy passing through the resonance circuit 19 is increased, and if the effective value of the voltage is decreased, the energy is decreased. For this purpose, a method of performing pulse width modulation or phase shift modulation on a bridge circuit including the MOSFET 12, the MOSFET 13, the MOSFET 14, and the MOSFET 15 is conceivable. Here, description will be made using phase shift modulation.

  For the sake of explanation, consider the case where a series resonant converter is applied to a battery charger. In the circuit shown in FIG. 3, the input voltage is 400V, the inductance Lr of the choke 17 is 8.5 μH, the inductance Lm of the choke 18 is 204.6 μH, the capacitance Cr of the capacitor 16 is 0.33 μF, and the turns ratio of the transformer is 110: 97. At a certain time, when the switching frequency f is 90 kHz, the phase shift amount is 0 degree, and the output voltage Vo is 350 V, 3300 W of power is output. The current waveform of the capacitor 16 at this time is shown in FIG. The effective value is 9.984 Arms.

  The battery is discharged and the output voltage drops to 300V, and then charged again. Here, the current waveform of the capacitor 16 when frequency modulation is applied is shown in FIG. The frequency was 192.1 kHz, and the current of the capacitor 16 was 11.40 Arms.

  FIG. 7 shows the current waveform of the capacitor 16 when the phase shift modulation is applied while the frequency f is not 90 kHz but the switching frequency f is 90 kHz. The current of the capacitor 16 was 14.98 Arms.

The current of the capacitor 16 not only flows through the resonance circuit 19 but also the bridge circuit including the MOSFET 12, MOSFET 13, MOSFET 14, and MOSFET 15, the bridge circuit including the transformer 20, the diode 21, the diode 22, the diode 23, and the diode 24, and the capacitor 25. Or a part flows. Therefore, if the effective value of this current is large, conduction loss increases in each component through which this current flows, and it is necessary to increase the size of the cooling component in order to improve the heat dissipation performance. Invite. If the peak value of this current is large, each component through which this current flows must use a component with a large rated current, which also leads to an increase in size.
Therefore, as apparent from the comparison between the frequency modulation of FIG. 6 and the phase shift modulation of FIG. 7, the phase shift modulation has a problem of increasing the size of the product.

  The example given here is only an output voltage fluctuation of only 50V. For example, when a 100-cell lithium ion battery is used at a discharge end of 2.5V / cell and a charge end of 4.2V / cell, the output voltage is from 250V. As much as 170V varies up to 420V. Therefore, the difference between frequency modulation and phase shift modulation is even greater.

  As described above, there is a problem that the countermeasure against noise shown in Patent Document 1 cannot be used for frequency modulation, and there is a problem that the size of the product is increased for phase shift modulation.

T. Duerbaum, “First harmonic approximation including design constraints,” in INTELEC-Twentieth International Telecommunications Energy Conference (Cat. No. 98CH36263), 1998, pp. 321-328.

JP 2003-18829 A

  The present invention solves the above problem by combining frequency modulation and other modulation methods and discretely executing frequency modulation.

  In the conventional modulation method, for example, in the case of frequency modulation, a method of setting the switching frequency to a multiple of the frequency interval of the carrier cannot be used in order to prevent the noise of the harmonic component of the switching frequency from affecting the AM radio. In the case of phase shift modulation, when the output voltage fluctuates, there is a problem that the effective value current and peak current of each part of the circuit are increased, resulting in an increase in the size of the product.

  The first switching power supply of the present invention determines a manipulated value from the amplified error, a means for setting a target value, a means for measuring a control target, a means for amplifying a deviation between the control target and the target value, and an amplified error Means, frequency variation determining means for determining one of the operations for raising / lowering / maintaining the frequency according to the operation amount, and the frequency is determined to be a multiple of a predetermined frequency in response to the determination. And means for generating a drive signal from the operation amount and the determined frequency. The means for determining the operation amount can be omitted.

  The second switching power supply according to the present invention determines the operation to increase or decrease the frequency according to the deviated direction when the manipulated variable or the amplified error value is out of the predetermined range, and manipulated quantity or amplified. It is characterized by having a frequency variation determining means for determining an operation for maintaining the frequency when the error value is within a predetermined range.

  The third switching power supply according to the present invention is characterized in that, at the upper limit or lower limit frequency, the means for generating the drive signal does not restrict the manipulated variable or the amplified error within a predetermined range.

  The fourth switching power supply of the present invention is characterized in that one of 9 kHz and 10 kHz can be selected as the predetermined frequency.

  The switching power supply of the present invention has the following effects.

First, by performing discrete frequency modulation so that the switching frequency is a multiple of the frequency interval of the AM radio carrier wave, it is possible to eliminate the influence of noise of harmonic components of the switching frequency on the AM radio receiver.
For example, in the case of a product for Japan, the switching frequency may be selected to be a multiple of 9 kHz (90 kHz, 99 kHz, 108 kHz ...), and in the case of a product for the United States, a multiple of 10 kHz (90 kHz, 100 kHz, 110 kHz ...). .

  Secondly, since another modulation such as phase shift modulation is performed from the state close to the frequency when the switching frequency is modulated, the amount of operation is small, and the effective value and peak value of the resonance current can be suppressed to a low state. it can. Therefore, an increase in the size of the product can be avoided.

FIG. 1 shows an embodiment of a modulation method according to the present invention. FIG. 2 shows another embodiment of the modulation method according to the present invention. FIG. 3 is a circuit diagram illustrating an example of a series resonant converter. FIG. 4 is a characteristic diagram showing how the input / output voltage ratio changes when the converter of FIG. 3 is frequency-modulated. FIG. 5 shows a current waveform of the capacitor 16 when the output voltage is 350 V, the output power is 3300 W, and the switching frequency is 90 kHz. FIG. 6 shows a current waveform of the capacitor 16 when the output voltage is 300 V, the output power is 3300 W, and the switching frequency is 192.1 kHz. FIG. 7 shows a current waveform of the capacitor 16 when the output voltage is 300 V, the output power is 3300 W, and the switching frequency is 90 kHz. FIG. 8 shows a current waveform of the capacitor 16 when the output voltage is 300 V, the output power is 3300 W, and the switching frequency is 100 kHz. FIG. 9 shows the current waveform of the capacitor 16 when the output voltage is 300 V, the output power is 3300 W, and the switching frequency is 110 kHz. FIG. 10 shows the current waveform of the capacitor 16 when the output voltage is 300 V, the output power is 3300 W, and the switching frequency is 120 kHz. FIG. 11 shows a current waveform of the capacitor 16 when the output voltage is 300 V, the output power is 3300 W, and the switching frequency is 130 kHz. FIG. 12 shows a current waveform of the capacitor 16 when the output voltage is 300 V, the output power is 3300 W, and the switching frequency is 140 kHz. FIG. 13 shows the current waveform of the capacitor 16 when the output voltage is 300 V, the output power is 3300 W, and the switching frequency is 150 kHz. FIG. 14 shows a current waveform of the capacitor 16 when the output voltage is 300 V, the output power is 3300 W, and the switching frequency is 160 kHz. FIG. 15 shows the current waveform of the capacitor 16 when the output voltage is 300 V, the output power is 3300 W, and the switching frequency is 170 kHz. FIG. 16 shows a current waveform of the capacitor 16 when the output voltage is 300 V, the output power is 3300 W, and the switching frequency is 180 kHz. FIG. 17 shows a current waveform of the capacitor 16 when the output voltage is 300 V, the output power is 3300 W, and the switching frequency is 190 kHz. FIG. 18 is a diagram showing temporal variation of the frequency and the phase shift amount when the load becomes heavy. FIG. 19 is a diagram showing the temporal variation of the frequency and the phase shift amount when the load is lightened. FIG. 20 is a diagram illustrating the temporal variation of the frequency and the phase shift amount when the phase shift amount for switching the frequency is changed from 180 degrees to 60 degrees.

  The mode for carrying out the present invention will be apparent from the following description of the preferred embodiments when read with reference to the accompanying drawings. However, the drawings are for explanation only and do not limit the technical scope of the present invention.

(Configuration of Example 1)
FIG. 1 is a block diagram showing a first switching power supply modulation method of the present invention. The control object 2 is measured by the measuring means 3, and the deviation from the output of the target value setting means 4 is amplified by the error amplifying means 5. An operation amount is determined by the operation amount determination means 6 from the amplified error, and the presence and direction of fluctuation is determined by the frequency variation determination means 7 in response to the result, and a drive signal is generated from the frequency and operation amount determined by the frequency determination means 8 The drive signal 10 is generated by the means 9.

(Operation of Example 1)
Although the control target 2 differs depending on the purpose of the apparatus, it is an output voltage in a general switching power supply, and an output current in the case of a battery charger.

Various examples of the measuring means 3 can be considered, but when the control target is a voltage, a voltage converting means for dividing and amplifying the voltage to convert it to an appropriate voltage level that can be handled by the error amplifying means 5 in the subsequent stage is conceivable. When the controlled object is current, current-voltage conversion means can be considered. In the case of digital control, A / D conversion means is included. When it is necessary to electrically insulate the control object 2 and the modulation means 1, an insulation means is also included.
In any case, the data of the controlled object 2 measured by the measuring means 3 is output in the form of voltage in the case of analog control and in the form of digital data in the case of digital control.

The target value set by the target value setting means 4 may be a fixed value or may change depending on the situation. For example, in the case of a fixed value in analog control, the target value setting means 4 serves as a reference voltage source. In the case of digital control, there are cases where a digital value read from the storage means is output, or a request value sent from the external device sent by the communication means is output. In addition, as an example in which the value changes depending on the situation, there is a case where the battery is charged with constant power. In this case, the output current as the target value is calculated by “output voltage ÷ battery voltage”.
In any case, the target value set by the target value setting means 4 is output in the form of voltage or the like in the case of analog control and in the form of digital data in the case of digital control.

  The error amplifying means 5 amplifies the deviation between the output of the measuring means 3 and the output of the target value setting means 4. In the case of analog control, an operation by a differential amplifier circuit may be performed, and in the case of digital control, a numerical operation using an arithmetic expression such as PI control or PID control may be executed.

The output of the error amplifying means 5 is input to the manipulated variable determining means 6. The manipulated variable that is output from the manipulated variable determining unit 6 has a value that varies depending on the modulation method used in the drive signal generating unit 9. become.
It is also possible to omit the manipulated variable determining means 6 and directly input the output of the error amplifying means 5 to the frequency fluctuation determining means 7 and the drive signal generating means 9. A block diagram in that case is shown in FIG.
Since the manipulated variable is determined from the amplified error, using the manipulated variable is equivalent to using the indirectly amplified error, and the essential operation of both is not changed. In the case of analog control, the operation amount determining means 6 is often omitted, and in the case of digital control, the calculated operation amount is output in the form of digital data.

The output of the error amplifying unit 5 or the output of the operation amount determining unit 6 is input to the frequency variation determining unit 7. The frequency variation determining means 7 determines one of the operations for increasing / decreasing / maintaining the frequency according to the manipulated variable or the value of the amplified error.
When the frequency is varied, the frequency is varied in a direction that decreases the absolute value of the error. For example, when the drive signal 10 drives the MOSFET 12, MOSFET 13, MOSFET 14, and MOSFET 15 of the series resonant converter shown in FIG. 3, the output voltage is too high because the operation is performed in a region where the output voltage decreases as the frequency is increased as described above. If the error is an error, the frequency is increased. If the output voltage is too low, the frequency is decreased.

The output of the frequency variation determining means 7 is input to the frequency determining means 8. In the case of an instruction to increase the frequency, the frequency determination means 8 adds a predetermined value to the current frequency to be the next frequency, and in the case of an instruction to decrease the frequency, the frequency determination means 8 subtracts the predetermined value from the current frequency to the next frequency. And In the case of an instruction to maintain the frequency, the current frequency is set as the next frequency.
In the case of analog control, the frequency is output in the form of a carrier triangular wave, for example, and in the case of digital control, the frequency is output in the form of digital data.

  The output of the error amplifying means 5 or the operation amount determining means 6 and the output of the frequency determining means 8 are input to the drive signal generating means 9. For example, when pulse width modulation is used in analog control, the drive signal generation means 9 is configured by a comparison circuit. In the case of digital control, digital data is given to a circuit that generates a predetermined drive signal 9 by combining a counting circuit and a comparison circuit.

(Effect of Example 1)
According to the first switching power supply modulation method of the present invention, the frequency fluctuates discretely rather than continuously. Therefore, by selecting an appropriate frequency, the harmonics of the switching frequency as shown in Patent Document 1 are used. The influence of the component noise on the AM radio receiver can be eliminated.

  Further, since another modulation such as phase shift modulation is applied from the state close to the frequency when the switching frequency is frequency-modulated, the operation amount is small, and the effective value and peak value of the resonance current can be suppressed to a low state. Therefore, an increase in the size of the product can be avoided. This will be described with reference to specific waveform examples.

In the circuit shown in FIG. 3, it is shown how the current waveform of the capacitor 16 is improved by unifying the input / output conditions to the input voltage 400V, the output voltage 300V, and the output power 3300W with the above-described circuit constants.
The effective value current in the case of frequency modulation is 11.40 Arms as shown in FIG. 6, and the switching frequency at this time is 192.1 kHz.
The effective value current when the phase shift modulation is performed at the switching frequency of 90 kHz is 14.98 Arms as shown in FIG.

  FIG. 8 shows a current waveform of the capacitor 16 when the phase shift modulation is performed with the switching frequency being 100 kHz. The rms current decreased to 14.33 Arms.

  FIG. 9 shows a current waveform of the capacitor 16 when the phase shift modulation is performed with the switching frequency being 110 kHz. The rms current decreased to 13.79 Arms.

  FIG. 10 shows a current waveform of the capacitor 16 when the phase shift modulation is performed with the switching frequency being 120 kHz. The rms current was reduced to 13.30 Arms.

  FIG. 11 shows a current waveform of the capacitor 16 when the phase shift modulation is performed with a switching frequency of 130 kHz. The rms current was reduced to 12.92 Arms.

  FIG. 12 shows a current waveform of the capacitor 16 when the phase shift modulation is performed with the switching frequency being 140 kHz. The rms current was reduced to 12.54 Arms.

  FIG. 13 shows a current waveform of the capacitor 16 when the phase shift modulation is performed with the switching frequency being 150 kHz. The rms current decreased to 12.22 Arms.

  FIG. 14 shows a current waveform of the capacitor 16 when phase shift modulation is performed with a switching frequency of 160 kHz. The rms current was reduced to 11.96 Arms.

  FIG. 15 shows a current waveform of the capacitor 16 when the phase shift modulation is performed with the switching frequency being 170 kHz. The rms current decreased to 11.69 Arms.

  FIG. 16 shows a current waveform of the capacitor 16 when phase shift modulation is performed with the switching frequency being 180 kHz. The rms current was reduced to 11.47 Arms.

  FIG. 17 shows a current waveform of the capacitor 16 when the phase shift modulation is performed with the switching frequency being 190 kHz. The rms current decreased to 11.37 Arms.

  As described above, even under the same input / output conditions, the current of the capacitor 16 when phase shift modulation is performed without changing the frequency is 14.98 Arms, whereas when discrete frequency modulation and phase shift modulation are combined, 11.37 Arms Can be reduced to. Since part or all of this current flows not only in the resonant circuit but also in other components, conduction loss of those components can be reduced, and an increase in size of the cooling component can be avoided. In addition, since the peak current is lowered, it is not necessary to increase the rated current of the component, thereby preventing an increase in size of the product.

  Here, an example in which phase shift modulation is combined is shown, but substantially the same effect can be obtained even in combination with pulse width modulation. The difference between the two is that since the time for the resonance current to return from the peak value to a value close to zero is slightly shorter in the pulse width modulation, the current waveform hardly changes.

(Configuration of Example 2)
The block diagram of the second switching power supply modulation method of the present invention is not different from the first modulation method, but the operation of the frequency variation determining means 7 is different. The frequency variation determining means 7 in the second switching power supply modulation method of the present invention increases or decreases the frequency according to the deviating direction when the manipulated variable or the amplified error value deviates from the predetermined range. The operation is determined, and the operation for maintaining the frequency is determined when the manipulated variable or the amplified error value is within a predetermined range.

(Operation of Example 2)
FIG. 18 is a diagram showing temporal variation of the frequency and the phase shift amount when the load becomes heavy.
Initially, it operates at a frequency of 130 kHz, and the amount of phase shift decreases with time to increase the output power. When the phase shift amount reaches 0 degrees, the output power cannot be increased any more, so the frequency is lowered to 120 kHz and the phase shift amount is changed to 180 degrees. Thereafter, control is performed by phase shift modulation, and when the phase shift amount reaches 0 degrees, the operation of decreasing the frequency is repeated to reach a steady state.

When the load becomes heavy, the operation of FIG. 18 is sufficient, but when the load becomes light, there is a problem that the frequency is not switched by this method. This will be described with reference to FIG.
FIG. 19 is a diagram showing temporal variation of the frequency and the phase shift amount when the load is lightened. Although operating at 90 kHz, the amount of phase shift increases with time and the output power is reduced. However, if the phase is shifted 180 degrees, no load can be accommodated, so that the steady state is reached without switching the frequency at all. This reduces the frequency but does not increase it.

This problem can be dealt with by changing the phase shift amount for switching the frequency. This will be described with reference to FIG.
FIG. 20 is a diagram illustrating the temporal variation of the frequency and the phase shift amount when the phase shift amount for switching the frequency is changed from 180 degrees to 60 degrees. Initially, it operates at 90 kHz, and the amount of phase shift increases with time to reduce output power. Since the steady state has not yet been reached when the phase shift amount reaches 60 degrees, the frequency increases to 100 kHz and the phase shift amount is returned to 0 degrees. Thereafter, the same process is repeated to reach a steady state at a frequency of 110 kHz.

(Effect of Example 2)
As described above, when the manipulated variable or amplified error value is out of the predetermined range, the operation to increase or decrease the frequency is determined according to the deviated direction, and the manipulated variable or amplified error is determined. By determining the operation of maintaining the frequency when the value of is within a predetermined range, it is possible to perform modulation by combining discrete frequency modulation and other modulation methods.

  As can be seen from FIG. 7 to FIG. 17, it is desirable that the frequency be closer to the frequency of 192.1 kHz when frequency modulation is performed in order to reduce the effective value and peak value of the current. For this purpose, the upper limit phase shift amount for switching the frequency may be further reduced from 60 degrees.

Phase shift modulation is an example of another modulation method combined with discrete frequency modulation, and can also be combined with other modulation methods such as pulse width modulation. The modulation amount in the case of pulse width modulation is the pulse width, but it is desirable to make the pulse width as large as possible in order to reduce the effective value of the current, so the lower limit pulse width should be constrained instead of the upper limit pulse width, etc. Arrangement according to the modulation method is necessary.
However, when the manipulated variable or amplified error value deviates from the predetermined range, the operation to increase or decrease the frequency is determined according to the deviated direction, and the manipulated variable or amplified error value falls within the predetermined range. The operation of determining the operation of maintaining the frequency when it is within is the same for either phase shift modulation or pulse width modulation.

(Configuration of Example 3)
The block diagram of the third switching power supply modulation method of the present invention is the same as the first modulation method, but the drive signal generation means 9 keeps the manipulated variable or amplified error within a predetermined range at the upper limit or lower limit frequency. The difference is that there are no restrictions.

(Operation of Example 3)
A case of combining with phase shift modulation will be described.
If the phase shift amount is constrained to 60 degrees instead of 180 degrees when the frequency reaches the upper limit frequency, it may not be possible to cope with no load depending on the circuit constants of FIG. In that case, the upper limit of the phase shift amount may be set to 180 degrees only when operating at the upper limit frequency.

(Effect of Example 3)
It is desirable that the frequency be closer to the frequency at the time of frequency modulation in order to reduce the effective value and peak value of the current. For this purpose, it is desirable that the upper limit phase shift amount for switching the frequency is small. However, if the upper limit phase shift amount is small, there is an increased possibility that no load can be handled when the frequency reaches the upper limit frequency.
If the upper limit of the phase shift amount is set to 180 degrees only when operating at the upper limit frequency, the upper limit phase shift amount can be reliably controlled to no load while sufficiently reducing the upper limit phase shift amount.

(Configuration of Example 4)
The block diagram of the fourth switching power supply modulation method of the present invention is the same as that of the first modulation method, except that the predetermined frequency of the frequency determining means 8 is limited to either 9 kHz or 10 kHz. .

(Operation of Example 4)
Since the predetermined frequency of the frequency determining means 8 is limited to any one of 9 kHz and 10 kHz, the determined frequency is limited to a multiple of 9 kHz or a multiple of 10 kHz.

(Effect of Example 4)
However, as shown in Patent Document 1, worldwide AM radio broadcast carriers are divided into two frequency intervals of 9 kHz or 10 kHz, so if one of these two can be selected, the whole world The purpose of preventing the noise of the harmonic component of the switching frequency from affecting the AM radio can be achieved.

  Although the series resonant converter has been described as an example in the above description, the present invention is not limited to this. For example, even if a converter using parallel resonance or a converter using a plurality of resonance circuits can be controlled by combining frequency modulation and other modulation methods, the present invention can be used to achieve the same effect. can get.

DESCRIPTION OF SYMBOLS 1 Modulation means 2 Control object 3 Measurement means 4 Target value setting means 5 Error amplification means 6 Operation amount determination means 7 Frequency fluctuation determination means 8 Frequency determination means 9 Drive signal generation means 10 Drive signal 11 DC power supply 12-15 MOSFET
16 Capacitor 17 Choke 18 Choke 19 Resonant Circuit 20 Transformer 21-24 Diode 25 Capacitor 26 Load

Claims (7)

Target value setting means for setting a target value;
A measuring means for measuring a control object;
Error amplifying means for amplifying the difference between the output value of the target value setting means and the output value of the measuring means;
A frequency variation determining means for determining any one of the operations of raising, lowering and maintaining the frequency based on the value of the error amplified by the error amplifying means;
Frequency determining means for determining a frequency so as to be a multiple of a predetermined frequency based on the determination of the frequency variation determining means;
Drive signal generation means for generating a drive signal based on the error value amplified by the error amplification means and the frequency determined by the frequency determination means;
A switching power supply characterized by having. Target value setting means for setting a target value;
A measuring means for measuring a control object;
Error amplifying means for amplifying the difference between the output value of the target value setting means and the output value of the measuring means;
An operation amount determining means for determining an operation amount based on the value of the error amplified by the error amplifying means;
A frequency variation determining means for determining any one of the operations for raising, lowering and maintaining the frequency based on the manipulated variable determined by the manipulated variable determining means;
Frequency determining means for determining a frequency so as to be a multiple of a predetermined frequency based on the determination of the frequency variation determining means;
Drive signal generation means for generating a drive signal based on the operation amount determined by the operation amount determination means and the frequency determined by the frequency determination means;
A switching power supply characterized by having. The frequency variation determining means determines an operation to increase or decrease the frequency according to the direction of the deviation when the value of the amplified error or the operation amount is out of a predetermined range, and the amplified error 3. The switching power supply according to claim 1, wherein an operation for maintaining a frequency is determined when the value or the operation amount is within a predetermined range. The drive signal generation means does not constrain the amplified error value or the operation amount within the predetermined range when the frequency determined by the frequency determination means is an upper limit frequency or a lower limit frequency. The switching power supply according to claim 1, claim 2, or claim 3. 5. The switching power supply according to claim 1, wherein the frequency determination means can select any one of 9 kHz and 10 kHz as the predetermined frequency. 6. The switching power supply according to claim 1, 2, 3, 4, or 5, wherein the driving signal generating means generates a driving signal using phase shift modulation. 6. The switching power supply according to claim 1, wherein the drive signal generating means generates a drive signal using pulse width modulation.

Images