JP2014195391A - Electric vehicle controller - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide an electric vehicle controller capable of reducing loss generated by a device driving an electric vehicle and achieving energy saving even in various driving conditions.SOLUTION: An electric vehicle controller according to an embodiment is an electric vehicle controller for PWM controlling an inverter and driving an electric motor. A control unit of the electric vehicle controller increases a carrier frequency in proportion to an increase in an output frequency of the inverter in a constant output controlled range of the electric motor.

Description

  Embodiments described herein relate generally to an electric vehicle control apparatus.

  In recent years, semiconductor devices capable of high-speed switching and semiconductor devices capable of operating at high temperatures have been developed. On the other hand, it has not been clarified how to use these devices effectively and to drive an electric vehicle applied to a railway or a new transportation system in an energy saving and stable manner.

  Under such circumstances, as an energy saving measure in electric vehicle drive systems, synchronous pulse (one pulse) control is not performed, asynchronous control is applied to the entire speed range, and the inverter output current is sine by increasing the carrier frequency. Controls approaching waves have been proposed. When the output current of the inverter is made close to a sine wave, the current flowing to the motor winding is optimized, so that it is possible to reduce the copper loss that is the loss of the motor winding.

  As a method of such a drive system, for example, there is a method of controlling the output current of the inverter with a certain high carrier frequency in the entire speed region.

JP 2008-220106 A

However, there are various driving conditions when driving an actual electric vehicle, and there is a possibility that energy saving cannot always be achieved with respect to changes in driving conditions.
Therefore, an object of the present invention is to provide an electric vehicle control device that can reduce loss generated by a device for driving an electric vehicle even under various driving conditions and can save energy.

The electric vehicle control device of the embodiment is an electric vehicle control device that drives an electric motor by PWM control of an inverter.
The control unit of the electric vehicle control device increases the carrier frequency in proportion to the increase of the output frequency of the inverter in the constant output control range of the electric motor.

FIG. 1 is a block diagram showing the overall configuration of the electric vehicle control device of the first embodiment. FIG. 2 is an explanatory diagram of the inverter output current loss ratio. FIG. 3 is an explanatory diagram of the relationship between the motor loss and the number of pulses per cycle of the inverter. FIG. 4 is an explanatory diagram of suitable control in the first embodiment. FIG. 5 is an explanatory diagram of a first modification of the control mode of the first embodiment. FIG. 6 is an explanatory diagram of a second modification of the control mode of the first embodiment. FIG. 7 is an explanatory diagram of a third modification of the control mode of the first embodiment. FIG. 8 is an explanatory diagram of a fourth modification of the control mode of the first embodiment. FIG. 9 is an explanatory diagram of a fifth modification of the control mode of the first embodiment. FIG. 10 is an operation explanatory diagram of the control unit of the electric vehicle control device of the first embodiment. FIG. 11 is an explanatory diagram for changing the carrier frequency digitally (discretely). FIG. 12 is an explanatory diagram of control in the emergency operation region. FIG. 13 is an explanatory diagram of output of torque TQ, exciting current id, and voltage v of the electric vehicle control apparatus of the first embodiment. FIG. 14 is an explanatory diagram (part 1) of output of torque TQ, exciting current id, and voltage v of the electric vehicle control apparatus of the second embodiment. FIG. 15 is an explanatory diagram of the optimum excitation current value (i1d0) of FIG. FIG. 16 is a diagram in which an iron loss curve and a copper loss + iron loss curve are added to FIG. FIG. 17 is an output explanatory diagram (part 2) of the torque TQ, the excitation current ild, and the voltage v of the electric vehicle control apparatus of the second embodiment. FIG. 18 is an explanatory diagram of a torque current command creation table of the electric vehicle control device of the second embodiment. FIG. 19 is an output explanatory diagram (part 3) of the torque TQ, the excitation current ild, and the voltage v of the electric vehicle control apparatus of the second embodiment. FIG. 20 is an explanatory diagram of the optimum excitation current value of FIG.

  Hereinafter, a control apparatus control apparatus according to an embodiment will be described with reference to the drawings.

[1] First Embodiment First, prior to the description of the first embodiment, the purpose of the first embodiment will be described.

  Asynchronous PWM (Pulse Width Modulation) control is applied to all regions using a semiconductor device capable of high-speed switching, and when the carrier frequency is constant, the inverter increases as the inverter frequency increases in the constant output region and the characteristic region. It has been found that the number of pulses per cycle of the frequency (carrier frequency / inverter frequency) decreases, and the loss reduction effect of copper loss by asynchronous PWM control decreases.

  Therefore, in the first embodiment, the electric vehicle capable of reducing the loss generated by the device for driving the electric vehicle even in the region where the frequency of the inverter is relatively high (constant output region and characteristic region). The object is to provide a control device.

Next, the first embodiment will be described in detail.
FIG. 1 is a block diagram showing the overall configuration of the electric vehicle control device of the first embodiment.
The electric vehicle control device 100 is roughly divided into an inverter 1, an induction motor 2, a current detector 3, a pulse generator (PG) 4, a filter capacitor (FC) 10, a filter reactor (FL) 11, a pantograph 12, an overhead line 13, The vehicle includes a wheel 14, a rail 15, a vehicle speed sensor 16, a carriage 17, a load detector 18, a train control device 19, and a motor control device 20.

  Here, the motor control device 20 includes an integrator 21, a PWM modulation 22, a first coordinate conversion unit (UVW / dq) 23, a coordinate conversion unit (dq / UVW) 24, a current control system 25, a current generation unit 30, and a slip. A calculation unit 27, a current command first-order lag filter 28, an adder 29, and a carrier frequency calculation unit 40 are included.

The pantograph 12 collects DC power from the overhead line 13 and is connected to one end of the filter capacitor 10 and the inverter 1 via the filter reactor 11.
The other end of the filter capacitor 10 is grounded to the rail 15 via the wheel 14 of the electric vehicle.

The DC power from the overhead wire 13 is supplied to the inverter 1 via the filter reactor 11.
The inverter 1 performs DC / AC conversion of the overhead line power by switching on / off of a built-in main circuit switching element (not shown) based on the gate command GC input from the motor control device 20 to the induction motor 2. Supply.

  The induction motor 2 receives the AC power supplied from the inverter 1, generates a magnetic field due to the three-phase AC current flowing in each excitation phase in the induction motor 2, and generates torque by magnetic interaction with the magnetic field due to the induction current of the rotor. Is driven to rotate.

  The current detector 3 has at least two phases (U-phase current, V-phase current, and W-phase current) flowing between the induction motor 2 and the inverter 1 (U-phase current and W-phase in the example of FIG. 1). Phase current).

The pulse generator 4 is installed near the induction motor 2, detects the angular velocity of the rotor of the induction motor 2, and outputs the rotor angular velocity ωm to the adder 29.
The vehicle speed sensor 16 is installed on the wheel 14 and outputs vehicle speed information vcar corresponding to the detected vehicle speed to the train control device 19.

  The load detector 18 is installed on a pillow spring that connects the vehicle body (not shown) of the carriage 17 and the carriage 17, measures the weight of the vehicle body (the weight of the vehicle body and the load of passengers, etc.), and loads the load information Mcar into the train control device. 19 output.

  As a result, the train controller 19 receives the vehicle speed information vcar from the vehicle speed sensor 16 and the load information Mcar from the load detector 18 as well as a notch command from an operation panel (not shown). The train information (control) TR such as a torque command is output to the motor control device 20 based on the above.

  When the train information TR is input from the train control device 19, the carrier frequency calculation unit 40 of the motor control device 20 generates a carrier frequency signal fc according to the input train information TR and outputs the carrier frequency signal fc to the PWM modulation unit 22. .

  The PWM modulation unit 22 generates a carrier wave (carrier wave) based on the carrier frequency signal fc input from the carrier frequency calculation unit 40. Then, the PWM modulation unit 22 compares the carrier wave with voltage commands vu *, vv *, vw * corresponding to the modulation wave, performs PWM modulation, and an inverter corresponding to each of the U phase, V phase, and W phase. 1 outputs a gate command GC for instructing on / off of the switching elements constituting 1 to the inverter 1.

Next, a method for generating voltage commands vu *, vv *, vw * corresponding to the modulated wave will be described.
When the train information TR from the train information device 19 is input, the current command generator 30 outputs a current command (d-axis current command) i1d * and a current command (q-axis current command) i1q * based on the train information TR. Generated and output to the current control unit 25.

  In parallel with the output of the current commands i1d * and i1q * to the current control unit 25, the current command generation unit 30 outputs the current command i1q * to the slip calculation unit 27, and also outputs the current command i1d * to the current command primary delay. Output to the filter 28.

Here, the reason why the current command generation unit 30 outputs the current command i1q * to the current command first-order lag filter 28 is to consider the response of the magnetic flux vector on the rotor side.
For this reason, the current command first-order lag filter 28 has a second-order time constant T2.

As a result, the current command first-order lag filter 28 generates a current command first-order lag i1qf based on the input current command i1q * and outputs it to the slip calculator 27.
The slip calculator 27 calculates a slip command value ωs * based on the current command primary delay i1qf and the current command i1q * input from the current command generator 30, and outputs the calculated slip command value ωs * to the adder 29.
As a result, the adder 29 adds the slip command value ωs * and the rotor angular velocity ωm of the induction motor 2 detected by the pulse generator 4 and outputs the result to the integrator 21.

On the other hand, the current detector 3 detects the U-phase current and the W-phase current out of the three-phase AC current flowing between the induction motor 2 and the inverter 1, and outputs the current detection values iu and iw to the first coordinate conversion unit (UVW / dq ) Is output to 23.
The first coordinate conversion unit (UVW / dq) 23 performs coordinate conversion from the three-phase fixed coordinate system to the dq-axis rotation coordinate system based on the current detection values iu and iv and the d-axis phase θ described later, and the d-axis The current value i1d and the q-axis current value i1q are calculated and output to the current control unit 25.

  The current control unit 25 compares the current detection values i1d and i1q output from the current detector 3 with the current command values i1d * and i1q * generated by the current command generation unit 30, and the voltage command values v1d *, v1q * is set and output to the second coordinate conversion unit 24.

  The addition result of the slip command value ωs * output from the adder 29 and the rotor angular velocity ωm is input to the integrator 21, and the integrator 21 integrates the addition result to generate the d-axis phase θ. The data is output to the first coordinate conversion unit 23 and the second coordinate conversion unit 24.

The voltage command values v1d * and v1q * generated by the current command generation unit 30 are output to the second coordinate conversion unit (dq / UVW) 24. Then, the second coordinate conversion unit 24 generates voltage commands vu *, vv *, vw * based on the voltage command values v1d *, v1q * and the d-axis phase θ, and outputs them to the PWM modulation unit 22.
As a result, as described above, the PWM modulation unit 22 generates the gate command GC based on the input carrier frequency signal fc and the voltage commands vu *, vv *, and vw *, and outputs them to the inverter 1. Become.

  The above is the description of the outline operation. Hereinafter, the detailed operation centering on the carrier frequency calculation unit 40 will be described.

First, the relationship between the control of the inverter 1 and the loss will be described.
FIG. 2 is an explanatory diagram of the inverter output current loss ratio.
In FIG. 2, the loss ratio 50 corresponding to the prior art is a loss when the output current of the inverter 1 is subjected to asynchronous PWM control with a constant carrier frequency / loss when 1 pulse control is performed.
The ideal loss ratio 51 is a loss when the output current of the inverter 1 is an ideal sine wave / a loss when the pulse is controlled.

  As shown in FIG. 2, in a so-called characteristic region (high-speed region) in which the speed improvement depends on the characteristics of the induction motor 2, the loss ratio 50 is almost “1”, and from the one-pulse control by performing the asynchronous PWM control. It can be seen that the loss reduction effect is reduced.

  This is because when the carrier frequency is subjected to constant out-of-sync PWM control, the number of pulses per one cycle of the inverter 1 (carrier frequency / inverter frequency) decreases as the inverter frequency increases in the high speed range.

FIG. 3 is a diagram illustrating the relationship between the switching harmonic copper loss (fundamental copper loss ratio) and the number of pulses per cycle of the inverter.
As shown in FIG. 3, the harmonic copper loss of the induction motor 2 tends to increase as the number of pulses decreases. This is because harmonics increase in the output current of the inverter 1 when the number of pulses decreases, and copper loss increases due to the harmonics.
For this reason, in the characteristic region (high speed region) where the frequency of the inverter 1 is high, increasing the carrier frequency (= fc) increases the number of pulses per cycle of the inverter, which is effective in reducing copper loss. all right.
Further, from FIG. 3, the harmonic copper loss can be reduced linearly up to the number of pulses per cycle of the inverter = 15 pulses, but the reduction effect is reduced beyond that. Therefore, the harmonic copper loss can be effectively reduced by setting the carrier frequency so that the constant output region and the characteristic region have 15 pulses or more.
If the number of pulses per cycle of the inverter exceeds 21 pulses, the harmonic copper loss cannot be reduced, and only the switching loss of the inverter increases.

  From the above, the carrier frequency is kept constant in the constant torque region, the inverter frequency is increased, and the number of pulses per cycle of the inverter is increased from 15 to 21 or the carrier frequency that coincides with 21 pulses. If synchronous PWM is performed so that the number of pulses is 21 or 21, system loss can be minimized.

FIG. 4 is an explanatory diagram of suitable control in the first embodiment.
In the present embodiment, based on the above consideration, as shown in FIG. 4, the carrier frequency is increased in a characteristic region (high speed region) where the frequency of the inverter 1 is high.

  More specifically, the increase in the output frequency of the inverter 1 in the constant output region of the induction motor 2, that is, the carrier frequency calculation unit 40 of the motor control device 20 is based on the train information TR or the output frequency of the inverter 1. The carrier frequency (fc) is increased in proportion to the increase in speed. The carrier frequency calculation unit sets the carrier frequency (fc) to a constant frequency higher than the carrier frequency in the constant torque control region at least in the characteristic region.

  By the way, in the constant output region, the current generally decreases as the torque decreases. Since the inverter loss is approximately proportional to the square of the current, the inverter loss is below the constant torque region even if the carrier frequency is increased in the constant output region where the current is decreasing.

Therefore, the carrier frequency can be increased. Here, an example is shown in which the carrier frequency is increased in accordance with the constant output region. However, if the carrier frequency is set to a higher frequency than the constant torque region in the region where the current is lower than the constant torque region, the first embodiment is performed. The same effect as the form can be obtained.
Thereby, in the first embodiment, it is possible to achieve higher efficiency in the operation of the electric vehicle.

FIG. 5 is an explanatory diagram of a first modification of the control mode of the first embodiment.
Based on the above-described consideration, in the first modification of the control mode shown in FIG. 5, asynchronous PWM control is performed with a constant carrier frequency in the constant torque region.

  Then, in the first modification of the control mode shown in FIG. 5, after the inverter frequency is increased and the number of pulses per cycle of the inverter reaches the carrier frequency (fc) that matches 15 pulses, one cycle of the inverter Synchronous PWM control is performed in which the carrier frequency (fc) is increased so that the number of pulses per pulse is maintained at 15 pulses.

FIG. 6 is an explanatory diagram of a second modification of the control mode of the first embodiment.
Also in the second modified example of the control mode shown in FIG. 6, as in the first modified example shown in FIG. 5, asynchronous PWM control is performed with a constant carrier frequency in the constant torque region.

  In the second modification of the control mode shown in FIG. 6, after the inverter frequency increases and the number of pulses per cycle of the inverter reaches the carrier frequency (fc) that matches 21 pulses, one cycle of the inverter Synchronous PWM control is performed in which the carrier frequency (fc) is increased so that the number of pulses per pulse is maintained at 21 pulses.

FIG. 7 is an explanatory diagram of a third modification of the control mode of the first embodiment.
FIG. 7 differs from the first modification of FIG. 5 in that, in synchronous PWM control, the carrier frequency is increased as a result of increasing the carrier frequency so that the number of pulses per period of the inverter is maintained at 15 pulses. This is the point where the maximum value fcmax has been reached. The maximum value fcmax is preferably set with a limit of the apparatus itself or a certain margin (for example, a margin considering safety) with respect to this limit, and is limited by heat and processing time.

  In the case of the third modification, after the carrier frequency reaches the maximum value fcmax, the number of pulses per cycle of the inverter is gradually increased from 15 pulses by maintaining the carrier frequency at the maximum value fcmax. Although it is reduced, the loss can be reduced as much as possible.

FIG. 8 is an explanatory diagram of a fourth modification of the control mode of the first embodiment.
FIG. 8 differs from the first modification of FIG. 5 in that, in the synchronous PWM control, the carrier frequency is increased as a result of increasing the carrier frequency so that the number of pulses per cycle of the inverter is maintained at 21 pulses. This is the point where the maximum value fcmax has been reached.

  Also in the case of the fourth modification, after the carrier frequency has reached the maximum value fcmax, the carrier frequency is maintained at the maximum value fcmax after the carrier frequency has reached the maximum value fcmax, as in the third modification. Although the number of pulses gradually decreases from 21 pulses, the loss can be reduced as much as possible.

FIG. 9 is an explanatory diagram of a fifth modification of the control mode of the first embodiment.
FIG. 9 differs from the first modification of FIG. 5 in that, in the synchronous PWM control, the carrier frequency is increased as a result of increasing the carrier frequency so that the number of pulses per period of the inverter is maintained at 21 pulses. If the maximum value fcmax has been reached, the carrier frequency is increased this time so that the number of pulses per period of the inverter is maintained at 15 pulses.

After the carrier frequency reaches the maximum value fcmax again, by maintaining the carrier frequency at the maximum value fcmax, the number of pulses per period of the inverter gradually decreases from 15 pulses, but the loss is small. Synchronous PWM control can be maintained in a wider area.
According to the modified examples of these control modes, it is possible to achieve higher efficiency in the operation of the electric vehicle.

Next, a description will be given on the assumption of a composition control system in which a plurality of train control devices 19 shown in FIG. 1 are connected and controlled.
Here, the formation control system is a vehicle configuration of a plurality of vehicles forming a train formation, a load state, an (on-vehicle) equipment operating state, a brake control, a torque control (or notch command, a constant speed travel command, etc.) A system that controls and manages (directive).

  Hereinafter, a case where the train control device 19 gives a torque command as the train information TR to the current command generation unit 30 and the carrier frequency calculation unit 40 at a certain current speed will be described.

FIG. 10 is an operation explanatory diagram of the control unit of the electric vehicle control device of the first embodiment.
FIG. 10A is an explanatory diagram of a state in which the current speed v and a torque command TA corresponding to the current speed are input to the motor control device 20. In FIG. 10A, A is a torque corresponding to the torque command TA, and torque B is a differential torque from the torque A to the maximum output torque. Torque A + torque B is the maximum output torque (torque maximum output curve) during normal operation (during normal operation) in the knitting control system.

  At this time, the carrier frequency calculation unit 40 calculates the carrier frequency (fc) from the ratio of the torque command to the predetermined torque performance. Instead of calculating the carrier frequency (fc), other configurations such as obtaining the carrier frequency (fc) with reference to a predetermined table may be employed.

FIG. 10B shows the ratio of the torque corresponding to the torque command to the maximum output torque.
For example, as shown in FIG. 10B, control is performed such that the carrier frequency (switching frequency) increases as A / (A + B) decreases. At this time, A / (A + B) = 1 is a normal torque output maximum value in the motor control device 20, and the inverter 1 is driven with the limit value of the cooling performance.
Therefore, when A / (A + B)> 1, the inverter is stopped.

By using such a control method, the carrier frequency can be made higher within the limit value of the cooling performance of the inverter 1.
Therefore, by increasing the carrier frequency (fc) together with the inverter frequency (speed) as shown in FIG. 4, the same control as the 15 pulses or 21 pulses in FIG. The harmonic copper loss can be reduced.

  Next, a case where the train control device 19 gives a train load (Mcar) to the current command generation unit 30 and the carrier frequency calculation unit 40 as the train information TR will be described.

The load detector 18 detects the load at a certain time or the total load.
Normally, since the torque command is set in proportion to the load, the torque command is output so that the torque increases if the load is large.

Accordingly, since the torque output maximum curve is provided according to the performance of the motor control device 20 as described above, the carrier frequency can be increased in accordance with the decrease in load (torque).
Also in this case, the copper loss of the induction motor 2 can be reduced by increasing the carrier frequency (fc) along with the increase of the inverter frequency as shown in FIG.

  As described above, according to the first embodiment, by changing the carrier frequency according to the speed (torque output) within the limit value of the output performance of the inverter, the electric vehicle can be operated in the entire speed range. It is possible to reduce the loss generated by the driving device.

[1.1] Modified Example of First Embodiment [1.1.1] First Modified Example In the above description, the carrier frequency (fc) is analog (continuously) according to the speed (inverter frequency). Although it is changed, it is also possible to configure it to change digitally (discretely) based on the torque command and the load condition.

FIG. 11 is an explanatory diagram for changing the carrier frequency digitally (discretely).
For example, as shown in FIG. 11, when the ratio of A / (A + B) exceeds the threshold value of the predetermined value D, the carrier frequency frequency is reduced by about 30%, and further when the threshold value of the next predetermined value E is exceeded. When the threshold value of the next predetermined value F is exceeded, it is set to zero (that is, the inverter is stopped). At this time, the predetermined value can be set as appropriate based on the train information TR such as a torque command and a load condition.

  Thus, if it is set as control which changes a carrier frequency (fc) discretely, control itself can be simplified and it becomes possible to provide an electric vehicle control apparatus with a high energy-saving effect easily.

[1.1.2] Second Modified Example In addition, although an example of an induction motor (IM) has been described in the first embodiment, a similar effect can be obtained with a permanent magnet synchronous motor (PMSM).

[1.1.3] Third Modified Example In the above description of the first embodiment, the case where the carrier frequency is changed according to the torque command and the load condition has been described. It is also possible to appropriately change according to the route conditions on which the train travels, the traveling curve (train type), and the knitting conditions (the number of knitting cars / MT ratio).

[1.1.4] Fourth Modification In addition, the carrier frequency calculation unit 40 illustrated in FIG. 1 has been described as being provided in the motor control device 20, but is provided outside the motor control device 20. It is also possible to configure. For example, in the train control device 19, data for changing the carrier frequency characteristic may be transmitted to the external motor control device 20, whereby the carrier frequency characteristic may be rewritten according to the aforementioned route condition, train type, composition condition, etc. .

[1.1.5] Fifth Modification Also, the carrier frequency calculation unit 40 is provided on the knitting control system side that controls the entire knitting of the vehicle, and sends a carrier frequency command to each motor control device 20 in the knitting. A configuration is also possible.

[1.1.6] Sixth Modification In the above description, control is performed so as not to exceed the maximum output torque (torque maximum output curve) during normal operation (normal operation) in the knitting control system. In the case of emergency (emergency operation) as in the case of rescue measures for trains that have failed (such as when connecting a healthy train and rescue operation of a faulty train), the maximum output torque during normal operation (normal operation) It may be necessary to control and operate beyond the (maximum torque output curve) (emergency operation region).

FIG. 12 is an explanatory diagram of control in the emergency operation region.
The emergency operation region shown in FIG. 12 is a region (mode) in which torque is output in a region exceeding the maximum torque output curve during normal operation (normal operation) and the vehicle is driven.

In the emergency operation region, the torque A exceeds the normal torque output maximum value, so that B <0 effectively.
Therefore, when the carrier frequency is changed from the ratio of the torque command TA, the ratio of the torque command TA {A / (A + B)}> 1, and in this case, the operation is performed to lower the carrier frequency (fc).
As a result, it is possible to obtain a torque A that exceeds the maximum torque output value at the normal time.

[1.1.7] Seventh Modification Further, the command transmitted from the train control device 19 to the motor control device 20 is not a torque command that is a command (continuous command) that can take a continuous value, but a notch. In the case of a discrete command such as a command (when the command value takes a discrete value), for example, when the emergency operation region is reached by some external trigger such as a high acceleration switch, the emergency operation region shown in FIG. It is also possible to control to operate in the switching frequency region.

[1.1.8] Eighth Modification Although the above description has been made on the assumption that the inverter 1 is operating normally, actually, the semiconductor device (semiconductor element: transistor) constituting the inverter 1 , Diodes, etc.) have a lifetime and lead to reduced reliability.

  Therefore, in the eighth modification, in the train control device 19, by estimating the lifetime of the semiconductor device constituting the inverter 1, the loss generated by the device that drives the electric vehicle in the entire speed range is reduced. This is a modified example for improving the reliability.

  That is, by estimating the lifetime of the semiconductor device constituting the inverter 1, the reliability is improved by performing maintenance such as replacement before the semiconductor device fails.

  In the eighth modification, the semiconductor device lifetime is estimated by integrating the heat cycle duration (duration) of the semiconductor device (thermal load information: thermal responsibility information), or the semiconductor device temperature is equal to or higher than a predetermined thermal temperature. In this case, it is performed based on the integrated value (thermal load information) of the continuation time (continuation period).

  As temperature information (heat load information) used for these estimations, the degree of temperature rise of the semiconductor device is preliminarily stored in association with the number of times of switching within a predetermined period, the duration of switching, the amount of current to the semiconductor device, etc. If stored as a table or the like and performed by referring to the stored data table, the temperature information can be easily obtained without providing a temperature measuring device.

  In this case, the heat load is calculated based on the value of the current supplied to the semiconductor device constituting the inverter 1, the carrier frequency supplied to the semiconductor device, and the loss of the semiconductor device calculated using the characteristics of the semiconductor device. It may be.

  Further, the calculation result of the thermal load calculated based on the value of the current flowing through the semiconductor device constituting the inverter 1, the carrier frequency supplied to the semiconductor device, and the loss of the semiconductor device calculated using the characteristics of the semiconductor device is obtained in advance. You may make it calculate based on the approximate expression which referred the stored table or approximated the calculation result of the thermal load.

  In these cases, the heat load may be corrected using information on element cooling conditions obtained from the traveling speed of the electric vehicle on which the electric vehicle control device is mounted.

  Further, a temperature measuring device such as a thermistor may be provided in the vicinity of the semiconductor device of the inverter 1 to detect the temperature.

  In these cases, the estimated lifetime of the semiconductor device constituting the inverter 1 is obtained by monitoring the state of the semiconductor device constituting the inverter 1 mounted on each motor control device by performing monitor output. The semiconductor device can be grasped and dealt with before it breaks down, and the reliability can be improved.

[1.1.9] Ninth Modification In addition, a torque command may be transmitted from the train control device 19 to the motor control device 20 to perform the power running operation and the regenerative operation of the drive system.

In this case, the actual power running torque and regenerative torque obtained by the current detector 3 of the induction motor 2 are fed back to the information control device 19.
Thereby, the information control device 19 manages the entire torque command based on the actual power running torque and the regenerative torque, and controls so that the acceleration / deceleration as the entire knitting does not change.

  In this case, it is conceivable that by increasing the carrier frequency, the temperature of the semiconductor device constituting the inverter 1 rises and the inverter current is controlled to be throttled.

  Therefore, the actual power running torque and the regenerative torque are transmitted for feedback to the train control device 19 that controls the entire train for feedback, and a load is applied to each motor control device 20 mounted on the same train from the overall control train control device 19. Control to distribute.

  As a result, when the carrier frequency (fc) is increased, even if the torque is reduced by any of the inverters according to the ninth modification, the target acceleration / deceleration can be maintained as a whole.

[1.1.10] Tenth Modification Also, when the device temperature of the inverter 1 rises during the regenerative operation of the inverter 1, the regenerative operation (regeneration cut) of the inverter 1 is stopped, and the inverter 1 device generates heat. It is possible to suppress it.

In such a case, conventionally, when the carrier frequency is lowered, the control performance of the induction motor 2 is deteriorated, and the DC voltage during light load regeneration may be increased.
Therefore, in the tenth modification, the switching frequency can be set high by applying a low-loss device / high-temperature operating device such as a SiC device having a high withstand voltage to the semiconductor device constituting the inverter.

Therefore, the carrier frequency is reduced before shifting to an operation mode (mode in which the inverter is stopped and the regenerative power is consumed as a mechanical brake) when the temperature of the inverter 1 rises. The inverter conversion operation can be continued without regenerative cut.
As a result, the possibility of shifting to the operation mode for cutting the regenerative operation is reduced, and energy saving can be stably maintained.

  In addition, after the estimated or measured temperature of the semiconductor device constituting the inverter 1 reaches the predetermined first threshold, the carrier frequency is decreased as the temperature of the semiconductor device increases, and the carrier frequency is further increased to a predetermined value. When the temperature of the semiconductor device to be estimated or measured rises after reaching the value, the current command at the time of regeneration may be decreased.

[1.2] Effects of First Embodiment As described above, according to the electric vehicle control device of the first embodiment, the carrier frequency is set according to the torque output within the limit value of the output performance of the inverter. By making it variable, it is possible to reduce the loss generated by the device that drives the electric vehicle in the entire speed range.

[2] Second Embodiment Next, a second embodiment will be described in detail with reference to the drawings.

The second embodiment is different from the first embodiment in the configuration of the current command generator 30 that generates the excitation current (d-axis current i1d). The second embodiment saves energy and has a response. The purpose is to ensure sex.
Hereinafter, the configuration of the current command generator 30 that generates the excitation current (d-axis current i1d) will be described in detail.

Here, in order to clarify the difference between the second embodiment and the first embodiment, the second embodiment will be described in comparison with the first embodiment.
FIG. 13 is an explanatory diagram of output of torque TQ, exciting current id, and voltage v of the electric vehicle control apparatus of the first embodiment.

In the electric vehicle control apparatus of the first embodiment, as shown in FIG. 13, the current command (i1d *) is generated so that the excitation current (d-axis current i1d) is constant up to the Z point where the voltage is limited. Yes.
This is because, in the electric vehicle control in the railway field, if the excitation current is changed by the torque command, if the excitation current changes suddenly due to the notch command, idling, etc., the magnetic flux will follow with a delay, This is because the actual torque response is delayed, which is not preferable.

  For this reason, in the electric vehicle control apparatus used for railways, the current command (i1d *) is generated so that the excitation current is constant.

  In the second embodiment, a value is selected so that the excitation current (d-axis current i1d) has a minimum loss while ensuring an actual torque response to a transient change such as a notch command peculiar to railways and idling. By doing so, high efficiency is achieved.

FIG. 14 is an explanatory diagram (part 1) of output of torque TQ, exciting current id, and voltage v of the electric vehicle control apparatus of the second embodiment.
That is, in the second embodiment, as shown in FIG. 14, the excitation current id is changed in accordance with the speed in the constant torque region (low speed region).

In this case, in order to cope with a transient change due to idling or sliding, in the second embodiment, the excitation current (d-axis current i1d) is changed based on the vehicle speed information vcar input from the train control device 19. Is adopted. According to this configuration, the change in speed is directly reflected in the excitation current. Therefore, the magnetic flux generated in response to the excitation current (d-axis current i1d) corresponds to the speed change.
For this reason, it is possible to prevent a delay in response at the time of a sudden transient change.

FIG. 15 is an explanatory diagram of the optimum excitation current value (i1d0) of FIG.
Here, a method for setting the excitation current will be described with reference to FIGS. As shown in FIG. 14, the exciting current at voltage v0 is set as exciting current i1d0. The exciting current i1d0 is set so that the copper loss is minimized. Specifically, the setting method will be described below.

  An equal torque curve TQeq in FIG. 15 is a function curve drawn by an excitation current and a torque current when the torque is constant. Further, the copper loss function corresponding to the equal torque curve in FIG. 15 is a copper loss curve. At this time, the exciting current value of the equal torque curve obtained from the same exciting current value as the minimum point of the copper loss curve is the optimum operating point (i1d0). That is, the exciting current i1d0 that minimizes the copper loss CuLoss is calculated by the following equation (1).

  Here, r1 is a stator resistance, r2 is a rotor resistance, M is a mutual inductance, and L2 is a rotor inductance parameter. As these parameters, design values may be used, or measured values may be used. If possible, it is better to use a value in consideration of magnetic saturation for the mutual inductance M.

  The exciting current i1d1 at the voltage v1 is set to an exciting current i1d that minimizes copper loss + iron loss. A specific setting method will be described below.

The copper loss CuLoss can be calculated by the expression (1) as in the case of the voltage v0.
Moreover, about iron loss CoreLoss, it can calculate by (2) Formula.

Here, rm represents the iron loss resistance.
In this case, a design value may be used as the iron loss resistance rm, or an actual measurement value may be used.

FIG. 16 is a diagram in which an iron loss curve and a copper loss + iron loss curve are added to FIG.
The exciting current i1d1 is set to i1d1 that minimizes the copper loss + iron loss curve.
The excitation current i1d between the voltage v0 and the voltage v1 is preferably set between the iron loss CoreLoss since the hysteresis loss is proportional to the speed and the Joule loss is proportional to the square of the speed.

FIG. 17 is an explanatory diagram (part 2) of the output of the torque TQ, the excitation current i1d, and the voltage v of the electric vehicle control device of the second embodiment.
For example, if the approximation is performed by a primary expression as shown in FIG. 14 or a secondary expression as shown in FIG. 17, the efficiency can be improved while the processing is simple.

FIG. 18 is an explanatory diagram of a torque current command creation table of the electric vehicle control device of the second embodiment.
For the torque current command, the current command generator 30 refers to the table TB from the excitation current command i1d *, the torque command, the vehicle speed, and the inverter DC voltage so as to satisfy the required torque as shown in FIG. The torque current command i1q * may be generated so as to satisfy the torque curve as shown in FIG. 13 or FIG. 14 regardless of the inverter DC voltage.

  Further, without using the table as shown in FIG. 18, the calculation may be performed using an approximate expression from the excitation current command i1d *, the torque command, the vehicle speed, and the inverter DC voltage.

  Thereby, it is possible to control the torque so as to satisfy a predetermined torque curve without depending on the inverter DC voltage.

The current command generation unit 30 can also be configured to input the load information Mcar as the train information TR from the train control device 19 and change the excitation current depending on the load.
FIG. 19 is an output explanatory diagram (part 3) of the torque TQ, the excitation current i1d, and the voltage v of the electric vehicle control apparatus of the second embodiment.

FIG. 20 is an explanatory diagram of the optimum excitation current value of FIG.
For example, as shown in FIG. 19, when the load is small, the exciting current i1d is reduced. Excitation current i1d that minimizes copper loss (high load = CuLossa, low load = CuLossb) is different between high load (= i1da) and light load (= i1db), as shown in FIG.

  Therefore, the efficiency is improved by increasing the excitation current i1d (= excitation current i1da) when the load is high and high (= excitation current i1da) and decreasing the excitation current i1d (= excitation current i1db) when the load is light. be able to.

  As described above, since the excitation current i1d is changed not by the torque command but by the load, the efficiency is improved while maintaining the responsiveness even when the notch is changed, the torque is narrowed down when idling, or the torque is narrowed down during light load regenerative control. Can be realized.

[2.1] Modified Example of Second Embodiment [2.1.1] First Modified Example In addition to the configuration of the second embodiment, the gradient information is obtained as train information TR from the train control device 19, and the gradient is obtained. It is also possible to exercise so as to change the excitation current according to the above.

  For example, the efficiency can be improved by increasing the excitation current because the load is high when the gradient is uphill, and by reducing the excitation current because the load is light when the gradient is downhill. At this time, the efficiency can be improved while maintaining the response of the torque by changing not by the torque command but by the gradient.

  Alternatively, a constant speed travel command is input from the train control device 19 as train information TR, and the excitation current is reduced when the constant speed travel command is issued. When a constant speed running command is issued, torque is generated so as to keep the vehicle speed constant. At this time, since the load is light, the efficiency can be improved by reducing the excitation current.

  Furthermore, during constant speed traveling, the load is lighter than when accelerating or decelerating, so the efficiency improvement effect by reducing the excitation current is great. Therefore, a large efficiency improvement effect can be easily expected by reducing the excitation current by the constant speed traveling command.

[2.2] Effects of the Second Embodiment As described above, according to the electric vehicle control device of the second embodiment, the control response speed can be changed by making the excitation current variable according to the characteristics of the vehicle. It is possible to achieve higher efficiency while securing the same level as in the past.

[3] Effects of Embodiments As described above, according to each embodiment, it is possible to reduce the loss generated by the device for driving an electric vehicle even under various driving conditions, and to save energy. It becomes.

[4] Other Aspects of Embodiments In the above description, the configuration of the first embodiment and the configuration of the second embodiment have been described as separate, but the configurations of both embodiments have been combined. A configuration is also possible.
According to this configuration, it is possible to achieve high efficiency and energy saving in the entire speed amount region while ensuring the control response speed equivalent to the conventional one.

  All the embodiments described above are presented by way of example and do not limit the scope of the invention. Therefore, the present invention can be implemented in various other forms, and various omissions, replacements, and changes can be made without departing from the spirit of the invention. These embodiments and modifications thereof are included in the scope of the invention described in the claims and equivalents thereof.

DESCRIPTION OF SYMBOLS 100 Electric vehicle control apparatus 1 Inverter 2 Induction motor 3 Current detector 4 Pulse generator (PG)
10 Filter capacitor (FC)
11 Filter reactor (FL)
12 Pantograph 13 Overhead Wire 14 Wheel 15 Rail 16 Vehicle Speed Sensor 17 Cart 18 Load Detector 19 Train Control Device 20 Motor Control Device 21 Integrator 22 PWM Modulation 23 First Coordinate Converter (UVW / dq)
24 Second coordinate converter (dq / UVW)
25 Current controller 27 Slip calculator 28 Current command first-order lag filter 29 Adder 40 Carrier frequency calculator

Claims (24)

In an electric vehicle control device that drives an electric motor by PWM control of an inverter,
An electric vehicle control device comprising a control unit that increases a carrier frequency in proportion to an increase in an output frequency of the inverter in a constant output control region of the electric motor. The control unit sets the carrier frequency to a constant frequency higher than the carrier frequency in the constant torque control region, at least in the characteristic region.
The electric vehicle control device according to claim 1. The control unit sets the increase rate of the carrier frequency based on carrier frequency-loss characteristics acquired in advance, or travel information acquired during travel of an electric vehicle,
The electric vehicle control device according to claim 1 or 2. Lowering the carrier frequency in response to an increase in the load on the electric vehicle,
The electric vehicle control device according to any one of claims 1 to 3. Increasing the carrier frequency in response to a decrease in electric vehicle load,
The electric vehicle control device according to any one of claims 1 to 4. The carrier frequency is increased when the ratio of the actual output torque with respect to the predetermined torque value set in advance or the ratio of the torque command value commanded from the external device such as the monitor device is decreased, and the carrier frequency is decreased when the ratio is increased,
Electric vehicle control device. The control unit, when a torque higher than the normal maximum output torque is set as the output torque, set to a carrier frequency lower than the carrier frequency used in the normal time,
The electric vehicle control device according to any one of claims 1 to 6. The control unit controls the excitation current to be supplied to the electric motor based on the excitation current-loss characteristic acquired in advance and the speed of the electric vehicle.
The electric vehicle control device according to any one of claims 1 to 7. The control unit controls the excitation current so that copper loss and iron loss of the electric motor are minimized.
The electric vehicle control device according to claim 8. The control unit sets an excitation current to be supplied to the electric motor according to a load of the electric vehicle.
The electric vehicle control device according to claim 8 or 9. The control unit sets a large excitation current for the electric motor when the load of the electric vehicle is large, and sets a small value when the load is small.
The electric vehicle control device according to claim 10. The controller sets the excitation current to be larger as the vehicle speed is accelerated according to the vehicle speed information, and sets the excitation current to be smaller as the vehicle speed is decelerated.
The electric vehicle control device according to any one of claims 8 to 11. The control unit corrects the torque current according to the input target torque command when setting the excitation current.
The electric vehicle control device according to any one of claims 8 to 12. In an electric vehicle control device that drives an electric motor by PWM control of an inverter,
An electric vehicle control device including a control unit that controls an excitation current to be supplied to the electric motor based on an excitation current-loss characteristic acquired in advance and a speed of the electric vehicle. The control unit controls the excitation current so that copper loss and iron loss of the electric motor are minimized.
The electric vehicle control device according to claim 14. The control unit sets an excitation current to be supplied to the electric motor according to a load of the electric vehicle.
The electric vehicle control device according to claim 14 or 15. The control unit sets a large excitation current for the electric motor when the load of the electric vehicle is large, and sets a small value when the load is small.
The electric vehicle control device according to claim 16. The controller sets the excitation current to be larger as the vehicle speed is accelerated according to the vehicle speed information, and sets the excitation current to be smaller as the vehicle speed is decelerated.
The electric vehicle control device according to any one of claims 14 to 17. The control unit corrects the torque current according to the input target torque command when setting the excitation current.
The electric vehicle control device according to any one of claims 14 to 18. The control unit calculates a thermal load of a semiconductor device that constitutes the inverter, and estimates a lifetime of the semiconductor device based on the calculated thermal load.
The electric vehicle control device according to any one of claims 1 to 19. The control unit obtains the thermal load as an integrated value of a heat cycle duration time of a semiconductor device constituting the inverter or an integrated value of a duration time when the semiconductor device temperature is equal to or higher than a predetermined thermal temperature.
The electric vehicle control device according to claim 20. The control unit calculates the thermal load based on a semiconductor device loss calculated using a value of a current flowing through a semiconductor device constituting the inverter, a carrier frequency supplied to the semiconductor device, and characteristics of the semiconductor device. To
The electric vehicle control device according to claim 20 or claim 21. The control unit is configured to calculate the thermal load calculated based on a value of a current supplied to the semiconductor device constituting the inverter, a carrier frequency supplied to the semiconductor device, and a loss of the semiconductor device calculated using characteristics of the semiconductor device. Refer to a table that stores the calculation results in advance, or calculate based on an approximate expression that approximates the calculation results of the thermal load,
The electric vehicle control device according to any one of claims 20 to 22. The control unit corrects the thermal load using information on element cooling conditions obtained from a traveling speed of a vehicle on which the electric vehicle control device is mounted.
The electric vehicle control device according to any one of claims 20 to 23.

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