JP2010206319A - Communication unit and coupler - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a communication unit where two lines are connected together by non-contact and that transmits communication signal, and to provide a coupler with the communication unit. <P>SOLUTION: The communication unit 101 includes a dielectric substrate 10 having a first principal plane (upper plane) and a second principal plane (lower plane), a ground conductor formed at substantially a whole surface of the second principal plane of the dielectric substrate 10, and a termination resistor RA connected to the outer circumference of the ground conductor and for suppressing reflection of electromagnetic waves which propagate at the dielectric substrate 10. The first principal plane of the dielectric substrate 10 is used as a combining plane for a mode of surface waves which propagate at the dielectric substrate. A center conductor 12 insulated from the ground conductor 11 is formed at an excitation area EA of the dielectric substrate 10. A conductor via 13 conducted to the center conductor 12 is formed at the interior of the dielectric substrate 10. <P>COPYRIGHT: (C)2010,JPO&INPIT

Description

  The present invention relates to a coupler coupled to each other in a proximity state and a communication body for a signal transmission device that performs communication in a proximity state.

The following prior art is listed as background art of the present invention.
[Patent Document 1]
An interface device that communicates with a sheet-like signal transmission device that propagates an electromagnetic field inside and leaches out the electromagnetic field from an opening, and is disposed in the vicinity of a region having an edge of the opening of the signal transmission device A first electrode, a second electrode disposed in the vicinity of the other region of the edge, a change in voltage or current between the first electrode and the second electrode, an electric field or a magnetic field in a certain region Interface device that communicates with the signal transmission device by the interaction between the change in the electric field and the change in the electric field or magnetic field in the other region.

[Patent Document 2]
FIG. 1 of Patent Document 2 discloses a structure for confining an electromagnetic field using air / dielectric / air interface reflection, and FIG. 2 of Patent Document 2 utilizes air / magnetic material / air interface reflection. A structure for confining an electromagnetic field is disclosed. Further, FIG. 7 of Patent Document 2 discloses a structure and operation explanation that enable planar propagation with a dielectric or magnetic plate-like structure.
[Non-Patent Document 1]
The same signal transmission device as that of Patent Document 1 is disclosed in p.

  Here, an interface device for a signal transmission device disclosed in Patent Document 1 will be described with reference to FIGS.

  1A and 1B are diagrams illustrating a schematic configuration of a signal transmission device according to Patent Document 1. FIG. 1A is a top view of a signal transmission device 501, and FIG. 1B is a cross-sectional view thereof. The signal transmission device 501 includes a mesh-like first conductor portion 111 and a flat plate-like second conductor portion 121 substantially parallel to the first conductor portion 111. A region sandwiched between the first conductor portion 111 and the second conductor portion 121 is a narrow space region 131, and a region on the upper side of the first conductor portion 111 is a leaching region 141.

  As described above, the first conductor portion 111 has a mesh-like structure and has an opening, so that the electromagnetic field leaches out to a height that is approximately the same as the mesh interval. A region where the electromagnetic wave is leached is a leaching region 141.

  FIG. 2 is an explanatory diagram showing a configuration of a directional interface device 601 according to Patent Document 1. As shown in FIG. The interface device 601 includes an inner conductor portion 602, an outer conductor portion 603, and a path conductor portion 604.

  The internal conductor portion 602 is a conductor close to the mesh-like first conductor portion 111 of the signal transmission device 501, and one end thereof is connected to the external conductor portion 603 and the other end is connected to the path conductor portion 604. The outer conductor portion 603 has a box shape and covers the inner conductor portion 602. The outer conductor portion 603 has an opening, and the path conductor portion 604 passes through the opening in a non-contact manner.

  With this configuration, when a signal transmission / reception circuit is coupled to the external conductor portion 603 and the path conductor portion 604 via a coaxial cable and the current flowing therethrough changes, electromagnetic waves are mainly emitted in the direction of the arrow in the figure.

JP 2007-150652 A JP 2008-99235 A

H. Shinoda, “Proximity Connector for Two-Dimentional Electromagnetic Wave Communication”, proceedings of the 23rd sensor symposium, pp. 397-402, 2006.

  Microwave electromagnetic waves have a wavelength in the order of several tens of centimeters in free space. The parallel plate structure used in Patent Document 1 and Non-Patent Document 1 has an advantage that the sheet thickness can be reduced with a dimension smaller than the wavelength when the electromagnetic wave is propagated in a plane (in a two-dimensional direction). When the millimeter wave band is used, the following problems occur in the devices disclosed in Patent Document 1 and Non-Patent Document 1.

(1) When the sheet thickness is made thicker than a half wavelength, a high-order mode having a wave number in the thickness direction is excited, and the single mode transmission characteristic is deteriorated. Therefore, when the frequency band is increased, the Q value in the transmission mode is deteriorated as the sheet thickness is reduced.

(2) As the frequency band becomes higher, the size of the aperture becomes smaller, and the processing accuracy of the electrode becomes severe.

(3) Since the evanescent mode induced on the surface through the aperture sharply attenuates, the allowable value of the gap between the first conductor portion 111 of the signal transmission device 501 and the interface device 601 is small when performing non-contact communication. Too much.

(4) In configuring an interface having no metal contact, it is necessary to dispose an insulating layer on a conductor layer having an opening, which causes a cost increase.

(5) The interface device is a resonator, and coupling is obtained by matching the frequency of the signal transmitted in the signal transmission device with the resonance frequency and matching the impedance. For this reason, the matching band tends to be narrowed, which is a problem when a wide band such as high-speed transmission is required.

  On the other hand, the communication device of Patent Document 2 uses a single plate of a dielectric or magnetic material as a means for propagating electromagnetic waves in a plane, and uses electromagnetic waves inside the dielectric or magnetic material using interface reflection with air. Although it confines the field, since it is an essential condition to have an air interface on both sides, the arrangement place is limited. Further, Patent Document 2 only shows a structure for inputting and outputting an electromagnetic field at the end face of the sheet, and does not disclose a structure for transmitting and receiving signals in a plane.

  SUMMARY OF THE INVENTION Accordingly, an object of the present invention is to provide a communication body that solves the above-described various problems and that can transmit a communication signal by connecting two lines in a non-contact manner and a coupler including the communication body. .

The communication body of the present invention is configured as follows.
(1) A dielectric substrate having a first main surface and a second main surface facing each other, a ground conductor formed on substantially the entire second main surface of the dielectric substrate, and an outer periphery of the ground conductor. A termination resistor that suppresses reflection of a signal propagating through the dielectric substrate, and the first principal surface of the dielectric substrate is a coupling surface for a surface wave mode propagating through the dielectric substrate.

(2) Further, the surface acoustic wave mode is excited by being disposed inside the dielectric substrate with a certain gap from the ground conductor, and by applying a potential difference to the ground conductor. An excitation conductor is provided.

(3) In addition to the dielectric substrate, the dielectric substrate has a first main surface and a second main surface facing each other, and the size of the plane substantially matches the wavelength of the surface wave mode. And an excitation conductor formed inside the dielectric substrate, and a surrounding conductor formed on the second main surface of the dielectric substrate with a certain gap around the excitation conductor. An input / output coupling unit may be provided.

(4) Further, based on the upper limit (fmax) of the operating frequency of the communication body and the relative dielectric constant (εr) of the dielectric substrate, the thickness dimension h of the dielectric substrate is a value represented by the following conditional expression: And the dielectric substrate is configured such that a single-mode surface wave mode propagates.

Conditional expression: h ≤ c0 / (2fmax * √ (εr-1)) (c0: speed of light)
(5) A part of the dielectric substrate may have a thickness portion different from the thickness dimension h.

The coupler of the present invention is configured as follows.
(6) The communication body and an input / output combination coupled to the communication body,
The input / output coupling body has a first main surface and a second main surface facing each other, and a dielectric substrate whose plane size substantially matches the wavelength of the surface wave mode, and the dielectric substrate A first excitation conductor formed inside the first excitation conductor and a second excitation conductor formed on the second main surface of the dielectric substrate with a certain gap around the first excitation conductor. ,
It arrange | positions so that the 1st main surface of the said input-output coupling body may oppose the 1st main surface of the said communication body.

According to the present invention, the following effects can be obtained.
(1) Since the surface wave mode is used as the transmission mode of the communication body, the transmission conductor loss can be reduced as compared with the case where the conventional parallel plate mode is used.

(2) When using the parallel plate mode, it is necessary to arrange a large number of openings for inputting / outputting signals on the surface. However, since the surface wave mode is used in the present invention, a ground conductor is formed on the back surface. There is no need to form an electrode pattern on the surface. Therefore, a continuous and smooth electromagnetic field distribution can be realized. In addition, since no pattern is formed, it can be manufactured at low cost.

(3) In order to finish as an interface device that does not have a metal contact in the communication body, it is necessary to form an insulating film on the uppermost layer in the parallel plate structure, but since the surface wave mode is used in this invention, the substrate is originally on the surface. It will be used in an exposed state, and it is not necessary to form a new insulating film.

(4) Increasing the size of the aperture in the parallel plate structure can increase the electromagnetic field leached from the surface, but it causes a standing wave in the substrate, and therefore ripples appear in the coupling characteristics. Since the surface wave mode is used in the present invention, the electromagnetic field leached from the surface can be increased by reducing the dielectric constant of the substrate or reducing the substrate thickness, so that standing waves are generated in the substrate. It is difficult and no ripple appears in the coupling characteristics.

(5) The surface wave mode allows the electric field energy density in the upper space of the dielectric substrate to be larger than the magnetic field energy density, and is suitable for capacitive coupling with an input / output coupling disposed close to the surface. Therefore, the conductor loss can be reduced.

(6) By providing a termination structure on the outer periphery of the dielectric substrate, standing waves in the dielectric substrate can be suppressed, and a broadband characteristic with suppressed ripples can be obtained as a coupling characteristic.

It is a figure which shows schematic structure of the signal transmission apparatus which concerns on patent document 1, FIG. 1 (A) is a top view of the signal transmission apparatus 501, and FIG. 1 (B) is the sectional drawing. 10 is an explanatory diagram showing a configuration of an interface device 601 having directivity according to Patent Document 1. FIG. FIG. 3A is a plan view of the communication body 101 according to the first embodiment, and FIG. 3B is a cross-sectional view taken along the line AA in FIG. 4A is a plan view of the communication body 102 according to the second embodiment, and FIG. 4B is a cross-sectional view taken along the line AA in FIG. 4A. FIG. 5A is a top view of the communication body 103 according to the third embodiment, and FIG. 5B is a cross-sectional view taken along the line AA in FIG. 5A. FIG. 6A is a top view of the coupler 301 according to the fourth embodiment, and FIG. 6B is a cross-sectional view taken along line AA in FIG. It is a figure which shows the structure and coordinate system which are used for the vector analysis of surface wave mode. It is a figure which shows the two functions shown to the simultaneous equations of Formula (7), and those intersections. It is a figure which shows the electromagnetic field distribution on az axis. It is a figure which shows the frequency dependence of various constants (k, (alpha), (beta)) of a surface wave mode. It is a figure which shows the frequency characteristic of the abundance ratio of the electric field energy and magnetic field energy in the area | region 2 (air area) using <design condition 1>. It is a figure which shows the frequency characteristic of the conductor Q calculated using <design condition 1>. It is a figure which shows two functions which appear in simultaneous equations (Formula (7)), and those intersections. It is a figure shown about the power density of a primary and secondary surface wave mode. It is a figure which shows the model of the electromagnetic field simulation containing the excitation structure of a surface wave mode. It is a figure which shows the electromagnetic field distribution in the simulation model of FIG. It is the figure which calculated | required the attenuation | damping characteristic (evanescent state) of the height direction in an air area | region when on the basis of the board | substrate surface. It is a figure which shows the frequency characteristic of a reflection loss. FIG. 19A is a top view of a communication body according to the sixth embodiment, and FIG. 19B is a cross-sectional view of the AA portion in FIG. 19A. It is a frequency characteristic figure of forward direction transmission coefficient S21 in an example. FIG. 21A is a diagram showing the inter-port distance dependency of the average value of the forward transmission coefficient S21 in the embodiment, and FIG. 21B is the S21 fluctuation range (dB of p ± σ shown in FIG. 21A). It is a figure which shows the distance dependency between ports of a difference. FIG. 22 is a diagram in which the horizontal axis (distance between ports (r)) in FIG. FIG. 23A is a diagram showing an average value of S21 in a band of 3 GHz centered on 60 GHz, and FIG. 23B is a diagram showing an average value of S21 in a band of 6 GHz centered on 60 GHz.

<< First Embodiment >>
FIG. 3 is a plan view of the communication body 101 according to the first embodiment, and FIG. 3B is a cross-sectional view taken along line AA in FIG. The communication body 101 includes a dielectric substrate 10 having a first main surface (upper surface) and a second main surface (lower surface), a ground conductor formed on substantially the entire second main surface of the dielectric substrate 10, A termination resistor RA which is connected to the outer periphery of the ground conductor and suppresses reflection of electromagnetic waves propagating through the dielectric substrate 10 (absorbs electromagnetic waves), and the first main surface of the dielectric substrate 10 is the dielectric substrate. This is the coupling surface for the surface wave mode propagating through the.

  The dielectric substrate 10 is made of a resin material having a relative dielectric constant of 4.0 and a thickness of 0.75 mm. The size of the ground conductor 11 on the second main surface of the dielectric substrate 10 is slightly smaller than that of the dielectric substrate 10, and a resistor having a thickness of 0.20 mm is provided on the outer periphery of the ground conductor 11 (peripheral region of the dielectric substrate 10). A (radio wave absorbing sheet material) 16 is formed, and a peripheral conductor 15 is attached to the surface thereof.

  A central conductor 12 that is insulated from the ground conductor 11 is formed in the excitation area EA of the dielectric substrate 10. Conductive vias 13 are formed in the dielectric substrate 10 so as to conduct to the central conductor 12. The conductor via 13 corresponds to an “excitation conductor” according to the present invention.

  With this structure, the surface wave mode having an axially symmetric structure centered on the conductor via 13 is excited by the conductor via 13, so that a surface wave having concentric equiphase surfaces propagates. Thereby, it can utilize as a two-dimensional communication body without an angle dependence in a coupling characteristic.

A feeding circuit is connected between the center conductor 12 and the ground conductor 11, and signals are input / output to / from the communication body 101 by this feeding circuit.
The configuration shown in FIGS. 3A and 3B provides the following operational effects.

By using a substrate having a relative dielectric constant of 4.0 and a substrate thickness of 0.75 mm, it is possible to provide a communication body (signal transmission device) in which the surface wave mode is a single transmission mode in a millimeter wave (for example, 60 GHz band).
Since the surface wave mode forms an evanescent region that decays exponentially with respect to the height direction of the air region, characteristics suitable for a non-contact two-dimensional communication body can be realized.

The surface wave mode excitation generates a potential difference between the excitation line pattern and the ground conductor by applying a millimeter wave signal voltage between the land and the ground conductor (gap) on the bottom surface of the substrate. Excited.
The excited surface wave mode propagates toward the outer periphery of the substrate, and is terminated when power is lost at the termination resistor disposed on the outer periphery.

  By terminating the surface wave at the outer periphery, a standing wave is less likely to be generated (suppressed) in the substrate, and the in-plane electromagnetic field distribution becomes flat.

  The flatness of the electromagnetic field distribution does not depend on the location of the input / output coupling device, and a frequency characteristic with a flat ripple can be realized.

<< Second Embodiment >>
FIG. 4 is a plan view of the communication body 102 according to the second embodiment, and FIG. 4B is a cross-sectional view taken along line AA in FIG. The communication body 102 includes a dielectric substrate 10 having a first main surface (upper surface) and a second main surface (lower surface), a ground conductor formed on substantially the entire second main surface of the dielectric substrate 10, A termination resistor RA which is connected to the outer periphery of the ground conductor and suppresses reflection of electromagnetic waves propagating through the dielectric substrate 10 (absorbs electromagnetic waves), and the first main surface of the dielectric substrate 10 is the dielectric substrate. This is the coupling surface for the surface wave mode propagating through the.

A difference from the communication body 101 shown in FIG. 3 is that a strip line 14 is provided on the first main surface of the dielectric substrate 10 so as to conduct to the conductor via 13.
In the communication body 101 shown in FIG. 3, the conductor vias 13 formed inside the dielectric substrate 10 are excited. In the communication body 102 shown in FIG. Exciting. Providing such a stripline 14 breaks the symmetry (axial symmetry) in the radial direction from the excitation area EA, but the mode conversion efficiency can be increased by resonating the line at a specific frequency.

<< Third Embodiment >>
FIG. 5A is a top view of the communication body 103 according to the third embodiment, and FIG. 5B is a cross-sectional view of the AA portion thereof. The communication body 103 includes a signal transmission unit 100 and an input / output coupling unit 200. In the signal transmission unit 100, a ground conductor 11 that is slightly smaller than the dielectric substrate 10 is formed on the second main surface (lower surface) of the dielectric substrate 10, and a resistor 16 and a peripheral conductor 15 are formed around the ground conductor 11.

  On the other hand, the input / output coupling unit 200 is obtained by forming various electrodes on a disk-shaped dielectric substrate 20 having a first main surface and a second main surface facing each other. A conductor via 23 is formed at the center of the dielectric substrate 20. On the second main surface (upper surface) of the dielectric substrate 20, there are formed a central conductor 22 conducting to the conductor via 23 and a peripheral conductor 21 around the central conductor 22 with a certain gap from the central conductor 22. Yes. The size of the plane of the dielectric substrate 20 of the input / output coupling unit 200 is approximately equal to the size of the wavelength of the surface wave mode.

  The input / output coupling unit 200 includes a first main surface (a surface on which no electrode is formed) of the dielectric substrate 10 of the signal transmission unit 100 and a dielectric of the input / output coupling unit 200 in the air region AA of the signal transmission unit 100. It arrange | positions so that the 1st main surface (surface in which the electrode is not formed) of the board | substrate 20 may oppose.

  With such a configuration, by applying a millimeter-wave signal (voltage) between the center conductor 22 and the surrounding conductor 21, a radial electric field is generated between the conductor via 23 and the surrounding conductor 21, and the electric field is generated. Can be coupled to the surface wave mode on the signal transmission unit 100 side to excite the surface wave mode.

  As described above, by independently configuring the signal transmission unit 100 and the input / output coupling unit 200, the input / output coupling unit 200 is coupled when the signal transmission unit 100 is brought close to the signal transmission unit 100, and is not coupled when pulled apart. Characteristics can be realized.

<< Fourth Embodiment >>
FIG. 6A is a top view of the coupler 301 according to the fourth embodiment, and FIG. 6B is a cross-sectional view taken along line AA in FIG. The coupler 301 includes a communication body 104 and an input / output coupling body 201. The configuration of the communication body 104 is basically the same as that of the communication body 101 shown in FIG. However, in this example, the dielectric substrate 10 has a rectangular parallelepiped plate shape, and the excitation area EA is disposed at a position shifted from the center of the dielectric substrate 10.

  The input / output coupler 201 has basically the same structure as the input / output coupler 200 shown in FIG. 5 in the third embodiment. That is, the input / output coupler 201 has a conductor via 23 formed at the center of a disk-shaped dielectric substrate 20 having a first main surface and a second main surface facing each other. On the main surface (upper surface), a central conductor 22 conducting to the conductor via 23 and a peripheral conductor 21 spaced from the central conductor 22 by a certain gap are formed around the central conductor 22. The size of the plane of the dielectric substrate 20 of the input / output coupler 201 substantially matches the wavelength of the surface wave mode.

  The input / output coupler 201 is formed in the air region AA of the communication body 104 on the first main surface (the surface on which no electrode is formed) of the dielectric substrate 10 of the communication body 104 and the dielectric substrate 20 of the input / output coupler 201. The first main surfaces (surfaces on which no electrodes are formed) are arranged so as to face each other.

  As described above, it is possible to transmit a signal using the surface wave mode as a medium between the input / output coupler 201 disposed in the air area AA of the communication body 104 and the excitation area EA on the dielectric substrate 10 side. That is, a coupler as a two-dimensional communication body having no metal contact can be configured.

<< Fifth Embodiment >>
In the fifth embodiment, the electromagnetic field vector of the two-dimensional signal transmission device using the surface wave mode is analytically obtained, and the design and characteristics in the millimeter wave band are specifically shown. Indicates that it can be used as a non-contact interface.

  This analysis shows that in the air region, a part of the power transmitted through the dielectric substrate is leached and becomes a region that does not radiate far away (evanescent region). Such a characteristic in the air region can be communicated when the non-contact interface comes close, and is suitable for realizing a characteristic of blocking in a remote arrangement. It also indicates that the attenuation constant and the existence ratio of electromagnetic energy in the air region can be controlled according to the dielectric constant and thickness of the substrate.

First, FIG. 7 shows a structure and a coordinate system used for vector analysis of the surface wave mode.
A region 1 in FIG. 7 is a dielectric substrate, a region 2 in contact with the dielectric substrate is air, and a ground conductor is formed on the bottom surface of the dielectric substrate. In the coordinate system, the x-axis is the propagation direction of the surface wave mode, and the z-axis is the thickness direction of the substrate. The remaining y-axis is perpendicular to the x- and z-axes, but the electromagnetic field vector can be analyzed only in the zx plane because the electromagnetic field distribution does not depend on the y-axis. In such coordinates, the electromagnetic field vector can be set as shown in Equation (1).

  When the phase constant of the surface wave mode is β, the attenuation constant in the air region is α, and the wave number in the z direction inside the substrate is k, the relational expression (dispersion relationship) that satisfies these according to the angular frequency (ω) is the region. Equations (2) and (3) are obtained for 1 and region 2, respectively.

  However, these two expressions are not independent but are expressions in each region of one dispersion relation. The electromagnetic field of Equation (1) is expressed as follows using variables that satisfy these dispersion relationships. However, the upper level represents region 1 (0 <z <h), and the lower level represents region 2 (z> h).

  The meanings of the constants in the formulas (4) to (6) are as shown below.

A: Amplitude (arbitrary constant)
k: wave number in z direction in region 1 α: attenuation constant in z direction in region 2 β: phase constant of surface wave mode (x direction)
ω: Angular frequency of surface wave mode The following simultaneous equations can be derived from the continuity condition and dispersion relation of the magnetic field in Equation (5) (Equation (2) and Equation (3)).

  However, dimensionless variables (p, q, r) were defined as follows.

  When the relative dielectric constant (εr) of the substrate is constant and the wave number (k), attenuation constant (α), and angular frequency (ω) are inversely proportional to the substrate thickness (h), the variables on the left side (x, y , r) is constant, such a combination is also a solution (surface wave mode) that satisfies the simultaneous equations.

Next, calculation examples of electromagnetic field distribution and various constants are shown.
In order to show a specific calculation example of the electromagnetic field distribution in the surface wave mode, the calculation is performed using the following <design condition 1>. As the substrate material, a low dielectric constant resin substrate having excellent low loss characteristics in the millimeter wave band was used.

<Design conditions 1>
・ Frequency: 60GHz
Relative permittivity εr: 3.5
・ Board thickness h: 0.75mm
The two functions shown in the simultaneous equations of equation (7) and their intersection are shown in FIG.
From the intersection coordinates (equation (9)) of the simultaneous equations, various constants (k, α, β) that determine the distribution of the surface wave mode are obtained as follows.

  In this way, the attenuation constant of the air region is 10.2 dB / mm, and it attenuates quickly.

  An amplitude (A) that is an undetermined coefficient is set to calculate the electromagnetic field distribution shown in equations (4) to (6). Here, the transmission power per unit width is set to 1W using the following formula.

  The electromagnetic field distribution when the amplitude (A) is set in this way is shown in FIGS. 9 (A) and 9 (B). FIG. 9C shows the power density distribution (the x component of the Poynting vector) calculated from these electromagnetic field distributions.

  Of the two electric field components (x component and z component) shown in FIG. 9A, the continuous condition at the interface is established only for the x component. The magnetic field component in FIG. 9B satisfies the continuous condition. The product of the z component of the electric field and the y component of the magnetic field is the power density shown in FIG. 9C, and the integral value in the z-axis direction becomes 1 W by normalization (formula (11)). Next, FIG. 10 shows the frequency dependence of various constants {attenuation constant (α), phase constant (β), wave number (k)} of the surface wave mode. In particular, the relationship between the frequency and the phase constant represents a dispersion relationship, which is an important characteristic for expressing the transmission characteristics of the surface wave mode.

  In this way, the attenuation constant (α) and the phase constant (β) monotonically increase, but the wave number (k) is a value obtained by dividing π / 2 by the substrate thickness (h) (2.09...) As the frequency increases. Asymptotically. From this characteristic, it can be seen that as the frequency increases, the cutoff in region 2 (air) increases and the wavelength in region 1 (dielectric) decreases. In addition, since the product of the wave number (k) in the z-axis direction of the region 1 and the substrate thickness (h) gradually approaches π / 2, it can be seen that half-wave resonance occurs in the thickness direction.

Next, the relationship between energy and the conductor Q will be described.
The energy that the surface wave mode accumulates per unit length is obtained by integration of the energy density, and is expressed as in equation (12).

  The results obtained by dividing the integration interval into region 1 (0 <z <h) and region 2 (z> h) are Equations (13) and (14).

  From the above results, it can be confirmed that the electric field energy and the magnetic field energy have the same total value in the region 1 and the region 2. That is, Expression (14) always holds.

  Using equations (13) and (14), the abundance ratio of the electric field energy and the magnetic field energy in region 2 (air region) is calculated as follows.

The frequency characteristics of the abundance ratio of electric field energy and magnetic field energy in region 2 (air region) were calculated using <Design Condition 1> (εr = 3.75, h = 0.75 mm) described above. These results are shown in FIG. (The noise seen below 20GHz is due to numerical analysis errors.)
From this result, it can be confirmed that the electric field energy ratio in the air region is always larger than the magnetic field energy ratio. If the frequency increases, the difference gradually approaches zero as the energy in the air region decreases. The energy existence ratio at 60GHz at the design center is about 28% for the electric field and about 10% for the magnetic field, and the electric field energy is about 18% larger. Such characteristics indicate that electric field coupling is more advantageous in the air region.

  Next, a calculation example of the conductor Q is shown. The conductor loss per wavelength in the surface wave mode is obtained by multiplying the surface resistance value of the ground conductor by the square of the surface magnetic field strength and integrating over one wavelength. The conductor Q is obtained by dividing the conductor loss per wavelength by the stored energy per wavelength. Equation (18) shows the calculation formula for the conductor Q.

  FIG. 12 shows the frequency characteristics of the conductor Q calculated using <Design Condition 1> (εr = 3.75, h = 0.75 mm) described above.

  When the frequency is lower than 30 GHz, the energy confinement inside the substrate is low, so that the conductor Q exceeds 10,000 and diverges toward zero Hz. As the frequency exceeds 50 GHz, the energy confinement within the substrate increases, and the conductor Q gradually approaches approximately 4,000. The conductor Q at 60GHz at the design center is about 3,800. Considering that the material Q of the substrate is on the order of about 100, it can be seen that the dominant term of the transmission loss is the material Q (dielectric loss).

Next, the conditions for single mode transmission will be described.
In order for the surface wave mode to be transmitted in a single mode, the simultaneous equations of the dimensionless variables (p, q) shown in Equation (7) do not have two or more solutions. This condition is equivalent to the condition that the radius r is less than or equal to p, as can be seen from the graph of FIG. Ie:

  From the equality condition, the cut-off frequency (fc) of the secondary mode is obtained as follows.

  Here, c0 is the speed of light. When <design condition 1> (εr = 3.75, h = 0.75 mm) shown above is used, the cutoff frequency (fc) is calculated to be 126.4 GHz. Therefore, it can be confirmed that the surface wave mode in the 60 GHz band sufficiently satisfies the conditions for single mode transmission.

  Here, based on the upper limit (fmax) of the operating frequency of the communication body and the relative dielectric constant (εr) of the dielectric substrate, the thickness dimension h of the dielectric substrate is set to a value represented by the following conditional expression: A single-mode surface wave mode propagates through the dielectric substrate.

Conditional expression: h ≤ c0 / (2fmax * √ (εr-1)) (c0: speed of light)
When the substrate is thick, the single mode is lost and a secondary mode is generated. An example of calculation when the secondary mode is generated using the following <design condition 2> is shown.

<Design condition 2>
・ Frequency: 60GHz
Relative permittivity εr: 3.5
・ Board thickness h: 2.00mm
FIG. 13 shows two functions appearing in the simultaneous equations (formula (7)) and their intersections. Here, r = π is a critical radius.

From the intersection coordinates of the simultaneous equations (Equations (21), (22)), various constants (k, α, β) of the first and second surface wave modes
Is obtained as follows.

FIG. 14 shows the power density of the primary and secondary surface wave modes.
From the distribution of FIG. 14, it can be seen that the primary mode has one antinode inside the substrate and the secondary mode has two antinodes. It can also be seen that the cutoff characteristics in the air region are steeper in the primary mode and gentler in the secondary mode. That is, assuming that excitation is performed from the air region, it can be seen that the secondary mode is more strongly coupled than the primary mode and is easily excited.
Under such conditions where the secondary mode exists, single-mode transmission is not possible, and thus it is necessary to reduce the substrate thickness.

Next, an example of electromagnetic field simulation including a surface wave mode excitation structure will be described.
Finite element analysis software was used for the simulator.
As the substrate design conditions, <design condition 1> shown above was used. Various characteristics when the surface wave mode propagates one-dimensionally under this design condition are as already described in the text.

  In the simulation, in order to realize a surface wave mode propagating concentrically in the substrate, the excitation part is composed of axisymmetric conductor vias, and ports connected to the conductor vias are arranged in the plane of the ground conductor. The inner conductor of the port matches the diameter of the conductor via, and there is ideally no land (ring-shaped pattern) on the end face of the conductor via. The outer conductor of the port is connected to the ground conductor through which the surface wave mode propagates. A simulation model is shown in FIG.

The setting conditions are as follows.
・ Dielectric substrate: 20 × 20 × t0.75mm, dielectric constant: 3.5, dielectric loss tangent: 0.01
-Conductor via diameter: φ0.5mm
・ Port inner diameter: φ0.5mm, port outer diameter: φ1.0mm (impedance: 22Ω)
・ Upper space (air): 20 × 20 × 20mm
・ Boundary condition (bottom surface): Conductor wall (5.3 × 10 ⁷ S / m)
・ Boundary conditions (top, side): Open boundary frequency 60 GHz
FIG. 16 shows the electromagnetic field distribution under these conditions. FIG. 16A shows an electric field distribution, FIG. 16B shows a magnetic field distribution, and the intensity is expressed by concentration. In FIGS. 16A and 16B, the left side is an overall cross-sectional view, and the right side is a cross-sectional view in the vicinity of a conductor via.

  As a result of the simulation, it was confirmed that the electromagnetic field propagated concentrically. Further, regarding the distribution in the cross section, it was confirmed that the electric field component was included in the cross section and the magnetic field component was perpendicular to the cross section.

  In addition, in order to confirm the attenuation characteristics (evanescent state) in the height direction in the air region, the attenuation in the height direction (dB display) with respect to the substrate surface is averaged in the section with a radius of 2 to 5 mm. Obtained by value. The result is shown in FIG. Thus, a smooth damping tendency is seen at a height of about 5 mm or less.

Finally, the frequency characteristics of reflection loss are shown in FIG.
The simulation was performed in 1GHz increments in the band 40GHz-80GHz. A reflection loss of 15 dB or more was obtained in the band from 57 GHz to 69 GHz (bandwidth 12 GHz).

  As described above, in the air region, a part of the power transmitted through the dielectric substrate is leached and becomes a region (evanescent region) that does not radiate far away. Such a characteristic in the air region can be communicated when the non-contact interface comes close, and is suitable for realizing a characteristic such as blocking in a remote arrangement. Further, the attenuation constant and the existence ratio of electromagnetic energy in the air region can be controlled according to the dielectric constant and thickness of the substrate. When a material with a relative dielectric constant of 3.5 and a thickness of 0.75 mm is used for the substrate, the attenuation constant in the air region is 10.2 dB at 60 GHz, and the abundance of energy is 28% for the electric field and 10% for the magnetic field, making it suitable for capacitive coupling. In the electromagnetic field simulation example using the vias in the excitation part, it was found that the electromagnetic field distribution propagating concentrically and the reflection loss of 15 dB or more were obtained in the 57 GHz to 69 GHz band.

<< Sixth Embodiment >>
FIG. 19A is a top view of a communication body according to the sixth embodiment, and FIG. 19B is a cross-sectional view of the AA portion in FIG. 19A. Here, the coordinate axes of x, y, and z are also shown.
In FIG. 19, the concave region is a region having a substrate thickness thinner than the peripheral substrate thickness, and the shielding effect of the surface wave mode on the air region AA is small. Therefore, the coupling is easy to obtain when the input / output coupling means is close to this region. For example, as illustrated in FIG. 5, when the communication body 103 is configured by the signal transmission unit 100 and the input / output coupling unit 200, the concave region illustrated in FIG. 19 is formed on the dielectric substrate of the signal transmission unit 100. To do. Then, the input / output coupling portion 200 may be brought close to or fitted in the recessed area.

  In FIG. 19, the convex region is a region having a substrate thickness thicker than the peripheral substrate thickness, and the shielding effect of the surface wave mode with respect to the air region AA is large. It acts as a resonator coupled to the surface wave mode due to the energy confinement effect by the cylindrical side surface. Therefore, a signal transmission system including a resonance system can be configured.

  The planar shape of the concave region and the convex region is not limited to a circle and may be a rectangle or a polygon. The cross-sectional shape may be a mortar shape or a dome shape.

  A coupler having the same basic structure as that of the coupler 301 shown in FIG. 6 and having no termination resistor RA was prototyped and its characteristics were measured. The dimensions of each part of the dielectric substrate of the communication body and the dielectric substrate of the input / output combination are as follows.

<Dielectric substrate of communication body>
・ Board dimensions: 90 × 90 × t0.75mm
・ Ground external dimensions: 70 × 70 × t0.035mm
・ Via diameter: φ0.3mm (through hole, inner wall plating)
・ Inner conductor diameter (land diameter of via connection): φ0.6mm
・ Ring gap between inner and outer conductors: 0.25mm
<Dielectric substrate for input / output coupling>
・ Via diameter: φ0.3mm (through hole, inner wall plating)
・ Inner conductor diameter: Via diameter + 0.3mm
・ Gap between inner conductor and outer conductor: 0.2mm
・ Outer conductor diameter: φ8.2mm
・ Board dimensions: 8.7 x 8.7mm (outer conductor diameter + 0.5mm square)
・ Board thickness: 0.75mm
The dielectric substrate of the input / output combination is brought into close contact with the dielectric substrate of the communication body, and the distance (r) between the ports is set to 0 mm at the position where the vias face each other. FIG. 20 shows the frequency characteristics of the forward transmission coefficient S21 when the inter-port distance (r) is set to several levels within the range of 0 to 36 mm.

  As shown in FIG. 20A, when the inter-port distance (r) is 0 mm, a maximum of −8 dBat 71 GHz is obtained, but a large attenuation pole is seen in the vicinity of 64 GHz. If it is 1 mm or more, the characteristics are relatively flat. The attenuation decreases from 1 to 2 mm, and increases from 2 to 4 mm. Since the outer dimension of the dielectric substrate of the input / output coupling body is 8.7 mm, 5 mm or less is inside the substrate and is highly likely to be affected by reflection at the end face.

  As shown in FIG. 20B, the attenuation gradually increases as the inter-port distance (r) increases from 5 mm to 9 mm. Looks relatively flat.

  As shown in FIG. 20C, the inter-port distance (r) is 10 mm, and S21 is about −25 dB. When the inter-port distance (r) exceeds 10 mm, the ripple amplitude further increases.

  As shown in FIG. 20D, when the inter-port distance (r) exceeds 20 mm, S21 is about −30 dB, and the amplitude of the ripple is further increased. The reason why the ripple amplitude becomes conspicuous is considered to be that the amplitude levels of the direct propagation and the standing wave are approximately the same.

  Next, the inter-port distance dependency of S21 was examined. Here, in order to reduce and evaluate the influence of ripple, a predetermined evaluation band was defined, and the power average and standard deviation within the band were calculated and evaluated. The evaluation band was set in the following two ways.

(I) 58.5 to 61.5 GHz (3 GHz width, specific bandwidth 5%)
(II) 57 to 63 GHz (6 GHz width, specific bandwidth 10%)
FIG. 21A shows the inter-port distance dependency of the average value of S21. The power dimension is averaged and displayed in dB. The vertical error range is obtained by calculating the standard deviation σ in the band, and displaying the range of p ± σ in dB with the average power as p.

  The distance between ports (r) is 0 mm, that is, at the position where the vias of the communication body and the input / output body face each other, S21 is the maximum, −12.4 dB in the 3 GHz band, and −13.4 dB in the 6 GHz band. Thereafter, the distance decreases as the inter-port distance (r) increases. Irregular behavior is seen when the distance (r) between the ports is 1 mm, but the range within 5 mm is inside the dielectric substrate of the input / output coupler and may be affected by reflection at the end face. high.

  FIG. 21B shows the inter-port distance dependency of the S21 fluctuation range (dB difference of p ± σ) shown in FIG. This value is an index that varies according to the ripple amplitude in the band. The inter-port distance (r) is relatively flat at 5 dB or less up to 15 mm, but if it exceeds 15 mm, the fluctuation range of S21 becomes large for both 3 GHz width and 6 GHz width, and the behavior becomes irregular.

  In order to verify that the measurement result this time is two-dimensional propagation, it is considered to evaluate the dependence of the inter-port distance by approximating it to a power function of (1 / inter-port distance (r)). In the case of ideal isotropic propagation (radiation), the signal power density is inversely proportional to the square of the inter-port distance (r) in three dimensions and inversely proportional to the square of the inter-port distance (r) in two dimensions. . Therefore, the exponent of the power function of (1 / port distance (r)) corresponds to the number of dimensions of propagation. That is, when the index is 1, it corresponds to two dimensions, and when it is 2, it corresponds to three dimensions.

  FIG. 22 shows logarithmic display of the horizontal axis (distance between ports (r)) in FIG. In such a display format, the power function is linear, and the index can be evaluated from the slope. It can be seen that both the bandwidths are 3 GHz and 6 GHz and the distance (r) between the ports is 2 mm or more and they are on the same straight line and can be approximated by a power function. The power exponent in section 1 where the inter-port distance (r) is 5 to 14 mm is 1.26 in the 3 GHz band and 1.23 in the 6 GHz band, and it is a characteristic that approximates two-dimensional propagation in consideration of loss. I understand.

  A reference line (power exponent = 1.25) is also shown in the figure, and it can be confirmed that the measurement result is almost parallel to this.

  When a simulation was performed using a coupler similar to the prototype coupler as a model, ripples were hardly seen unlike the experiment. In the simulation, it is considered that the model is covered with an absorption boundary and the influence of reflection is small. Therefore, the cause of the ripple in the experiment can be assumed to be the influence of reflection at the outer end of the dielectric substrate of the communication body.

  Next, as in the case of the experiment, evaluation was performed based on the average value of S21 in a band of 3 GHz and 6 GHz centered on 60 GHz. Comparison between the experimental results and the simulation results is shown in FIG. 23A for a bandwidth of 3 GHz and FIG. 23B for a bandwidth of 6 GHz. From these figures, it can be seen that the simulation results and the experimental results have similar attenuation tendencies. The simulation results are on a straight line in a section of 1 to 12 mm. When approximated by a power function of (1 / port distance (r)), the power exponent is 1.02 in the 3 GHz bandwidth and 6 GHz bandwidth. 1.03. These values indicate almost ideal two-dimensional propagation characteristics.

  As described above, the power exponent in the experiment is 1.25, which is about 0.2 larger than the simulation result. This indicates deterioration from ideal two-dimensional propagation, and transmission loss and radiation loss are considered as causes.

DESCRIPTION OF SYMBOLS 10 ... Dielectric substrate 11 ... Ground conductor 12 ... Center conductor 13 ... Conductor via (excitation conductor)
14 ... Stripline (Excitation conductor)
DESCRIPTION OF SYMBOLS 15 ... Peripheral conductor 16 ... Resistor 20 ... Dielectric board 21 ... Peripheral conductor 22 ... Center conductor 23 ... Conductor via (excitation conductor)
DESCRIPTION OF SYMBOLS 100 ... Signal transmission part 101-104 ... Communication body 200 ... Input / output coupling part 201 ... Input / output coupling body 301 ... Coupler AA ... Air area EA ... Excitation area RA ... Termination resistance part

Claims (6)

  A dielectric substrate having a first main surface and a second main surface facing each other; a ground conductor formed on substantially the entire second main surface of the dielectric substrate; and an outer periphery of the ground conductor, And a termination resistor that suppresses reflection of a signal propagating through the body substrate, and the first main surface of the dielectric substrate is a coupling surface for a surface wave mode that propagates through the dielectric substrate.   Provided inside the dielectric substrate is an excitation conductor that is arranged with a certain gap from the ground conductor and that excites the surface wave mode by applying a potential difference to the ground conductor. The communication body according to claim 1.   A dielectric substrate having a first main surface and a second main surface facing each other, the size of the plane of which substantially matches the wavelength of the surface wave mode; and an excitation formed inside the dielectric substrate 2. The input / output coupling portion according to claim 1, further comprising: a conductor, and a peripheral conductor formed on the second main surface of the dielectric substrate with a certain gap around the excitation conductor. Communication body. Based on the upper limit (fmax) of the operating frequency of the communication body and the relative dielectric constant (εr) of the dielectric substrate, the thickness dimension h of the dielectric substrate is set to a value represented by the following conditional expression: The communication body according to claim 1, wherein a single-mode surface wave mode propagates through the dielectric substrate.
Conditional expression: h ≤ c0 / (2fmax * √ (εr-1)) (c0: speed of light)   The communication body according to claim 4, wherein a part of the dielectric substrate is provided with a thickness portion different from the thickness dimension h. A communication body according to any one of claims 1 to 5 and an input / output coupling body coupled thereto,
The input / output coupling body has a first main surface and a second main surface facing each other, and a dielectric substrate whose plane size substantially matches the wavelength of the surface wave mode, and the dielectric substrate An excitation conductor formed inside, and a surrounding conductor formed on the second main surface of the dielectric substrate with a certain gap around the excitation conductor,
A coupler disposed such that the first main surface of the input / output coupler faces the first main surface of the communication body.

Images