JP2010130752A - Phase current estimator for motor - Google Patents

Abstract

A phase current estimation accuracy is appropriately improved.
An electric motor phase current estimating apparatus 10 applies a DC-DC converter 12 to an inverter 13 in accordance with a deviation (modulation rate deviation ΔRT) between a modulation rate RTc of the inverter 13 and a predetermined modulation rate lower limit value RTL. A DC-DC converter output voltage control unit that outputs an output voltage command Vdcc instructing the magnitude of the DC voltage (the input voltage Vdc of the inverter 13). The predetermined modulation factor lower limit value RTL is an inverter that can ensure a pulse width of a desired length (pulse width in pulse width modulation) for appropriately detecting the DC side current Idc of the inverter 13 by the DC side current sensor. 13 is a lower limit value of the modulation rate.
[Selection] Figure 1

Description

  The present invention relates to a phase current estimation device for an electric motor.

Conventionally, for example, two first and second basic voltage vector components having different 60-degree phases capable of generating a command voltage vector are output within a first period forming one PWM period (one period of a carrier wave period), and By outputting two third and fourth basic voltage vector components that are 180 degrees out of phase with respect to each of the first and second basic voltage vector components within a second period that forms one PWM cycle that is continuous with the first period. For example, an inverter device that secures a pulse width of a desired length in a state where the modulation degree is small or the phase of the output voltage vector is close to a single basic voltage vector is known (for example, Patent Document 1). reference).
Further, conventionally, when the command voltage vector is small, for example, the command voltage vector is increased to the positive side and the negative side by superimposing the pulse voltage with an average value of zero in a predetermined period on the command voltage vector, so that the sensorless A control device that calculates a rotor position angle and a rotational speed from a voltage equation of a motor in control is known (see, for example, Patent Document 2).
JP 2005-12934 A JP 09-233900 A

By the way, in the inverter device according to the above prior art, when the command voltage vector is decomposed into the vectors in the PWM periods of the two first and second periods in order to secure a desired pulse width. Due to the generated harmonic components (that is, the difference between the vector obtained by the decomposition and the command voltage vector), there arises a problem that noise and torque pulsation increase and current control stability deteriorates.
Further, in the control device according to the above prior art, there arises a problem that harmonic components are generated due to superimposition of the pulse voltage on the command voltage vector, and the stability of the current control is deteriorated.
In addition, in the inverter device and the control device according to the above prior art, when the rotational speed and torque of the motor are small, the modulation rate of the inverter decreases, and the phase current is estimated from the DC side current of the inverter, for example, during sensorless control. In this case, there arises a problem that it is difficult to secure a pulse width having a desired length required for appropriately detecting the DC side current (that is, a pulse width in pulse width modulation). And the area | region where this modulation factor is small raises the problem that it will expand with the input voltage of an inverter increasing.
The present invention has been made in view of the above circumstances, and suppresses an increase in noise and torque pulsation caused by the generation of harmonic components, while ensuring the desired stability of current control and appropriately increasing the estimation accuracy of the phase current. It is an object of the present invention to provide an electric motor phase current estimation device that can be improved to a high level.

  In order to solve the above-described problems and achieve the object, a phase current estimation apparatus for an electric motor according to a first aspect of the present invention uses a three-phase AC electric motor (for example, the motor 11 in the embodiment) by a pulse width modulation signal. ) In turn (for example, the inverter 13 in the embodiment) and pulse width modulation signal generating means (for example, the PWM signal in the embodiment) for generating the pulse width modulation signal from a carrier wave signal. Generator 25), DC power variable supply means (for example, DC-DC converter 12 in the embodiment) that variably supplies DC power to the inverter, and DC-side current sensor that detects the DC-side current of the inverter (For example, the DC current sensor 31 in the embodiment) and the phase current estimation for estimating the phase current based on the DC current detected by the DC current sensor Stage (for example, phase current estimation unit 23 in the embodiment) and applied voltage control means for controlling the magnitude of the DC voltage applied to the inverter from the DC power variable supply means according to the modulation rate of the inverter (For example, the DC-DC converter output voltage control unit 48 in the embodiment).

  Furthermore, in the motor phase current estimating apparatus according to the second aspect of the present invention, the applied voltage control means is a predetermined lower limit modulation rate that defines a modulation rate range in which the phase current can be estimated by the phase current estimating means. As the modulation factor deviation obtained by subtracting from the modulation factor of the inverter decreases, the magnitude of the DC voltage changes in a decreasing trend.

  According to the phase current estimating apparatus for an electric motor according to the first aspect of the present invention, the DC voltage of the inverter is controlled by controlling the magnitude of the DC voltage applied to the inverter from the DC power variable supply means according to the modulation rate of the inverter. When estimating the phase current from the side current, it is possible to secure a pulse width having a desired length required for appropriately detecting the DC side current (that is, the pulse width in the pulse width modulation). As a result, the operation region where generation or application of harmonic voltage is required is reduced, for example, the operation region where noise and torque pulsation increase due to the harmonic voltage or the stability of current control deteriorates. The phase current estimation accuracy can be appropriately improved.

  Furthermore, according to the phase current estimating apparatus for an electric motor according to the second aspect of the present invention, since the modulation factor of the inverter is inversely proportional to the input voltage, the magnitude of the DC voltage applied to the inverter from the DC power variable supply means is reduced. By changing to a decreasing tendency, the modulation rate of the inverter can be increased. As a result, when estimating the phase current from the DC side current of the inverter, the pulse width of a desired length required for properly detecting the DC side current (that is, the pulse width in the pulse width modulation) is secured, and the phase current The estimation accuracy can be improved appropriately.

DESCRIPTION OF EMBODIMENTS Hereinafter, an embodiment of an electric motor phase current estimating apparatus according to the present invention will be described with reference to the accompanying drawings.
An electric motor phase current estimation device 10 (hereinafter simply referred to as a phase current estimation device 10) according to this embodiment is energized by, for example, a three-phase AC brushless DC motor 11 (hereinafter simply referred to as a motor 11). The motor 11 includes a rotor (not shown) having a permanent magnet used for a field, and a stator (not shown) that generates a rotating magnetic field that rotates the rotor. It is configured with.
And the phase current estimation apparatus 10 is provided with the inverter 13 which makes the DC-DC converter 12 DC power supply, and the motor control apparatus 14, as shown, for example in FIG.

The three-phase (for example, U-phase, V-phase, and W-phase) AC motor 11 is driven by the inverter 13 in response to a control command output from the motor control device 14.
The inverter 13 includes a bridge circuit 13a formed by bridge connection using a plurality of switching elements (for example, MOSFETs: Metal Oxide Semi-conductor Field Effect Transistors) and a smoothing capacitor C, and the bridge circuit 13a performs pulse width modulation ( It is driven by the PWM signal.

  In this bridge circuit 13a, for example, a high-side and low-side U-phase transistor UH, UL paired for each phase, a high-side and low-side V-phase transistor VH, VL, a high-side and low-side W-phase transistor WH, WL is bridge-connected. The drains of the transistors UH, VH, and WH are connected to the positive terminal of the DC-DC converter 12 to form a high side arm, and the sources of the transistors UL, VL, and WL are grounded to the DC-DC converter 12. The low side arm is configured by being connected to the negative terminal. For each phase, the sources of the high-side arm transistors UH, VH, WH are connected to the drains of the low-side arm transistors UL, VL, WL, and the transistors UH, UL, VH, VL, WH, WL. Each of the diodes DUH, DUL, DVH, DVL, DWH, DWL is connected between the drain and the source so as to be in the forward direction from the source to the drain.

  The inverter 13 is, for example, a gate signal (that is, a PWM signal) that is a switching command that is output from the motor control device 14 when driving the motor 11 and is input to the gates of the transistors UH, VH, WH, UL, VL, WL. ), The DC power supplied from the DC-DC converter 12 is converted into three-phase AC power by switching the on (conductive) / off (cutoff) state of each transistor paired for each phase. By sequentially commutating energization to the stator windings of the phases, AC U-phase current Iu, V-phase current Iv and W-phase current Iw are passed through the stator windings of each phase.

  As will be described later, the motor control device 14 performs feedback control (vector control) of current on the γ-δ coordinates forming the rotation orthogonal coordinates, and calculates the command γ-axis current Iγc and the command δ-axis current Iδc. The phase voltage commands Vu, Vv, and Vw are calculated based on the command γ-axis current Iγc and the command δ-axis current Iδc, and a PWM signal that is a gate signal for the inverter 13 is calculated according to the phase voltage commands Vu, Vv, and Vw. Output. Then, the γ-axis current Iγ and the δ-axis current Iδ obtained by converting the phase currents Iu, Iv, Iw actually supplied from the inverter 13 to the motor 11 on the γ-δ coordinates, the command γ-axis current Iγc, and Control is performed so that each deviation from the command δ-axis current Iδc becomes zero.

The motor control device 14 includes, for example, a current sensor I / F (interface) 21, an overcurrent protection device 22, a phase current estimation unit 23, a control device 24, and a PWM signal generation unit 25. .
The current sensor I / F (interface) 21 is a DC-side current sensor that detects a DC-side current Idc of the bridge circuit 13 a of the inverter 13 between the bridge circuit 13 a of the inverter 13 and the negative-side terminal of the DC-DC converter 12. The detection signal output from the DC-side current sensor 31 is output to the overcurrent protection device 22 and the phase current estimation unit 23.
The direct current sensor 31 may be disposed between the bridge circuit 13a of the inverter 13 and the positive terminal of the DC-DC converter 12.
Further, it is output from the angle sensor 32 that detects the rotation angle of the rotor of the motor 11 (that is, the rotation angle of the rotor magnetic pole from the predetermined reference rotation position and the rotation position of the rotation shaft of the motor 11). The detection signal is input to the control device 24.
Note that the angle sensor 32 may be omitted, and a device for estimating the magnetic pole position of the rotor may be provided instead.

The overcurrent protection device 22 performs a predetermined overcurrent protection operation according to the DC side current Idc detected by the DC side current sensor 31.
The phase current estimation unit 23 actually uses the inverter 13 based on the DC side current Idc detected by the DC side current sensor 31 at the detection timing according to the gate signal (that is, PWM signal) output from the PWM signal generation unit 25. The phase currents Iu, Iv, and Iw supplied to the motor 11 are estimated. Details of the operation of the phase current estimation unit 23 will be described later.

The control device 24 is based on the rotation angle (rotation angle of the magnetic pole of the rotor from a predetermined reference rotation position) θe output from the angle sensor 32, so that d− of the rotation orthogonal coordinates that the actual motor 11 has. Current feedback control (vector control) is performed on this γ-δ coordinate on the γ-δ axis of the rotation orthogonal coordinate equivalent to the q-axis.
The control device 24 generates a command γ-axis current Iγc and a command δ-axis current Iδc, calculates each phase voltage command Vu, Vv, Vw based on the command γ-axis current Iγc and the command δ-axis current Iδc, and generates a PWM signal. To the unit 25.
Further, the control device 24 converts the phase currents Iu, Iv, and Iw output from the phase current estimation unit 23 onto the γδ coordinates, the γ-axis current Iγ and the δ-axis current Iδ, and the command γ-axis current Iγc. Current feedback control (vector control) is performed so that each deviation from the command δ-axis current Iδc becomes zero.
Details of the operation of the control device 24 will be described later.

  The PWM signal generation unit 25 compares each phase voltage command Vu, Vv, Vw with a carrier signal such as a triangular wave in order to pass a sinusoidal current to the three-phase stator winding, and A gate signal (that is, a PWM signal) for driving the transistors UH, VH, WH, UL, VL, WL on / off is generated. In the inverter 13, the DC power supplied from the DC-DC converter 12 is changed to the three-phase AC power by switching the on (off) / off (cut off) state of each pair of transistors for each of the three phases. By converting and sequentially energizing the stator windings of the three-phase motor 11, AC U-phase current Iu, V-phase current Iv, and W-phase current Iw are passed through each stator winding.

The gate signal input from the PWM signal generation unit 25 to the inverter 13 is, for example, according to the combination of the on / off states of the transistors UH, UL and VH, VL and WH, WL that are paired for each phase. 1 and FIGS. 2A to 2H, the PWM (pulse width) corresponding to each of the eight switching states S1 to S8 (that is, the states of the basic voltage vectors V0 to V7 having different phases by 60 degrees). Modulation) signal. In Table 1 below, the transistors that are turned on among the high-side (High) and low-side (Low) transistors are shown, and the transistors that are turned on in FIGS. Is shaded.
Then, phase currents Iu, Iv, Iw are intermittently generated on the DC side of the bridge circuit 13a of the inverter 13 according to the switching states S1 to S8, and the DC side current Idc detected by the DC side current sensor 31. Is one of the phase currents Iu, Iv, Iw, or one of the phase currents Iu, Iv, Iw inverted, or zero.

  The phase current estimation unit 23, for example, in each of the switching states S2 to S7 (that is, the states of the basic voltage vectors V1 to V6 having different phases by 60 degrees) in one period of a carrier signal such as a triangular wave. The two-phase phase currents of the three-phase phase currents are acquired from the DC-side current Idc detected by the DC-side current sensor 31 in two predetermined sets of states. Based on these two-phase phase currents, the other one-phase phase current among the three-phase phase currents is estimated. Then, each estimated value of the three-phase current obtained by estimating from the DC-side current Idc detected by the DC-side current sensor 31 is output to the control device 24.

For example, as shown in FIG. 3, in the case of three-phase modulation using a triangular carrier signal, the carrier signal with a voltage pattern symmetrical to the peak (carrier vertex) on the valley side of the triangular carrier signal is obtained. In the period of one cycle Ts, the detected value of each phase current for two phases can be acquired twice.
In other words, the phase current estimation unit 23 is symmetric with respect to the carrier vertex at times tu1 and tu2 (that is, at the time tp of the carrier vertex on the valley side) in the state of the two basic voltage vectors V1 symmetrical with respect to the carrier vertex. On the other hand, the first U-phase current Iu1 and the second U-phase current Iu2 are obtained from the DC-side current Idc detected by the DC-side current sensor 31 at the time having the same time interval T1, and further, with respect to the carrier vertex In the state of two symmetrical basic voltage vectors V3, direct current is generated at times tw1 and tw2 that are symmetrical with respect to the carrier apex (that is, the time having the same time interval T2 with respect to the time tp of the carrier apex on the valley side). The first W-phase current Iw1 and the second W-phase current Iw2 are acquired from the DC-side current Idc detected by the side current sensor 31.

Then, the phase current estimating unit 23 calculates an average value for each phase based on the phase currents Iu1, Iu2 and Iw1, Iw2, and sets each average value as a current value at the time tp of the carrier peak on the valley side. .
Then, the phase current estimation unit 23 uses the fact that the sum of the current values of the respective phase currents at the same timing is zero, so that the current values of the two-phase currents (for example, the U-phase current and the W-phase current) ( That is, the current value of the other one-phase phase current (for example, V-phase current) is calculated from the current value at the time tp at the carrier apex on the valley side.
The phase current estimation unit 23 calculates the average value based on the phase currents Iu1, Iu2 and Iw1, Iw2, and estimates the phase current of the other one phase from the two-phase phase current. However, the present invention is not limited to this. Instead, each phase current may be estimated by another estimation method.

  For example, as illustrated in FIG. 4, the control device 24 includes a speed control unit 41, a command current generation unit 42, a current control unit 43, a γδ-3 phase conversion unit 44, a three phase-γδ conversion unit 45, A rotation speed calculation unit 46, a command voltage-modulation rate conversion unit 47, and a DC-DC converter output voltage control unit 48 are provided.

The speed control unit 41 performs the closed loop control so that the deviation between the rotational speed command value ωrc input from the outside and the rotational speed ωr output from the rotational speed calculation unit 46 becomes zero, and the torque command Tc is set. Calculate. Then, a torque command Tc is output.
The control device 24 may include a torque control unit instead of the speed control unit 41, and execute torque control.

  The command current generator 42 calculates a command δ-axis current Iδc based on the torque command Tc output from the speed controller 41. Based on the command δ-axis current Iδc, the command γ-axis current Iγc is calculated, and the command δ-axis current Iδc and the command γ-axis current Iγc are output.

  The current control unit 45 calculates a deviation ΔIγ between the command γ-axis current Iγc output from the command current generation unit 42 and the γ-axis current Iγ output from the three-phase-γδ conversion unit 45, and the command current generation unit 42 A deviation ΔIδ between the output command δ-axis current Iδc and the δ-axis current Iδ output from the three-phase-γδ converter 45 is calculated. Then, for example, by PI (proportional / integral) operation, the deviation ΔIγ is controlled and amplified to calculate the γ-axis voltage command value Vγ, and the deviation ΔIδ is controlled and amplified to calculate the δ-axis voltage command value Vδ. Then, the γ-axis voltage command value Vγ and the δ-axis voltage command value Vδ are output.

The γδ-3 phase conversion unit 44 uses the rotation angle θe of the motor 11 output from the angle sensor 32, and the γ-axis voltage command value Vγ and the δ-axis voltage command value Vδ on the γ-δ coordinates are stationary coordinates. The voltage is converted into a U-phase voltage command Vu, a V-phase voltage command Vv and a W-phase voltage command Vw which are voltage command values on the three-phase AC coordinates.
The three-phase-γδ conversion unit 45 converts the estimated values of the phase currents Iu, Iv, Iw output from the phase current estimation unit 23 into γ-δ coordinates based on the rotation angle θe of the motor 11 output from the angle sensor 32. The γ-axis current Iγ and the δ-axis current Iδ are converted into the above.

  The rotation speed calculation unit 46 calculates the rotation speed ωr according to the rotation angle θe of the motor 11 output from the angle sensor 32, and outputs the rotation speed ωr.

The command voltage-modulation rate conversion unit 47 outputs each phase voltage command Vu, Vv, Vw output from the γδ-3 phase conversion unit 44 and the input voltage Vdc of the inverter 13 (that is, from the DC-DC converter 12 to the inverter 13). The modulation rate RTc of the inverter 13 is calculated based on the applied DC voltage) and the modulation rate RTc is output.
The command voltage-modulation rate converter 47 is based on the γ-axis voltage command value Vγ and the δ-axis voltage command value Vδ output from the current control unit 43 and the input voltage Vdc of the inverter 13. RTc may be calculated.

The DC-DC converter output voltage controller 48 includes a modulation factor deviation calculator 51 and an output voltage command generator 52, as shown in FIG.
The modulation factor deviation calculating unit 51 outputs a value obtained by subtracting a predetermined modulation factor lower limit value RTL from the modulation factor RTc output from the command voltage-modulation factor converter 47 as a modulation factor deviation ΔRT.
The predetermined modulation factor lower limit value RTL ensures a pulse width having a desired length for appropriately detecting the DC-side current Idc of the inverter 13 by the DC-side current sensor 31 (that is, a pulse width in pulse width modulation). This is the lower limit value of the modulation rate of the inverter 13.

Based on the modulation factor deviation ΔRT output from the modulation factor deviation calculating unit 51, the output voltage command generation unit 52 performs a command value (output) for the output voltage of the DC-DC converter 12 by a map search for a predetermined map set in advance. Voltage command Vdcc) is calculated. Then, an output voltage command Vdcc is output.
In this predetermined map, for example, the output voltage command Vdcc changes stepwise with hysteresis with respect to a change in the modulation factor deviation ΔRT. For example, when the modulation factor deviation ΔRT increases to a predetermined value, the output voltage command Vdcc Is set to a predetermined lower limit value, and when the modulation factor deviation ΔRT increases beyond the predetermined value, the output voltage command Vdcc is set to a predetermined upper limit value, and the modulation factor deviation ΔRT decreases to zero. The output voltage command Vdcc is set to a predetermined upper limit value, and when the modulation factor deviation ΔRT decreases to less than zero, the output voltage command Vdcc is set to a predetermined lower limit value.

As described above, according to the motor phase current estimation apparatus 10 according to the present embodiment, the DC voltage applied from the DC-DC converter 12 to the inverter 13 in accordance with the modulation rate RTc of the inverter 13 (that is, the input voltage of the inverter 13). The magnitude of Vdc) is controlled. That is, since the modulation rate RTc of the inverter 13 is inversely proportional to the input voltage Vdc, the modulation rate RTc of the inverter 13 can be increased by changing the magnitude of the input voltage Vdc of the inverter 13 in a decreasing tendency. Therefore, as the modulation factor deviation ΔRT decreases, the modulation factor RTc of the inverter 13 decreases below the predetermined modulation factor lower limit value RTL by changing the magnitude of the input voltage Vdc in a decreasing tendency. Can be prevented.
As a result, a pulse width of a desired length required for appropriately detecting the DC side current Idc when estimating the phase current from the DC side current Idc of the inverter 13 (that is, a pulse width in pulse width modulation) is ensured. Can do. For example, even when the rotational speed of the motor 11 is low, the operation region where generation or application of the harmonic voltage is required is reduced, and noise and torque pulsation increase due to the harmonic voltage, for example. In addition, it is possible to reduce the operation region where the stability of the current control is deteriorated and appropriately improve the estimation accuracy of the phase current.

  In the above-described embodiment, the phase current estimation unit 23 uses the detected values of the respective phase currents for two phases in the period of one cycle Ts of the carrier signal for three-phase modulation using a triangular wave carrier signal. An average value is calculated for each phase, and each average value is defined as a current value at the time Ts / 2 at the peak of the carrier on the valley side. However, the present invention is not limited to this. The average value may be calculated for each phase from the detected values of the phase currents for two phases in the period of one cycle Ts, and each average value may be used as the current value at the time Ts / 2 of the carrier peak on the valley side. The phase current estimation unit 23 calculates the average value for each phase from the detected values of the phase currents for two phases, and estimates the phase current of the other one phase from the phase currents of the two phases. However, the present invention is not limited to this, and each phase current may be estimated by another estimation method.

  In the above-described embodiment, the output voltage command generation unit 52 of the DC-DC converter output voltage control unit 48 outputs the output voltage command Vdcc that changes stepwise with respect to the change in the modulation factor deviation ΔRT. For example, as shown in FIG. 6, an output voltage command Vdcc that changes smoothly with respect to the change in the modulation factor deviation ΔRT may be output. In this case, the output voltage command generator 52 changes the output voltage command Vdcc from a predetermined lower limit value to a predetermined upper limit value, for example, as the modulation factor deviation ΔRT increases from zero to a predetermined value, and When the modulation rate deviation ΔRT is less than zero, the output voltage command Vdcc is set to a predetermined lower limit value, and when the modulation rate deviation ΔRT is greater than the predetermined value, the output voltage command Vdcc is set to a predetermined upper limit value. Command Vdcc is calculated.

In the above-described embodiment, the DC-DC converter output voltage control unit 48 includes the output voltage command generation unit 52. However, the present invention is not limited to this. For example, as shown in FIG. Instead of 52, a PI control unit 53 and a limit processing unit 54 may be provided. In this case, the PI control unit 53 performs PI (proportional / integral) control, for example, and outputs an output voltage command Vdcc * obtained by controlling and amplifying the modulation factor deviation ΔRT. Thus, the output voltage command Vdcc * increases as the modulation rate RTc of the inverter 13 becomes larger than the predetermined modulation rate lower limit value RTL.
The limit processing unit 54 calculates the output voltage command Vdcc by, for example, a map search for a predetermined map indicating the correspondence between the output voltage command Vdcc * and the output voltage command Vdcc, and outputs the output voltage command Vdcc.
In the predetermined map showing the correspondence between the output voltage command Vdcc * and the output voltage command Vdcc, for example, as shown in FIG. 7, the output voltage command Vdcc * is output voltage with respect to a range from zero to the first predetermined value. As the output voltage command Vdcc * increases from the first predetermined value to the second predetermined value (> first predetermined value) so that the command Vdcc becomes a certain predetermined lower limit value, the output voltage command Vdcc becomes The output voltage command Vdcc becomes a constant predetermined upper limit value so that the output voltage command Vdcc * changes in an increasing trend from the predetermined lower limit value to the upper limit value and the output voltage command Vdcc * is larger than the second predetermined value. Is set.
In this case, the PI control unit 53 may be omitted, and the output voltage command Vdcc may be calculated by a method other than PI (proportional / integral) control.

  In the above-described embodiment, the DC power source of the inverter 13 is the DC-DC converter 12, but the present invention is not limited to this, and other power sources that can change the input voltage Vdc of the inverter 13, such as an AC-DC converter. It may be.

It is a block diagram of the phase current estimation apparatus of the electric motor which concerns on embodiment of this invention, and the magnetic pole position estimation apparatus of an electric motor. It is a figure which shows each switching state S1-S8 of the inverter shown in FIG. It is a figure which shows the example of the detection timing of the on-off pattern and each phase current of the carrier wave and each transistor UH, UL and VH, VL and WH, WL which concern on embodiment of this invention. It is a block diagram of the phase current estimation apparatus of the electric motor which concerns on embodiment of this invention. It is a block diagram of the DC-DC converter output voltage control part shown in FIG. It is a block diagram of the DC-DC converter output voltage control part which concerns on the 1st modification of embodiment of this invention. It is a block diagram of the DC-DC converter output voltage control part which concerns on the 2nd modification of embodiment of this invention.

Explanation of symbols

DESCRIPTION OF SYMBOLS 10 Electric motor phase current estimation apparatus 11 Motor 13 Inverter 23 Phase current estimation part (phase current estimation means)
24 Control device (control means)
25 PWM signal generator (pulse width modulation signal generator)
31 DC-side current sensor 48 DC-DC converter output voltage control unit (applied voltage control means)

Claims (2)

An inverter that sequentially commutates energization of a three-phase AC motor using a pulse width modulation signal, pulse width modulation signal generation means that generates the pulse width modulation signal using a carrier wave signal, and variably supplies DC power to the inverter DC power variable supply means;
A DC current sensor for detecting a DC current of the inverter; a phase current estimating means for estimating a phase current based on the DC current detected by the DC current sensor;
An apparatus for estimating a phase current of an electric motor, comprising: applied voltage control means for controlling the magnitude of a DC voltage applied to the inverter from the DC power variable supply means in accordance with a modulation rate of the inverter. The applied voltage control unit reduces a modulation rate deviation obtained by subtracting a predetermined lower limit modulation rate defining a modulation rate range in which the phase current can be estimated by the phase current estimation unit from the modulation rate of the inverter. Accordingly, the phase current estimating device for an electric motor according to claim 1, wherein the magnitude of the DC voltage is changed in a decreasing tendency.

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