JP2009182928A - Tuner - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a tuner capable of canceling noise components by a third harmonic among higher harmonic components of a local oscillation signal in the tuner which converts a signal of a carrier band into a base band signal or an intermediate frequency signal using a local oscillator. <P>SOLUTION: The noise components by the third harmonic are canceled among the higher harmonic components of the local oscillation signal by multiplying a three-phase signal having phase differences which are different by 120°, respectively, by a received signal as the local oscillation signal, and canceling common-mode signals of each demodulation wave. <P>COPYRIGHT: (C)2009,JPO&INPIT

Description

  The present invention relates to an image receiver or an audio receiver for digital broadcasting. More specifically, the present invention relates to a tuner having a function of removing an interference received signal due to harmonics of a local oscillation signal, a receiver circuit including peripheral circuits thereof, a television receiver and an audio receiver including them, and a method for manufacturing the tuner It is.

  Digital broadcasting, which has been spreading in recent years, has the advantage of being able to receive high-quality and multi-channel services, and is expected to improve reception performance on mobile devices by taking advantage of the characteristics of digital signal processing. Along with this, it is expected that receivers (hereinafter referred to as “digital broadcast receivers”) that receive digital television broadcasts and digital radio broadcasts in portable receiver terminals such as mobile phones and moving automobiles are expected.

  In such a receiver, a desired signal transmitted in a high frequency band is converted into a baseband or an intermediate frequency by using a local oscillator and a mixer.

  In this frequency conversion, since the local oscillation signal is not accurately multiplied in the mixer, a signal having a frequency corresponding to a harmonic of the local oscillation signal is also multiplied with the input signal. Therefore, the mixer output signal includes an interference wave having a frequency corresponding to the harmonic frequency of the local oscillation signal.

  In particular, when the desired signal is a TV signal at 450 MHz to 750 MHz, the 1 GHz to 2 GHz band, which is the third harmonic, has a mobile phone transmission band, and for the TV tuner, the third harmonic of the local oscillator, The simultaneous demodulation of signals other than the desired signal is a major problem as a factor that degrades the SNR (Signal to Noise Ratio) of the signal. Conventionally, this interference wave was sufficiently blocked by using a tracking filter before inputting the signal to the mixer. However, it is difficult to use a tracking filter having sufficient interference wave blocking characteristics in a tuner for mobile use. is there.

  In order to solve such a problem, a technique has been proposed in which a signal with respect to harmonics of a local oscillation signal is canceled to avoid demodulation of a signal in an unnecessary frequency band.

  Patent Document 1 is a tuner that mixes signals having different phases and received signals from two local oscillators with a mixer, and the phase of the local oscillator is adjusted in advance so as to cancel the third harmonic of the received signal. The invention is disclosed.

Non-Patent Document 1 discloses a technique for canceling high-frequency components by combining a large number of local oscillators and mixers.
USP 6,766,158 IEEE JSSC 37-12, pp. 1710-1720, 2002

  However, when this method is used to obtain a conventional I / Q signal, there is a problem that the number of phases required for the local oscillator increases. For example, if an I / Q signal is to be obtained while canceling the frequency component corresponding to the third harmonic of the local oscillator, a local oscillation signal having the number of phases as many as 12 is required.

  In Non-Patent Document 1, a large number of mixers are required and the gain ratio needs to be adjusted to the set value with high accuracy.

  The present invention has been conceived in view of the above problems. That is, a three-phase signal having a phase difference of 2π / 3 is mixed as a first local oscillation signal with a high-frequency signal in the first mixer, and the obtained three-phase signal, which is a baseband or intermediate frequency signal, has a phase of 2π / 3. Different three-phase signals are mixed as a second local oscillation signal with a three-phase baseband signal or intermediate frequency signal in the second mixer, and as a result, a one-phase intermediate frequency signal is obtained to cancel out the third harmonic component. While obtaining an intermediate frequency signal. The one-phase intermediate frequency signal can be directly demodulated to obtain an I / Q baseband signal.

  Since the tuner of the present invention cancels the third harmonic component, even if there is another signal in a frequency band three times higher than the frequency targeted for demodulation, it will not be demodulated, so a good S / N signal Can be demodulated.

  As a specific example, the frequency band used by a mobile phone in the vicinity of about 2 GHz has a three-fold relationship with the frequency of a signal of about 400 MHz to 700 MHz used in television broadcasting. Therefore, by using the tuner of the present application when demodulating a signal in the television broadcast band, it is possible to avoid the signal in the cellular phone band being demodulated at the same time and the S / N of the signal being deteriorated.

  A tuner according to the present invention includes a first mixer that down-converts a desired signal modulated in a high-frequency band with a three-phase local oscillation signal, and a cancel unit that cancels an in-phase signal included in the output of the first mixer Have Further, a polyphase filter may be provided to remove a signal adjacent to the output of the cancel unit. Moreover, you may have the frequency conversion part which frequency-converts the output of a cancellation part further. Further, the frequency conversion unit may have a function of canceling the in-phase signal. Preferred embodiments will be described below.

(Embodiment 1)
FIG. 1 shows the configuration of the tuner according to the first embodiment of the present invention. This assumes a television tuner that receives an OFDM signal with a bandwidth of 8 MHz. The output signal of the first mixer is a baseband signal whose center frequency is zero. The output signal of the tuner is an intermediate frequency (hereinafter referred to as “IF signal”) signal having a center frequency of 4.571 MHz.

  The tuner 1 according to the present embodiment includes a low noise amplifier 10 (hereinafter referred to as “LNA”), a first mixer 15, a first local oscillator 70, and a baseband signal filter 20 (hereinafter referred to as “BB signal filter”). ), Second mixer 30, second local oscillator 75, IF-PPF 40 (Intermediate Frequency-Poly Phase Filter: hereinafter referred to as “IF-PP filter”), three-phase / one-phase converter 60 (hereinafter referred to as “3P / 1P converter)) and a buffer 65. In the present embodiment, the cancel unit is the second mixer 30, and the frequency converter is also used as the second mixer.

  The LNA 10 amplifies a high frequency signal (hereinafter referred to as “RF signal”) received from an antenna or a cable. The amplified RF signal Srf is mixed with the signal Slo1 from the first local oscillator 70 by the first mixer 15, and converted into a three-phase baseband signal Sd3. Although details of these baseband signals will be described later, when the RF signal is a sine wave, each is the same signal and has a phase difference of 2π / 3. In addition, these baseband signals include components related to signals transmitted in a frequency band higher than the band in which the desired signal was transmitted. These signals are included in the baseband signal Sd3 as signals having the same frequency.

  From these baseband signals output from the first mixer 15, the BB signal filter 20 blocks the interference wave and the second mixer 30 generates a frequency component that causes reception interference.

  In the second mixer 30, the baseband signal Sd3f that has passed through the BB signal filter 20 is frequency-mixed with the signal Slo2 of the second local oscillator 75. At the time of mixing, the coefficients of the three-phase baseband signal are manipulated to cancel and remove the down-converted signal from a band higher than the desired signal included in the baseband signal. At this time, the baseband signal is converted to the IF signal Sif.

  Since an image signal is generated from the IF signal Sif, it is removed by the IF-PP filter 40. Since the IF-PP filter can remove negative frequency components, an IF signal without an image signal can be obtained.

  Since the output Sifp of the IF-PP filter 40 is a three-phase IF signal, it is converted into a one-phase complementary signal by the 3P / 1P converter 60. The one-phase (single-phase) complementary signal Sif1 obtained in this way is an IF signal without a signal down-converted from a higher frequency band than the band in which the desired signal in the RF signal was transmitted. This single-phase IF signal can be appropriately amplified by a buffer or the like to be output Sot.

Each component will be described in detail below.
The LNA 10 is a first-stage amplifier that amplifies the RF signal Srf, and is not particularly limited. However, a configuration with less intermodulation distortion is preferably used. Further, in the later signal processing, a circuit configuration in which distortion is unlikely to occur during signal processing can be used for complementary signals, so if it has a configuration capable of non-equilibrium-balance conversion, it should be used more suitably. Can do.

  The first mixer 15 mixes the signal Slo1 from the first local oscillator 70 with the RF signal, which is the output signal from the LNA 10, and down-converts it into a three-phase baseband signal Sd3. In this embodiment, since the RF signal that is an output signal from the LNA is a complementary signal, an example in which a six-phase oscillator is used as the first local oscillator is shown.

  The first mixer 15 is mainly composed of the frequency mixer 13, but the transconductance amplifier 12 may be disposed in the preceding stage. The transconductance amplifier is an amplifier having a flat frequency characteristic. The frequency mixer 13 is a multiplier.

  FIG. 2 shows a circuit example of the frequency mixer 13 and the first local oscillator 70. The input signal Srf is applied to the input terminal IN. The first local oscillator 85 includes local oscillators from V1 to V6, and generates signals having different phases every 60 °. Output signals from V1 to V6 are represented by s0 to s300, and the numbers represent phases. The input terminal IN is connected to the gates of the two transistors M1 and M2. These two transistors have a common source and are connected to the current source I1. On the other hand, the drains of these two transistors are connected to the voltage Vsup via resistors R1 and R2, respectively. That is, the input part constitutes a transconductance amplifier that is a differential amplifier.

  Between the drains of the two transistors M1 and M2 in the input stage and the resistors R1 and R2, a gate is connected to a local oscillator (V1 to V6) whose phases are 0 °, 60 °, 120 °, 180 °, 240 °, and 300 °. Are connected to the sources of the transistors (M3 to M14). Since the transistors M1 and M2 in the input stage are differential amplifiers, there are two transistors on the left and right. Since a transistor for each phase is connected to the drain of each transistor, there are two transistors of one phase on each of the left and right.

  In FIG. 2, M3 and M4 are transistors in which a local oscillator signal V1 having a phase of 0 ° is connected to the gate. Thereafter, M5 and M6 are 60 °, M7 and M8 are 120 °, M9 and M10 are 180 °, M11 and M12 are 240 °, and M13 and M14 are 300 ° local oscillator signals (V6, V2, V4, and V3, respectively). , V5) are connected to the gate.

  The drains of the transistors M3 to M14 are connected to the respective output terminals. Specifically, M3, M4, M9, and M10, which are connected to the gates of signals from local oscillators with phases of 0 ° and 180 °, are connected to the output terminal OP1. Similarly, M5, M6, M11, and M12, which are connected to the gates of signals from local oscillators having phases of 60 ° and 240 °, are connected to the output terminal OP2. Further, M7, M8, M13, and M14, which are connected to the gates of signals from the local oscillators whose phases are 120 ° and 300 °, are connected to the output terminal OP3. The output terminals OP1, OP2, and OP3 are down-converted from the RF signals, respectively, and output baseband signals Sd3 having different phases every 120 °.

  FIG. 3 shows the state of signals from these six local oscillators. A signal having a phase of 0 ° is represented by a thick solid line, a signal having a phase difference of 60 ° is represented by a thin dotted line, and a signal having a phase difference of 120 ° is represented by a thick dashed line. Further, a signal whose phase is delayed by 180 ° is represented by a thin solid line, a signal that is delayed by 240 ° is represented by a thick dotted line, and a signal that is delayed by 300 ° is represented by a thin one-point difference line.

  In the first mixer shown in FIG. 2, only the transistor having the largest output among these six signals is operated. For example, referring to FIG. 3, the signal having the phase of 60 ° is the largest during the period from time t1 to t2. Therefore, during this period, the two transistors M5 and M6 in FIG. 2 are turned on to output signals. The output is output from OP2.

  In this way, a three-phase baseband signal can be obtained. Since this signal directly down-converts the RF signal into a baseband signal, it does not include an image signal. However, the signal modulated in the corresponding band in the RF signal by the third harmonic contained in the signal of the local oscillator is simultaneously down-converted.

  This will be described in more detail with reference to FIG. FIG. 4 shows the relationship between frequency and signal. FIG. 4A shows an RF signal. In addition to the signals shown here, many signals are superimposed on the RF signal, but here, as a desired wave, a signal having a frequency f1 and its adjacent wave and a signal having a frequency three times that of f1 (3f1). Indicates. The desired wave 80 is indicated by oblique diagonal lines, the adjacent wave 81 is indicated by vertical lines, and the interference wave 82 is indicated by horizontal lines.

  FIG. 4B shows the output signal Sd3 of the first mixer. The local oscillators V1 to V6 directly downconvert the signal of frequency f1 to the baseband signal. Similarly, the adjacent wave is down-converted by the frequency of f1 and exists as an adjacent signal of the baseband signal. On the other hand, the signal of the frequency 3f1 in the RF signal is simultaneously down-converted into a baseband signal by the third harmonic of the signal of the local oscillator. The signal transmitted in this high frequency band becomes noise for the signal of the desired wave and degrades the S / N of the signal. In the present invention, the signal transmitted in the 3f1 band (hereinafter referred to as “3f1 signal”) is canceled to improve the SNR of the desired wave.

  Referring again to FIG. 1, the description of the tuner of the present invention will be continued. The three-phase baseband signal Sd3 output from the first mixer 15 passes through the BB signal filter 20, which is a low-pass filter whose passband is slightly wider than 4 MHz. By passing this filter, the interference wave is blocked to some extent, and the frequency component that causes reception interference is blocked by the harmonic of the signal of the second local oscillator. That is, in the relationship between the frequency and the signal in FIG. 4B, when there is an extra signal in the harmonic band of the signal of the local oscillator 75 of the second mixer, the signal becomes a baseband signal as an interference wave. Since it is superimposed, such a signal is deleted.

・・・（１） Here, the output 15 of the first mixer will be described in more detail. The RF signal Srf is expressed by equation (1).

... (1)

・・・（２）

・・・（３）

・・・（４）

ここで、ｋおよびｈはミキサのゲインである。 Here, ωs represents the angular frequency of the desired wave, and φ represents the phase. a is the amplitude. When the angular frequency of the signal of the first local oscillator is ω _L and the three types of output signals having different phases from the first mixer are x ₁ (t), x ₂ (t), and x ₃ (t), respectively, It is expressed as (2), (3), and (4).

... (2)

... (3)

... (4)

Here, k and h are mixer gains.

In each equation, the first term on the right side represents the result of down-converting the desired wave, and the second term on the right side is down-converted from a frequency band three times the desired wave by the third harmonic component of the signal of the local oscillator. It is a disturbing wave. The phase of the desired wave of x ₁ (t) is φ, the phase of the desired wave of x ₂ (t) and x ₃ (t) is −120 ° and −240 °, and the three kinds of desired waves are 120 respectively. ° Out of phase. However, it can be seen that all the disturbing waves due to the harmonic components have the same phase.

Accordingly, when the three signals of the first mixer output are subjected to primary conversion, by appropriately determining the coefficients of x ₁ (t), x ₂ (t), and x ₃ (t), in-phase interference waves due to harmonic components Can only be offset.

  Next, the second mixer 30 for canceling out-of-phase interference waves will be described. The second mixer 30 receives the three-phase signal Sd3f, frequency-mixes the six-phase second local oscillator signal Slo2, and converts it to an intermediate frequency signal having a center frequency of 4.571 MHz. The second mixer outputs a three-phase signal Sif.

  FIG. 5 shows a mixer 100 for one phase output of the second mixer. In the entire second mixer, three mixers 100 are combined. The mixer 100 includes analog switches 104 and 105 and a differential amplifier 106. The signal Slo2 of the second local oscillator 75 is a 6-phase digital signal, and controls on / off of the analog switches 104 and 105. An example of the second local oscillator signal Slo2 is shown in FIG. The signals of L1 to L6 turn on / off the analog switches 104 and 105, switch the input signal, and obtain an output.

・・・（５）

・・・（６）

・・・（７） Here, y ₁ (t), y ₂ (t), and y ₃ (t) are input signals to the second mixer. These are the output Sbbf of the BB signal filter 20. More specifically, it may be considered as x ₁ (t), x ₂ (t), and x ₃ (t) given by equations (2) to (4). Assuming that the output of the second mixer is z ₁ (t), z ₂ (t), z ₃ (t), the respective signals are expressed by the following equations (5) to (7).

... (5)

... (6)

... (7)

・・・（８） y ₁ (t), y ₂ (t), and y ₃ (t) are desired signals having different phases by 120 °, and each signal has a carrier frequency 3f1 that is an in-phase interference wave component due to the third harmonic. It has a signal component. Therefore, by setting c ₁ (t), c ₂ (t), and c ₃ (t) in the relationship shown in the following equation (8), the in-phase disturbance wave component of the third harmonic can be canceled out. .

... (8)

・・・（９）

ここでＴ_２はｃ_１（ｔ）の周期である。 Further, in the second mixer 30, c ₁ (t), c ₂ (t), c so as to satisfy the condition of the following expression (9) in order to suppress the interference wave with respect to the even harmonics of the second local oscillator 90. It is desirable to set ₃ (t).

... (9)

Here, T ₂ is the period of c ₁ (t).

・・・（１０）

・・・（１１）
なお、後述するがｃ_２（ｔ）、ｃ_３（ｔ）はｃ_１（ｔ）と同じ信号で位相が異なるだけである。従って（１０）式、（１１）式はｃ_２（ｔ）、ｃ_３（ｔ）にも同様に適用できる。 Furthermore, by making c ₁ (t), c ₂ (t), and c ₃ (t) into the following expressions (10) and (11), c ₁ (t), c ₂ (t), c ₃ (t) may not include the third harmonic.

... (10)

(11)
As will be described later, c ₂ (t) and c ₃ (t) are the same signals as c ₁ (t) but only have different phases. Therefore, the equations (10) and (11) can be similarly applied to c ₂ (t) and c ₃ (t).

The fact that c ₁ (t), c ₂ (t), and c ₃ (t) do not include the third harmonic means that z ₁ (t), z ₂ (t), and z ₃ that are the outputs of the second mixer. Any of (t) can suppress the interference wave with respect to the third harmonic of the second local oscillator.

Here, the operation of the mixer of FIG. 5 will be described in more detail. The mixer of FIG. 5 includes input terminals 101, 102, and 103 to which y ₁ (t), y ₂ (t), and y ₃ (t) are input, and analog switches 104 and 105 that switch the output of each signal, A differential amplifier 106 that outputs a difference between outputs of these analog switches is included. Outputs y ₁ (t), y ₂ (t), and y ₃ (t) are input to the two analog switches, respectively. The switching operation of the analog switch is performed by digital signals L1 to L6. Here, digital signals L1, L2, and L3 are input to the analog switch 104, and digital signals L4, L5, and L6 are input to the analog switch 105. Therefore, L1, L2, and L3 are not simultaneously turned on, and L4, L5, and L6 are not simultaneously turned on.

Moreover, focusing on the output of the analog switch 104, L1 _y 1 (t) is output by, _y 3 (t) is output by _y 2 (t) is output L3 by L2. In the analog switch 105 in the same manner, L4 _y 2 (t) is output by, L5 _y 3 (t) is output by, _y 1 (t) is output by L6.

FIG. 6 shows an example of digital signals L1 to L6. For example, in section 110, the L1 and L4 signals are turned on. The signal L1 outputs y ₁ (t), and the signal L4 outputs y ₂ (t). Therefore, in this section 110, the mixer outputs a signal y ₁ (t) −y ₂ (t).

FIG. 7 summarizes the output of the mixer from the viewpoint of the output of y ₁ (t), y ₂ (t), and y ₃ (t). The horizontal axis is the time axis, and the vertical direction represents the output voltage. On each axis, the output is zero, the upper direction indicates a positive output, and the lower direction indicates a negative voltage output. The output of the mixer 100 is represented as the sum of three outputs at a specific time. For example, in section 111, y ₁ (t) has a positive output, y ₂ (t) has a negative output, and y ₃ (t) has an output of zero. This is the state when the L1 and L4 signals illustrated in FIG. 6 are turned on.

In addition, this relationship is a relationship in which one of the remaining two signals is subtracted from any one of y ₁ (t), y ₂ (t), and y ₃ (t) at any time. Yes. In y ₁ (t), y ₂ (t), and y ₃ (t), the interference wave caused by the third harmonic of the first local oscillator is superimposed in phase, so that the third harmonic is always output from the mixer output. The interference wave due to the wave is canceled and is not output.

  The second local oscillator signal shown in FIG. 6 further satisfies the expressions (9) to (11). Therefore, it does not have even-order (second and fourth) harmonics and third harmonic components. Accordingly, reception interference due to these harmonics is suppressed. The output Sif of the second mixer 30 is shown in FIG. A signal whose center frequency is f2 and the frequency at RF which is an interference wave is 3f1 is canceled out and does not exist. The adjacent wave is somewhat attenuated by the BB signal filter 20.

  Returning to FIG. 1, the output of the second mixer 30 is a three-phase IF signal Sif having a phase difference of 120 °. However, since the signals of the respective phases are complementary signals, the output of the second mixer is 3 A balanced-non-equilibrium conversion is made for each phase. Then, the IF-PP filter 40 performs filter processing for removing adjacent signals with respect to the three-phase signal in which the signals of each phase are unbalanced. At that time, a negative frequency component is removed by using a polyphase filter that performs filtering by dividing the positive frequency and the negative frequency.

  In the IF-PP filter 40, a switched capacitor filter can be used, and a highly accurate filter can be realized. The characteristic of the IF-PP filter is shown in FIG. FIG. 4E shows the frequency allocation of the IF signal Sifp selected by the IF-PP filter. Here, it is possible to obtain a signal 82 of the desired wave 80 from which the signal 82 having the frequency 3f1 in RF that is the interference wave and the adjacent wave 81 are removed.

  The frequency-selected three-phase IF signal Sifp is converted into a pair of complementary signals Sif1 and output as an output signal Sot. The output signal is then A / D converted and demodulated. A buffer 65 may be provided at the final output stage.

  As described above, when down-converting a desired signal modulated in a high frequency band, the tuner 1 according to the present embodiment transmits a signal transmitted in a higher band than the desired signal due to the harmonic component of the local oscillator. Are simultaneously down-converted to solve the problem of degrading the S / N for the desired signal. Further, the problem that it is difficult to obtain an accurate I / Q signal when the first mixer is a three-phase mixer is solved by generating an IF signal that does not include an image signal.

(Embodiment 2)
FIG. 8 shows the configuration of the tuner 2 according to the second embodiment of the present invention. This assumes a television tuner that receives an OFDM signal with a bandwidth of 8 MHz. The output signal is an IF signal with a center frequency of 4.571 MHz. The output signal of the first mixer is an IF signal having a center frequency of −4.571 MHz.

  The tuner 2 of the present embodiment includes an LNA 10, a first mixer 15, a first local oscillator 71, a low-pass filter filter 21, an in-phase signal removal amplifier 33, a first IF filter 41, a second mixer 50, a second local oscillator 75, A second IF filter 61 and a buffer 65 are included. In the present embodiment, the cancellation unit is an in-phase signal removal amplifier and a second mixer, and the frequency conversion unit is a second mixer.

  In the present embodiment, the first local oscillator 71 outputs a signal Slo1 having a frequency of f1 + f2 with respect to the center frequency f1 of the desired wave. Thus, after the RF signal Srf is amplified by the LNA 10, the frequency is mixed by the first mixer 15 and converted into the three-phase IF signal Sd3. Since the RF signal is converted to the IF signal, the image signal at this point is a signal having a center frequency of −4.571 MHz. FIG. 11A shows the spectrum of the output of the first mixer. The display of the desired wave 80, the adjacent wave 81, and the interference wave 82 in FIG. 11 is the same as the display in FIG.

  The three-phase IF signal Sd3, which is the output of the first mixer 15, passes through the low-pass filter 21 having a balanced-unbalanced conversion function, thereby blocking the interference wave to some extent. At this time, an interference wave due to the third harmonic of the first local oscillator 71 is included as an in-phase signal. The three-phase signal Sd3f that has passed through the low-pass filter 21 is converted into a three-phase signal Sdf from which the in-phase signal component has been removed by passing through the amplifier 33 having the in-phase signal removal function. At this time, by removing the in-phase signal of the three-phase signal, a signal having a frequency 3f1-f2 at RF which is an interference wave component due to the third harmonic of the first local oscillator is removed.

  The output signal Sdf of the amplifier 33 with the in-phase signal removal function removes unnecessary waves to some extent in the first IF filter 41 which is a polyphase filter. In particular, the frequency component that causes reception interference is sufficiently blocked by the fifth and higher harmonics of the second local oscillator signal 75.

  FIG. 9 shows an example of the configuration of the common-mode signal cancellation amplifier 33. The in-phase signal rejection amplifier 33 includes transistors Q1 to Q3, resistors R1 to R6, and a current source I2. Input terminals Ina, Inb, and Inc are connected to the bases of Q1, Q2, and Q3, respectively. The emitters of the three transistors are connected to one current source I2 via resistors R1, R2, and R3, respectively. The collectors of the three transistors are connected to the voltage power supply Vcc via resistors R4, R5, and R6, respectively. Then, three output terminals of Outa, Outb, and Outc are provided from the collectors of Q1, Q2, and Q3.

  Three outputs Sd3f from the low-pass filter 21 are input to input terminals Ina, Inb, and Inc, respectively. Since the common current source I2 is connected to the emitters of the three transistors Q1 to Q3, the voltages of the output terminals Outa, Outb, and Outc change ideally when the signals applied to the three input terminals are in phase. do not do. Therefore, in-phase signal components are removed. Actually, since the in-phase signal rejection ratio is finite, the in-phase signal component is attenuated to some extent.

  A spectrum of the output Sdff of the first IF filter 41 is shown in FIG. The interference wave 82 indicated by the horizontal line is canceled and removed.

  Returning to FIG. 8 again, in the second mixer 50, the IF signal Sif is frequency-mixed with the signal Slo2 of the six-phase second local oscillator 71 and converted into an intermediate frequency signal having a center frequency of 4.571 MHz. The second mixer 50 outputs a single-phase complementary signal. For example, the configuration shown in FIG. 5 can be used. Therefore, the second mixer 50 includes analog switches 104 and 105 and a differential amplifier 106.

The second local oscillator signal Slo2 is a 9.142 MHz six-phase digital signal, and controls the on / off of the analog switches 104 and 105. As the second local oscillator signal Slo2, for example, the same signal as that shown in FIG. 6 can be used. C ₁ (t), c ₂ (t), and c ₃ (t) by this signal are as shown in FIG. This is the same as that shown in FIG. Therefore, since these satisfy the condition of the equation (8), the interference wave due to the third harmonic of the first local oscillator 71 is removed. That is, the second mixer in the present embodiment also has a function of canceling the in-phase signal. Note that “cancel in-phase signal” is treated as an agreement with “removing in-phase signal” in this specification.

Even in the amplifier 33 with the in-phase signal removal function, this interference wave is removed to some extent, but since the in-phase signal removal rate is finite, it cannot be completely removed. Since c ₁ (t), c ₂ (t), and c ₃ (t) satisfy the expressions (9) to (11), they do not have the second, third, and fourth harmonic components, and their harmonics. Reception disturbance due to waves is suppressed. Therefore, an interference wave with respect to higher harmonics higher than the fifth harmonic is mainly considered. The spectrum of the output Sif1 of the second mixer 50 is shown in FIG.

  The output of the second mixer 50 is a single-phase balanced IF signal Sif1. Then, the second IF filter 61 performs filter processing for sufficiently removing adjacent signals. In the present embodiment, since the output of the second mixer 50 is a single-phase signal, it is impossible to separate the positive frequency component and the negative frequency component, so the signal component corresponding to the negative frequency component is the first frequency component. The 1IF filter 41 needs to be sufficiently blocked. The spectrum of the output Sif1f of the second IF filter 61 is shown in FIG.

  As the second IF filter 61, a switched capacitor filter can be used, and a highly accurate filter can be realized. The output signal Sif1f of the second IF filter 61 is output as the final output Sot, and is then demodulated after being amplified or A / D converted. Note that the buffer 65 may be disposed at the final stage.

  The advantage of this embodiment is that, by using the two IF filters 41 and 61, a signal having a frequency immediately adjacent to the reception band can be well removed, so that the desired wave can be received well even if the adjacent interference wave is strong. A tuner that can be realized. Since the signal bandwidth is 7.61 MHz, the signal bandwidth of IF is −8.376 MHz to −0.766 MHz and 0.766 MHz to 8.376 MHz, respectively. In both cases, the value with a small absolute value of the frequency of the IF signal band is lower than the center frequency of the center frequency. Therefore, both the first and second IF filters cut off the frequency near 0 to make adjacent signals on both sides efficient. Can block well.

  The second advantage of the present embodiment is that the output of the first mixer 15 is an IF signal, so that it is less susceptible to flicker noise generated by the first mixer. Furthermore, when the output of the first mixer 15 is a baseband signal, it is difficult to receive subcarriers at the center frequency, but in this embodiment, all subcarriers can be received.

Another advantage of the present embodiment is that the configuration of the second mixer 50 is simple.
Although the present embodiment is a tuner that receives an OFDM signal, the received signal may not be OFDM, may be of another digital modulation method, or may receive an analog signal. Also good.

(Embodiment 3)
FIG. 12 shows a tuner 3 according to the third embodiment of the present invention.
The tuner 2 of the present embodiment includes an LNA 10, a first mixer 15, a first local oscillator 71, a low-pass filter filter 21, an in-phase signal removal amplifier 33, a first IF filter 41, a second mixer 51, a second local oscillator 76, A second IF filter 62 and a buffer 65 are included. In the present embodiment, the cancel unit is the in-phase signal removal amplifier 33 and the second mixer, and the frequency conversion unit is the second mixer.

  The tuner 3 of the present embodiment is a low-IF double superheterodyne system (a reception system having two-stage IF signals) as in the second embodiment, but the output of the second mixer 51 is three-phase. The second IF filter 62 is a three-phase polyphase filter.

  FIG. 13 shows an example of the configuration of the second mixer 51. The second mixer 51 includes input terminals 151 to 153, analog switches 154 to 156, and output terminals 157 to 159. The analog switch performs switching operation according to the digital signals L1 to L3. This digital signal is a three-phase signal supplied from the second local oscillator 76.

The three outputs of the _first IF filter 41 are input to input terminals 153, 152, 151 as x ₁ (t), x ₂ (t), x ₃ (t), respectively. These input signals x ₁ (t), x ₂ (t), and x ₃ (t) are connected to all of the analog switches 154 to 156. On the other hand, the digital signals L1 to L3 are input to the analog switches 154 to 156 to switch the input signals. The analog switch outputs are z ₁ (t), z ₂ (t), and z ₃ (t), respectively, and are output from output terminals 159, 158, and 157, respectively.

Focusing on the analog switch, the analog switch 154 outputs x ₃ (t) as a digital signal L1, x ₂ (t) as L2, and x ₁ (t) as L3. Similarly, the analog switch 155 outputs x ₂ (t) at L1, x ₃ (t) at L2, and x ₁ (t) at L3. The analog switch 156 outputs x ₁ (t) at L1, x ₂ (t) at L2, and x ₃ (t) at L3.

FIG. 14 shows an output example of L1, L2, and L3. By switching the input signal by this digital signal, z ₁ (t), z ₂ (t), and z ₃ (t) can be obtained in the same manner as the equations (5), (6), and (7). However, in the expressions (5), (6), and (7), y ₁ (t), y ₂ (t), and y ₃ (t) are respectively x ₁ (t), x ₂ (t), and x ₃ ( Replace with t). That is, the output signals z ₁ (t), z ₂ (t), and z ₃ (t) are signals that are the same signal and have phases that differ by 120 °. Therefore, the second mixer 51 in the present embodiment also has a function of canceling the in-phase signal.

  Since the signal Slo2 of the second local oscillator 76 is a three-phase signal, it is easier to generate than the case of generating a six-phase signal. For example, when an accurate three-phase signal is generated by dividing a high-frequency signal, a signal having a frequency three times that of the output signal is required. To obtain an accurate six-phase signal, the output signal 6 A signal having a double frequency is required, and its value is 27.426 MHz. Having a very high frequency signal inside the tuner has the potential for interference with other signals and is not very desirable.

  Since the second local oscillator signal Slo2 is a three-phase signal, the second mixer 51 generates a mixing output for the second harmonic of the second local oscillator signal Slo2. This is because the condition of formula (9) is not satisfied. However, since the frequency of the second harmonic of the second local oscillator signal Slo2 is different in sign from the frequency of the fundamental wave, the disturbing wave generated thereby is far away from the desired wave. This is shown in FIG.

  In FIG. 15, the display of the desired wave 80, the adjacent wave 81, and the interference wave 82 is the same as in FIG. The output Sd3 of the first mixer is an IF signal having −f2 as the center frequency, as in the second embodiment. This is shown in FIG. FIG. 15B shows the output Sdff of the first IF filter 41. Further, when the center frequency of the output of the first IF filter is converted from −f2 to f2, the region 83 that becomes an interference wave with respect to the desired wave is f2 by the frequency of the second harmonic of the second local oscillator signal Slo2. It shows that the point is the frequency of the fifth harmonic.

  This is because Slo2 outputs a frequency of 2f2, but its second harmonic, that is, a frequency of -4f2, down-converts the signal in the band of 5f2 to the band of f2. Here, the sign of the second harmonic of 2f2 is negative because the frequency of the second harmonic of the three-phase signal is opposite to the sign of the fundamental wave. That is, since the frequency component of + 4f2 is not included as the second harmonic wave, the signal in the band of −3f2 does not become an interference wave. Therefore, in the configuration of the present embodiment, the first IF filter 41 can sufficiently block the frequency component of the interference wave.

  The output Sif of the second mixer 51 removes adjacent waves by the second IF filter 53. Since the output Sif is a three-phase signal, the adjacent wave can be accurately cut off by configuring the second IF filter 62 with a polyphase filter. The signal Sif1f from which the adjacent wave has been removed is output as an output Sot through the buffer 65 as appropriate. FIG. 15C shows a state where the output Sif of the second mixer is up-converted with the center frequency f2. FIG. 15D shows the signal Sif1f from which the adjacent wave 81 has been removed by the second IF filter 62. FIG.

  One advantage of the third embodiment of the present invention is that since the second IF filter 62 is a three-phase polyphase filter, a frequency selective filter that distinguishes positive and negative frequencies is used. It is easy to block more adjacent signals.

  Further, by allowing the second mixer 51 to mix the second local oscillator signal 76 with respect to the second harmonic, there is an advantage that the structure of the second mixer can be simplified.

(Embodiment 4)
FIG. 16 shows the configuration of the tuner 4 according to the fourth embodiment of the present invention.
The tuner 4 of the present embodiment includes an LNA 10, a first mixer 15, a first local oscillator 70, a low-pass filter filter 21, an in-phase signal removal amplifier 33, a BB signal polyphase filter 42, a second mixer 50, and a second local part. An oscillator 75, a second IF filter 61, and a buffer 65 are included. In the present embodiment, the cancel unit is the in-phase signal removal amplifier 33 and the second mixer, and the frequency conversion unit is the second mixer.

  The tuner 4 according to the present embodiment is a down-up conversion method in which the RF signal Srf is once dropped into a three-phase baseband signal and then converted into an IF signal, like the tuner 1 according to the first embodiment. The difference is that a BB signal polyphase filter 43 serving as a polyphase filter is used as a filter in the baseband signal.

  The first mixer 15 mixes the input signal from the LNA 10 with the three-phase signal Slo1 from the six-phase first local oscillator 70 to obtain a baseband signal Sd3. This Sd3 is a signal having the same waveform and different phases by 120 °. In this signal, a component from the third harmonic of the RF signal Srf exists as an in-phase signal.

  The output Sd3 of the first mixer 15 is passed through the low-pass filter 21 to remove other interference waves as much as possible. The output Sd3f of the low-pass filter 21 is input to the in-phase signal removal amplifier 33. At this time, the frequency component three times that of the RF signal superimposed as an in-phase signal is canceled.

  The output Sdf of the in-phase signal removal amplifier 33 is a three-phase baseband signal. The adjacent wave is removed from this signal by using the BB signal polyphase filter 42 to obtain a signal Sdff. Thereafter, the frequency is converted into a single-phase IF signal Sif1 by the second mixer 50, and an extra component is removed by a normal IF filter 61 to obtain a signal Sif1f, which is then output Sot. Note that the second mixer in the present embodiment can use the same mixer as the second mixer 50 in FIG. 8, and thus has a function of canceling the in-phase signal.

  One advantage of the present embodiment is that the circuit in the baseband is a polyphase filter, so that the circuit can be simplified compared to the case where filters are provided for each of the three phases. In addition, since the first mixer output is a baseband signal, a simple low-pass filter can be used as a filter for blocking adjacent interference as compared with the IF signal, so that the circuit can be simplified correspondingly. .

(Embodiment 5)
FIG. 17 shows a configuration of a tuner according to the fifth embodiment of the present invention.
The tuner 5 of this embodiment includes an LNA 10, a first mixer 15, a first local oscillator 71, a low-pass filter filter 21, an in-phase signal removal amplifier 33, an IF-PP filter 43, a three-phase one-phase converter 53, and a buffer 65. including. In the present embodiment, the cancel unit is an in-phase signal removal amplifier 33 and a three-phase / one-phase converter 53.

  The signal received from the antenna or cable is amplified by the LNA, and then the frequency is mixed by the first mixer 15 and converted into a three-phase IF signal S3d. Since the RF signal is converted into the IF signal, the image signal at this time is mixed as an interference wave. The three-phase IF signal Sd3 blocks the interference wave to some extent by passing through a low-pass filter 21 having a balanced-unbalanced conversion function.

  At this time, an interference wave due to the third harmonic of the first local oscillator 71 is included as an in-phase signal. The three-phase signal S3df that has passed through the low-pass filter 21 is converted into a three-phase signal Sdf from which the in-phase signal component has been removed by passing through the amplifier 33 having the in-phase signal removal function. At this time, the in-phase signal of the three-phase signal is removed, so that the interference wave component due to the third harmonic of the first local oscillator is removed.

（１２） The output signal Sdf of the amplifier 33 with the in-phase signal removal function removes unnecessary waves in the first IF filter 43 (IF-PPF), which is a polyphase filter, to be a signal Sdf. In particular, it is necessary to sufficiently remove signal components in which the absolute value of the frequency of the image signal is the same as the desired wave and the sign is different. At this time, since the signal is a three-phase signal, it is converted into a one-phase IF signal Sdffx. At that time, it is possible to take out only one phase, but the following is performed. Now, assuming that three-phase signals are x ₁ (t), x ₂ (t), and x ₃ (t), and one phase to be extracted is y ₁ (t), the relationship satisfying Expression (12) is satisfied.

(12)

  Then, since the condition of the expression (12) is satisfied, the interference wave component due to the third harmonic of the first local oscillator signal that could not be removed by the amplifier 33 with the in-phase signal removal function is further suppressed here. Can do. Thereafter, the output signal is set as appropriate through the buffer 65.

（１３） In the present embodiment, the output signal is an IF signal, but it may be a baseband signal. In that case, the baseband signal needs to be a two-phase signal, and the three-phase to two-phase conversion may be performed by the three-phase one-phase converter in FIG.

(13)

(Embodiment 6)
FIG. 18 shows a configuration diagram of a tuner according to the sixth embodiment.
The tuner 6 of this embodiment includes an LNA 10, a first mixer 15, a first local oscillator 71, a low-pass filter filter 21, a three-phase three-phase converter 35, an IF-PP filter 43, a three-phase one-phase converter 53, and a buffer. 65. In the present embodiment, the cancel unit is a three-phase three-phase converter 35.

  In the fifth embodiment, the interfering wave with respect to the third harmonic of the first local oscillator in the first mixer is removed by the in-phase signal removal amplifier. However, in the sixth embodiment, in the three-phase three-phase converter 35 Do. The three-phase three-phase converter 35 corresponds to performing a primary conversion operation on the three-phase output Sd3 of the first mixer 15.

FIG. 19 shows an example of a partial configuration of the three-phase / three-phase converter 35. A part 120 of the three-phase three-phase converter 35 receives a three-phase input signal, and when each input signal is x ₁ (t), x ₂ (t), x ₃ (t), b ₁ , Amplifiers 121, 122, and 123 that multiply b ₂ and b ₃ , an adder 125 that adds the outputs of the amplifier 122 and the amplifier 123, and an adder 124 that adds the output of the adder 125 and the output of the amplifier 121 are included. The final output is the output of the adder 124. The three-phase / three-phase converter 35 has three converters 120 shown in FIG.

・・・・・（１４）
このとき、（１５）式の関係が成り立つように係数ｂ_１、ｂ_２、ｂ_３を設定することにより同相信号を除去することができる。

・・・・・（１５） The three-phase / three-phase converter 35 uses the signals before the primary conversion as x ₁ (t), x ₂ (t), and x ₃ (t), and the signals after the primary conversion as y ₁ (t) and y _2. When (t) and y ₃ (t) are set, primary conversion is performed as shown in equation (14).

(14)
At this time, the in-phase signal can be removed by setting the coefficients b ₁ , b ₂ , and b ₃ so that the relationship of the expression (15) is satisfied.

(15)

Assuming that the in-phase signal component of the three-phase signal is the average of the three-phase signals and subtracting the in-phase signal component, b ₁ : b ₂ : b ₃ = 1−1 / 3: −1/3: −1 / 3, that is, _{b 1 = 2, b 2 =} -1, b 3 = -1. Further, assuming that the in-phase signal is superimposed on the ideal three-phase signal, b ₁ = 1, b ₂ = −1, and b ₃ = 0.

  In-phase signal rejection amplifiers have a limit on the in-phase signal rejection ratio, but by performing such three-phase to three-phase conversion, a high rejection rate of interference waves can be expected. By using this three-phase / three-phase converter and an in-phase signal rejection amplifier in combination, a higher rejection rate of interference waves can be expected. At that time, the common-mode signal rejection amplifier may be installed in front of the three-phase / three-phase converter, in the rear, or in both the front and rear.

  In addition, although the case where the primary conversion of the signal is performed on the output signal of the first mixer has been described, the same applies to the signal after the output signal of the first mixer is passed through a frequency selection filter or the like. The operation can be performed.

  The tuner of the present invention can be used for a receiver that receives a radio signal.

FIG. 3 shows a configuration of a tuner according to the first embodiment. FIG. 3 shows a configuration of a multiplier of the tuner according to the first embodiment. FIG. 3 shows a first local transmitter signal according to the first embodiment. FIG. 3 is a diagram illustrating frequency allocation of signals at each element in the tuner according to the first embodiment. FIG. 3 shows a configuration of a second mixer according to the first embodiment. FIG. 3 shows a second local transmitter signal according to the first embodiment. FIG. 5 is a diagram illustrating timing for outputting an input signal of the second mixer according to the first embodiment. FIG. 5 shows a configuration of a tuner according to the second embodiment. FIG. 5 shows a configuration of an amplifier with a common-mode signal removal function according to a second embodiment. FIG. 10 is a diagram illustrating signal output timing in the second mixer according to the second embodiment. FIG. 6 is a diagram illustrating frequency allocation of signals in each part of the tuner according to the second embodiment. FIG. 5 shows a configuration of a tuner according to a third embodiment. FIG. 6 is a diagram showing a configuration of a second mixer according to a third embodiment. FIG. 10 shows a second local transmitter signal according to the third embodiment. FIG. 10 is a diagram illustrating frequency allocation of signals in each part of the tuner according to the third embodiment. FIG. 6 shows a configuration of a tuner according to a fourth embodiment. FIG. 10 shows a configuration of a tuner according to a fifth embodiment. FIG. 11 shows a configuration of a tuner according to a sixth embodiment. The figure which shows an example of a one part structure of a three-phase three-phase converter.

Explanation of symbols

1 to 6 tuners 10 LNA
DESCRIPTION OF SYMBOLS 15 1st mixer 20 Baseband signal filter 30 2nd mixer 40 IF-PP filter 60 3P / 1P converter 70 1st local oscillator

Claims (15)

A first mixer that mixes a three-phase local oscillation signal that is 120 degrees out of phase with the received signal;
A tuner having a cancel unit that cancels an in-phase signal included in the output using the output of the first mixer. 2. The tuner according to claim 1, wherein the cancel unit is a second mixer that mixes three-phase local transmission signals having phases different from each other by 120 ° with respect to the first mixer output. The tuner according to claim 2, further comprising a polyphase filter that inputs an output of the second mixer. The second mixer has coefficients c ₁ (t), c ₂ (t), and c _{3 where} y ₁ (t), y ₂ (t), and y ₃ (t) are input signals, respectively. (T) and signals z ₁ (t), z ₂ (t), z ₃ (t) satisfying the relations of the expressions (5), (6), and (7) whose phases are different from each other by 120 °, The tuner according to any one of claims 2 and 3, wherein the tuner satisfies the relationship of the expression (8).

... (5)

... (6)

... (7)

... (8)
5. The tuner according to claim 4, wherein each of the coefficients satisfies a relationship of Expression (9).

... (9)
Where T ₂ is the period of c ₁ (t) 6. The tuner according to claim 4, wherein each of the coefficients satisfies a relationship of Expressions (10) and (11).

... (10)

(11)
Where T ₂ is the period of c ₁ (t) The second mixer is
A local oscillator that generates a six-phase digital signal;
First and second analog switches for inputting all of the outputs of the first mixer and switching the outputs with signals for three phases of the six-phase digital signals;
4. The tuner according to claim 2, further comprising a differential amplifier that uses the outputs of the first and second analog switches as differential inputs. 5. The tuner according to claim 1, wherein the cancel unit is an in-phase signal removal amplifier. The tuner according to claim 8, further comprising a polyphase filter that inputs an output of the common-mode signal cancellation amplifier. 10. The tuner according to claim 8, further comprising a frequency conversion unit configured to frequency-convert an output of the common-mode signal removal amplifier. A three-phase one-phase converter for converting the output of the polyphase filter into a one-phase output;
When the output of the polyphase filter is x ₁ (t), x ₂ (t), x ₃ (t) and the coefficients are b ₁ , b ₂ , b ₃ , the output is y ₁ (t) The tuner according to claim 9, further comprising a three-phase one-phase converter that satisfies the expression (12).

(12)
A filter to which the output of the common-mode signal cancellation amplifier is input;
A mixer that mixes a three-phase digital signal with the output of the filter and outputs a signal having a phase difference of 120 °;
The tuner according to claim 8, further comprising a polyphase filter that inputs an output of the mixer.
The cancel unit has input signals x ₁ (t), x ₂ (t), x ₃ (t) and output signals y ₁ (t), y ₂ (t), y ₃ (t) The tuner according to claim 1, wherein the tuner is a three-phase three-phase converter that performs the primary conversion of the equation (14).

(14)
Here, the coefficients b ₁ , b ₂ , and b ₃ satisfy the expression (15).

(15)
The tuner according to claim 13, further comprising a polyphase filter that inputs an output of the cancel unit. A three-phase one-phase converter for converting the output of the polyphase filter into a one-phase output;
When the output of the polyphase filter is x ₁ (t), x ₂ (t), x ₃ (t) and the coefficients are b ₁ , b ₂ , b ₃ , the output is y ₁ (t) The tuner according to claim 14, comprising a three-phase one-phase converter that satisfies the expression (12).

(12)

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