JP2009005103A - Radio communication base station apparatus and signal spreading method - Google Patents

Abstract

The present invention improves throughput by obtaining frequency diversity gain while avoiding multi-access interference.
In a radio communication base station apparatus, an FDE / despreading unit 205 has a weight corresponding to a cyclic delay amount for each of a plurality of radio communication mobile stations, and an error between a delay time domain signal of a superimposed signal and a reference signal. Frequency domain equalization and despreading are performed on the superimposed signal using a weight that minimizes the frequency, and the demultiplexing / combining unit 207 applies the superimposed signal after frequency domain equalization to a plurality of radio communication mobile station apparatuses. Are divided into a plurality of delay time domain signals, and a synthesis process is performed for each of the plurality of delay time domain signals.
[Selection] Figure 5

Description

  The present invention relates to a radio communication base station apparatus and a signal spreading method.

  The radio channel is composed of a number of paths having different delay times. Such a radio channel is called a frequency selective channel and is an obstacle to high-quality signal transmission. Wideband DS-CDMA (Direct-Sequence Code Division Multiple Access) is used as multi-access in third to third-generation mobile communication systems using a 5 MHz bandwidth. In DS-CDMA, a Rake reception technique that improves transmission characteristics by using frequency selectivity of a radio channel is used. In Rake reception, a path diversity gain is obtained by decomposing each path by despreading and coherent combining.

  In the fourth generation, ultra-wideband wireless access utilizing a 20 MHz bandwidth or a bandwidth exceeding it is expected. However, since the number of paths of such an ultra-wideband radio channel becomes very large, there is a problem that frequency selectivity becomes too strong and BER (Bit Error Rate) characteristics deteriorate. Therefore, recently, attention has been focused on multi-carrier (MC) -CDMA that performs parallel transmission using a large number of orthogonal subcarriers. Also, attention is paid to OFDMA (Orthogonal Frequency Division Multiple Access) in which different subcarrier groups are assigned to each terminal. In MC-CDMA, frequency domain equalization (FDE) is used. In MC-CDMA, it is possible to perform multi-access while obtaining frequency diversity gain by applying FDE (MMSE-FDE) based on the Minimum Mean Square Error (MMSE) standard. Also in DS-CDMA, excellent BER characteristics can be obtained similarly to MC-CDMA by using MMSE-FDE instead of Rake synthesis. Therefore, DS-CDMA using MMSE-FDE has also attracted attention as one of the fourth generation access technologies.

  However, since the fading state of each mobile station is different in the uplink in which a large number of radio communication mobile station apparatuses (hereinafter referred to as mobile stations) access simultaneously, the radio communication base station apparatus (hereinafter referred to as the base station) has an MMSE-FDE. Even if is used, the channel cannot be completely converted to a frequency flat. For this reason, there has been a problem that throughput is reduced due to occurrence of multi-access interference (MAI) or other-user interference (MUI).

As a conventional technique for this, for example, a transmission method is proposed in which a plurality of spreading codes are generated by giving different cyclic delays to one spreading code, and signals from each mobile station are code-division multiplexed and orthogonalized using them. (For example, refer to Patent Document 1). As a result, transmission signals between mobile stations can be orthogonalized on the time axis, and MAI can be gradually reduced while obtaining a frequency diversity effect, thereby improving throughput.
JP-A-8-09749

  However, in the related art, when the conventional MMSE-FDE optimized in the frequency domain is applied to the code multiplex transmission method using the cyclic delay using the delay time domain, a diversity gain in the frequency domain can be obtained. However, since the orthogonality of the signals on the time axis is broken, there is a problem that MAI increases and throughput decreases.

  The present invention has been made in view of such points, and by performing MMSE-FDE in the frequency domain while maintaining orthogonality on the time axis, throughput is improved by obtaining frequency diversity gain while avoiding MAI. It is an object to provide a base station and a signal spreading method that can be used.

  The base station of the present invention provides a superimposed signal in which a plurality of signals spread by a second spreading code generated by cyclically delaying the first spreading code by a different cyclic delay amount for each of a plurality of wireless communication mobile station apparatuses are superimposed. And a weight according to the cyclic delay amount for each of the plurality of radio communication mobile stations, and the weight that minimizes an error between the delay time domain signal of the superimposed signal and the reference signal. And equalizing means for performing frequency domain equalization on the superimposed signal, and separating means for separating the superimposed signal after frequency domain equalization into a plurality of delay time domain signals for each of the plurality of radio communication mobile station apparatuses And combining means for performing combining processing for each of the plurality of delay time domain signals.

  According to the present invention, it is possible to improve throughput by obtaining frequency diversity gain while avoiding MAI.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the accompanying drawings.

(Embodiment 1)
First, the principle of delay division multi-access according to the present embodiment will be described.

<Principle of delay division multi-access>
In delay division multiple access, a cyclic code is generated by giving different cyclic delays (Cyclic Delay) for each mobile station to one spread code. Specifically, as shown in FIG. 1, the mobile station n gives a cyclic delay nΔ to the spreading code c (t) = ± 1 (t = 0 to SF-1) of the spreading factor SF, and the cyclic delay A spreading code c ((t−nΔ) mod SF) (t = 0 to SF−1) is generated. In other words, in this delay division multi-access, a frequency diversity gain is obtained while avoiding MAI, and an SF / Δ mobile station can perform multi-access. That is, the throughput of the system can be improved. Here, from the viewpoint of frequency domain equalization at the base station, it is desirable that the spreading code has a substantially uniform frequency spectrum. Therefore, in this embodiment, a PN (Pseudo Noise) code having randomness is used as the spreading code.

  Also, a transmission chip sequence generated by spreading transmission data symbols with a cyclic delay spread code is divided into SF chip blocks equal to the spreading factor at the base station in order to equalize the frequency domain at the base station. A guard interval (GI) is inserted, and a cyclic prefix (CP) is added to the GI for transmission.

ここで、Ｓ_ｎは送信信号電力である。また、送信データシンボルｄ_ｎはＥ［｜ｄ_ｎ｜^２］＝１を満たすものとする。 <Transmission signal>
In the mobile station obtains the transmitted data symbols _{d n} cyclic delay spread code c is diffused by ((t-nΔ) mod SF ) sends shown in the following equation (1) chip sequence _s n (t).

Here, _Sn is transmission signal power. Further, the transmitted data symbols _{d n} are _E shall meet the ^{_{[| 2 | d n] =}} 1.

<Channel>
The radio propagation path is composed of L independent paths. When the path gain of path l is h _{n, l} and the delay time is τ _{n, l} , the channel impulse response h _n (τ) is expressed by the following equation (2). Given.

ここで、η（ｔ）は零平均で分散２σ^２の白色複素ガウス雑音であり、Ｓ_ｎは平均受信電力である。また、複数の移動局が同時アクセスしているものとすると、基地局には、次式（４）のように式（３）に示す信号が重畳して受信される。

ここで、基地局と各移動局との位置が異なること、およびシャドウイング損失が異なることから、Ｓ_ｎは一般的には等しくない。 <Received signal>
A signal propagated through the radio channel expressed by the above equation (2) and received by the antenna of the base station can be expressed by the following equation (3).

Here, eta (t) is a white complex Gaussian noise of variance 2 [sigma] ² zero mean, is S _n is the average received power. Also, assuming that a plurality of mobile stations are accessing simultaneously, the base station receives the signal shown in Equation (3) superimposed as shown in Equation (4) below.

Here, since the positions of the base station and each mobile station are different and the shadowing loss is different, _Sn is generally not equal.

ここで、

であるため、式（６）を式（５）に代入すると、周波数領域信号Ｒ（ｋ）は次式（７）のようになる。

ここで、Π（ｋ）は零平均で分散２σ^２・ＳＦの複素ガウス雑音であり、次式（８）のようになる。

<Frequency domain equalization>
In the base station, an SF point FFT is performed on the signal r (t) (t = 0 to SF-1) shown in Expression (4), and a frequency domain signal R (k) configured by SF frequency components is obtained. (K = 0 to SF-1). The frequency domain signal R (k) is expressed by the following equation (5).

here,

Therefore, when the equation (6) is substituted into the equation (5), the frequency domain signal R (k) is expressed by the following equation (7).

Here, Π (k) is a complex Gaussian noise with zero mean and variance of 2σ ² · SF, and is given by the following equation (8).

ここで、ｗ_ｎ（ｋ）は拡散変調を取り除くためおよびＦＤＥのための複合ＦＤＥ重みである。遅延分割マルチアクセスでは、信号分離を遅延時間領域で行う。そのため、基地局では、ＳＦポイントＩＦＦＴを用いて、ＦＤＥ後の周波数領域信号Ｒ_ｎ＾（ｋ）（ｋ＝０〜ＳＦ−１）を遅延時間領域信号ｈ_ｎ＾（τ）（τ＝０〜ＳＦ−１）に変換する。 Then, the base station performs FDE on the frequency domain signal R (k) (k = 0 to SF-1) obtained by FFT. Specifically, the frequency domain signal R (k) (k = 0 to SF-1) is multiplied by an equalization weight w (k) (k = 0 to SF-1). Then, the frequency domain signal R _n ^ (k) after FDE is expressed by the following equation (9).

Where w _n (k) is the composite FDE weight for removing spread modulation and for FDE. In delay division multiple access, signal separation is performed in the delay time domain. Therefore, the base station uses the SF point IFFT to convert the frequency domain signal R _n ^ (k) (k = 0 to SF-1) after FDE into the delay time domain signal h _n ^ (τ) (τ = 0 to 0). SF-1).

<Signal separation>
Here, for ease of explanation, only equalization weights for removing spread modulation are considered. Then, after performing signal separation, path diversity combining is performed in the delay time domain, and the data symbol of mobile station n is demodulated. A detailed description of equalization weights for FDE will be described later.

よって、ＦＤＥ後の周波数領域信号Ｒ_ｎ＾（ｋ）は次式（１１）のようになる。

First, the frequency domain signal R (k) (k = 0 to SF-1) is multiplied by an equalization weight w _n (k) shown in the following equation (10). Here, the equalization weight w _n (k) shown in Expression (10) is a kind of zero forcing (ZF) weight.

Therefore, the frequency domain signal R _n ^ (k) after FDE is expressed by the following equation (11).

Next, the SF point IFFT is applied to the frequency domain signal R _n ^ (k) (k = 0 to SF−1), and the delay time domain signal h _n ^ (τ) (τ = 0 to SF− of the SF symbol). Convert to 1). The delay time domain signal h _n ^ (τ) (τ = 0 to SF-1) is expressed by the following equation (12).

Here, the state of the delay time domain signal h _n ^ (τ) (τ = 0 to SF-1) is shown in FIG. As shown in Equation (12) and FIG. 2, transmission data symbols of each mobile station appear at equal intervals Δ in the delay time domain. Also, since the impulse response has a spread, the transmission data symbols of each mobile station in the delay time region appear to spread. In addition, since the spread of the impulse response is within GI, if the cyclic delay unit amount Δ is determined based on the GI length, the signals of each mobile station are not overlapped, and the transmission data symbols of each mobile station are easily separated. can do.

また、図２に示すように、移動局ｎの送信データシンボルが遅延時間区間［０，Δ］で広がって分布しているため、これを遅延時間領域整合フィルタで合成する。すなわち、

を得ると、送信データシンボルの軟判定値ｄ_ｎ＾が次式（１４）のように得られる。

ここで、パスダイバーシチ合成を表す式（１４）と送信信号を表わす式（１）とを比較して、式（１４）中の

を等価チャネル利得と呼ぶ。ここで、Ｌ個のパスは独立に変動するため、等価チャネル利得の変動は浅くなる。また、一様電力遅延プロファイル

であれば、レイリーフェジング環境下での

の分布は自由度２Ｌのχ２乗分布になる。 <Path diversity synthesis>
As shown in FIG. 2, the delay time domain signal of mobile station n appears in the delay time interval [0, Δ]. Therefore, the base station separates and extracts the delay time domain signal h _n ^ (τ) (τ = 0 to SF-1) of the delay time interval [0, Δ] from the entire delay time domain, and performs path diversity combining. . Specifically, in the base station, the delay time domain signal h _n ^ (τ) (τ = 0 to SF-1) of the mobile station n is separated from the entire delay time domain, as shown in the following equation (13): It can be taken out.

Further, as shown in FIG. 2, since the transmission data symbols of mobile station n are spread and distributed in the delay time interval [0, Δ], they are synthesized by the delay time domain matched filter. That is,

Is obtained, the soft decision value d _n ^ of the transmission data symbol is obtained as in the following equation (14).

Here, the expression (14) representing the path diversity combining and the expression (1) representing the transmission signal are compared, and the expression (14)

Is called the equivalent channel gain. Here, since the L paths vary independently, the variation of the equivalent channel gain becomes shallow. Also uniform power delay profile

Then, in Rayleigh fencing environment

Is a χ-square distribution with 2 L degrees of freedom.

  As shown in Expression (14), in delay division multi-access, the same path diversity gain (or frequency diversity gain) as Rake combining in DS-CDMA can be obtained while removing MAI.

<MMSE-FDE>
Here, the equalization weight for FDE will be described in detail. Hereinafter, three MMSE-FDEs will be described.

ここで、√（２Ｓ_ｎ）ｄ_ｎＨ_ｎ（ｋ）は参照信号である。この参照信号は、遅延時間領域の［０，Δ−１］区間における移動局ｎのチャネルインパルス応答と送信データシンボルとの積ｄ_ｎｈ_ｎ（τ）が誤差最小で現れるようにする。 <1. First weight>
Here, equalization weights that minimize the error signal in the delay time region assigned to each mobile station are used. Specifically, an equalization error e (k) defined by the following equation (15) is used.

Here, √ (2S _n ) d _n H _n (k) is a reference signal. This reference signal causes the product d _n h _n (τ) of the channel impulse response of the mobile station n and the transmission data symbol in the [0, Δ−1] section of the delay time domain to appear with a minimum error.

そして、δＥ［｜ｅ（ｋ）^２｜］／δｗ_ｎ（ｋ）＝０を解いて、平均２乗等化誤差を最小とするＭＭＳＥ等化重みが、次式（１７）で与えられる。

ここで、移動局ｎのみが通信しており、さらに雑音の影響が無視できるとき、式（１７）に示すＭＭＳＥ等化重みは、次式（１８）のようになる。

上式（１８）は、式（１０）に示す等化重みと同一である。 From the fact that the data symbols of the respective mobile stations are independent from each other and the equation (7), the mean square equalization error E [| e (k) ² |] is expressed by the following equation (16).

Then, the MMSE equalization weight that solves δE [| e (k) ² |] / δw _n (k) = 0 and minimizes the mean square equalization error is given by the following equation (17).

Here, when only the mobile station n is communicating and the influence of noise can be ignored, the MMSE equalization weight shown in the equation (17) is expressed by the following equation (18).

The above equation (18) is the same as the equalization weight shown in equation (10).

Then, the base station performs FDE using the obtained MMSE equalization weight, and delays the generated frequency domain signal R _n ^ (k) (k = 0 to SF-1) using the SF point IFFT. By converting to a time domain signal h _n ^ (τ) (τ = 0 to SF-1), path diversity combining can be performed as shown in equation (14).

Thus, by using the first weight, the mean square equalization error can be minimized within the delay time region assigned to each mobile station, and the signals of the mobile station are collected in that region. be able to. That is, MAI can be avoided when viewed in the entire delay time region.

この参照信号は、遅延時間領域全体にわたって

が誤差最小で現れるようにする。ここで、遅延時間領域の［０，Δ−１］区間に移動局ｎのチャネルインパルス応答と送信データシンボルとの積ｄ_ｎｈ_ｎ（τ）が現れる。このように、各移動局のチャネルインパルス応答と送信データシンボルの積は互いに重なることなく現れるため、ＭＡＩを避けて移動局ｎのチャネルインパルス応答と送信データシンボルとの積ｄ_ｎｈ_ｎ（τ）だけを取り出すことができる。 <2. Second weight>
Here, equalization weights that minimize the error signal in the entire delay time region are used. Specifically, the above <1. The reference signal X (k) represented by the following equation (19) is used instead of the reference signal d _n H _n (k) of the first weight>.

This reference signal is used throughout the delay time domain.

Appears with minimum error. Here, the product d _n h _n (τ) of the channel impulse response of mobile station n and the transmission data symbol appears in the [0, Δ−1] section of the delay time region. Thus, to appear without the product of the channel impulse response and transmission data symbols of each mobile station overlap each other, the product of the transmitted data symbols and the channel impulse response of the mobile station n to avoid the MAI d _{n h n (τ)} Can only take out.

そして、各移動局のデータシンボルが互いに独立であることおよび式(７)より、平均２乗等化誤差Ｅ［｜ｅ（ｋ）^２｜］は次式（２１）のようになる。

よって、δＥ［｜ｅ（ｋ）^２｜］／δｗ_ｎ（ｋ）＝０を解いて、平均２乗等化誤差を最小とするＭＭＳＥ等化重みが、次式（２２）で与えられる。

ここで、移動局ｎのみが通信しており、さらに雑音の影響が無視できるとき、式（２２）に示すＭＭＳＥ等化重みは、次式（２３）のようになる。

The equalization error e (k) defined using the reference signal X (k) shown in the equation (19) is expressed by the following equation (20).

Then, from the fact that the data symbols of each mobile station are independent from each other and from the equation (7), the mean square equalization error E [| e (k) ² |] is expressed by the following equation (21).

Therefore, MMSE equalization weight that solves δE [| e (k) ² |] / δw _n (k) = 0 and minimizes the mean square equalization error is given by the following equation (22).

Here, when only the mobile station n is communicating and the influence of noise can be ignored, the MMSE equalization weight shown in the equation (22) is expressed by the following equation (23).

Then, the base station performs FDE using the obtained MMSE equalization weight, and delays the generated frequency domain signal R _n ^ (k) (k = 0 to SF-1) using the SF point IFFT. The time domain signal h _n ^ (τ) (τ = 0 to SF-1) is converted. Thus, the product d _n h _n (τ) of the channel impulse response of mobile station n and the transmission data symbol appears with a minimum error in the [0, Δ−1] section of the delay time domain. Therefore, path diversity combining can be performed in the same manner as Expression (14). Further, a path diversity gain can be obtained in the same manner as when the first weight is used.

  In this way, by using the second weight, the mean square equalization error can be minimized over the entire delay time region, and the influence of the MMSE-FDE weight of the own mobile station on the signals of other mobile stations is affected. In consideration, the signal of the mobile station can be collected in the allocated delay time region. Therefore, the influence of MAI can be further avoided as compared with the first weight.

この参照信号は、遅延時間領域にｄ_ｎδ（τ）が誤差最小で現れるようにする。 <3. Third weight>
Here, equalization weights that minimize the error signal in the cyclic delay time assigned to each mobile station are used. Specifically, a reference signal X (k) represented by the following equation (24) is used.

This reference signal causes _dn δ (τ) to appear in the delay time region with a minimum error.

ここで、式(２５)は次式（２６）のように書き表せるため、ＩＦＦＴ処理は不要となる。すなわち、ＦＤＥ自体がパスダイバーシチ合成処理、つまり、周波数ダイバーシチ合成処理と逆拡散処理とを兼ねることができる。

Here, using the SF point IFFT, the frequency domain signal R _n ^ (k) (k = 0 to SF-1) is converted into the delay time domain signal h _n ^ (τ) (τ = 0 to SF-1). Then, soft-decision value _{d n} of the transmitted data symbol ^ is given by the following equation (25).

Here, since the expression (25) can be expressed as the following expression (26), the IFFT processing is not necessary. That is, the FDE itself can serve as path diversity combining processing, that is, frequency diversity combining processing and despreading processing.

そして、各移動局のデータシンボルが互いに独立であることおよび式(７)より、平均２乗等化誤差Ｅ［｜ｅ（ｋ）^２｜］は次式（２８）のようになる。

よって、δＥ［｜ｅ（ｋ）^２｜］／δｗ_ｎ（ｋ）＝０を解いて、平均２乗等化誤差を最小とするＭＭＳＥ等化重みが次式（２９）で与えられる。

ここで、移動局ｎのみが通信しており、さらに雑音の影響が無視できるとき、式（２９）で示すＭＭＳＥ等化重みは、次式（３０）のようになる。

上式（３０）は良く知られたＺＦ−ＦＤＥである。ＺＦ等化重みは雑音強調を引き起こすが、上式（３０）のＭＭＳＥ等化重みは雑音強調を避けることができる。 The equalization error e (k) defined using the reference signal X (k) shown in Expression (24) is expressed by the following Expression (27).

Then, from the fact that the data symbols of each mobile station are independent from each other and from the equation (7), the mean square equalization error E [| e (k) ² |] is expressed by the following equation (28).

Therefore, δSE [| e (k) ² |] / δw _n (k) = 0 is solved and the MMSE equalization weight that minimizes the mean square equalization error is given by the following equation (29).

Here, when only the mobile station n is communicating and the influence of noise can be ignored, the MMSE equalization weight represented by the equation (29) is as the following equation (30).

The above equation (30) is a well-known ZF-FDE. Although the ZF equalization weight causes noise enhancement, the MMSE equalization weight of the above equation (30) can avoid noise enhancement.

  In this way, by using the third weight, it is possible to perform processing combining both path diversity combining and despreading, and minimize the mean square equalization error with the cyclic delay time assigned to each mobile station. And MAI can be avoided.

<Configuration of mobile station and base station>
Next, configurations of the mobile station and the base station according to the present embodiment will be described.

  FIG. 3 shows the configuration of mobile station 100 according to the present embodiment, and FIG. 4 shows the configuration of base station 200 according to the present embodiment. In order to avoid complicated explanation, FIG. 3 shows components related to transmission of uplink transmission data and reception of control information data in downlink, which are closely related to the present invention. Illustration and description of components related to reception of transmission data in the downlink are omitted. Similarly, FIG. 4 shows components related to reception of transmission data on the uplink and transmission of control information data on the downlink, which are closely related to the present invention, for transmission of transmission data on the downlink. The illustration and explanation of the related components are omitted.

  In mobile station 100 shown in FIG. 3, transmission data is input to encoding section 101. Encoding section 101 encodes transmission data and outputs the encoded transmission data to modulation section 102.

  Modulation section 102 modulates the transmission data input from encoding section 101 to generate a transmission data symbol, and outputs the transmission data symbol to spreading section 104.

  The spreading code generation unit 103 converts the cyclic delay spreading code c ((t−nΔ) mod SF) (t = 0 to SF−1) into the decoding unit 110 as described in <Principle of delay division multi-access> above. Is generated based on the cyclic delay amount nΔ included in the control information data input from. Then, spreading code generating section 103 outputs generated cyclic delay spreading code c ((t−nΔ) mod SF) to spreading section 104.

Spreading unit 104, the as described in <transmit signal>, cyclic delay spread code c input transmission data symbols _{d n} received as input from modulating section 102 from the spread code generating unit 103 ((t-nΔ) mod The transmission chip sequence s _n (t) shown in Expression (1) is generated by spreading with SF). Then, spreading section 104 outputs the generated transmission chip sequence s _n (t) to GI insertion section 105.

The GI insertion unit 105 inserts a GI at the head of the transmission chip sequence s _n (t) input from the spreading unit 104 and adds a cyclic prefix to the GI as described in <Principle of delay division multi-access> above. To do.

  Radio transmission section 106 performs transmission processing such as D / A conversion, amplification and up-conversion on the signal after GI insertion, and transmits the signal from antenna 107 to base station 200.

  On the other hand, radio receiving section 108 receives control information data symbols transmitted from base station 200 via antenna 107, and performs reception processing such as down-conversion and A / D conversion on the control information data symbols.

  Demodulation section 109 demodulates the control information data symbol after reception processing, and outputs the demodulated control information data to decoding section 110.

  Decoding section 110 decodes the demodulated control information data and outputs it to spreading code generation section 103.

  On the other hand, in base station 200 shown in FIG. 4, radio reception section 202 receives a signal transmitted from mobile station 100 via antenna 201, and performs reception processing such as down-conversion and A / D conversion on the received signal. I do.

  The GI removal unit 203 removes the GI from the signal after reception processing, and obtains a reception signal r (t) (t = 0 to SF-1) shown in Expression (4).

  The FFT unit 204 performs FFT on the reception signal input from the GI removal unit 203 for each transmission chip sequence, and converts the time domain signal into a frequency domain signal. Specifically, the FFT unit 204 performs an SF point FFT on the received signal r (t) (t = 0 to SF-1) of the SF chip shown in Equation (4), and is shown in Equation (7). The frequency domain signal R (k) (k = 0 to SF-1) is converted. Then, the FFT unit 204 outputs the obtained frequency domain signal R (k) (k = 0 to SF−1) to the FDE / despreading unit 205.

The FDE / despreading unit 205, as described above in <Frequency domain equalization> and <Signal separation>, receives the frequency domain signal R (k) (k = 0 to SF-1) input from the FFT unit 204. FDE and despreading are performed. Specifically, the FDE / despreading unit 205 performs the equalization weight w (k) shown in the equation (10) with respect to the frequency domain signal R (k) (k = 0 to SF-1) shown in the equation (7). ) (K = 0 to SF-1) to generate a frequency domain signal R _n ^ (k) (k = 0 to SF-1) shown in Equation (11). As described in <MMSE-FDE> above, the FDE / despreading unit 205 uses the equalization weight represented by any one of the equations (17), (22), or (29) as the equalization weight. May be used. Then, the FDE / despreading unit 205 outputs the generated frequency domain signal R _n ^ (k) to the IFFT unit 206.

As described in <Signal Separation> above, IFFT section 206 performs IFFT for each transmission chip sequence on the frequency domain signal input from FDE / despreading section 205 and converts it to a delay time domain signal. Specifically, the IFFT unit 206 performs SF point IFFT on the frequency domain signal R _n ^ (k) (k = 0 to SF−1) shown in Expression (11), and shows in Expression (12). Delay time domain signal h _n ^ (τ) (τ = 0 to SF-1). Then, IFFT section 206 outputs delay time domain signal h _n ^ (τ) to separation / synthesis section 207.

As described in the above <Path diversity combining>, the separating / combining unit 207 determines that the delay time domain signal h _n ^ (τ) (τ = 0 to SF-1) input from the IFFT unit 206 is expressed by the equation (13). , The delay time domain signal of the mobile station n is separated and extracted, and path diversity combining is performed. Separation / combination section 207 then outputs soft decision value d _n ^ of the transmission data symbol shown in Expression (14) obtained by path diversity combining to demodulation section 208.

Demodulation section 208 demodulates the soft decision value d _n ^ of the transmission data symbol input from demultiplexing / combining section 207 to obtain demodulated data. Demodulation section 208 then outputs the obtained demodulated data to decoding section 209.

  The decoding unit 209 decodes the demodulated data input from the demodulating unit to obtain received data.

  On the other hand, uplink estimation section 210 estimates the channel estimation value of each mobile station from the pilot signal of each mobile station input from GI removal section 203. Then, uplink estimation section 210 outputs the obtained channel estimation value of each mobile station to delay time estimation section 211.

  Delay time estimation section 211 estimates the maximum delay time of each mobile station from the channel estimation value of each mobile station input from uplink estimation section 210. Then, the delay time estimation unit 211 outputs the obtained maximum delay time of each mobile station to the delay amount determination unit 212.

  The delay amount determination unit 212 determines the largest delay time among the maximum delay times of each mobile station input from the delay time estimation unit 211 as the cyclic delay unit amount Δ for all mobile stations. Then, the delay amount determination unit 212 outputs the determined cyclic delay amount unit Δ to the control information generation unit 213. Note that the cyclic delay unit amount Δ determined by the delay amount determination unit 212 is the same as the GI length in the GI insertion unit 105 (FIG. 3) of the mobile station 100.

  The control information generation unit 213 generates control information data including the cyclic delay unit amount Δ input from the delay amount determination unit 212. Then, the control information generation unit 213 outputs the generated control information data to the encoding unit 214.

  The encoding unit 214 encodes the control information data input from the control information generation unit 213 and outputs the control information data to the modulation unit 215.

  Modulation section 215 modulates control information data input from coding section 214 to generate control information data symbols. Modulation section 215 then outputs the generated control information data symbol to radio transmission section 216.

  Radio transmission section 216 performs transmission processing such as D / A conversion, amplification and up-conversion on the control information data symbol input from modulation section 215 and transmits the result from antenna 201 to mobile station 100.

  Thus, according to the present embodiment, the mobile station uses a cyclic delay spreading code generated by giving a cyclic delay nΔ, which is an integral multiple of the cyclic delay unit amount Δ, to one spreading code. Thereby, in the base station, since the data of each mobile station appears orthogonally in the delay time region, the data of each mobile station can be easily separated. In addition, since the data of each mobile station separated by the base station is received with a spread within the delay time region assigned to each mobile station, frequency diversity gain can be obtained by performing path diversity combining. Excellent BER characteristics can be obtained. Therefore, according to the present embodiment, frequency diversity gain can be obtained while avoiding MAI, and throughput can be improved.

  In the present embodiment, the maximum delay time of each mobile station is calculated from the signal before FDE and determined as the cyclic delay unit amount Δ, but the maximum delay time of each mobile station is calculated from the signal after FDE, The longest delay time may be determined as the cyclic delay unit amount Δ for all mobile stations. As a result, the influence of the side lobe (MAI leaking to the adjacent delay time region) caused by the FDE can be reflected in the determination of the cyclic delay unit amount Δ, so that MAI can be appropriately avoided and throughput can be further improved. The effect that it can be obtained.

(Embodiment 2)
The present embodiment is different from the first embodiment in that a different cyclic delay amount is determined for each mobile station.

  Only differences from the first embodiment will be described below.

4 determines the cyclic delay amount for each mobile station based on the maximum delay time of each mobile station input from the delay time estimation unit 211. The delay amount determination unit 212 illustrated in FIG. Specifically, as shown in FIG. 5, the maximum delay time at mobile station n−1 is τ _{max, n−1} , the maximum delay time at mobile station n is τ _{max, n} , and the maximum delay time at mobile station n + 1 is When the maximum delay time at τ _{max, n + 1} and the mobile station n + 2 is τ _{max, n + 2} , the delay amount determining unit 212 determines that the cyclic delay amount Δ _n−1 of the mobile station n ₋₁ is 0, the mobile station n the cyclic delay amount delta _n of determining tau _max, the _n-1, the mobile station n + 1 of the cyclic delay amount delta _{n + 1} to determine _{τ max, n-1 + τ} max, to _n, the mobile station n + 2 of the cyclic delay amount delta _{n + 2} Is determined as τ _{max, n−1} + τ _{max, n} + τ _{max, n + 1} . Then, the delay amount determination unit 212 outputs the determined cyclic delay amount of each mobile station to the control information generation unit 213.

  The control information generation unit 213 generates control information data including the cyclic delay amount of each mobile station input from the delay amount determination unit 212. Then, the control information generation unit 213 outputs the generated control information data to the encoding unit 214.

On the other hand, the spreading code generator 103 (FIG. 3) of each mobile station generates a cyclic delay spreading code based on the cyclic delay amount of the own station among the cyclic delay amounts included in the input control information data. Specifically, as shown in FIG. 6, the spreading code generation unit 103 of the mobile station n−1 circulates the spreading code as it is without shifting the spreading code because the cyclic delay amount Δ _n−1 = 0. Generated as a delay spread code. Further, spreading code generator 103 of the mobile station n, as shown in FIG. 6, to produce a cyclic delay spread code obtained by shifting the spread code by delta _n Since cyclic delay amount delta _{n =} tau _max, with _n-1 . Similarly, as shown in FIG. 6, the spreading code generation unit 103 of the mobile station n + 1 has a cyclic delay obtained by shifting the spreading code by Δn _{+ 1} because the cyclic delay amount is Δn _{+ 1} = τmax _{, n−1} + τmax _{, n.} The spreading code is generated by the spreading code generation unit 103 of the mobile station n + 2 because the cyclic delay amount Δ _{n + 2} = τ _{max, n−1} + τ _{max, n} + τ _{max, n + 1} , and the spreading code is shifted by Δ _{n + 2.} Generate a spreading code.

When each mobile station spreads the transmission data symbols of each mobile station using the spreading code shown in FIG. 6, the base station 200 obtains a delay time domain signal as shown in FIG. Here, the delay time domain signal shown in FIG. 7 is based on the mobile station n. In addition, the cyclic delay amount Δ _{n + 3} shown in FIG. 7 is assumed to be Δ _{n + 2} = τ _{max, n−1} + τ _{max, n} + τ _{max, n + 1} + τ _{max, n + 2} . As shown in FIG. 7, the delay time interval of mobile station n−1 is [Δ _n−1 −Δ _n , 0], and the delay time interval of mobile station n is [0, Δ _{n + 1} −Δ _n ], The delay time interval of the station n + 1 is [Δ _{n + 1} −Δ _n , Δ _{n + 2} −Δ _n ], and the delay time interval of the mobile station n + 2 is [Δ _{n + 2} −Δ _n , Δ _{n + 3} −Δ _n ].

In the mobile station n + 1 having a small maximum delay amount shown in FIG. 6, the interval of the delay time interval [Δ _{n + 1} −Δ _n , Δ _{n + 1} −Δ _n ] is narrowed as shown in FIG. 7, while the maximum delay amount shown in FIG. In mobile station n with a large delay, the interval of the delay time interval [0, Δ _{n + 1} −Δ _n ] is widened as shown in FIG. That is, in the delay time domain, the delay time domain signals of each mobile station appear at unequal intervals according to the maximum delay amount of each mobile station. This minimizes the overall delay time of the delay time domain signal.

  Thus, according to the present embodiment, the cyclic delay amount for each mobile station is made different from each other based on the maximum delay amount of each mobile station. Thereby, in the delay time region, the delay time region signal of each mobile station appears at an interval corresponding to the cyclic delay amount of each mobile station. Therefore, delay division multi-access can be performed with a minimum delay time, so that a frequency diversity gain can be obtained more efficiently than in the first embodiment.

  In this embodiment, the cyclic delay amount for each mobile station is determined based on the maximum delay amount of each mobile station from the signal before FDE, but based on the maximum delay amount of each mobile station from the signal after FDE. You may decide. As a result, the fact that the side lobe spread caused by the FDE differs for each mobile station can be reflected in the determination of the cyclic delay amount, so that the frequency diversity gain can be obtained more efficiently.

(Embodiment 3)
In the present embodiment, a case will be described in which the number of mobile stations exceeding the number of mobile stations SF / Δ accessible using one spreading code accesses the base station.

  Only differences from the first embodiment will be described below.

  In the present embodiment, a PN code (spreading factor SF) and two orthogonal codes (spreading factor SF) orthogonal to each other are used.

The spreading code generation unit 103 shown in FIG. 3 generates a product code of a PN code and an orthogonal code as a new spreading code. Then, spreading code generation section 103 generates a cyclic delay spreading code from the product code based on the cyclic delay amount, and outputs the generated cyclic delay spreading code to spreading section 104 as in the first embodiment. Specifically, as illustrated in FIG. 8, the spreading code generation unit 103 converts the product code of the PN code c (t) and the orthogonal code c _m (t) (m = 0, 1) to a new spreading code c. 'Generate as _m (t). That is, a new spreading code c ′ _m (t) is obtained from c (t) × c _m (t).

Thus, as shown in FIG. 8, the spread code c _'0 is generated from the PN code c (t) orthogonal code _c 0 and (t) (t), and, an orthogonal code c and PN code c (t) ₁ (t) and 'even when the same cyclic delay amount Enuderuta against ₁ (t), cyclic delay spread code c' spreading code c generated from cyclic delay and _{0 ((t-nΔ) mod} SF) Since the spread codes c ′ ₁ ((t−nΔ) mod SF) are orthogonal to each other, two different cyclic delay spread codes can be generated with the same cyclic delay amount. For example, eight cyclic delay spread codes can be generated by using two orthogonal codes for PN codes that can generate four (SF / Δ = 4) cyclic delay spread codes.

  Therefore, even when the number of mobile stations exceeding SF / Δ = 4 accesses the base station, the spreading code generator 103 of each mobile station uses a plurality of orthogonal codes to exceed the number SF / Δ = 4. A cyclic delay spreading code can be generated. For example, when the number of mobile stations is 5 to 8, two orthogonal spreading codes are required.

  Thus, according to the present embodiment, even when the number of mobile stations that can be accessed by using one spreading code is accessed, the same effect as in the first embodiment can be obtained. it can.

  The embodiment of the present invention has been described above.

であることのみ上りリンクで用いた場合と相違し、上記＜ＭＭＳＥ−ＦＤＥ＞の式（１７）、式（２２）および式（２９）の３つのＭＭＳＥ等化重みは次式（３２）で得られる。

Note that although cases have been described with the above embodiments where delay division multiple access is used in the uplink, similar effects can be obtained when the present invention is applied to the downlink. Specifically, when delay division multiple access is used in the downlink, the channel impulse response is

Unlike the case of being used in the uplink, the three MMSE equalization weights of the above <MMSE-FDE> equations (17), (22), and (29) are obtained by the following equation (32). It is done.

を一定とした最適電力割り当て問題に帰着する。つまり、基地局は、総送信電力を一定に保ちつつ、移動局毎に独立に送信電力を割り当てる最適電力割り当てを行う。 Further, when the present invention is used in the downlink, power control may be performed on each mobile station independently. However, the total power in all mobile stations

This results in an optimal power allocation problem with a constant. That is, the base station performs optimal power allocation that allocates transmission power independently for each mobile station while keeping the total transmission power constant.

  Further, although cases have been described with the above embodiment as examples where the present invention is configured by hardware, the present invention can also be realized by software.

  Each functional block used in the description of each of the above embodiments is typically realized as an LSI which is an integrated circuit. These may be individually made into one chip, or may be made into one chip so as to include a part or all of them. The name used here is LSI, but it may also be called IC, system LSI, super LSI, or ultra LSI depending on the degree of integration.

  Further, the method of circuit integration is not limited to LSI's, and implementation using dedicated circuitry or general purpose processors is also possible. An FPGA (Field Programmable Gate Array) that can be programmed after manufacturing the LSI, or a reconfigurable processor that can reconfigure the connection and setting of circuit cells inside the LSI may be used.

  Furthermore, if integrated circuit technology comes out to replace LSI's as a result of the advancement of semiconductor technology or a derivative other technology, it is naturally also possible to carry out function block integration using this technology. Biotechnology can be applied.

  The present invention can be applied to a mobile communication system or the like.

The figure which shows the cyclic delay spreading code which concerns on Embodiment 1 of this invention. The figure which shows the delay time-domain signal which concerns on Embodiment 1 of this invention. Block configuration diagram of a mobile station according to Embodiment 1 of the present invention Block configuration diagram of a base station according to Embodiment 1 of the present invention The figure which shows maximum delay amount estimation of each mobile station which concerns on Embodiment 2 of this invention. The figure which shows the cyclic delay spreading code of each mobile station which concerns on Embodiment 2 of this invention. The figure which shows the delay time-domain signal which concerns on Embodiment 2 of this invention. The figure which shows the cyclic delay spreading code of each mobile station which concerns on Embodiment 3 of this invention.

Explanation of symbols

DESCRIPTION OF SYMBOLS 100 Mobile station 200 Base station 101,214 Encoding part 102,215 Modulation part 103 Spreading code production part 104 Spreading part 105 GI insertion part 106,216 Radio transmission part 107,201 Antenna 108,202 Radio reception part 109,208 Demodulation part 110, 209 Decoding unit 203 GI removal unit 204 FFT unit 205 FDE / despreading unit 206 IFFT unit 207 Separation / combination unit 210 Uplink estimation unit 211 Delay time estimation unit 212 Delay amount determination unit 213 Control information generation unit

Claims (5)

Receiving means for receiving a superimposed signal on which a plurality of signals spread with a second spreading code generated by cyclically delaying the first spreading code by a different cyclic delay amount for each of a plurality of wireless communication mobile station devices;
A weight corresponding to the cyclic delay amount for each of the plurality of radio communication mobile stations, and for the superimposed signal using the weight that minimizes an error between the delay time domain signal of the superimposed signal and a reference signal Equalization means for performing frequency domain equalization,
Separating means for separating the superimposed signal after frequency domain equalization into a plurality of delay time domain signals for each of the plurality of radio communication mobile station devices;
Combining means for performing a combining process for each of the plurality of delay time domain signals;
A wireless communication base station apparatus comprising: The equalization means performs the frequency domain equalization using the weights that minimize the error in each of the delay time domains for each of the plurality of radio communication mobile station devices.
The radio communication base station apparatus according to claim 1. The equalization means performs the frequency domain equalization using the weights that minimize the error in the delay time domain of the plurality of radio communication mobile station devices as a whole.
The radio communication base station apparatus according to claim 1. The equalization means performs the frequency domain equalization using the weights that minimize the error in delay times corresponding to the cyclic delay amounts for the plurality of radio communication mobile station devices, respectively.
The radio communication base station apparatus according to claim 1. A second spreading code is generated by cyclically delaying the first spreading code by a different cyclic delay amount for each of the plurality of radio communication mobile station apparatuses;
Spreading the transmission signal with the second spreading code;
Signal spreading method.

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