JP2007043895A - Stepping motor driving apparatus and driving method - Google Patents

Abstract

In a stepping motor driving device, waveform distortion is prevented from occurring particularly in the vicinity of a zero cross due to detection delay.
A driving device for a stepping motor includes detection means for detecting a current supplied to a winding of the stepping motor, first offset addition means (40) for adding an offset to the output of the detection means, and a first Amplifying means (22) for amplifying the output of the offset adding means, reference signal generating means (14) for generating a reference signal representing the current limit value, and second offset addition for adding an offset to the output of the reference signal generating means The means (41), the switching means (5) for supplying power to the winding in the conductive state, and stopping the power supply to the winding in the non-conductive state; Comprises a pulse width modulation control means (15) for making the switching means non-conductive when the output of the second offset voltage adding means exceeds the output.
[Selection] Figure 1

Description

  The present invention relates to a technique for driving a stepping motor with low noise and low vibration.

  Conventionally, stepping motors are used for various position control and the like. The stepping motor includes a rotor and a stator including a plurality of phases of windings, and the rotor is configured to rotate by a unit angle and stop. Further, the rotor is rotated by a target angle and stopped without feedback control by controlling the number of rotation steps. Such characteristics in the operation of the stepping motor are suitable for position control applications.

  In recent years, stepping motors are used for adjustment of aperture, focus, zoom, etc. as optical system actuators in photographing electronic devices such as DSC (Digital Still Camera) and DVC (Digital Video Camera). .

  The operation of a stepping motor used in a moving image photographing electronic device is particularly required to have low noise and low vibration. This is because the noise generated by the stepping motor is caught by the built-in microphone of the device and recorded as noise, and the vibration causes blurring and the like to deteriorate the recorded image quality. In response to this requirement, for example, Patent Document 1 discloses a driving technique for reducing the noise and vibration of the operation of the stepping motor.

  FIG. 15 is a configuration diagram of a conventional stepping motor driving device described in Patent Document 1. In FIG. Here, only the components necessary for explanation of the principle are described, and the stepping motor includes a plurality of phases of windings. However, since the components provided for each winding are the same, the windings of one phase and the windings are concerned. Only the components provided in the line are described.

  Hereinafter, the conventional stepping motor driving apparatus will be described with reference to FIG. In addition, since the component provided for every winding is the same, the other winding which comprises a stepping motor, and the component provided in another winding are abbreviate | omitted, and the component provided in a winding is demonstrated. .

  The pulse width modulation control unit 15 includes a comparator 16, a flip-flop 17, a reference pulse generation unit 18, and an energization logic unit 19. The reference pulse generation unit 18 sets the flip-flop 17 for each pulse width modulation period (hereinafter referred to as “PWM period”), so that the energization logic unit 19 causes the transistors 6 and 9 constituting the switching unit 5 to Any one of the transistors 7 and 8 is turned on at regular intervals with a combination and timing at which no penetration occurs. The current direction switching signal (PHASE) input to the energization logic unit 19 determines which of the transistors 6 and 9 and the transistors 7 and 8 is to be conducted, and the direction of the current flowing through the winding 3 is determined.

  During the conduction period of the transistors 6 to 9, power is supplied from the power source 1 to the winding 3, and the current flowing through the winding 3 increases. Hereinafter, a period in which the flip-flop 17 is set and the current flowing through the winding 3 is increased by supplying power to the winding 3 is referred to as a “PWM ON period”.

  The current supplied from the power source 1 to the winding 3 by the conduction of the transistors 6 to 9 is detected by the feeding current measuring means 20 and output to the comparator 16 as a detected current value. The feeding current measuring means 20 includes a detection resistor 21, a sense amplifier 22, and gain setting resistors 23 and 24. The sense amplifier 22 and the gain setting resistors 23 and 24 constitute an amplifier circuit 25. The gain of the amplifier circuit 25, that is, the gain from the input to the output of the sense amplifier 22, is set by the gain setting resistors 23 and 24. The current supplied to the winding 3 flows into the detection resistor 21, and the voltage generated at the end of the detection resistor 21 is input to the sense amplifier 22. The sense amplifier 22 outputs a voltage obtained by multiplying the input voltage by a gain to the comparator 16 as a detected current value.

  In the following description of the operation, the current flowing through the winding 3 detected by the feeding current measuring means 20 is simply referred to as “detected current value”. The reference signal generating means 14 generates a step wave that increases and decreases on the step, and outputs it to the comparator 16 as a reference signal representing a current limit value. The reference signal representing the current limit value generated by the reference signal generation unit 14 is a current target value for the winding current 3.

  The comparator 16 compares the input detected current value with the current target value, and resets the flip-flop 17 when the detected current value exceeds the current target value. By resetting the flip-flop 17, the energization logic unit 19 cuts off both the transistors 7 and 8 constituting the switching means 5. During the period when the flip-flop 17 is reset and both the transistors 7 and 8 are cut off, the power supply from the power source 1 to the winding 3 is cut off, and the current flowing through the winding 3 is reduced by the regenerative operation.

  When both transistors 6 and 9 are off during the period when both transistors 7 and 8 are off, the current flowing through winding 3 is either one of the flywheel diodes 11 and 12 or the flywheel. Regeneration is performed with either of the diodes 10 and 13. When both the transistors 6 and 9 are turned on during the period when both the transistors 7 and 8 are cut off, the current flowing through the winding 3 is regenerated by the transistors 6 and 9.

  When only one of the transistors 6 and 9 is turned on while both the transistors 7 and 8 are cut off, the flywheel diode connected to the non-conductive transistor is forward biased. Regeneration is performed by one of the wheel diodes 11 and 12 and one of the transistors 6 and 9. If the flywheel diode connected to the non-conductive transistor is reverse-biased, it is regenerated by either the flywheel diode 10 or 13 and the transistor 6 or 9.

  Hereinafter, a period in which the flip-flop 17 is reset and the current flowing through the winding 3 is reduced by the regenerative operation is referred to as a “PWM off period”. During the PWM off period, the current flowing through the winding 3 decreases, but again, the output signal of the reference pulse generator 18 changes to the PWM on period again by setting the flip-flop 17, and the winding 3 is turned on. The flowing current increases again.

  With the above operation, the average current supplied to the winding 3 operates so as to approach the current target value. By increasing or decreasing the current target value stepwise, the average current supplied to the winding 3 increases or decreases stepwise. The windings of the phases other than the winding 3 operate in the same manner, so that the step progresses. The stepping motor 2 rotates at a rotational speed corresponding to the speed to be performed.

  Next, the current target value generated by the reference signal generation unit 14 will be described. FIG. 16 is a diagram showing a relationship between a reference signal and a current direction switching signal in a conventional stepping motor driving apparatus.

  The reference signal generation means 14 generates a step wave that increases and decreases in a step-like manner, and outputs it to the comparator 16 as a current target value. As the current target value increases or decreases stepwise, the stepping motor rotates by a unit angle. The progress of the step of the current target value is performed by inputting a CLK (clock signal) instructing the progress of the step, but may be configured by measuring the step progress interval by a timer. The period in which the step of the current target value proceeds is determined by the period of the input CLK period or the timer that determines the step progression interval, and the period in which the stepping motor rotates the unit angle by the period in which the step of the current target value proceeds. As a result, the rotation period of the stepping motor is determined. The target current value is preferably a sinusoidal signal from the viewpoint of low noise and low vibration. The reference signal generation means 14 generates a staircase wave obtained by sampling a sine wave.

  In FIG. 16, the staircase wave sampled corresponding to 64 steps is shown as a current target value. As the steps progress, each value of the staircase wave obtained by sampling the sine wave at each step is output in order, so that a staircase wave obtained by sampling the sine wave is output.

  Note that the direction of the current flowing through the winding 3 is specified by a current direction switching signal as shown in FIG. That is, the magnitude of the staircase wave indicates the magnitude of the current target value, and the current direction switching signal indicates the direction of the current. Furthermore, in order to avoid a steep current change due to a step-like level change, a staircase wave smoothed by an integrating means such as a low-pass filter is output to the comparator 16 as a current target value.

  Also, it is not always necessary to use a staircase that samples a sine wave. From the viewpoint of mounting area, a staircase that samples an approximate sine wave or a staircase that deviates from a sine wave may be used. If a steep current change due to is acceptable, an unsmoothed staircase wave may be output to the comparator 16.

JP 2004-215385 A

  According to the conventional stepping motor driving device, there is a problem that the waveform distortion of the current flowing through the winding 3 occurs due to the response delay of the sense amplifier 22.

  Hereinafter, the problem will be described with reference to FIGS.

  FIG. 17 is a general sense amplifier configuration diagram and a circuit diagram showing an operating point of a PWM off period. The sense amplifier 22 includes P-channel MOS transistors 30a, 30b, 30c, N-channel MOS transistors 31a, 31b, 31c, differential transistors 32a, 32b, a current source 33, and a phase compensation capacitor 34. The gain setting resistors 23 and 24 have the same resistance value “R” and double the gain.

  18 is a diagram showing a general sense amplifier configuration diagram and a PWM on-period operating point, and FIGS. 19A to 19C are diagrams showing current paths at the time of phase switching (“35” in the figure is a current path). Is shown). FIGS. 20A to 20E are current waveform diagrams when the current target value in the conventional stepping motor driving apparatus is large, and FIGS. 21A to 21E are small current target values in the conventional stepping motor driving apparatus. FIG. 22A and FIG. 22B are waveform diagrams showing current waveform distortion in the conventional stepping motor driving apparatus.

  During the PWM off period, since the above-described regenerative operation is performed, no current flows through the detection resistor 21. Therefore, the ground voltage is input to the non-inverting input terminal (+) of the sense amplifier 22. In FIG. 17, it is shown as Vin + = 0V.

  Here, the sense amplifier 22 cannot output a voltage equal to or lower than the voltage (minimum voltage) determined by the constant current flowing from the P channel MOS transistor 30c and the ON resistance of the N channel MOS transistor 31c. Even if an amplifier of the type called Rail-to-Rail is used, if the minimum voltage of the sense amplifier 22 is 0V, 0V cannot be output.

  In FIG. 17, the minimum voltage is 20 mV, and Vout = 0.02V. At this time, half of 10 mV is input to the inverting input terminal (−) of the sense amplifier 22 from the configuration. FIG. 17 shows Vin− = 0.01V. In the state shown in FIG. 17, the virtual ground relationship of the sense amplifier 22 is broken and the differential transistors 32 a and 32 b are not in an equilibrium state, so that a voltage substantially equal to the voltage of the power supply 1 is applied to the phase compensation capacitor 34. FIG. 17 shows Vc = VCC.

  Hereinafter, the state in which the virtual ground relationship is broken as described above is referred to as “the sense amplifier loop is out”. When the capacitance value of the phase compensation capacitor 34 is Ccomp, the charge of [Ccomp × (VCC−20 mV)] is accumulated in the phase compensation capacitor 34.

  FIG. 18 shows the operating point of the sense amplifier during the PWM ON period. Since a current flows through the detection resistor 21 during the PWM ON period, a voltage determined by the current flowing through the detection resistor 21 and the resistance value of the detection resistor 21 is input to the non-inverting input terminal of the sense amplifier 22. In FIG. 18, it is shown as Vin + = 0.2V. 0.2V is input to the inverting input terminal of the sense amplifier 22, and the sense amplifier 22 outputs 0.4V.

In FIG. 18, Vin− = 0.2V and Vout = 0.4V are shown. At this time, the virtual ground relationship of the sense amplifier 22 is maintained, and the differential transistors 32a and 32b are in an equilibrium state. For this reason, the phase compensation capacitor 34 has a gate voltage of the N-channel MOS transistor 31c such that the voltage determined by the constant current flowing from the P-channel MOS transistor 30c and the ON resistance of the N-channel MOS transistor 31c is 0.4V. Vgs1 is applied. In FIG. 18, it is shown as Vc = Vgs1.

  Hereinafter, the state in which the virtual ground relationship as described above is maintained is referred to as “the sense amplifier loop is maintained”. In the phase compensation capacitor 34, [Ccomp × (Vgs1−0.4V)] is accumulated.

  Naturally, no matter how much voltage is input from the end of the detection resistor 21, the sense amplifier 22 does not respond and cannot correctly determine the detected current value in a state where the loop of the sense amplifier is removed. In order to correctly determine the detected current value after the transition from the PWM off period to the PWM on period, it is necessary to transition from the operating point shown in FIG. 17 to the operating point shown in FIG. Charge becomes a challenge.

  As described above, the phase compensation capacitor 34 has a charge of [Ccomp × (VCC−20 mV)] at the operating point shown in FIG. 17, and a charge of [Ccomp × (Vgs1−0.4V)] at the operating point shown in FIG. Accumulated. When VCC = 5.02V and Vgs1 = 1.0V, the detected current value cannot be determined correctly unless the charge difference at both operating points, that is, [Ccomp × 4.4V] is discharged. The time required for this discharge is the time until the sense amplifier 22 can correctly determine the detected current value after the transition from the PWM off period to the PWM on period, that is, “detection delay”.

  The discharge is performed by the difference between the current flowing through the N-channel MOS transistor 31b and the current flowing through the differential transistor 32b. As the differential transistor 32b is completely cut off (the higher the voltage input to the non-inverting terminal of the sense amplifier 22 after the transition to the PWM on period, the more the differential transistor 32b is cut off), the time required for discharging Becomes shorter and the detection delay becomes shorter.

  Conversely, the smaller the voltage input to the non-inverting terminal of the sense amplifier 22 after the transition to the PWM ON period, that is, the smaller the current flowing through the detection resistor 21, the more the differential transistor 32 b is cut off. Low, the time required for discharge becomes longer, and the detection delay becomes longer. Therefore, the detection delay appears more conspicuously as the drive step has a smaller current target value, such as drive steps = 0, 31 to 33, 63 shown in FIG. The drive step with a small current target value is hereinafter referred to as “zero cross” in the sense that it is in the vicinity of the point where the polarity of the current is reversed. In FIG. 16, zero crossing is indicated by point A.

  As described above, the state in which the loop of the sense amplifier is disconnected during the PWM off period has been described. However, the loop of the sense amplifier may be disconnected during the PWM on period.

  Hereinafter, a case where the sense amplifier loops out during the PWM ON period will be described with reference to FIGS. 16 and 19.

  In the PWM ON period, as shown in FIG. 19A, power is supplied to the winding 3 and current flows into the power supply current measuring means 20. In FIG. 19A, the transistors 8 and 9 are conducting and the transistors 6 and 7 are shut off. On the other hand, in the PWM off period, as shown in FIG. 19B, the regenerative operation is performed and no current flows to the feeding current measuring means 20. In FIG. 19B, the transistor 9 is conductive and the transistors 6, 7 and 8 are cut off. During the PWM off period shown in FIG. 19B, the current flowing through the winding 3 is attenuated. However, in FIG. 16, in the driving step shown by driving step = 32 or 0, the current itself is small, so the PWM off. During the period, the voltage applied across the winding 3 is small, and as a result, the current flowing through the winding 3 is difficult to attenuate.

  Here, when the driving step proceeds for a short time, that is, when the rotational speed of the stepping motor is high, the current of the winding 3 is not attenuated to 0 at the time of transition to the next driving step. In the state where the current of the winding 3 remains, when the drive step transitions from 32 to 33, or from 0 to 1, the current direction switching signal is switched, and the current of the winding 3 is reversed. As shown in (), a transistor different from that in the previous driving step is turned on. In FIG. 19C, the transistors 8 and 9 are cut off and the transistors 6 and 7 are conductive. At this time, the current of the winding 3 flows from the ground to the power source. The current also flows back from the ground to the feeding current measuring means 20, and as a result, the current flows back from the ground to the detection resistor 21.

  Therefore, a negative potential is generated at the end of the detection resistor 21 and the negative potential is also input to the sense amplifier 22. When a negative potential is input, the sense amplifier 22 is out of the sense amplifier loop for the same reason as that when the ground voltage is input, and a detection delay occurs. Therefore, immediately after the switching of the current direction switching signal PHASE, that is, immediately before the current of the winding 3 is reversed, as indicated by A in FIG. 16, even after the transition to the PWM ON period. , The sense amplifier loop goes out and a detection delay occurs.

  Next, a current waveform when a detection delay occurs will be described with reference to FIGS. 20 and 21, the part indicated by A is a detection delay part. In FIG. 20, the detected current value does not exceed the current target value during the detection delay. In this case, even if there is a detection delay, the detection operation is not adversely affected.

  When the amount of attenuation during the PWM off period is large, it takes time to reach the current target value after the transition to the PWM on period. Therefore, the current target value is not reached during the detection delay, and is shown in FIG. So there is a high possibility that there is no adverse effect. The higher the current target value, the greater the amount of attenuation in the regenerative operation during the PWM off period. Therefore, there is a high possibility that there will be no adverse effect or a small effect in the driving step with a high current target value.

  In FIG. 21, the detected current value exceeds the target current value during the detection delay. In this case, since the detection is not performed during the detection delay, the PWM ON period is continued even though the current target value is already exceeded, and as a result, the current target value is deviated. When the amount of attenuation during the PWM off period is small, it does not take time to reach the current target value after the transition to the PWM on period, so the current target value is reached during the detection delay. There is a high possibility that there will be an adverse effect.

  The lower the current target value, the smaller the amount of attenuation in the regenerative operation during the PWM off period. Therefore, there is a high possibility that the driving step with a low current target value will have an adverse effect. This means that the waveform deviates significantly from the target current, particularly near the zero cross. That is, as shown in part A of FIG. 22, the current deviates from the current target value near the zero cross, and the current waveform is distorted.

  As described above, according to the conventional stepping motor driving apparatus, there is a problem that the waveform distortion is noticeably generated particularly in the vicinity of the zero cross due to the detection delay of the sense amplifier. Due to the waveform distortion, the effect of reducing vibration and noise is not sufficient particularly in application to electronic equipment for photographing, and there is still a demand for further reduction in vibration and noise in the stepping motor operation.

  The present invention has been made in view of the above-described conventional problems, and an object of the present invention is to provide a stepping motor driving device and a driving method that can reduce vibration and noise in the stepping motor operation.

  The stepping motor driving apparatus according to the first aspect of the present invention includes a detecting means for detecting a current supplied to a winding provided in the stepping motor, a first offset adding means for adding an offset to the output of the detecting means, and a first Amplifying means for amplifying the output of the offset adding means; reference signal generating means for generating a reference signal representing a current limit value; second offset adding means for adding an offset to the output of the reference signal generating means; Switching means for supplying power to the winding in a non-conducting state and stopping the power supply to the winding in a non-conducting state, and after the switching means is turned on every predetermined cycle, the output of the amplifying means is added to the second offset voltage Pulse width modulation control means for switching the switching means to a non-conductive state when the output of the means is exceeded.

  The stepping motor drive apparatus according to the second aspect of the present invention includes a detection unit that detects a current supplied to a winding included in the stepping motor, a first offset addition unit that adds an offset to the output of the detection unit, An amplifying means for amplifying the output of the offset adding means, an offset subtracting means for subtracting the offset from the output of the amplifying means, a reference signal generating means for generating a reference signal representing a current limit value, and supplying power to the winding in the conductive state And switching means for stopping the power supply to the winding in the non-conducting state, and after the switching means is turned on every predetermined cycle, the output of the offset subtracting means has a current limit value represented by the reference signal. And pulse width modulation control means for switching the switching means to a non-conducting state at the time of exceeding.

  The stepping motor drive apparatus according to the third aspect of the present invention includes a detection unit that detects a current supplied to a winding included in the stepping motor, a first offset addition unit that adds an offset to the output of the detection unit, Amplifying means for amplifying the output of the offset adding means, reference signal generating means for generating a reference signal representing a current limit value, and supplying power to the winding in the conductive state and stopping power supply to the winding in the non-conductive state And a pulse for making the switching means non-conductive when the output of the amplifying means exceeds the current limit value represented by the reference signal after the switching means is turned on at predetermined intervals. Width modulation control means.

  A stepping motor drive apparatus according to a fourth aspect of the present invention includes a detection unit that detects a current supplied to a winding included in a stepping motor, a first offset addition unit that adds an offset to the output of the detection unit, and a detection unit. Selection means for selecting and outputting the output of the first offset addition means, amplification means for amplifying the output of the selection means, and reference signal generation means for generating a reference signal representing a current limit value And switching means for supplying power to the winding in the conductive state and stopping power supply to the winding in the non-conductive state, and after the switching means is turned on every predetermined cycle, the output of the amplification means is determined by the reference signal. When the current limit value shown is exceeded, the pulse width modulation control means for switching the switching means to the non-conductive state and the selector drive for controlling the selection means And a signal generator.

  The selector drive signal generation unit determines that the pulse width modulation control unit has made the switching unit non-conductive, and outputs the determination result. The selection unit receives the determination result from the selector drive signal generation unit, and selects and outputs either the output of the detection unit or the output of the first offset addition unit according to the determination result.

  In the stepping motor driving apparatus according to the fourth aspect, the selecting means selects the output of the first offset adding means and switches the switching means during the entire period when the switching means is non-conductive according to the determination result of the selector driving signal generation unit. You may make it select the output of a detection means in the whole period when a means is a conduction | electrical_connection state.

  Alternatively, the selection unit selects the output of the first offset addition unit in a partial period in which the switching unit is in the non-conduction state, and in the remaining period in which the switching unit is in the non-conduction state and in the entire period in which the switching unit is in the conduction state. You may make it select the output of a detection means.

  Alternatively, the selecting means selects the output of the first offset adding means during a part of the period when the switching means is in the conducting state and the whole period when the switching means is in the non-conducting state, and the detecting means is in the remaining period when the switching means is in the conducting state. May be selected.

  Alternatively, the selection unit selects the output of the first offset addition unit in the partial period in which the switching unit is in the conductive state and the partial period in which the switching unit is in the non-conductive state, and the switching unit is in the remaining period in the conductive state. You may make it select the output of a detection means in the remaining period when a switching means is a non-conduction state.

  The selector drive signal generation unit may further determine that the direction of winding current has been commanded and output the determination result. In this case, the selection means selects the output of the first offset addition means in the entire period in which the switching means is in a non-conducting state and a certain period after the direction switching of the winding current is commanded according to the determination result, You may select the output of a detection means in the period except the fixed period after the direction change of winding current was instruct | indicated from the whole period when the switching means is a conduction | electrical_connection state.

  Alternatively, the selection unit selects the output of the first offset addition unit during a partial period in which the switching unit is in a non-conduction state and a certain period after the direction switching of the winding current is commanded. The output of the detection means may be selected in a period excluding a certain period after the command to switch the direction of the winding current from the remaining period in the non-conduction state and the entire period in which the switching means is in the conduction state. Good.

  Alternatively, the selection means may include the first offset adding means in a partial period in which the switching means is in a conducting state, in a whole period in which the switching means is in a non-conducting state, and in a certain period after the direction switching of the winding current is commanded. The output of the detection means may be selected in a period excluding a certain period after the switching of the direction of the winding current is commanded from the remaining period in which the switching means is in the conductive state.

  Alternatively, the selection unit may add the first offset in a partial period in which the switching unit is in a conductive state, a partial period in which the switching unit is in a non-conducting state, and a certain period after the direction switching of the winding current is commanded. The output of the detecting means is selected in a period excluding a certain period after the switching of the direction of the winding current is commanded from the remaining period in which the switching means is in the conductive state and the remaining period in which the switching means is in the non-conductive state. May be selected.

  A stepping motor driving method according to a fifth aspect of the present invention includes a step of detecting a current supplied to a winding included in a stepping motor, a step of adding a first offset to the detected current value, Amplifying the current value to which the offset has been added; generating a reference signal representing the current limit value; adding a second offset to the reference signal; Controlling the conduction of the switching means for stopping the power supply to the winding in the state. In the controlling step, the switching means is turned on every predetermined period, and when the amplified current value exceeds the value of the reference signal to which the second offset is added, the switching means is turned off. .

  A stepping motor driving method according to a sixth aspect of the present invention includes a step of detecting a current supplied to a winding included in a stepping motor, a step of adding a first offset to the detected current value, Amplifying the current value to which the offset has been added; subtracting the second offset from the amplified current value; generating a reference signal representing a current limit value; And controlling the conduction of the switching means for stopping the power supply to the winding in the non-conduction state. In the controlling step, the switching means is turned on every predetermined period, and the switching means is made non-conductive when the current value obtained by subtracting the second offset exceeds the current limit value represented by the reference signal. Put it in a state.

  A stepping motor drive method according to a seventh aspect of the present invention includes a step of detecting a current supplied to a winding provided in a stepping motor, a step of adding an offset to the detected current value, and a current obtained by adding the offset A step of amplifying the value, a step of generating a reference signal representing a current limit value, a step of controlling the conduction of the switching means for supplying power to the winding in the conductive state and stopping the power supply to the winding in the non-conductive state; including. In the controlling step, the switching unit is turned on every predetermined cycle, and when the amplified current value exceeds the current limit value represented by the reference signal, the switching unit is turned off.

  A stepping motor driving method according to an eighth aspect of the present invention includes a step of detecting a current supplied to a winding included in a stepping motor, a step of adding an offset to the detected current value, and a detected current value. Selecting a current value to which the offset is added; amplifying the selected current value; generating a reference signal representing a current limit value; When the switching means for stopping the power supply to the winding in the non-conduction state is turned on every predetermined cycle, the switching means is turned on when the amplified current value exceeds the current limit value represented by the reference signal. A non-conducting state. In the selecting step, it is determined that the switching means has been turned off, and a current value is selected according to the determination result.

  According to the stepping motor driving apparatus and the driving method of the present invention, detection delay can be eliminated by adding an offset to the input of the detecting means, and generation of waveform distortion particularly near the zero cross can be prevented. Further, by adding the second offset so as to cancel the offset added to the input of the detection means, it is possible to prevent the occurrence of a detection current deviation due to the added offset. Further, by preventing the occurrence of waveform distortion due to detection delay and preventing the occurrence of a detection current deviation when an offset is added, it is possible to reduce noise and vibration in the stepping motor drive device.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings. In the following description of each drawing, members corresponding to those already described are assigned the same reference numerals, and detailed description thereof is omitted.

(First embodiment)
A stepping motor driving apparatus according to a first embodiment of the present invention will be described with reference to FIGS. 1 and 16 and FIGS.

  FIG. 1 is a block diagram illustrating a configuration example of the stepping motor driving device according to the first embodiment. The stepping motor has a plurality of phase windings, and the components provided for each winding are the same for each phase. Therefore, only the components provided in one phase winding will be described.

  FIG. 16 is a diagram showing the relationship between the reference signal and the current direction switching signal PHASE in the conventional stepping motor driving apparatus. The reference signal and the current direction switching signal PHASE are not different from the conventional example described above in this embodiment.

  In FIG. 1, a stepping motor driving apparatus receives power from a power source 1 and drives a stepping motor 2. A stepping motor 2 to be controlled includes a winding 3 and a rotor 4.

  The stepping motor driving apparatus includes a switching unit 5 that controls power supply to the winding 3, a reference signal generation unit 14 that generates a reference signal representing a current limit value, a pulse width modulation control unit 15, and a supply current measurement unit 20. Including.

  The switching means 5 includes transistors 6 to 9 serving as a power supply path to the winding 3 and flywheel diodes 10 to 13. The pulse width modulation control unit 15 includes a comparator 16, a flip-flop 17, a reference pulse generation unit 18, and an energization logic unit 19.

  The stepping motor driving apparatus according to the first embodiment is configured to perform pulse width modulation control (hereinafter referred to as PWM control) so that the average current supplied to the winding 3 gradually approaches the current limit value generated by the reference signal generation unit 14. More specifically, PWM control is performed by a current chopper method. In the following description of the operation, the current flowing through the winding 3 detected by the feeding current measuring means 20 is simply referred to as “detected current value”, and the reference signal representing the current limit value generated by the reference signal generating means 14 is referred to as “current detection value”. This is simply called “current target value”.

  First, the current target value generated by the reference signal generation unit 14 will be described. The operation of the reference signal generation means 14 is the same as that of the conventional stepping motor driving device, generates a step wave that increases and decreases in steps, and outputs it as a current target value. As the current target value increases or decreases stepwise, the stepping motor rotates by a unit angle. The progress of the step of the current target value is made by the input of CLK instructing the progress of the step, but the same effect can be obtained even if the configuration is made by measuring the step progress interval by the timer.

  The period in which the step of the current target value proceeds is determined by the period of the input CLK period or the timer that determines the step progression interval, and the period in which the stepping motor rotates the unit angle by the period in which the step of the current target value proceeds. As a result, the rotation period of the stepping motor is determined. The current target value is preferably a sinusoidal signal from the viewpoint of low noise and low vibration. The reference signal generation means 14 generates a staircase wave obtained by sampling a sine wave.

  In FIG. 16, the staircase wave sampled corresponding to 64 steps is shown as a current target value. As each step progresses, each value of the staircase wave obtained by sampling the sine wave is output in order at each step, thereby outputting a staircase wave obtained by sampling the sine wave.

  The current direction of the current flowing through the winding 3 is specified by a current direction switching signal PHASE as shown in FIG. That is, the magnitude of the staircase wave indicates the magnitude of the current target value, and the current direction switching signal PHASE indicates the current direction. Furthermore, in order to avoid a steep current change due to a step-like level change, a staircase wave smoothed by an integrating means such as a low-pass filter is output as the current target value.

  Also, it is not always necessary to use a staircase wave sampled from a sine wave. From the viewpoint of mounting area, a staircase wave sampled from an approximate sine wave or a staircase wave deviating from a sine wave can be used. When a steep current change due to a level change is allowable, an unsmoothed step wave can be output.

  The pulse width modulation control unit 15 includes a comparator 16, a flip-flop 17, a reference pulse generation unit 18, and an energization logic unit 19, and PWM-controls the current in the winding 3.

  Hereinafter, the operation related to the PWM control will be described in detail with reference to FIG. FIG. 2 shows time changes of main signals related to the PWM control operation in addition to the current waveform of the stepping motor driving device.

  The reference pulse generator 18 outputs a signal with a fixed period instructing the start of power supply to the winding 3 to the set terminal of the flip-flop 17 and sets the flip-flop 17 at a fixed period. Transistors constituting the switching means 5 when the flip-flop 17 is set so that the energizing logic unit 19 that has received the output signal of the flip-flop 17 supplies the transistors 6 to 9 with a gate signal for conducting or blocking the transistors. 6 and 9 and any of transistors 7 and 8 are made conductive at a timing and a combination in which no penetration from the power source to the ground occurs, thereby starting power feeding to winding 3 and current flowing through winding 3 Begins to increase.

  The current direction switching signal PHASE input to the energization logic unit 19 determines which of the transistors 6 and 9 and the transistors 7 and 8 is to be conducted, and the direction of the current flowing through the winding 3 is determined. A period in which the flip-flop 17 is set and the current flowing through the winding 3 is increased by supplying power to the winding 3 is referred to as a “PWM ON period”. The power supply to the winding 3 is started and the transition to the PWM ON period is performed for each signal with a fixed period generated by the reference pulse generator 18, so that the fixed period generated by the reference pulse generator 18 is set as the PWM period. Works.

  The transistors 6 and 7 are supplied with a gate signal for turning on the transistors, and the transistors 8 and 9 are supplied with a gate signal for shutting off the transistors. A current path in the PWM ON period in this case is shown as a current path 42 in FIG. In FIG. 3A, a current flows from the power source 1 to the ground via the transistor 6, the winding 3, the transistor 7, and the feeding current measuring means 20, whereby the power is fed from the power source 1 to the winding 3. The

  In the present embodiment, the feeding current measuring means 20 is arranged between the ground and the switching means 5 to detect the current flowing to the ground via the feeding current measuring means 20, but the feeding current measurement is performed. By disposing the means 20 between the power supply 1 and the switching means 5, it is possible to detect the current flowing from the power supply 1 via the feeding current measuring means 20, and even in this case, the present embodiment The same effect can be obtained.

  However, in this case, the detected current value and the current target value are not the ground reference but a reference signal of the power source 1, and the magnitude relationship between the detected current value and the current target value is opposite to that at the ground reference.

  In FIG. 3A, the current flowing through the winding 3 flows through the current path 42 and is detected by the feeding current measuring means 20. The feeding current measuring means 20 outputs a detected current value flowing through the winding 3. Immediately after the transition to the PWM ON period, there may be an overshoot in the detected current value.

  The overshoot mainly occurs when a discharge current of the parasitic capacitance of the switching unit 5, for example, a current that discharges the charge of the parasitic capacitance between the drain and gate of the transistor 7 flows into the feeding current measuring unit 20. . Therefore, if the feeding current measuring means 20 and the comparator 16 follow the overshoot, even if the current of the winding 3 does not actually exceed the current target value, the detected current value exceeds the current target value due to the overshoot. There is a case where it is erroneously detected.

  In this case, current detection by the feeding current measuring means 20 and the comparator 16 is masked during a certain time during which overshoot may occur (hereinafter referred to as “mask time”). In the present embodiment, the current detection can be masked by using a set priority flip-flop as the flip-flop 17 and setting the pulse width of the signal output from the reference pulse generator 18 to a pulse width corresponding to the mask time. . That is, while the reference pulse generation unit 18 outputs a pulse width corresponding to the mask time, the flip-flop 17 is reset because the flip-flop 17 has set priority even if the comparator 16 detects erroneously due to overshoot. There is no. Further, the same effect can be obtained by fixing the output of the feeding current measuring means 20 or the output of the comparator 16 during the mask time.

  The comparator 16 receives a signal indicating the detected current value and a signal indicating the current target value. In the present embodiment shown in FIG. 1, the signal indicating the detected current value is the output of the feeding current measuring means 20, the signal indicating the current target value is the current target value output from the reference signal generating means 14, and the second target value. It is an addition result with the offset by the offset addition means 41. Detailed operation and utility of the second offset adding means 41 will be described later.

  The comparator 16 compares the input signal indicating the detected current value with the signal indicating the current target value, and when the signal indicating the detected current value exceeds the signal indicating the current target value, the flip-flop 17 is reset and regenerative operation is started. A period in which the flip-flop 17 is reset and the current flowing through the winding 3 decreases due to the regenerative operation is referred to as a “PWM off period”.

  In the present embodiment, the relationship between the set and reset of the flip-flop 17 and the PWM on-period and the PWM off-period transitions to the PWM on-period by the set of the flip-flop 17 and transitions to the PWM off-period by resetting the flip-flop 17. Although it is configured as described above, it is possible to configure by changing the relationship, and even in this case, the same effect as in the present embodiment can be obtained.

  By resetting the flip-flop 17, the energization logic unit 19 supplies a gate signal for shutting off the transistor to the transistors 7 and 8. When both the transistors 7 and 8 are cut off, a transition is made to the PWM off period, the power supply to the winding 3 is cut off, and the current flowing through the winding 3 starts to decrease due to the regenerative operation. In the case where the transistors 6 and 7 are conducting immediately before the transition to the PWM off period, a current path in the PWM off period is shown as a current path 42 in FIG.

  In FIG. 3B, the current flowing through the winding 3 is regenerated and reduced via the flywheel diode 11 and the transistor 6. It is also possible to make both transistors 6 and 9 conductive for the purpose of reducing the amount of decrease in the current flowing through the winding 3 and reducing the ripple of the current flowing through the winding 3 during the PWM off period. The power consumption due to the flywheel diode 11 is replaced with the power consumption due to the on-resistance of the transistor 9 and is reduced, so that the amount of current flowing through the winding 3 during the PWM off period is reduced. The current path in this case is shown as a current path 42 in FIG.

  In FIG. 3C, the current flowing through the winding 3 is regenerated and reduced via the transistors 6 and 9. It is also possible to cut off both the transistors 6 and 9 for the purpose of quickly reducing the current flowing through the winding 3 during the PWM off period. A current path in this case is shown as a current path 42 in FIG.

  In FIG. 3D, the current flowing through the winding 3 is regenerated and reduced via the flywheel diodes 10 and 11. In this embodiment, the flywheel diodes 10 to 13 are provided, but a body diode composed of the back gate and the drain of the transistors 6 to 9 can be substituted. Further, for the purpose of reducing the amount of decrease in the current flowing through the winding 3 during the PWM off period, a Schottky barrier diode can be used as the flywheel diodes 10 to 13.

  After the transition to the PWM off period due to the reset of the flip-flop 17, the reference pulse generator 18 sets the flip-flop 17 at regular intervals to repeat the above operation. By repeating this increase in current during the PWM ON period and decrease in current during the PWM OFF period, the average current supplied to the winding 3 gradually approaches the current target value. By increasing / decreasing the current target value on the step, the average current supplied to the winding 3 increases / decreases on the step, and the windings of other phases other than the winding 3 operate in the same manner, The stepping motor 2 rotates at a rotational speed corresponding to the traveling speed.

  Next, the configuration and operation of the feeding current measuring means 20 will be described. The feeding current measuring means 20 detects the current supplied from the power source 1 to the winding 3 by the conduction of the transistors 6 to 9, and outputs the detected current value.

  The feeding current measuring unit 20 in the present embodiment includes a detection resistor 21 that is a detection unit, an amplification circuit 25 that is an amplification unit, and a first offset addition unit 40. The amplifier circuit 25 includes a sense amplifier 22 and gain setting resistors 23 and 24. The gain of the amplifier circuit 25, that is, the gain from the input to the output of the sense amplifier 22, is determined by the gain setting resistors 23 and 24. Is set.

  In FIG. 1, the detection resistor 21 is used as detection means. However, as shown in FIG. 4, the ON resistance of the MOS transistor 44 when the gate application voltage 45 is applied is used to detect the detection resistor 21 in FIG. It is possible to obtain the same effect. The current supplied to the winding 3 flows to the ground through the detection resistor 21, and a voltage determined by the resistance value of the detection resistor 21 and the flowing current is generated at both ends of the detection resistor 21. The voltage generated across the detection resistor 21 is added with an offset by the first offset adding means 40 and then input to the non-inverting input terminal (+) of the sense amplifier 22 constituting the amplifier circuit 25. The sense amplifier 22, that is, the amplifying unit 25 outputs a voltage obtained by amplifying the input voltage by a gain multiple to the comparator 16 as a detected current value.

  A specific example of offset addition by the first offset addition means 40 will be described with reference to FIGS.

  In FIG. 5A, the first offset adding means 40 is constituted by a resistor 47 and a current source 48. The voltage determined by the resistance value of the resistor 47 and the current value of the current source 48 is an offset to be added. A diode may be used in place of the resistor 47.

  In FIG. 5 (b), the first offset adding means 40 includes a current source 48, a gate applied voltage 49, and a MOS transistor 50. The voltage determined by the ON resistance of the MOS transistor 50 determined by the gate applied voltage 49 and the current value of the current source 48 is an offset to be added. MOS transistor 50 can be realized by either a P-channel MOS transistor or an N-channel MOS transistor.

  In FIG. 5C, the first offset adding means 40 is composed of a source follower composed of a MOS transistor 51 and a current source 52, and serves as an offset for adding a gate-source voltage. Instead of the source follower by the MOS transistor 51, an emitter follower by a bipolar transistor can be used.

  5D, P-channel MOS transistors 30a to 30c, N-channel MOS transistors 31a to 31c, differential transistors 32a and 32b, and a current source 33 constitute a sense amplifier 22. The first offset adding means 40 is composed of differential transistors 32 a and 32 b constituting the sense amplifier 22. The offset caused by the difference in size or number between the differential transistors 32a and 32b is an offset to be added. The offset is not generated by the difference in size or number, but the current flowing through the differential transistor 32a and the differential transistor 32b is unbalanced by adjusting one current of the differential transistor 32a and the differential transistor 32b. It is also possible to generate it.

  Although FIG. 5 shows the first offset addition means 40, the same applies to the second offset addition means. However, in the specific example corresponding to FIG. 5D, the sense amplifier 22 is replaced with the comparator 16 for the second offset adding means.

  6 shows the PWM off period operating point when the offset is added by the first offset adding means 40, and FIG. 7 shows the PWM on period operating point. The operation of the sense amplifier when the offset is added by the first offset adding means will be described with reference to FIGS.

  6 and 7, the gain setting resistors 23 and 24 have the same resistance value R, that is, the gain is doubled. In the PWM off period, since the regenerative operation is performed, no current flows through the detection resistor 21. At this time, the voltage at the end of the detection resistor 21 becomes the ground voltage.

  In FIG. 6, the offset by the first offset adding means 40 is 20 mV. Therefore, 20 mV obtained by adding the voltage at the end of the detection resistor 21 and the offset by the first offset adding means 40 is input to the non-inverting input terminal of the sense amplifier 22, and Vin + = 0. It is shown as 02V. 0.02V is input to the inverting input terminal of the sense amplifier 22, and the sense amplifier 22 outputs 0.04V. In FIG. 6, Vout = 0.04V.

  Here, the sense amplifier 22 cannot output a voltage equal to or lower than the lowest voltage determined by the constant current flowing from the P channel MOS transistor 30c and the ON resistance of the N channel MOS transistor 31c. Even if an amplifier of the type called Rail-to-Rail is used, if the lowest potential of the sense amplifier 22 is 0V, 0V cannot be output.

  In the conventional stepping motor driving apparatus described above shown in FIG. 17, the output of the sense amplifier 22 is 20 mV, which is the lowest voltage, during the PWM off period. At this time, the relationship between the virtual grounds of the sense amplifier 22 collapses and the differential transistors 32a and 32b are not in an equilibrium state. Therefore, a voltage substantially equal to the voltage of the power supply 1 is applied to the phase compensation capacitor 34.

  On the other hand, in FIG. 6 of the present embodiment, because of the offset by the first offset adding means 40, the output of the sense amplifier 22 is 40 mV exceeding 20 mV, which is the lowest voltage, and the relationship of the virtual ground of the sense amplifier 22 Is maintained. At this time, since the differential transistors 32a and 32b are in an equilibrium state, the phase compensation capacitor 34 has a voltage determined by the constant current flowing from the P-channel MOS transistor 30c and the on-resistance of the N-channel MOS transistor 31c of 0.04V. The gate voltage Vgs2 of the N channel MOS transistor 31c is applied. In FIG. 6, it is shown as Vc = Vgs2. Hereinafter, the state in which the virtual ground relationship is maintained is referred to as “sense amplifier loop is maintained”, and the state in which the virtual ground relationship is lost is referred to as “sense amplifier loop is out”. In FIG. 6, the charge of [Ccomp × (Vgs2−0.04V)] is accumulated in the phase compensation capacitor 34.

  Next, the PWM on-period operating point is shown in FIG. Since a current flows through the detection resistor 21 during the PWM ON period, the voltage across the detection resistor 21 is a voltage determined by the current flowing through the detection resistor 21 and the resistance value of the detection resistor 21. When the voltage across the detection resistor 21 is 0.2 V and the offset by the first offset adding means 40 is 20 mV, the non-inverting input terminal (+) of the sense amplifier 22 is connected to the voltage across the detection resistor 21 and the first voltage. 0.22V obtained by adding the offset by the offset adding means 40 of 1 is input, and Vin + = 0.22V is shown in FIG.

  0.22V is input to the inverting input terminal (−) of the sense amplifier 22, and the sense amplifier 22 outputs 0.44V. In FIG. 7, Vin− = 0.22V and Vout = 0.44V are shown. At this time, the virtual ground relationship of the sense amplifier 22 is maintained, and the differential transistors 32a and 32b are in an equilibrium state. Therefore, the phase compensation capacitor 34 includes a constant current flowing from the P-channel MOS transistor 30c and an N-channel. The gate voltage Vgs3 of the N-channel MOS transistor 31c is applied such that the voltage determined by the on-resistance of the MOS transistor 31c is 0.44V. In FIG. 7, it is shown as Vc = Vgs3. In the phase compensation capacitor 34, [Ccomp × (Vgs3−0.44V)] is accumulated.

  In the present embodiment shown in FIG. 6, since the sense amplifier loop is maintained even during the PWM off period, the sense amplifier loop is maintained out of the state at the transition from the PWM off period to the PWM on period. No transition to state occurs. As in this embodiment, in order to maintain the sense amplifier loop even during the PWM off period, the amplification factor of the amplifier circuit 25 is α, the offset by the first offset addition means 40 is OFFSET, and the sense amplifier 22 outputs When the lowest possible voltage is Vmin, OFFSET needs to satisfy the condition of the following formula (1) at the minimum.

OFFSET ≧ Vmin / α (1)
More specifically, it is necessary to add a margin that allows for the variation in each value in equation (1) and the offset of the sense amplifier 22 itself.

  As described above, the charge of [Ccomp × (Vgs2−0.04V)] is accumulated at the operating point shown in FIG. 6, and the charge of [Ccomp × (Vgs3−0.44V)] is accumulated at the operating point shown in FIG. Yes. From the square characteristics of the input gate voltage and current in the MOS transistor, there is no significant difference between Vgs2 and Vgs3, so Vgs2 = Vgs3 for simplicity.

  At this time, the charge of the phase compensation capacitor 34 that must be discharged at the transition from the PWM off period to the PWM on period is [Ccomp × 0.4 V], and the discharge in the conventional example shown in FIGS. 17 and 18 is performed. Compared with [Ccomp × 4.4V] of the power charge, it is as small as about 1/10. In addition, when the current is smaller and the voltage across the detection resistor 21 is smaller, the charge to be discharged is further smaller.

  For example, when the voltage across the detection resistor 21 is 0.1 V, the charge to be discharged in the conventional example is [Ccomp × 4.2 V], and in this embodiment, [Ccomp × 0.2 V]. It is smaller than the conventional example by about 1/20. As described above, the time required for this discharge is the time until the sense amplifier 22 can correctly determine the detected current value after the transition from the PWM off period to the PWM on period, that is, a detection delay. In the present embodiment, there are few charges to be discharged, and as a result, no detection delay occurs. Therefore, detection delay can be eliminated, and waveform distortion particularly near the zero cross can be prevented.

  As already described in the description of the conventional stepping motor driving device, the current direction switching signal PHASE is switched while the current of the winding 3 remains, and when the current of the winding 3 is reversed, Current flows from ground to the power source. At this time, the current also flows back to the sense amplifier 22 from the ground, and as a result, the current also flows back to the detection resistor 21 from the ground. For this reason, a negative potential is generated at the end of the detection resistor 21. The condition shown by the equation (1) is for the case where the voltage at the end of the detection resistor 21 is the ground voltage. In order to eliminate the detection delay even for a negative potential, the following equation (Equation 2) is used. It is necessary to satisfy the conditions.

OFFSET ≧ (Vmin / α) + Vneg (2)
The amplification factor of the amplifier circuit 25 is α, the offset by the first offset adding means 40 is OFFSET, the minimum voltage that can be output by the sense amplifier 22 is Vmin, and the maximum negative potential that can be generated at the end of the detection resistor 21 is Vneg. To do.

  When the first offset adding means 40 adds an offset satisfying the expression (2), the current direction switching signal PHASE is switched, and even when the current of the winding 3 is inverted, the detection delay is eliminated and the waveform distortion is eliminated. Can be prevented.

  Next, the second offset adding means 41 connected to the pulse width modulation control unit 15 will be described. Due to the offset added by the first offset adding means 40, the detected current value output from the feeding current measuring means 20 actually deviates from a value corresponding to the amount of current flowing through the detecting resistor 21.

  In FIG. 2E, the solid line is the detected current value after the offset addition, and the broken line is the detected current value before the offset addition. FIG. 2E shows a state where the output of the feeding current measuring unit does not become the ground voltage due to the offset even in the PWM off period in which no current flows.

  In the present embodiment, in order to prevent the current flowing through the winding 3 from deviating from the current target value due to the offset added by the first offset adding means 40, the offset by the second offset adding means 41 is set to the current target value. to add. As shown in FIG. 2 (e), the output from the feeding current measuring means 20, that is, the output from the amplifier circuit 25 is a value obtained by multiplying the offset added by the first offset adding means 40 and the amplification factor of the amplifier circuit 25. However, it deviates from the value corresponding to the amount of current actually flowing through the detection resistor 21 as indicated by the broken line.

  By making the offset of the current target value provided by the second offset adding means 41 equal to the value obtained by multiplying the offset added by the first offset adding means 40 and the amplification factor of the amplifier circuit 25, the comparator 16 actually Both the current target value and the current detection value input to the same value are shifted by the same value, and the difference voltage input to the comparator 16 is equal to the case where there is no offset. For this reason, the magnitude determination of the current target value and the current detection value by the comparator 16 is the same as the case where both offsets are not provided, and it is possible to prevent the detection current from being shifted due to the added offset.

  As described above, according to the stepping motor driving apparatus according to the first embodiment, detection delay can be eliminated by adding an offset to the input of the sense amplifier 22, and waveform distortion especially in the vicinity of the zero cross can be prevented. Can do. Further, by adding the second offset in order to cancel the offset added to the input of the sense amplifier 22, it is possible to prevent a detection current shift due to the added offset. From this, in this embodiment, the noise reduction and the vibration reduction of the stepping motor driving device are realized.

(Second Embodiment)
The stepping motor driving apparatus according to the second embodiment of the present invention is provided with an offset subtracting means instead of the second offset adding means in the first embodiment, so that the stepping motor driving according to the first embodiment is provided. Different from the device. Hereinafter, the difference from the first embodiment will be mainly described with reference to FIGS. 8 and 9, and detailed description of the same operation as that of the first embodiment will be omitted.

  FIG. 8 is a block diagram illustrating a configuration example of a stepping motor driving apparatus according to the second embodiment, and FIG. 9 is a diagram illustrating an example of an offset subtraction unit according to the second embodiment. The stepping motor has a plurality of phase windings, and the components provided for each winding are the same for each phase. Therefore, only the components provided in one phase winding 3 will be described.

  The stepping motor driving apparatus according to this embodiment shown in FIG.

  In the present embodiment shown in FIG. 8, the signal indicating the current target value is the current target value output from the reference signal generating unit 14, and the signal indicating the detected current value is offset from the output of the feeding current measuring unit 20. This is the result of subtracting the offset by 55. Detailed operation and utility of the offset subtracting means 55 will be described later. The comparator 16 compares the input signal indicating the detected current value with the signal indicating the current target value, and resets the flip-flop 17 when the signal indicating the detected current value exceeds the signal indicating the current target value. Transition to the PWM off period.

  9A to 9D show specific examples of offset subtraction by the offset subtracting means 55. FIG.

In FIG. 9A, the offset subtracting means 55 includes a resistor 47 and a current source 48.
The voltage determined by the resistance value of the resistor 47 and the current value of the current source 48 is an offset to be subtracted. A diode may be used in place of the resistor 47.

  In FIG. 9B, the offset subtracting means 55 includes a current source 48, a gate applied voltage 49, and a MOS transistor 50. The voltage determined by the ON resistance of the MOS transistor 50 determined by the gate applied voltage 49 and the current value of the current source 48 is an offset to be subtracted. The MOS transistor 50 can be realized by either a P-channel MOS transistor or an N-channel MOS transistor.

  In FIG. 9C, the offset subtracting means 55 is composed of a source follower composed of a MOS transistor 51 and a current source 52, and serves as an offset for subtracting the gate-source voltage. Instead of the source follower by the MOS transistor 51, an emitter follower by a bipolar transistor can be used.

  9D, P-channel MOS transistors 56a to 56c, N-channel MOS transistors 57a to 57c, differential transistors 58a and 58b, and a current source 59 constitute a comparator 16. The offset subtracting means 55 includes differential transistors 58 a and 58 b that constitute the comparator 16. An offset caused by the difference in the size or number of the differential transistors 58a and 58b is an offset to be subtracted. The offset may not be generated by the difference in size or number, but may be generated by adjusting the current of one of the differential transistors 58a and 58b to make the current flowing through the differential transistors 58a and 58b unbalanced. Is possible.

  Next, the configuration and operation of the supply current measuring unit 20 of the present embodiment will be described. The feeding current measuring means 20 detects a current supplied from the power source 1 to the winding 3 by the conduction of the transistor 6 to the transistor 9 and outputs it as a detected current value. The feeding current measuring unit 20 in the present embodiment includes a detection resistor 21 that is a detection unit, an amplification circuit 25 that is an amplification unit, and a first offset addition unit 40.

  The amplifier circuit 25 includes a sense amplifier 22 and gain setting resistors 23 and 24. The gain of the amplifier circuit 25, that is, the gain from the input to the output of the sense amplifier 22, is set by the gain setting resistors 23 and 24. The

  In FIG. 8, the detection resistor 21 is used as the detection means. However, as shown in FIG. 4, the detection resistance in FIG. 8 is obtained by using the on-resistance of the MOS transistor 44 when the gate applied voltage 45 is applied. It is also possible to obtain the same action as 21. The current supplied to the winding 3 flows to the ground through the detection resistor 21, and a voltage determined by the resistance value of the detection resistor 21 and the flowing current is generated at the end of the detection resistor 21. The voltage generated at the end of the detection resistor 21 is added to the offset by the first offset adding means 40 and then input to the non-inverting input terminal of the sense amplifier 22 constituting the amplifier circuit 25. The sense amplifier 22, that is, the amplifier circuit 25 outputs a voltage obtained by amplifying the input voltage by a gain multiple to the offset subtracting unit 55.

  Also in the present embodiment shown in FIG. 8, since the sense amplifier loop is maintained by the first offset adding means 40 during the PWM off period as in the first embodiment, the PWM on period is changed from the PWM off period. In the transition to, the transition from the state where the loop of the sense amplifier is out of the maintained state does not occur.

  As already described in the first embodiment, the current direction switching signal PHASE is switched by setting the offset by the first offset adding means 40 to be an offset that satisfies the above-described equations (1) and (2). Even when the current of the winding 3 is reversed, the detection delay can be eliminated, and waveform distortion particularly near the zero cross can be prevented.

  Next, the offset subtracting means 55 will be described. In the present embodiment, the second offset adding means in the first embodiment does not exist, and an offset subtracting means 55 is provided instead. Also in the present embodiment, the output value from the feeding current measuring unit 20 corresponds to the amount of current actually flowing through the detection resistor 21 because of the offset added by the first offset adding unit 40 as in the first embodiment. It will deviate from the value to be. The output from the feeding current measuring means 20 is input to the offset subtracting means 55, and the value obtained by subtracting the offset by the offset subtracting means 55 is detected current value in order to prevent the current flowing through the winding 3 from deviating from the current target value. To the comparator 16.

  As described in the first embodiment, the output from the feeding current measuring unit 20, that is, the output from the amplifier circuit 25 is obtained by multiplying the offset added by the first offset adding unit 40 by the amplification factor of the amplifier circuit 25. This value deviates from the value corresponding to the amount of current actually flowing through the detection resistor 21 shown by the broken line in FIG.

  By making the offset by the offset subtracting means 55 equal to the value obtained by multiplying the offset added by the first offset adding means 40 by the amplification factor of the amplifier circuit 25, the offset added by the first offset adding means 40 is The offset subtracted by the offset subtracting means 55 is canceled to ± 0. For this reason, the magnitude determination of the current target value and the current detection value by the comparator 16 is the same as when both offsets are not provided, and a detection current deviation due to the added offset can be prevented.

  As described above, according to the stepping motor driving apparatus according to the second embodiment, detection delay can be eliminated by adding an offset to the input of the sense amplifier 22, and waveform distortion particularly near the zero cross can be prevented. . Further, by subtracting the offset to cancel the offset added to the input of the sense amplifier 22, it is possible to prevent a detection current shift due to the added offset. From this, in this embodiment, the noise reduction and the vibration reduction of the stepping motor driving device are realized.

(Third embodiment)
The stepping motor driving apparatus according to the third embodiment of the present invention is different from the stepping motor driving apparatus according to the first embodiment in that the second offset adding means in the first embodiment is not provided.

  Hereinafter, the difference from the first embodiment will be mainly described with reference to FIG. 10, and detailed description of the same operations as those of the first embodiment will be omitted.

  FIG. 10 is a block diagram showing a configuration example of a stepping motor driving apparatus according to the third embodiment of the present invention. The stepping motor has a plurality of phase windings, and the components provided for each winding are the same for each phase. Therefore, only the components provided in one phase winding will be described.

  A signal indicating the detected current value and a signal indicating the current target value are input to the comparator 16. In the present embodiment illustrated in FIG. 10, the signal indicating the current target value is the current target value output from the reference signal generating unit 14, and the signal indicating the detected current value is the output of the feeding current measuring unit 20. The comparator 16 compares the input signal indicating the detected current value with the signal indicating the current target value. When the signal indicating the detected current value exceeds the signal indicating the current target value, the comparator 16 Reset and transition to PWM off period.

  Next, the configuration and operation of the feeding current measuring unit 20 will be described.

  The feeding current measuring means 20 detects the current supplied from the power source 1 to the winding 3 by the conduction of the transistor 6 to the transistor 9 and outputs it as a detected current value. The feeding current measuring unit 20 in the present embodiment includes a detection resistor 21 that is a detection unit, an amplification circuit 25 that is an amplification unit, and a first offset addition unit 40.

  The amplifier circuit 25 includes a sense amplifier 22 and gain setting resistors 23 and 24. The gain of the amplifier circuit 25, that is, the gain from the input to the output of the sense amplifier 22, is determined by the gain setting resistors 23 and 24. Is set.

  In FIG. 10, the detection resistor 21 is used as the detection means. However, as shown in FIG. 4, the detection resistance in FIG. 10 is obtained by using the ON resistance of the MOS transistor 44 when the gate applied voltage 45 is applied. It is also possible to obtain the same action as 21. The current supplied to the winding 3 flows to the ground through the detection resistor 21, and a voltage determined by the resistance value of the detection resistor 21 and the flowing current is generated at the end of the detection resistor 21. The voltage generated at the end of the detection resistor 21 is added to the offset by the first offset adding means 40 and then input to the non-inverting input terminal of the sense amplifier 22 constituting the amplifier circuit 25. The sense amplifier 22, that is, the amplifier circuit 25 outputs a voltage obtained by amplifying the input voltage by a gain multiple to the comparator 16 as a detected current value.

  Also in the third embodiment shown in FIG. 10, since the sense amplifier loop is maintained by the first offset adding means 40 during the PWM off period as in the first embodiment, the PWM is started from the PWM off period. In the transition to the ON period, the transition from the state where the loop of the sense amplifier is out of the maintained state does not occur.

  As already described in the first embodiment, the current direction switching signal PHASE is switched by setting the offset by the first offset adding means 40 to an offset that satisfies the above-described formulas (1) and (2). Even when the current of the line 3 is reversed, detection delay can be eliminated, and waveform distortion particularly near the zero cross can be prevented.

  Also in this embodiment, since the offset added by the first offset adding means 40 is the same as in the first embodiment, the output value from the feeding current measuring means 20 is the current amount actually flowing through the detection resistor 21. It will deviate from the corresponding value. When the offset added by the first offset adding means 40 is offset and the resistance value of the detection resistor 21 is Rcs, the deviation of the detection current is [offset / Rcs], and the amount of current actually flowing through the detection resistor 21 is A deviation occurs in a direction smaller than the current target value.

  In the present embodiment, the second offset addition means in the first embodiment does not exist, and the offset subtraction means in the second embodiment does not exist, so the current flowing through the winding 3 from the current target value The deviation cannot be canceled. In the case where the deviation of the detection current is small and the deviation is permissible, this embodiment is useful from the viewpoint of reducing the number of components.

  Further, the fact that the amount of current actually flowing through the detection resistor 21 is shifted in a direction smaller than the current target value means that when the current target value indicates 0 A, the current flowing through the winding 3 depends on variations and the like. It is surely to be 0A. The offset added by the first offset adding means 40 can be used as an offset for guaranteeing that the current flowing through the winding 3 is 0A. When the current target value indicates 0A, the winding This embodiment is also useful when it is guaranteed that the current flowing through 3 is surely 0A.

  As described above, according to the stepping motor driving apparatus according to the third embodiment, by adding an offset to the input of the sense amplifier 22, detection delay can be eliminated, and waveform distortion particularly near the zero cross can be prevented. Can do. Although a detection current shift occurs due to the offset added to the input of the sense amplifier 22, on the other hand, it can be ensured that the current flowing through the winding 3 is surely 0 A by the added offset. From this, in this embodiment, the noise reduction and the vibration reduction of the stepping motor driving device are realized.

(Fourth embodiment)
In the stepping motor driving apparatus according to the fourth embodiment of the present invention, the second offset adding means does not exist, and either the output of the detecting means or the output of the first offset adding means is selected and output to the subsequent stage. The first means is that a selection means (selector) that performs the determination and that the pulse width modulation control means determines that the switching means has been made non-conductive, and a selector drive signal generator that controls the selection means according to the determination result. This is different from the stepping motor driving apparatus according to the embodiment.

  Hereinafter, the difference from the first embodiment will be mainly described with reference to FIGS. 11 and 12, and detailed description of the same operation as that of the first embodiment will be omitted.

  FIG. 11 is a block diagram showing a configuration example of a stepping motor driving apparatus according to the fourth embodiment of the present invention. The stepping motor drive device includes a selector 65 and a selector drive signal generator 66. The stepping motor has a plurality of phase windings, and the components provided for each winding are the same for each phase. Therefore, only the components provided in one phase winding will be described.

  12A to 12C are waveform diagrams of the stepping motor driving apparatus according to the fourth embodiment.

  In the fourth embodiment shown in FIG. 11, a signal indicating the detected current value and a signal indicating the current target value are input to the comparator 16. The signal indicating the current target value is the current target value output from the reference signal generating unit 14. The signal indicating the detected current value is the output of the selector 65 that selects either the output of the detection resistor 21 or the output of the first offset adding means 40 and outputs it to the subsequent stage. Detailed operation and utility of the selector 65 will be described later. The comparator 16 compares the input signal indicating the detected current value with the signal indicating the current target value, and resets the flip-flop 17 when the signal indicating the detected current value exceeds the signal indicating the current target value. , Transition to the PWM off period.

  Next, the configuration and operation of the feeding current measuring unit 20 will be described. The feeding current measuring means 20 detects the current supplied from the power source 1 to the winding 3 by the conduction of the transistor 6 to the transistor 9 and outputs it as a detected current value. The feeding current measuring unit 20 in the present embodiment includes a detection resistor 21 as a detection unit, an amplification circuit 25 as an amplification unit, a first offset addition unit 40, and a selector 65.

  The amplifier circuit 25 includes a sense amplifier 22 and gain setting resistors 23 and 24. The amplification factor of the amplifier circuit 25, that is, the gain from the input to the output of the sense amplifier 22, is the gain setting resistors 23 and 24. Set by.

  11, the detection resistor 21 is used as the detection means. However, as shown in FIG. 4, the detection resistor 21 in FIG. 11 is used by using the on-resistance of the MOS transistor 44 when the gate application voltage 45 is applied. It is also possible to obtain the same action.

  The current supplied to the winding 3 flows to the ground through the detection resistor 21, and a voltage determined by the resistance value of the detection resistor 21 and the flowing current is generated at the end of the detection resistor 21. The voltage generated at the end of the detection resistor 21 is added with an offset by the first offset adding means 40 and then input to one terminal of the selector 65. The voltage generated at the end of the detection resistor 21 is input to the other terminal of the selector 65.

  In response to a command from the selector drive signal generator 66, the selector 65 is either a signal with an offset added or a signal without an offset added to the non-inverting input terminal of the sense amplifier 22 constituting the amplifier circuit 25. Is output. The sense amplifier 22, that is, the amplifier circuit 25 outputs a voltage obtained by amplifying the input voltage by a gain multiple to the comparator 16 as a detected current value. As long as the selector 65 makes the output of the first offset adding means 40 conductive to the sense amplifier 22 in the transition from the PWM off period to the PWM on period, the first offset adding means is the same as in the first embodiment. Since the loop of the sense amplifier 22 is maintained by 40, the transition from the state where the loop of the sense amplifier 22 is removed to the state where it is maintained does not occur. That is, detection delay can be eliminated and waveform distortion can be prevented.

  The timing at which the selector drive signal generator 66 controls the selection operation of the selector 65 will be described with reference to FIGS. 12 (a) to 12 (c).

  FIG. 12A shows a waveform when the selector 65 conducts the output of the first offset adding means 40 in the PWM off period and conducts the output of the detection resistor 21 in the remaining period.

  In FIG. 12A, the output of the selector drive signal generator 66 when the selector 65 selects the output of the first offset addition means 40 and turns it on is “A”. Further, when the selector 65 selects the output of the detection resistor 21 and makes it conductive, the output of the selector drive signal generation unit 66 is “B”. During the PWM off period, the output of the selector drive signal generator 66 outputs “A”, and the output of the first offset adding means 40 is made conductive. Therefore, as described in the first embodiment, the loop of the sense amplifier 22 is Maintained. During the PWM ON period, the output of the selector drive signal generator 66 outputs “B”, and the output of the detection resistor 21 is made conductive. Since a current flows through the detection resistor 21 during the PWM ON period, the loop of the sense amplifier 22 is maintained, and the transition from the state where the loop of the sense amplifier 22 is removed to the state where it is maintained does not occur. That is, detection delay can be eliminated and waveform distortion can be prevented.

  Further, since the output of the detection resistor 21 is input to the sense amplifier 22 without going through the first offset addition means 40 during the PWM ON period, the detection current deviation by the first offset addition means 40 does not occur. However, when the current direction switching signal PHASE is switched and the current of the winding 3 is inverted, a detection delay occurs if a negative potential is generated in the detection resistor 21, but this detection delay disappears if the negative potential disappears.

  In FIG. 12B, the output of the first offset adding means 40 is turned on for a predetermined time before the selector 65 transitions from the PWM off period to the PWM on period, and the output of the detection resistor 21 is turned on for the remaining period. The waveform when this is done is shown.

  Also in FIG. 12B, the output of the selector drive signal generator 66 when the selector 65 selects the output of the first offset addition means 40 and makes it conductive is “A”. Further, when the selector 65 selects the output of the detection resistor 21 and makes it conductive, the output of the selector drive signal generation unit 66 is “B”. Since the output of the selector drive signal generation unit 66 outputs “A” for a predetermined time before the transition from the PWM off period to the PWM on period, the output of the first offset adding means 40 is made conductive. As described in the embodiment, the loop of the sense amplifier 22 is maintained.

  During the subsequent PWM ON period, the output of the selector drive signal generator 66 outputs “B”, and the output of the detection resistor 21 is made conductive. Since a current flows through the detection resistor 21 during the PWM ON period, the loop of the sense amplifier 22 is maintained, and the transition from the state where the loop of the sense amplifier 22 is removed to the state where it is maintained does not occur. That is, detection delay can be eliminated and waveform distortion can be prevented.

  However, since the sense amplifier 22 is out of the loop during the PWM off period and the output of the selector drive signal generation unit 66 outputs “B”, it is before the transition from the PWM off period to the PWM on period. The predetermined time needs to be secured more than the time required for the transition from the state in which the sense amplifier 22 is out of the loop to the maintained state. Otherwise, the transition from the PWM off period to the PWM on period is reached with the sense amplifier 22 being out of the loop, and a detection delay occurs.

  During the PWM ON period, the output of the detection resistor 21 is input to the sense amplifier 22 without going through the first offset addition means 40, so that a detection current deviation by the first offset addition means 40 does not occur. . However, when the current direction switching signal PHASE is switched and the current of the winding 3 is inverted, a detection delay occurs if a negative potential is generated in the detection resistor 21, but this detection delay disappears if the negative potential disappears.

  In FIG. 12C, the selector 65 makes the output of the first offset adding means 40 conductive during the PWM off period and a predetermined time after transition from the PWM off period to the PWM on period, and the remaining period. Shows the waveform when the output of the detection resistor 21 is conducted.

  Also in FIG. 12C, the selector drive signal generation unit 66 output when the selector 65 selects the output of the first offset addition means 40 and makes it conductive is “A”. Further, when the selector 65 selects the output of the detection resistor 21 and makes it conductive, the output of the selector drive signal generation unit 66 is “B”. In the PWM off period and a predetermined time after transition from the PWM off period to the PWM on period, the output of the selector drive signal generator 66 outputs “A”, and the output of the first offset adding means 40 is output. In order to conduct, the loop of the sense amplifier 22 is maintained as described in the first embodiment. After the predetermined time has elapsed after the transition from the PWM off period to the PWM on period, the output of the selector drive signal generator 66 outputs “B”, and the output of the detection resistor 21 is made conductive. Since a current flows through the detection resistor 21 during the PWM ON period, the loop of the sense amplifier 22 is maintained, and the transition from the state where the loop of the sense amplifier 22 is removed to the state where it is maintained does not occur. That is, detection delay can be eliminated and waveform distortion can be prevented.

  Note that, during the PWM ON period, since the output of the detection resistor 21 is input to the sense amplifier without passing through the first offset addition means 40, a detection current deviation by the first offset addition means 40 does not occur. During the PWM ON period and the period during which the output of the selector drive signal generation unit 66 outputs “A”, a detection current deviation by the first offset adding means 40 occurs. On the other hand, when the current direction switching signal PHASE is switched and the current of the winding 3 is reversed, the output of the selector drive signal generation unit 66 is “A” for a time longer than the time until the negative potential generated in the detection resistor 21 disappears. By outputting, since the loop of the sense amplifier 22 is maintained even for the negative potential generated in the detection resistor 21, detection delay can be eliminated and waveform distortion can be prevented.

  In the example of FIG. 12C, the output of the first offset adding means 40 is selected by the selector 65 in the entire PWM off period, but the first offset addition is performed in a part of the PWM off period. The output of the means 40 may be selected. That is, the output of the first offset adding means 40 may be selected in a predetermined period before the transition from the PWM off period to the PWM on period, and the output of the detection resistor 21 may be selected in the remaining period of the PWM off period. .

  As described above, according to the stepping motor drive apparatus according to the fourth embodiment, the offset is added to the input of the sense amplifier 22 in the transition from the PWM off period to the PWM on period, so that the PWM on period is changed from the PWM off period. Detection delay at the time of transition to a period can be eliminated, and waveform distortion particularly near the zero cross can be prevented. In addition, when a current is detected during the PWM on period, an offset is not added to the input of the sense amplifier 22, thereby preventing a detection current shift due to the offset. From this, in this embodiment, the noise reduction and the vibration reduction of the stepping motor driving device are realized.

(Fifth embodiment)
In the stepping motor drive device according to the fifth exemplary embodiment of the present invention, in addition to the selector drive signal generation unit in the fourth exemplary embodiment, the switching unit is made non-conductive by the pulse width modulation control unit, It differs from the stepping motor drive device according to the fourth embodiment in that it is determined that the direction switching of the winding current has been commanded. Hereinafter, the difference from the fourth embodiment will be mainly described with reference to FIGS. 13 and 14, and detailed description of the same operation as that of the fourth embodiment will be omitted.

  FIG. 13 is a block diagram showing a configuration example of a stepping motor driving apparatus according to the fifth embodiment of the present invention. The stepping motor has a plurality of phase windings, and the components provided for each winding are the same for each phase. Therefore, only the components provided in one phase winding will be described. FIG. 14 is a waveform diagram of the stepping motor driving apparatus according to the fifth embodiment.

  As shown in FIG. 13, in the stepping motor drive apparatus according to the fifth embodiment, the current direction switching signal PHASE is input to the selector drive signal generation unit 66, and in addition to the control in the fourth embodiment, the current direction The selector 65 is controlled based on the switching signal PHASE.

  The timing at which the selector drive signal generation unit 66 controls the selection operation of the selector 65 will be described with reference to FIGS. 14 (a) to 14 (c).

  14A, the selector 65 makes the output of the first offset adding means 40 conductive during a certain period after the PWM off period and the direction switching of the winding current are commanded, and the detection resistor 21 in the remaining period. The waveform when the output of is turned on is shown.

  In FIG. 14A, the output of the selector drive signal generator 66 when the selector 65 selects the output of the first offset adding means 40 and makes it conductive is “A”. Further, the selector drive signal generation unit 66 output when the selector 65 selects the output of the detection resistor 21 and makes it conductive is set to “B”. During a certain period after the PWM off period and the direction switching of the winding current are commanded, the output of the selector drive signal generator 66 outputs “A”, and the output of the first offset adding means 40 is made conductive. As described in the first embodiment, the loop of the sense amplifier 22 is maintained.

  During the subsequent PWM ON period, the output of the selector drive signal generator 66 outputs “B”, and the output of the detection resistor 21 is made conductive. Since a current flows through the detection resistor 21 during the PWM ON period, the loop of the sense amplifier 22 is maintained, and the transition from the state in which the loop of the sense amplifier 22 is removed to the state in which the sense amplifier 22 is maintained does not occur. That is, detection delay can be eliminated and waveform distortion can be prevented.

  Further, since the output of the detection resistor 21 is input to the sense amplifier 22 without going through the first offset addition means 40 during the PWM ON period, the detection current deviation by the first offset addition means 40 does not occur. In the present embodiment, the selector drive signal generator 66 outputs “A” for a certain time after the current direction switching signal PHASE is switched, and the output of the first offset adding means 40 is made conductive. Even when the switching signal PHASE is switched and the current of the winding 3 is inverted, detection delay can be eliminated and waveform distortion can be prevented.

  In FIG. 14B, the first offset adding means 40 has a selector 65 for a predetermined period before the transition from the PWM OFF period to the PWM ON period and for a fixed period after the direction switching of the winding current is commanded. A waveform when the output is conducted and the output of the detection resistor 21 is conducted during the remaining period is shown.

  Also in FIG. 14B, when the selector 65 selects the output of the first offset addition means 40 and makes it conductive, the output of the selector drive signal generation unit 66 is “A”. Further, when the selector 65 selects the output of the detection resistor 21 and makes it conductive, the output of the selector drive signal generation unit 66 is “B”. The selector drive signal generator 66 outputs “A” for a predetermined time before the transition from the PWM off period to the PWM on period and for a certain period after the direction switching of the winding current is commanded. In order to make the output of the offset adding means 40 conductive, the loop of the sense amplifier 22 is maintained as described in the first embodiment.

  During the subsequent PWM ON period, the output of the selector drive signal generator 66 outputs “B”, and the output of the detection resistor 21 is made conductive. Since a current flows through the detection resistor 21 during the PWM ON period, the loop of the sense amplifier 22 is maintained, and the transition from the state where the loop of the sense amplifier 22 is removed to the state where it is maintained does not occur. That is, detection delay can be eliminated and waveform distortion can be prevented.

  However, since the sense amplifier 22 is out of the loop during the PWM off period and the output of the selector drive signal generation unit 66 outputs “B”, it is before the transition from the PWM off period to the PWM on period. The predetermined time needs to be secured more than the time required for the transition from the state in which the sense amplifier 22 is out of the loop to the maintained state. Otherwise, the transition from the PWM off period to the PWM on period is reached while the sense amplifier 22 is out of the loop, and a detection delay occurs.

  Note that, during the PWM ON period, since the output of the detection resistor 21 is input to the sense amplifier 22 without going through the first offset addition means 40, the detection current deviation by the first offset addition means 40 does not occur.

  In the present embodiment, the selector drive signal generator 66 outputs “A” for a certain time after the current direction switching signal PHASE is switched, and the output of the first offset adding means 40 is made conductive. Even when the current direction switching signal PHASE is switched and the current of the winding 3 is reversed, detection delay can be eliminated and waveform distortion can be prevented.

  In FIG. 14C, the selector 65 includes a PWM off period, a predetermined period after transition from the PWM off period to the PWM on period, and a certain period after the command to switch the direction of the winding current is given. A waveform when the output of the first offset adding means 40 is conducted and the output of the detection resistor 21 is conducted during the remaining period is shown.

  Also in FIG. 14C, the selector drive signal generation unit 66 output when the selector 65 selects the output of the first offset addition means 40 and makes it conductive is “A”. Further, the selector drive signal generation unit 66 output when the selector 65 selects the output of the detection resistor 21 and makes it conductive is set to “B”.

  The output of the selector drive signal generator 66 is “A” for a PWM off period, a predetermined time after the transition from the PWM off period to the PWM on period, and a certain period after the direction of winding current is commanded. , And the output of the first offset adding means 40 is conducted, so that the loop of the sense amplifier 22 is maintained as described in the first embodiment. After a lapse of a predetermined time after transition from the PWM off period to the PWM on period, the output of the selector drive signal generation unit 66 outputs “B”, and the output of the detection resistor 21 is made conductive. Since a current flows through the detection resistor 21 during the PWM ON period, the loop of the sense amplifier 22 is maintained, and the transition from the state where the loop of the sense amplifier 22 is removed to the state where it is maintained does not occur. That is, detection delay can be eliminated and waveform distortion can be prevented.

  During the PWM ON period, the output of the detection resistor 21 is input to the sense amplifier 22 without going through the first offset addition means 40, so that a detection current deviation by the first offset addition means 40 does not occur. However, during the PWM ON period and the period when the output of the selector drive signal generation unit 66 outputs “A”, a detection current deviation by the first offset adding means 40 occurs.

  In the present embodiment, the selector drive signal generator 66 outputs “A” for a certain time after the current direction switching signal PHASE is switched, and the output of the first offset adding means 40 is made conductive. Even when the current direction switching signal PHASE is switched and the current of the winding 3 is reversed, detection delay can be eliminated and waveform distortion can be prevented.

  In the example of FIG. 14C, the output of the first offset adding means 40 is selected by the selector 65 during the entire PWM off period, but the first offset addition is performed during a part of the PWM off period. The output of the means 40 may be selected. That is, the output of the first offset adding means 40 may be selected in a predetermined period before the transition from the PWM off period to the PWM on period, and the output of the detection resistor 21 may be selected in the remaining period of the PWM off period. .

  As described above, according to the stepping motor drive device of the present invention, the offset from the PWM off period to the PWM on period is added by adding an offset to the input of the sense amplifier 22 in the transition from the PWM off period to the PWM on period. Detection delay at the time of transition can be eliminated, and waveform distortion particularly near the zero cross can be prevented. In addition, an offset is added to the input of the sense amplifier 22 for a certain time after the current direction switching signal PHASE is switched. Thereby, the current direction switching signal PHASE is switched, the detection delay when the current of the winding 3 is inverted can be eliminated, and the waveform distortion particularly near the zero cross can be prevented.

  In addition, when a current is detected during the PWM on period, an offset is not added to the input of the sense amplifier 22, thereby preventing a detection current shift due to the offset. Thereby, in this embodiment, the noise reduction and the vibration reduction of the stepping motor driving device are realized.

  Although the present invention has been described with respect to particular embodiments, many other variations, modifications, and other uses will be apparent to those skilled in the art. Accordingly, the invention is not limited to the specific disclosure herein, but can be limited only by the scope of the appended claims.

  As described above, the present invention is applied to a stepping motor driving device, and particularly useful as a device for reducing vibration and noise by preventing occurrence of waveform distortion and detection current deviation due to detection delay. .

The block diagram which shows the structural example of the stepping motor drive device which concerns on the 1st Embodiment of this invention. FIG. 4 is a current waveform diagram of the stepping motor driving apparatus according to the first embodiment of the present invention ((a) output of the reference pulse generation unit, (b) comparator output, (c) flip-flop output, and (d) reference signal generation means. Output (current target value) (J) and winding current (K), (e) Output of reference signal generation means (current target value) (X) after offset addition, Output of reference signal generation means (current target value) (X '), power supply current measuring means output (detected current value) (Y)) Current path diagram of stepping motor driving apparatus according to first embodiment of the present invention The figure which shows an example of the detection means which concerns on the 1st Embodiment of this invention The figure which shows an example of the 1st offset addition means which concerns on the 1st Embodiment of this invention. FIG. 3 is a diagram illustrating a sense amplifier configuration diagram and a PWM off period operating point according to the first embodiment of the present invention; FIG. 3 is a diagram illustrating a sense amplifier configuration diagram and a PWM on-period operating point according to the first embodiment of the present invention; The block diagram which shows the structural example of the stepping motor drive device which concerns on the 2nd Embodiment of this invention. The figure which shows an example of the offset subtraction means which concerns on the 2nd Embodiment of this invention. The block diagram which shows the structural example of the stepping motor drive device which concerns on the 3rd Embodiment of this invention. The block diagram which shows the structural example of the stepping motor drive device which concerns on the 4th Embodiment of this invention. Waveform diagram of the stepping motor driving apparatus according to the fourth embodiment of the present invention ((i) output of the reference pulse generator, (ii) comparator output, (iii) flip-flop output (output of the PWM control means), (iv) ) Current direction switching signal PHASE, (v) Output of selector drive signal generation unit, (vi) Output of reference signal generation means (current target value) (X) and output of feed current measurement means (detection current value) (Y) ) The block diagram which shows the structural example of the stepping motor drive device which concerns on the 5th Embodiment of this invention. Waveform diagram of a stepping motor driving apparatus according to the fifth embodiment of the present invention ((i) output of a reference pulse generation unit, (ii) comparator output, (iii) flip-flop output (output of PWM control means), (iv) ) Current direction switching signal PHASE, (v) Output of selector drive signal generation unit, (vi) Output of reference signal generation means (current target value) (X) and output of feed current measurement means (detection current value) (Y) ) Configuration diagram of conventional stepping motor drive device The figure which shows the reference signal and current direction switching signal in the conventional stepping motor drive device General sense amplifier configuration diagram and diagram showing the PWM off period operating point General sense amplifier configuration diagram and diagram showing PWM on-period operating point Current path diagram during phase switching Current waveform diagram when current target value is large in conventional stepping motor driving device ((a) output of reference pulse generator, (b) comparator output, (c) flip-flop output, (d) output of reference signal generating means (Current target value) (J) and winding current (K), (e) Output of reference signal generating means (current target value) (X), Output of feeding current measuring means (detected current value) (Y)) Current waveform diagram when current target value is small in conventional stepping motor driving device ((a) output of reference pulse generator, (b) comparator output, (c) flip-flop output, (d) output of reference signal generating means (Current target value) (J) and winding current (K), (e) Output of reference signal generating means (current target value) (X), Output of feeding current measuring means (detected current value) (Y)) The figure which shows the current waveform distortion in the conventional stepping motor drive device ((a) ideal current waveform, (b) current waveform with distortion)

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Power supply 2 Stepping motor 3 Winding 4 Rotor 5 Switching means 6-9 Transistors 10-13 Flywheel diode 14 Reference signal generation means 15 Pulse width modulation control means 16 Comparator 17 Flip-flop 18 Reference pulse generation part 19 Energization logic part 20 Feeding current measuring means 21 Detection resistor 22 Sense amplifier 23 Gain setting resistor 24 Gain setting resistor 25 Amplifier circuits 30a to 30c P channel MOS transistors 31a to 31c N channel MOS transistors 32a and 32b Differential transistor 33 Current source 34 Phase compensation capacitor 35 Current Path 40 First offset addition means 41 Second offset addition means 42 Current path 44 MOS transistor 45 Gate application voltage 47 Resistance 48 Current source 49 Gate application voltage 50 MOS transistor 51 OS transistor 52 current source 55 offset subtraction means 56 a to 56 c P-channel MOS transistor 57a-57c N-channel MOS transistors 58a, 58b the differential transistor 59 current source 65 the selector 66 selector drive signal generator

Claims (32)

Detection means for detecting a current supplied to the winding provided in the stepping motor;
First offset addition means for adding an offset to the output of the detection means;
Amplifying means for amplifying the output of the first offset adding means;
Reference signal generating means for generating a reference signal representing a current limit value;
Second offset adding means for adding an offset to the output of the reference signal generating means;
Switching means for supplying power to the winding in a conductive state and stopping power supply to the winding in a non-conductive state;
Pulse width modulation control means for making the switching means conductive at every predetermined period, and for making the switching means non-conductive when the output of the amplification means exceeds the output of the second offset voltage adding means; ,
A stepping motor driving device comprising:   2. The stepping motor drive according to claim 1, wherein a value obtained by multiplying an offset added by the first offset adding means by an amplification factor of the amplifying means is used as an offset added by the second offset adding means. apparatus. Detection means for detecting a current supplied to the winding provided in the stepping motor;
First offset addition means for adding an offset to the output of the detection means;
Amplifying means for amplifying the output of the first offset adding means;
Offset subtracting means for subtracting an offset from the output of the amplifying means;
Reference signal generating means for generating a reference signal representing a current limit value;
Switching means for supplying power to the winding in a conductive state and stopping power supply to the winding in a non-conductive state;
Pulse width modulation control means for making the switching means conductive at every predetermined period, and for making the switching means non-conductive when the output of the offset subtracting means exceeds the current limit value represented by the reference signal. When,
A stepping motor driving device comprising:   4. The stepping motor driving apparatus according to claim 3, wherein a value obtained by multiplying an offset added by the first offset adding means by an amplification factor of the amplifying means is used as an offset to be subtracted by the offset subtracting means. Detection means for detecting a current supplied to the winding provided in the stepping motor;
First offset addition means for adding an offset to the output of the detection means;
Amplifying means for amplifying the output of the first offset adding means;
Reference signal generating means for generating a reference signal representing a current limit value;
Switching means for supplying power to the winding in a conductive state and stopping power supply to the winding in a non-conductive state;
Pulse width modulation control means for bringing the switching means into a conductive state at a predetermined cycle and setting the switching means to a non-conductive state when the output of the amplifying means exceeds a current limit value represented by the reference signal; ,
A stepping motor driving device comprising: Detection means for detecting a current supplied to the winding provided in the stepping motor;
First offset addition means for adding an offset to the output of the detection means;
Selecting means for selecting and outputting either the output of the detecting means or the output of the first offset adding means;
Amplifying means for amplifying the output of the selecting means;
Reference signal generating means for generating a reference signal representing a current limit value;
Switching means for supplying power to the winding in a conductive state and stopping power supply to the winding in a non-conductive state;
Pulse width modulation control means for bringing the switching means into a conductive state at a predetermined cycle and setting the switching means to a non-conductive state when the output of the amplifying means exceeds a current limit value represented by the reference signal; ,
A selector drive signal generator for controlling the selection means,
The selector drive signal generation unit determines that the pulse width modulation control unit has made the switching unit non-conductive, and outputs the determination result,
The selection unit receives the determination result from the selector drive signal generation unit, and selects and outputs either the output of the detection unit or the output of the first offset addition unit according to the determination result. Stepping motor drive device characterized by the above.   The selection means selects the output of the first offset addition means during the entire period in which the switching means is in a non-conduction state according to the determination result of the selector drive signal generation unit, and the switching means is in the whole conduction state. 7. The stepping motor driving apparatus according to claim 6, wherein an output of the detecting means is selected in a period.   The selection unit selects an output of the first offset addition unit in a partial period in which the switching unit is in a non-conduction state, and the switching unit is in a non-conduction state according to a determination result of the selector drive signal generation unit. 7. The stepping motor driving device according to claim 6, wherein the output of the detecting means is selected in the remaining period of time and in the whole period in which the switching means is in a conductive state.   9. The stepping motor driving apparatus according to claim 8, wherein the partial period is a predetermined time before the switching unit transitions from a non-conducting state to a conducting state.   The selection means outputs the output of the first offset addition means during a part of the period when the switching means is in a conducting state and the whole period when the switching means is in a non-conducting state, according to the determination result of the selector drive signal generation unit. 7. The stepping motor driving device according to claim 6, wherein the switching means selects the output of the detection means in the remaining period of the conduction state.   11. The stepping motor driving apparatus according to claim 10, wherein the partial period is a predetermined time after the switching unit transitions from a non-conducting state to a conducting state.   In accordance with the determination result of the selector drive signal generation unit, the selection unit includes the first offset adding unit in a partial period in which the switching unit is in a conductive state and a partial period in which the switching unit is in a non-conductive state. 7. The stepping motor driving apparatus according to claim 6, wherein an output is selected, and the output of the detection means is selected in a remaining period in which the switching means is in a conductive state and a remaining period in which the switching means is in a non-conductive state. The partial period in which the switching means is in a conductive state is a predetermined time after the switching means has transitioned from a non-conductive state to a conductive state,
13. The stepping motor driving apparatus according to claim 12, wherein the partial period in which the switching unit is in the non-conducting state is a predetermined time before the switching unit transitions from the non-conducting state to the conducting state. The selector drive signal generation unit further determines that direction switching of the winding current has been commanded, and outputs the determination result,
In accordance with the determination result of the selector drive signal generation unit, the selection unit adds the first offset addition in the entire period in which the switching unit is in a non-conduction state and in a certain period after the command to switch the direction of the winding current is issued. 7. The stepping motor driving apparatus according to claim 6, wherein the output of the detecting means is selected, and the output of the detecting means is selected in a period in which the switching means excludes the predetermined period from the entire period in the conductive state. The selector drive signal generation unit further determines that direction switching of the winding current has been commanded, and outputs the determination result,
In accordance with the determination result of the selector drive signal generation unit, the selection unit is configured to perform the first operation in a partial period in which the switching unit is in a non-conduction state and a certain period after the direction switching of the winding current is commanded. The output of the offset adding means is selected, and the output of the detecting means is selected in a period excluding the predetermined period from the remaining period in which the switching means is in a non-conducting state and the entire period in which the switching means is in a conducting state. A stepping motor driving apparatus according to claim 6.   16. The stepping motor driving apparatus according to claim 15, wherein the partial period is a predetermined time before the switching unit transitions from a non-conducting state to a conducting state. The selector drive signal generation unit further determines that direction switching of the winding current has been commanded, and outputs the determination result,
After the selection means is instructed to switch the direction of the winding current in accordance with the determination result of the selector drive signal generation unit, the switching means is in a part of the conduction state, the switching means is in the non-conduction state, and the winding current direction is switched. The output of the first offset adding means is selected during a certain period of time, and the output of the detecting means is selected during a period in which the switching means excludes the certain period from the remaining period of conduction. The stepping motor driving device according to claim 6.   18. The stepping motor driving apparatus according to claim 17, wherein the partial period is a predetermined time after the switching unit transitions from a non-conducting state to a conducting state. The selector drive signal generation unit further determines that direction switching of the winding current has been commanded, and outputs the determination result,
According to the determination result of the selector drive signal generation unit, the selection means instructs a partial period in which the switching means is in a conductive state, a partial period in which the switching means is in a non-conductive state, and a direction change of winding current. In a certain period after being selected, an output of the first offset adding means is selected, and in a period obtained by removing the certain period from a remaining period in which the switching means is in a conducting state and a remaining period in which the switching means is in a non-conducting state. 7. The stepping motor driving apparatus according to claim 6, wherein an output of the detecting means is selected. The partial period of the conductive state that is the switching means is a predetermined time after the switching means transitions from the non-conductive state to the conductive state,
20. The stepping motor driving device according to claim 19, wherein the partial period in which the switching means is in a non-conductive state is a predetermined time before the switching means transitions from the non-conductive state to the conductive state. A stepping motor driving method,
Detecting a current supplied to a winding provided in the stepping motor;
Adding a first offset to the detected current value;
Amplifying the detected current added with the first offset;
Generating a reference signal representing a current limit value;
Adding a second offset to the reference signal;
Controlling the conduction of the switching means for supplying power to the winding in a conductive state and stopping the power supply to the winding in a non-conductive state,
The controlling step makes the switching means conductive at a predetermined cycle, and when the amplified current value exceeds the value of the reference signal to which the second offset is added, the switching means is turned on. To turn off,
And a stepping motor driving method. A stepping motor driving method,
Detecting a current supplied to a winding provided in the stepping motor;
Adding a first offset to the detected current value;
Amplifying the current value to which the first offset is added;
Subtracting a second offset from the amplified current value;
Generating a reference signal representing a current limit value;
Controlling the conduction of the switching means for supplying power to the winding in a conductive state and stopping the power supply to the winding in a non-conductive state,
The controlling step makes the switching means conductive at a predetermined cycle, and when the current value obtained by subtracting the second offset exceeds a current limit value represented by the reference signal, the switching means Make the means non-conductive,
And a stepping motor driving method. A stepping motor driving method,
Detecting a current supplied to a winding provided in the stepping motor;
Adding an offset to the detected current value;
Amplifying the current value to which the offset is added;
Generating a reference signal representing a current limit value;
Controlling the conduction of the switching means for supplying power to the winding in a conductive state and stopping the power supply to the winding in a non-conductive state,
In the controlling step, the switching unit is turned on every predetermined period, and when the amplified current value exceeds a current limit value represented by the reference signal, the switching unit is turned off. To
And a stepping motor driving method. A stepping motor driving method,
Detecting a current supplied to a winding provided in the stepping motor;
Adding an offset to the detected current value;
Selecting one of a current value to which the offset is added and a detected current value to which the offset is not added;
Amplifying the selected current value;
Generating a reference signal representing a current limit value;
Switching means for supplying power to the winding in the conductive state and stopping the power supply to the winding in the non-conductive state is turned on every predetermined cycle, and the amplified current value is represented by the reference signal. When the current limit value is exceeded, the switching means is turned off.
The step of selecting a stepping motor characterized in that, in the selecting step, it is determined that the switching means is in a non-conductive state, and a current value is selected according to the determination result.   The selecting step selects a current value to which the offset is added during the entire period in which the switching unit is in a non-conduction state according to the determination result, and the offset is added in the entire period in which the switching unit is in a conduction state. 25. The stepping motor driving method according to claim 24, wherein the detected current value to which no is added is selected.   The selecting step selects a current value to which the offset is added during a partial period in which the switching unit is in a non-conduction state, and the switching unit is in a non-conduction state remaining period and the switching in accordance with the determination result. 25. The stepping motor driving method according to claim 24, wherein the detected current value to which the offset is not added is selected during the entire period when the means is in the conductive state.   The selecting step selects a current value obtained by adding the offset in a partial period in which the switching unit is in a conductive state and in a whole period in which the switching unit is in a non-conductive state, according to the determination result, and the switching unit. 25. The stepping motor driving method according to claim 24, wherein the detected current value, to which the offset is not added, is selected in the remaining period of the conduction state.   The selecting step selects a current value obtained by adding the offset in a partial period in which the switching unit is in a conductive state and a partial period in which the switching unit is in a non-conductive state according to the determination result, 25. The stepping according to claim 24, wherein the detected current value to which the offset is not added is selected in a remaining period in which the switching means is in a conducting state and a remaining period in which the switching means is in a non-conducting state. Motor drive method. The step of selecting includes
Furthermore, it is determined that the direction switching of the winding current has been commanded,
According to the determination result, a current value added with the offset is selected during the entire period in which the switching means is in a non-conduction state and a certain period after the direction of switching of the winding current is commanded, and the switching means is turned on. 25. The stepping motor driving method according to claim 24, wherein a detected current value to which the offset is not added is selected in a period obtained by excluding the predetermined period from the entire period of the state.   The selecting step further determines that the direction switching of the winding current has been commanded, and further, the selecting step determines a partial period in which the switching means is in a non-conductive state and the winding current according to the determination result. In a certain period after the direction switching is commanded, the current value to which the offset is added is selected. The stepping motor driving method according to claim 24, wherein a detected current value to which the offset is not added is selected in a period excluding a period.   The selecting step further determines that the direction switching of the winding current is instructed, and further, the selecting step is performed according to the determination result, and the switching means is in a partial period in which the switching means is in a conductive state. The current value to which the offset is added is selected in the entire period of non-conduction state and the constant period after the command to switch the direction of the winding current, and the switching means changes the constant period from the remaining period of the conduction state. 25. The stepping motor driving method according to claim 24, wherein a detected current value to which the offset is not added is selected in a period excluded. The selecting step further determines that the direction switching of the winding current is instructed, and the selecting step further includes a partial period in which the switching unit is in a conductive state and the switching unit according to the determination result. Is selected for a certain period after the switching of the direction of the winding current is commanded and a current value to which the offset is added is selected, and the switching means switches between the remaining period of the conduction state and the switching 25. The stepping motor driving method according to claim 24, wherein the means selects the detected current value to which the offset is not added in a period excluding the predetermined period from the remaining period in the non-conduction state.

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