JP2006191707A - Dc converter - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To achieve the efficientization and downsizing by simplification of a circuit and reduction of loss. <P>SOLUTION: This DC converter has a switch Q1 which converts the DC voltage of a DC power source Vdc1 into high frequency voltage, being switched on or off via primary winding 5a, a series circuit which is composed of a switch Q2 and a capacitor C2, being connected to both ends of the primary winding, a smoothing capacitor Co which smoothes the rectified output of synchronous rectification circuits Q3 and Q4 which synchronize and rectify the higher frequency voltage of secondary winding 5b coupled loosely with the primary winding, feedback winding 5d which is tightly coupled with the primary winding and besides is connected in series with the secondary winding and feeds back the energy accumulated in the leakage inductance between the two pieces of the winding to Co at off of Q1, a capacitor C5 which accumulates, via a diode D5, the energy accumulated by the current flowing reversely to the leakage inductance at light load, a switch Q5 which feeds back the energy of C5 via feedback winding, being switched on synchronously with Q1, and a control circuit 10a which switches on or switches off Q1 and Q2 alternately. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a DC converter such as a high-efficiency, small-sized DC / DC converter.

  FIG. 10 shows a circuit configuration diagram of a conventional DC converter. In the DC converter shown in FIG. 10, in order to reduce the loss, a synchronous rectifier including a MOSFET having a small on-resistance is used on the secondary side (output side) of the transformer.

  In FIG. 10, a switch Q1 made of a MOSFET (field effect transistor) or the like is connected to a DC power source Vdc1 through a primary winding 5a (number of turns n1) of a transformer T1, and a resistor R1 and a capacitor are connected to both ends of the switch Q1. A series circuit with C1 is connected. A series circuit of a diode D2 and a capacitor C2 is connected to both ends of the primary winding 5a of the transformer T1, and a resistor R2 is connected to both ends of the capacitor C2. The switch Q1 is turned on / off by PWM control of the control circuit 10.

  Further, the primary winding 5a of the transformer T1 and the secondary winding 5b (the number of turns n2) of the transformer T1 are wound so as to generate an in-phase voltage, and are formed at both ends of the secondary winding 5b of the transformer T1. The switch Q3 made of MOSFET and the switch Q4 made of MOSFET are connected in series. One end (● side) of the secondary winding 5b of the transformer T1 is connected to the gate of the switch Q4, and the other end of the secondary winding 5b of the transformer T1 is connected to the gate of the switch Q3. A diode D3 is connected in parallel to the switch Q3, and a diode D4 is connected in parallel to the switch Q4. A synchronous rectifier circuit is constituted by these elements. This synchronous rectifier circuit rectifies and outputs a voltage (pulse voltage on / off controlled) generated in the secondary winding 5b of the transformer T1 in synchronization with the on / off of the switch Q1.

  Further, a smoothing reactor Lo and a smoothing capacitor Co are connected in series at both ends of the switch Q3 to constitute a smoothing circuit. The smoothing circuit smoothes the rectified output of the synchronous rectifier circuit and outputs a DC output to the load RL.

  The control circuit 10 performs on / off control of the switch Q1, and controls the on width of the pulse applied to the switch Q1 to be narrowed when the output voltage of the load RL becomes equal to or higher than the reference voltage. That is, when the output voltage of the load RL becomes equal to or higher than the reference voltage, the output voltage is controlled to a constant voltage by narrowing the ON width of the pulse of the switch Q1.

  Next, the operation of the DC converter configured as described above will be described with reference to a timing chart at light load shown in FIG. In FIG. 11, Q1v is the drain-source voltage of the switch Q1, Q1i is the drain current of the switch Q1, Q3i is the drain current of the switch Q3, Q4i is the drain current of the switch Q4, and Q3v is the drain-source of the switch Q3. A voltage Q1g indicates a gate voltage signal of the switch Q1.

  First, the operation under heavy load will be described. When the switch Q1 is turned on by the gate voltage signal Q1g, Vdc1 → 5a → Q1 → Vdc1 and current Q1i flow. This current Q1i increases linearly with time.

At this time, since a voltage is also generated in the secondary winding 5b of the transformer T1, the switch Q4 is turned on, the current Q4i flows through 5b → Lo → Co → Q4 → 5b, and power is supplied to the load RL. The The current Q4i increases linearly with time. In this case, Lo (Io) ^2/2 of energy is stored in the smoothing reactor Lo. Io is a current flowing through the smoothing reactor Lo.

  Next, when the switch Q1 is turned off, the voltage of the secondary winding 5b of the transformer T1 becomes a reverse voltage, so that the switch Q4 is turned off and the switch Q3 is turned on. For this reason, Lo → Co → Q3 → Lo and the current Q3i flow due to the energy stored in the smoothing reactor Lo, and power is continuously supplied to the load RL.

  Next, when the switch Q1 is turned on, the voltage of the secondary winding 5b is reversed again, the switch Q4 is turned on, the switch Q3 is turned off, and the same operation is performed. This state is called a continuous mode because the current of the smoothing reactor Lo continuously flows in the same direction.

On the other hand, when the load current decreases (during light load), the current of the smoothing reactor Lo when the switch Q1 is turned off (for example, at time t ₃₂ ) becomes zero during the off period of the switch Q1, Q3 is on. For this reason, the electric charge accumulated in the smoothing capacitor Co is discharged, a current flows through Co → Lo → Q3 → Co, and energy is stored in the smoothing reactor Lo.

Then, at time _{t 33} (time _{t 31} same), the switch Q1 is turned on, the switch Q4 is turned on, the switch Q3 is turned off. For this reason, the current of the smoothing reactor Lo flows in the order of Lo → 5b → Q4 → Co → Lo. For this reason, energy is fed back to the DC power supply Vdc1 on the primary side (input side) via the primary winding 5a of the transformer T1.

  For example, Patent Document 1 is known as a related art of a conventional switching power supply device.

    JP 2002-10636 A

  Thus, in the conventional DC converter shown in FIG. 10 and the switching power supply described in Patent Document 1, when synchronous rectification is applied, in the heavy load state in which the current of the smoothing reactor Lo continues, the loss It works less and lightly.

  However, in a light load state as shown in FIG. 11, the current of the smoothing reactor Lo is not continuous, and reversely flows in the feedback mode, and when the switch Q3 is off, the energy stored in the smoothing reactor Lo is changed to the switch Q4. And is fed back to the DC power supply Vdc1 on the input side via the transformer T1.

  At this time, when the turn-on timing of the switch Q4 is delayed or the leakage inductance of the transformer T1 is large, a large spike voltage SP is generated as shown in FIG. 11, and the element (synchronous rectifier) is destroyed.

  As a countermeasure, a spike voltage absorption circuit such as an absorber of the resistor R1, the capacitor C1, the resistor R2, and the capacitor C2 is mounted. In addition, a reverse current is detected to stop synchronous rectification, or an element with a high breakdown voltage is used. This complicates the circuit and increases the loss.

  An object of the present invention is to provide a DC converter capable of achieving high efficiency and miniaturization by simplifying a circuit and reducing loss.

  The present invention employs the following means in order to solve the above problems. According to the first aspect of the present invention, there is provided a first switch for converting a DC voltage of a DC power source into a high frequency voltage by turning on / off via a primary winding of the transformer, and both ends of the primary winding of the transformer or the first switch. Synchronous rectification of a high frequency voltage generated in a series circuit connected to both ends of one switch, a second switch and a clamp capacitor connected in series, and a secondary winding loosely coupled to the primary winding of the transformer A synchronous rectifier circuit, a smoothing capacitor for smoothing the rectified output of the synchronous rectifier circuit, and a primary winding of the transformer, which are tightly coupled and connected in series to the secondary winding, the primary winding and the secondary winding A feedback winding of the transformer that feeds back the energy stored in the leakage inductance between windings to the smoothing capacitor when the first switch is off, and the leakage inductance when the load is light. A capacitor that stores the energy stored by the flowing current through a diode and the first switch is turned on in synchronization with the first switch, and the energy stored in the capacitor is fed back to the primary side through the feedback winding of the transformer. And a control circuit for alternately turning on / off the first switch and the second switch.

  According to a second aspect of the present invention, in the DC converter according to the first aspect, the transformer includes a tertiary winding connected in series to a feedback winding of the transformer, and the third switch is the first switch. Is turned on by the voltage generated in the tertiary winding of the transformer when turned on, and the energy stored in the capacitor is fed back to the primary side via the feedback winding of the transformer.

  According to a third aspect of the present invention, in the DC converter according to the second aspect, one end of the diode is connected to a connection point between the feedback winding and the tertiary winding of the transformer, and the other end of the diode is connected to the capacitor. The smoothing capacitor is connected to one end or the other end of the smoothing capacitor.

  According to a fourth aspect of the present invention, in the DC converter according to any one of the first to third aspects, the primary winding and the secondary winding of the transformer are wound so that a common-mode voltage is generated. The primary winding and the feedback winding of the transformer are wound so as to generate a reverse phase voltage.

  According to a fifth aspect of the present invention, in the DC converter according to any one of the first to fourth aspects, the on-time setting means for setting the on-time of the third switch to be shorter than the on-time of the first switch. It is characterized by having.

  According to the present invention, the energy stored in the leakage inductance between the primary winding and the secondary winding of the transformer due to the backflow of current at the time of light load is stored in the capacitor via the diode, and is synchronized with the ON of the first switch. The spike voltage can be eliminated by returning energy to the primary side without loss through the feedback winding of the transformer by the third switch that is turned on. As a result, the breakdown voltage of the element used for the synchronous rectifier is reduced, and a low breakdown voltage element can be used. Further, a circuit that causes a loss such as a surge absorber becomes unnecessary, and the loss can be reduced. Therefore, it is possible to achieve high efficiency and miniaturization of the DC converter.

  Hereinafter, embodiments of a direct-current converter according to the present invention will be described in detail with reference to the drawings. In the DC converter according to the embodiment, the energy stored in the leakage inductance between the primary winding and the secondary winding of the transformer due to the backflow of current at light load is stored in the capacitor via the diode, and the first switch is turned on. The spike voltage is eliminated by returning the energy to the primary side without loss through the feedback winding of the transformer by the third switch that is turned on synchronously.

  FIG. 1 is a circuit configuration diagram of the DC converter according to the first embodiment. In FIG. 1, a DC power source Vdc1 is connected to a switch Q1 comprising a MOSFET as a first switch via a primary winding 5a (number of turns n1) of a transformer T2, and a diode D1 and a capacitor C1 are connected to both ends of the switch Q1. Are connected in parallel. The diode D1 and the capacitor C1 may be a parasitic diode and a parasitic capacitance of the switch Q1.

  One end of a switch Q2 made of a MOSFET as a second switch is connected to a connection point between one end of the primary winding 5a of the transformer T2 and one end of the switch Q1, and the other end of the switch Q2 is connected to the DC via the clamp capacitor C2. The power supply Vdc1 is connected to the positive electrode. Note that the other end of the switch Q2 may be connected to the negative electrode of the DC power supply Vdc1 via the clamp capacitor C2.

  A diode D2 is connected in parallel to both ends of the switch Q2. The diode D2 may be a parasitic diode of the switch Q2. The switches Q1 and Q2 both have a period (dead time) in which they are turned off, and are alternately turned on / off by PWM control of the control circuit 10a.

  The transformer T2 includes a primary winding 5a, a secondary winding 5b (number of turns n2) that is loosely coupled to the primary winding 5a and wound so as to generate an in-phase voltage, and a primary winding 5a. The feedback winding 5d (number of turns n4) wound so as to generate a reverse-phase voltage in close coupling and the tertiary winding wound so as to generate an in-phase voltage in close coupling with the primary winding 5a. Line 5c (number of turns n3). A feedback winding 5d is connected in series to the secondary winding 5b, and a tertiary winding 5c is connected in series to the feedback winding 5d.

  Here, the common-mode voltage of the tertiary winding 5c means that the tertiary winding 5c is turned to the primary winding 5a so that the switch Q5 is turned on when the switch Q1 is turned on and the switch Q5 is turned off when the switch Q1 is turned off. That is, it is wound around. The reverse phase voltage of the feedback winding 5d means that the feedback winding 5d is wound around the primary winding 5a so that the switch Q3 is turned off when the switch Q1 is turned on and the switch Q3 is turned on when the switch Q1 is turned off. It has been done.

  A saturable reactor SL1 is connected to both ends of the secondary winding 5b. The saturable reactor SL1 uses the saturation characteristic of the core of the transformer T2. Since currents of equal magnitude flow through the saturable reactor SL1, the magnetic flux increases and decreases equally in the first quadrant and the third quadrant, and the magnetic flux is saturated with a predetermined magnetic field. By turning off the switch Q2 when the saturable reactor SL1 is saturated, the charge of the capacitor C1 is sharply discharged, and the zero voltage switch of the switch Q1 is achieved.

  The connection point between one end of the secondary winding 5b of the transformer T2 and one end of the feedback winding 5d is connected to the gate of the switch Q3 made of MOSFET, one end (drain) of the switch Q4 made of MOSFET, and the cathode of the diode D4. Yes. The other end (● side) of the secondary winding 5b of the transformer T2 is connected to one end of the smoothing capacitor Co.

  The connection point between the other end (● side) of the feedback winding 5d of the transformer T2 and one end of the tertiary winding 5c is one end (drain) of the switch Q3, the gate of the switch Q4, the cathode of the diode D3, and the third switch. Are connected to one end (source) of a switch Q5 made of a MOSFET and the anode of a diode D5. The anode of the diode D4, the other end (source) of the switch Q4, the anode of the diode D3, and the other end (source) of the switch Q3 are connected to the other end of the smoothing capacitor Co and one end of the capacitor C5. The cathode of the diode D5 and the other end (drain) of the switch Q5 are connected to the other end of the capacitor C5. The other end (● side) of the tertiary winding 5c is connected to the gate of the switch Q5. These elements (Q3, Q4, D3, D4) constitute a synchronous rectifier circuit.

  The diode D6 is connected between a connection point between the secondary winding 5b and the cathode of the diode D4 and a connection point between the cathode of the diode D5 and the capacitor C5. The diode D6 is provided in order to suppress the spike voltage by absorbing energy due to the spike voltage at the time of recovery of the diode D4 to the capacitor C5. That is, during the recovery of the diode D4, a recovery current flows in the order of D3 → 5d → D4 → D3.

  The control circuit 10a alternately turns on / off the switch Q1 and the switch Q2, and when the output voltage of the load RL becomes equal to or higher than the reference voltage, the on width of the pulse applied to the switch Q1 is reduced. Control is performed so as to widen the ON width of the pulse applied to Q2. That is, when the output voltage of the load RL becomes equal to or higher than the reference voltage, the output voltage is controlled to a constant voltage by narrowing the ON width of the pulse of the switch Q1.

  Further, when the control circuit 10a turns on the switch Q1, the leakage inductance (not shown) between the primary winding and the secondary winding of the capacitor C1 and the transformer T2 in which the voltage of the switch Q1 is connected in parallel with the switch Q1. The switch Q1 is turned on during a predetermined period from when the voltage becomes zero due to resonance.

(Transformer configuration)
A configuration example of the transformer T2 is shown in FIG. The transformer T2 shown in FIG. 2 has a Japanese character-shaped core 30, and a primary winding 5a, a feedback winding 5d, and a tertiary winding 5c are wound close to the core portion 30a of the core 30. It has been turned. Thus, a slight leakage inductance is provided between the primary winding 5a, the feedback winding 5d, and the tertiary winding 5c. In addition, a pass core 30c and a gap 31 are formed in the core 30, and a secondary winding 5b is wound around the outer peripheral core. That is, the leakage inductance is increased by loosely coupling the primary winding 5a and the secondary winding 5b by the pass core 30c.

  In addition, two recesses 30b are formed on the outer core and between the primary winding 5a and the secondary winding 5b. Due to the recess 30b, the cross-sectional area of a part of the magnetic path of the outer peripheral core becomes narrower than the other part and only that part is saturated, so that the core loss can be reduced. Further, this saturated portion is also used as the saturable reactor SL1.

  Next, the operation of the direct-current converter according to Embodiment 1 configured as described above will be described with reference to a timing chart at light load shown in FIG. In FIG. 3, Q1v is the drain-source voltage of the switch Q1, Q1i is the drain current of the switch Q1, Q3i is the drain current of the switch Q3, Q4i is the drain current of the switch Q4, Q5i is the drain current of the switch Q5, Q3v Is a drain-source voltage of the switch Q3, and Q1g is a gate voltage signal of the switch Q1. First, the operation under heavy load will be described.

  First, when the switch Q2 is turned off, resonance occurs due to the leakage inductance between the primary and secondary windings of the transformer T2 and the capacitor C1, and the voltage Q1v of the switch Q1 decreases. When the voltage Q1v of the switch Q1 is zero and the switch Q1 is turned on, a zero voltage switch of the switch Q1 is realized.

  Next, when the switch Q1 is turned on, Vdc1 → 5a → Q1 → Vdc1 and current Q1i flow. This current Q1i increases linearly with time.

At this time, since the voltage is also generated in the secondary winding 5b and the feedback winding 5d of the transformer T2, the switch Q4 is turned on, the current Q4i flows through 5b → Co → Q4 → 5b, and power is supplied to the load RL. Is done. The current Q4i increases linearly with time. In addition, since the primary winding 5a and the secondary winding 5b are loosely coupled, a large leakage inductance Lr (not shown) is provided between the primary and secondary windings. Therefore, the leakage inductance ^{Lr, Lr (Ir) 2/} 2 of energy is stored. Ir is a current flowing through the leakage inductance Lr.

  Next, when the switch Q1 is turned off, a current flows due to the excitation energy stored in the primary winding 5a, and the capacitor C1 is charged. At this time, resonance occurs due to the leakage inductance between the primary and secondary windings of the transformer T2 and the capacitor C1, and the voltage Q1v of the switch Q1 increases.

  When the potential of the switch Q1 becomes the same as the potential of the clamp capacitor C2, the diode D2 is turned on and the clamp capacitor C2 is charged. At this time, by turning on the switch Q2, the switch Q2 becomes a zero voltage switch.

  When the switch Q2 is turned on, the voltages of the secondary winding 5b and the feedback winding 5d of the transformer T2 are reversed, so that the switch Q4 is turned off and the switch Q3 is turned on. A voltage is generated in the feedback winding 5d, and the current Q3i flows as 5d → 5b → Co → Q3 → 5d due to the energy stored in the leakage inductance Lr. Is supplied.

  In this state, when the switch Q1 is turned on, the voltages of the secondary winding 5b and the feedback winding 5d are reversed again, the switch Q3 is turned off, the switch Q4 is turned on, and the same operation is performed. In this state, the current flowing in the leakage inductance Lr (same as the current flowing in the secondary winding 5b) continuously flows in the same direction, and is therefore referred to as a continuous mode.

On the other hand, when the load current decreases (when the load is light), the current of the leakage inductance Lr when the switch Q1 is turned off (for example, at time t ₂ ) becomes zero during the off period of the switch Q1. Q3 is on. For this reason, the electric charge accumulated in the smoothing capacitor Co is discharged, current flows through Co → 5b → 5d → Q3 → Co, and energy is stored in the leakage inductance Lr.
Next, at time t _{3 (time} t ₁ is also the same), the switch Q1 is turned on, the switch Q4 voltage is applied to the gate of the switch Q4 end (● side) of the feedback winding 5d of the transformer T2 is turned on . Further, the switch Q3 is turned off. For this reason, the energy stored in the leakage inductance Lr is stored in the capacitor C5 via the diode D5. That is, since the diode D5 is turned on and energy is stored in the capacitor C5, the spike voltage is absorbed.

  Further, since a voltage is generated at one end (● side) of the tertiary winding 5c of the transformer T2, this voltage is applied to the gate of the switch Q5 and the switch Q5 is turned on.

  For this reason, since current flows from C5 → Q5 → 5d → Q4 → C5, the energy stored in the capacitor C5 is fed back to the DC power supply Vdc1 on the primary side via the feedback winding 5d and the primary winding 5a. The

  As shown in FIG. 3, the voltage Q3v of the switch Q3 is clamped and no spike voltage is generated. For this reason, the breakdown voltage of the switch Q3 can be lowered. Therefore, the loss can be further reduced by using an element having a low on-resistance.

  As described above, according to the DC converter of the first embodiment, the energy stored in the leakage inductance Lr between the primary and secondary windings of the transformer T2 due to the backflow of current at the time of light load is stored in the capacitor C5 via the diode D5. The spike voltage can be removed by returning energy to the primary side without loss through the feedback winding 5d of the transformer T2 by the switch Q5 that is turned on in synchronization with the switch Q1 being turned on. Thereby, the withstand voltage of the rectifying element is reduced, and the on-resistance can be reduced by using a low withstand voltage element. In addition, since the CR absorber for preventing spike voltage can be removed, the circuit can be simplified.

  FIG. 4 is a circuit configuration diagram of the DC converter according to the second embodiment. The direct-current converter according to the second embodiment shown in FIG. 4 is further different from the direct-current converter according to the first embodiment shown in FIG. 1 between one end of the tertiary winding 5c of the transformer T2 and the gate of the switch Q5. A waveform shaping circuit 11 is provided as on-time setting means.

  The other configuration shown in FIG. 4 is the same as the configuration shown in FIG. 1, and the same reference numerals are given to the same parts, and the details thereof are omitted.

  The waveform shaping circuit 11 shapes the waveform of the voltage generated in the tertiary winding 5c of the transformer T2, and sets the drive voltage to the gate of the switch Q5 to be shorter than the ON time of the switch Q1. FIG. 5 shows an example of the waveform shaping circuit 11. The waveform shaping circuit 11 includes a resistor RT having one end connected to the tertiary winding 5c and the other end connected to the gate of the switch Q5, and a capacitor CT connected between the other end of the resistor RT and the ground. The time constant circuit.

As shown in FIG. 6, the time constant circuit of the resistor RT and the capacitor CT has a linear waveform of the voltage V _CT across the capacitor CT with respect to the input rectangular voltage waveform V _{n3 of the} tertiary winding 5c. It is raised, by applying the voltage to the gate of the switch Q5, thereby turning on the switch Q5 only the threshold voltage V _TH or more portions of the gate of the switch Q5. That is, since the on-time of the switch Q5 is set shorter than the on-time of the switch Q1, the current of the switch Q5 under heavy load can be reduced.

  FIG. 7 is a timing chart of signals in the respective portions when the DC converter is heavy loaded when the waveform shaping circuit 11 is not provided. FIG. 8 is a timing chart of signals in the respective parts when the DC converter is heavy loaded when the waveform shaping circuit 11 is provided. Comparing FIG. 7 and FIG. 8, it can be seen that the current of the switch Q5 at the time of heavy load is greatly reduced.

  Further, as in the first embodiment, when the switch Q5 is turned on only for the same time as the on time of the switch Q1, the leakage inductance between the capacitor C5 and the primary winding 5a and the feedback winding 5d of the transformer T2 is used. A resonant circuit is formed, and a resonant current flows. In the second embodiment, the resonance circuit is cut off by shortening the ON time of the switch Q5, so that the resonance current can be reduced.

  FIG. 9 is a circuit configuration diagram of the DC converter according to the third embodiment. The direct current converter of the third embodiment shown in FIG. 9 is different from the direct current converter of the first embodiment shown in FIG. 1 in that the connection of the capacitor C5 is changed. That is, one end of the capacitor C5 is connected to the other end of the smoothing capacitor Co in the first embodiment, but the third embodiment is different in that one end of the capacitor C5 is connected to one end of the smoothing capacitor Co.

  Even such a DC converter according to the third embodiment operates in the same manner as the DC converter according to the first embodiment, and therefore, detailed description thereof is omitted here.

  In addition to the configuration of the DC conversion apparatus according to the third embodiment shown in FIG. 9, a waveform shaping circuit 11 as shown in FIG. 4 is further connected between one end of the tertiary winding 5c of the transformer T2 and the gate of the switch Q5. May be provided. Thereby, the effect similar to the effect of Example 2 is acquired.

  In the first to third embodiments, the drive signals for the switches Q3, Q4, and Q5 are self-excited and generated by the voltages of the secondary winding 5b, the feedback winding 5d, and the tertiary winding 5c. The control circuit 10a can operate in the same manner even by a separately excited type in which a drive signal for driving the switches Q3, Q4, and Q5 is generated.

  The present invention is applicable to switching power supply devices such as a DC-DC converter and an AC-DC converter.

It is a circuit block diagram of the direct-current converter of Example 1. FIG. 2 is a structural diagram of a transformer provided in the direct-current converter of Example 1. It is a timing chart of the signal in each part at the time of the light load of the direct-current converter of Example 1. It is a circuit block diagram of the direct-current converter of Example 2. It is a figure which shows an example of the waveform shaping circuit provided in the direct-current converter of Example 2. It is a figure which shows the operation | movement waveform of the waveform shaping circuit shown in FIG. It is a timing chart of the signal in each part at the time of heavy load of a direct-current converter when there is no waveform shaping circuit. It is a timing chart of the signal in each part at the time of heavy load of a direct-current converter when there is a waveform shaping circuit. 6 is a circuit configuration diagram of a DC converter according to Embodiment 3. FIG. It is a circuit block diagram of the conventional DC converter. It is a timing chart of the signal in each part at the time of light load of the DC converter shown in FIG.

Explanation of symbols

Vdc1 DC power supply 10, 10a Control circuit Q1-Q5 Switch RL Load
R1, R2, RT Resistance Lo Smoothing reactor Co Smoothing capacitor C1, C2, C5, CT Capacitor T1, T2 Transformer SL1 Saturable reactor 5a Primary winding (n1)
5b Secondary winding (n2)
5c Tertiary winding (n3)
5d Feedback winding (n4)
D1 to D6 Diode 11 Waveform shaping circuit 30 Core 30a Core portion 30b Recessed portion 30c Path core 31 Gap

Claims (5)

A first switch that converts a DC voltage of a DC power source into a high-frequency voltage by turning it on / off via a primary winding of a transformer;
A series circuit connected to both ends of the primary winding of the transformer or both ends of the first switch, and a second switch and a clamp capacitor connected in series;
A synchronous rectifier circuit for synchronously rectifying a high-frequency voltage generated in a secondary winding loosely coupled to the primary winding of the transformer;
A smoothing capacitor that smoothes the rectified output of the synchronous rectifier circuit;
When the first switch is turned off, the energy stored in the leakage inductance between the primary winding and the secondary winding is tightly coupled to the primary winding of the transformer and connected in series to the secondary winding. A feedback winding of the transformer that feeds back to the smoothing capacitor;
A capacitor for storing the energy stored by the current flowing back to the leakage inductance at a light load via a diode;
A third switch that is turned on in synchronization with the first switch and returns the energy stored in the capacitor to the primary side via a feedback winding of the transformer;
A control circuit for alternately turning on and off the first switch and the second switch;
A DC converter characterized by comprising: Having a tertiary winding of the transformer connected in series with a feedback winding of the transformer;
The third switch is turned on by a voltage generated in the tertiary winding of the transformer when the first switch is turned on, and the energy stored in the capacitor is transferred to the primary side through the feedback winding of the transformer. The DC converter according to claim 1, wherein the DC converter is fed back.   One end of the diode is connected to a connection point between the feedback winding and the tertiary winding of the transformer, and the other end of the diode is connected to one end or the other end of the smoothing capacitor via the capacitor. The DC converter according to claim 2.   The primary and secondary windings of the transformer are wound so as to generate an in-phase voltage, and the primary winding and feedback winding of the transformer are wound so as to generate a reverse-phase voltage. The DC converter according to any one of claims 1 to 3, wherein:   5. The DC converter according to claim 1, further comprising an on-time setting unit configured to set an on-time of the third switch to be shorter than an on-time of the first switch.

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