JP2006060687A - Oscillator circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To suppress a decrease in negative resistance and selectively increase the negative resistance near an oscillation frequency.
In an oscillation circuit that oscillates by feedback from an output of an amplifier to an input, a second feedback double that performs amplitude and phase transition is provided separately from the first feedback for oscillation that is normally provided in the oscillation circuit. By performing feedback, the negative resistance of the oscillation circuit is increased at a desired frequency, and the equivalent reactance component is adjusted. The oscillation circuit may be configured by a first feedback circuit that generates a negative resistance included in the Colpitts oscillation circuit, and a second feedback circuit that performs current feedback from the emitter terminal of the oscillation stage transistor of the Colpitts oscillation circuit to the base terminal. it can. Current feedback from the emitter terminal to the base terminal is performed via a feedback capacitor.
[Selection] Figure 1

Description

  The present invention relates to an oscillation circuit, and more particularly to a Colpitts oscillation circuit.

  With an increase in network speed and capacity, demand for a reference frequency generation source in an optical digital communication network such as SONET / SDH is increasing. As such reference frequency generation sources, oscillators and SAW oscillators that multiply 150 MHz fundamental crystal oscillation have been commercialized. In the multiplication method, deterioration of jitter characteristics is a problem, and in the SAW oscillator, frequency temperature characteristics are a problem.

  In an oscillator using a resonator in the 100 MHz to GHz band, a Colpitts oscillation circuit is mainly used. In Colpitts oscillation circuits of 100 MHz or higher, the capacitance value that determines the oscillation frequency must be reduced in order to obtain sufficient negative resistance, making it difficult to design and variable frequency when configuring a voltage-controlled oscillator. There are problems such as insufficient range.

  Further, when the negative resistance is increased by increasing the bias current, there is a problem that the power consumption increases and the current flowing through the resonator at the time of oscillation increases, leading to deterioration of characteristics.

  To solve this problem, a circuit (Non-patent Document 1) that increases the negative resistance of the oscillation circuit by capacitively coupling back from the collector terminal to the emitter terminal of the Colpitts oscillation circuit, or the output of the Colpitts oscillation circuit is used as the emitter. A circuit that increases the negative resistance by feeding back through a follower has been proposed (Non-Patent Documents 2 and 3).

  FIG. 27 shows an oscillation circuit of Non-Patent Document 3. A Colpitts oscillation circuit is constituted by the transistor Q1, the capacitors CA and CB and the crystal resonator. The portion surrounded by the broken line in the figure is an emitter-follower, which detects the signal voltage at the emitter terminal of the Colpitts oscillation circuit, and an almost equal voltage is applied to the collector terminal of the transistor Q1, contributing to an increase in negative resistance. Yes.

  In general, a circuit in which a capacitor Cp is connected in parallel to a series circuit of a negative resistor -R and a capacitor Ci can be equivalently expressed by a series circuit of a negative resistor -R 'and a capacitor Ci'. At this time, the absolute value of the negative resistance −R ′ is smaller than the absolute value of the negative resistance −R.

  On the other hand, the Colpitts oscillation circuit includes an inductance L and an active circuit. The active circuit is composed of two capacitors connected in series and a voltage controlled current source (transistor) connected to these capacitors, and is represented by an equivalent circuit composed of a series circuit of a negative resistance -R and a capacitor Ci. Can do. Here, since the capacitance Cbc (capacitance between the base and the collector of the transistor) exists between the base and collector of the transistor included in the Colpitts oscillation circuit, the Colpitts oscillation circuit includes the negative resistance −R and the capacitance Ci in series. The circuit can be represented as a circuit in which a capacitor Cbc ′ obtained by converting the capacitor Cbc into a capacitor viewed from the base terminal of the transistor is connected in parallel.

Therefore, the net negative resistance −R ′ of this Colpitts oscillation circuit is
−R ′ = − R / ((1 + Cbc ′ / Ci) ² + (ωCbc′R) ² )
Therefore, it is smaller than the original negative resistance -R. This effect becomes larger when a high-frequency oscillator is used, since Cbc 'becomes a value close to Ci.

  Here, when a Colpitts oscillation circuit is configured by connecting a load resistor Rc between the collector terminal of the transistor and the power source, the capacitance Cbc ′ viewed from the input terminal (base terminal) of the transistor is And Cbc ′ = (1 + A) Cbc, where A is the gain of the amplifier constituted by the load resistor Rc. Since the gain A = gmRc and is sufficiently larger than 1, Cbc ′ is larger than Cbc. The above phenomenon is called a mirror effect. Therefore, in the Colpitts oscillation circuit having the load resistance Rc, the influence of the capacitance Cbc between the base and collector of the transistor appears remarkably. The influence of the capacitance Cbc between the base and collector of the transistor appears as a decrease in negative resistance, which degrades the oscillation characteristics.

  In the oscillation circuit of Non-Patent Document 3, in order to remove the influence of Cbc, the current flowing through Cbc may be set to zero. As a result, Cbc becomes the same as in the open state, and the influence of Cbc can be removed.

  In order to reduce the current flowing through Cbc to 0, the voltage across Cbc may be equalized. In the oscillation circuit of Non-Patent Document 3, the emitter follower is used to return a voltage equal to the base terminal to the collector terminal of the oscillator. , The voltage across Cbc is made equal.

Matsuoka, Sato, Oshima; “High Negative Resistance, Low Drive 600MHz Band Crystal Oscillator”, 32nd EM Symposium, pp.45-48, 2003-05 Zhou, Nomura, Aoyagi, Sekine; "A Study on High Frequency of Colpitts Oscillators", 2003 IEEJ Electronics, Information and Systems Division Conference, GS3-6, pp.728-730, 2003-09 Zhou, Nomura, Aoyagi, Sekine; "Study on high-frequency Colpitts oscillation circuit with emitter-follower circuit", IEEJ Electronic Circuits, ECT-04-20, pp.17-21, 2004

  In the Colpitts oscillation circuit for high frequency to which the emitter-follower circuit is added, as described above, the influence of the capacitance Cbc between the base and collector of the transistor is removed by returning a voltage equal to the base terminal to the collector terminal of the oscillator. Although an attempt is made to suppress the decrease in negative resistance, in an actual circuit, the influence of Cbc remains due to various factors, and thus there is a problem that suppression of the decrease in negative resistance is insufficient. .

  Factors that remain without removing the influence of the Cbc include, for example, a difference in the magnitude and phase of the AC voltage between the voltage of the base terminal and the input voltage of the emitter-follower circuit, and the emitter-follower. Since the output impedance of the circuit does not become zero, the collector terminal voltage is smaller than the output voltage of the emitter-follower circuit due to the relationship between the load resistance Rc and Ro (output resistance of the emitter-follower circuit). Due to the influence of these factors, the voltage across the capacitor Cbc between the base and collector of the transistor is not equal, and the influence of Cbc remains.

  An object of the present invention is to solve the above-described conventional problems, to suppress a decrease in negative resistance, and to increase the negative resistance selectively near the oscillation frequency.

  The present invention provides an oscillation circuit that oscillates by feedback amplification from an output to an input, and a second feedback double feedback that performs amplitude and phase transition separately from the first feedback for oscillation normally provided in the oscillation circuit. As a result, the negative resistance of the oscillation circuit is increased at a desired frequency, and the equivalent reactance component is adjusted.

  The feedback circuit (first feedback) normally provided in the oscillation circuit is configured to perform positive feedback to the amplifier by a parallel circuit of two capacitors and an inductance connected in series. In the present invention, another feedback circuit (the above-described second feedback) is provided in this feedback circuit, and the current in phase with the current Ix from the inductance of the first feedback circuit is fed back, so that the current entering the amplifier is This increases the negative resistance. At this time, in the second feedback, the feedback current is adjusted by performing amplitude and phase transition.

  In this current feedback, the equivalent reactance component of the oscillation circuit can be adjusted by positively adding a phase difference without being completely in phase with the current Ix.

  Since the oscillation circuit of the present invention has a double feedback configuration, the second feedback circuit is provided as a circuit design element in addition to the oscillation stage and the first feedback circuit included in the normal oscillation circuit. The frequency characteristics of the amplitude and phase of the gain can be adjusted, and the degree of freedom in circuit design can be greatly increased as compared with the conventional Colpitts oscillation circuit.

  When the oscillation circuit of the present invention is configured by a Colpitts oscillation circuit, the oscillation circuit includes a first feedback circuit that generates a negative resistance included in the Colpitts oscillation circuit, and a base from the emitter terminal of the oscillation stage transistor of the Colpitts oscillation circuit. And a second feedback circuit that feeds back current to the terminal. Current feedback from the emitter terminal to the base terminal is performed via a feedback capacitor.

  The second feedback circuit of the present invention is not limited to a passive circuit but can be configured by an active circuit, and the negative resistance and / or the equivalent reactance component is changed according to the amplification characteristic and phase transition characteristic of the active circuit.

  By changing the amplitude characteristic and phase characteristic of the voltage gain of the second feedback circuit of the present invention, the frequency characteristic of the negative resistance can be changed. That is, the frequency and magnitude of the negative resistance peak can be changed.

  Further, by changing the amplitude characteristic and phase characteristic of the voltage gain of the second feedback circuit, the capacitance value of the equivalent reactance component of the oscillation circuit can be changed and further changed to be inductive.

  Since the change of the negative resistance and / or equivalent reactance component by the second feedback circuit and the capacitance determining the frequency of the Colpitts oscillation circuit and the resistance and load resistance of the bias circuit can be made independent, the circuit of the oscillation stage The negative resistance and equivalent reactance component can be changed without changing the element value.

  In addition, since the negative resistance is inversely proportional to the capacity of the oscillation stage, in a normal Colpitts oscillation circuit, if the oscillation stage capacity is increased, the negative resistance will decrease, making it difficult to start oscillation, or due to fluctuations in circuit elements. Since it is easy for oscillation to stop, these capacitance values cannot be increased. However, in the oscillation circuit of the present invention, the negative resistance can be increased independently of the oscillation stage capacitance value by the second feedback circuit, so that the capacitance of the oscillation stage can be increased.

  The oscillation circuit of the present invention can also be applied to a configuration in which an inverter is used as an active element that oscillates.

  Further, the oscillation element can include an inductance or a resonator, and the resonator includes a crystal resonator, a ceramic resonator, a surface acoustic wave resonator, an end surface reflection resonator, a piezoelectric resonator of a bulk resonator, or A dielectric resonator can be used.

  The oscillation circuit of the present invention can be applied to a voltage controlled oscillation circuit. In general, in a voltage controlled oscillator circuit, a variable capacitor Cs is connected between an inductance of an oscillation stage and an active circuit, and the frequency is changed by changing the variable capacitor Cs. The oscillation frequency is determined by the series capacitance of the inductance L, the variable capacitance Cs, and the equivalent capacitance Ci of the active circuit. Since the variable capacitance Cs and the equivalent capacitance Ci are connected in series, the oscillation frequency is mainly determined by the smaller capacitance value. Become. In a high-frequency oscillation circuit, if the capacitance is reduced to increase the frequency, the equivalent capacitance Ci is reduced. Therefore, as described above, the oscillation frequency mainly depends on the small equivalent capacitance Ci, and the variable capacitance Cs is changed. It is difficult to change the frequency.

  In the oscillation circuit of the present invention, since the negative resistance can be increased by the second feedback circuit, the capacitance of the oscillator can be increased and the equivalent capacitance Ci can be increased. Therefore, the variable range of the oscillation frequency due to the change of the variable capacitance Cs. Can be taken greatly. Further, by making the equivalent capacitance Ci variable, the oscillation frequency can be changed without using the variable capacitance Cs. The variable capacitor Cs is usually configured using a varicap, but this varicap is not suitable for integration because of problems such as volume and characteristics, and is also unsuitable for frequency control from the viewpoint of linearity of oscillation frequency control. . On the other hand, according to the oscillation circuit of the present invention, the frequency can be changed without using a variable capacitor such as a varicap by changing the equivalent capacitance Ci by the second feedback circuit.

  Further, in the voltage controlled oscillation circuit including a resonator such as a piezoelectric vibrator instead of the inductance, it is necessary to reduce the influence of Ci in order to increase the variable range of the oscillation frequency. Usually, in order to reduce the influence of Ci, an extension coil is inserted in series with the vibrator. However, a voltage controlled oscillation circuit can be obtained by making the equivalent reactance of the active circuit (second feedback circuit) of the present invention inductive. Since the extension coil can be replaced with a part or all of the extension coil, the configuration in which the extension coil is not required, or the extension coil can be made smaller. Thereby, the circuit can be reduced in size.

  As the inductance of the voltage controlled oscillator, a piezoelectric vibrator such as a crystal vibrator, a ceramic vibrator, a surface acoustic wave vibrator, an end surface reflecting vibrator, a bulk vibrator, or a dielectric resonator can be used.

  The oscillation circuit of the present invention can be applied to a temperature compensation oscillation circuit. According to the temperature compensated oscillation circuit of the present invention, the frequency-temperature characteristic is compensated by making the equivalent capacitance of the active circuit of the oscillation circuit variable according to the temperature.

  Further, the oscillation stage transistor of the oscillation circuit of the present invention can be a bipolar transistor or a MOS transistor.

  Furthermore, an inverter oscillator can be used as an oscillation stage for performing the oscillation of the present invention. The oscillation element of the oscillation stage can be an inductance or a resonator.

  The oscillation circuit of the present invention can be configured in a plurality of forms.

  In the first form of the oscillation circuit of the present invention, the output terminal of the second feedback circuit and the base terminal of the oscillation stage transistor are used as feedback capacitors used for current feedback from the emitter terminal to the base terminal of the oscillation stage transistor of the Colpitts oscillation circuit. A capacitive element is connected between the output terminal and current feedback from the output terminal to the base terminal via the capacitive element.

  In the second form of the oscillation circuit according to the present invention, the collector-base capacitance of the oscillation stage transistor is used as the feedback capacitance, and the output terminal of the second feedback circuit has a different capacitance (a sufficiently large coupling capacitance). Is connected to the collector terminal of the oscillation stage transistor, and current is fed back from the collector terminal to the base terminal via the collector-base capacitance.

  In the first embodiment, a configuration in which the collector terminal of the oscillation transistor and the power supply are directly connected or a configuration in which a load resistor is connected can be employed.

  Since the load resistance reduces the negative resistance due to the mirror effect, a high negative resistance can be obtained by directly connecting the collector terminal and the power source to eliminate the load resistance.

  In the second embodiment, a load resistor is connected between the collector terminal of the oscillation transistor and the power supply.

  In the first and second embodiments, the second feedback circuit can be composed of a two-stage grounded-emitter amplifier.

  In the third form of the oscillation circuit of the present invention, the second feedback circuit can be configured to have a resonance characteristic, and thereby have a narrow band characteristic.

  In addition, the second feedback circuit can be provided with a plurality of resonance characteristics so as to be switchable, and can switch multi-frequency by switching the oscillation frequency.

  In the fourth form of the oscillation circuit of the present invention, the second feedback circuit is constituted by a differential feedback circuit. As a result, a differential output can be obtained, and in-phase noise can be removed. Further, phase noise can be improved as compared with a circuit in which two stages of grounded emitters are connected.

  In the oscillation circuit of the present invention, the frequency band in which the negative resistance is generated can be switched by switching the parameter of the second feedback circuit.

  As described above, according to the present invention, it is possible to suppress a decrease in negative resistance.

  Further, the negative resistance can be selectively increased near the oscillation frequency.

  Further, the circuit capacity of the oscillation stage can be increased, and when the voltage controlled oscillator is configured, the variable range of the oscillation frequency can be increased.

  In addition, an oscillator of a high frequency band (for example, 100 MHz to several tens GHz band) can be easily configured.

  Further, the negative resistance is greatly increased with the same or low power consumption as that of the conventional circuit, and the oscillation can be easily started. It is possible to provide a large margin for stopping oscillation due to resonator and circuit fluctuations, and to prevent oscillation from occurring, and thus a reliable oscillator can be configured.

  In addition, in the current feedback, the magnitude of the equivalent reactance of the oscillation circuit can be changed by positively adding a phase difference without being completely in phase with the oscillation current, or inductive equivalent reactance can be obtained. it can.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

  First, the basic configuration of the oscillation circuit of the present invention will be described with reference to FIGS. In the following, a Colpitts oscillation circuit will be described as the oscillation circuit.

  FIG. 1 shows a basic circuit configuration of the oscillation circuit of the present invention, FIG. 2 shows a circuit configuration focusing on only an AC signal in the circuit of FIG. 1, and FIG. 3 is a rewrite of the circuit configuration of FIG. 4 shows a simplified equivalent circuit of the basic circuit configuration of the oscillation circuit of the present invention, and FIG. 5 shows a two-port simplification of the basic circuit configuration of the oscillation circuit of the present invention. An equivalent circuit is shown.

  1A is a configuration example in which the load resistance Rc is not connected, FIG. 1B shows a configuration example in which the load resistance Rc is connected, and other configurations are common. It should be noted that the generation of the negative resistance in the oscillation circuit does not require the load resistance Rc, and the circuit configuration that does not include the load resistance Rc is less in the circuit. Therefore, the configuration example of FIG. 1A will be mainly described below.

  In FIG. 1A, the Colpitts oscillation circuit includes a transistor Q1, capacitors CA and CB, and an inductance L, and a circuit in which a series circuit of capacitors CA and CB and an inductance L are connected in parallel to the base terminal of the transistor Q1. Is connected. The capacitors CA and CB and the inductance L constitute a feedback circuit (first feedback circuit) that feeds back the signal at the collector terminal of the transistor Q1 to the base terminal. Note that a voltage obtained by dividing the power supply voltage VCC by the resistors RA and RB is applied to the base terminal. In addition, a resistor RE for determining a bias current is connected to the emitter terminal. The first feedback circuit constitutes a feedback circuit for performing oscillation by feedback from the output of the oscillation stage amplifier to the input. Here, the transistor Q1 is an oscillation stage transistor of the Colpitts oscillation circuit, and constitutes a current control source of the oscillation circuit.

  The oscillation circuit of the present invention includes the second feedback circuit H2 in the Colpitts oscillation circuit described above. The second feedback circuit H2 feeds back a current proportional to the voltage at the emitter terminal of the transistor Q1 to the base terminal of the transistor Q1, thereby increasing the negative resistance of the oscillation circuit at a predetermined frequency and equivalently. Adjust reactance components.

  Here, current feedback from the second feedback circuit H2 to the base terminal is performed via the feedback capacitor CF. The feedback capacitor CF is replaced by a collector-base capacitor provided in the transistor Q1 in addition to a configuration in which one end of the capacitor is connected to the base terminal of the transistor Q1 as a capacitor provided in the second feedback circuit H2. Alternatively, the output terminal of the second feedback circuit H2 may be connected to the collector terminal of the transistor Q1 via a capacitor (a sufficiently large coupling capacitor) different from the previous feedback capacitor. The current connected to the collector terminal of the transistor Q1 is fed back to the base terminal of the transistor Q1 via the collector-base capacitance.

  With this configuration, the current Ix from the inductance L is input to the transistor Q1, and in-phase current is fed back from the second feedback circuit H2, and the current entering the transistor Q1 increases, thereby increasing the negative resistance. To do.

  As described above, the circuit shown in FIG. 1A has a configuration in which the collector terminal of the transistor Q1 is directly connected to the power supply, whereas the circuit shown in FIG. The load resistance Rc is connected between the two. Therefore, the circuit configuration of FIG. 1A can be the same as that of FIG. 1B in which the load resistance is zero.

  2 shows a circuit configuration in which attention is paid only to an AC signal in the circuit of FIG. 1, and FIG. 3 shows a circuit configuration in which the circuit configuration of FIG. 2 is rewritten to a 2-port type. The oscillation circuit is classified into a 2-port type and a 1-port type. The 1-port type is a one-terminal pair circuit (two-terminal circuit) that uses the vibration of the resonator to compensate for the loss of the resonator with the negative resistance of the active circuit to maintain the vibration. Yes, the 2-port type is a two-terminal pair circuit (four-terminal circuit), and is an oscillation method in which oscillation is generated by feedback from output to input. The 1-port type oscillator can be rewritten into a 2-port circuit to apply feedback analysis.

  FIG. 2 shows a single-port type only for an AC signal in the oscillation circuit of the present invention. When the circuit of FIG. 2 is rewritten to a 2-port type, the oscillation circuit of the present invention can be represented by the 2-port type circuit configuration shown in FIG.

  In FIG. 3, the first feedback circuit H1 is represented by an LC circuit having an inductance L and capacitors CA and CB, and the second feedback circuit H2 is fed back to the oscillation stage A through the feedback capacitor CF.

  4 and 5 are equivalent circuits, FIG. 4 is the one-port equivalent circuit of FIG. 2, and FIG. 5 is an equivalent circuit when rewritten to the two-port type of FIG.

  4 and 5, the transistor is represented by a current control source, and multiplies the input terminal voltage VA by a mutual conductance gm to output a current VAgm. The second feedback circuit H2 outputs a current based on the terminal voltage VB of the oscillation stage by the current control source and the first feedback circuit H1, and feeds back to the input terminal of the current control source A via the feedback capacitor CF. To do.

  FIG. 6 is a vector diagram for explaining the effect of the second feedback circuit H2 of the present invention.

  In FIG. 6, the terminal voltage of the capacitor CA of the Colpitts oscillation circuit is delayed by 90 degrees with respect to the transducer current Ix. The collector current of the transistor is gm times in phase with the terminal voltage VA of the capacitor CA.

  Since the voltage generated by the collector current gm · VA of the transistor flowing through the capacitor CB is obtained by multiplying the impedance of CB by gm · VA, the phase is further delayed by 90 degrees from the collector current. Therefore, the phase is exactly 180 degrees behind the transducer current Ix. That is, a negative voltage is generated in the capacitor CB and becomes a negative resistance -R.

  FIG. 6A shows the negative resistance when only the current Ix flowing through the inductance flows through the Colpitts oscillation circuit without performing current feedback according to the present invention. On the other hand, FIG. 6B shows a negative resistance when the feedback current IF flows through the Colpitts oscillation circuit in addition to the current Ix flowing through the inductance by performing current feedback according to the present invention.

When the feedback current IF having the same phase as the vibrator current Ix is fed back to the Colpitts oscillation circuit by the second feedback circuit, the current Ib flowing through the transistor increases as (Ix + IF). Due to the increase in the current Ib, a negative voltage is generated in the capacitor CB, as described above, and its negative resistance is -R '(= gmVA' / jωCB). The negative resistance -R 'increases by feeding back the current because VA' of the collector current gmVA 'increases as the current Ib increases.
In the first embodiment, the feedback capacitor CF plays a role of adjusting the magnitude and phase of the current fed back from the second feedback circuit to the oscillation stage amplifier circuit, and connecting the second feedback circuit and the transmission stage amplifier circuit to a direct current. It also has the role of separating and returning only the alternating current. In the second embodiment, the feedback capacitor CF plays a role of adjusting the magnitude and phase of the current fed back from the second feedback circuit to the oscillation stage amplifier circuit.

  FIG. 6 will be described in detail below. Here, description will be made using the simplified equivalent circuit shown in FIG. An ideal case where the output impedance R0 of the second feedback circuit is infinite and the feedback capacitance is sufficiently large will be described.

  The current flowing from the coil (inductor) into the active circuit of the oscillation circuit is Ix.

  First, the case of a normal Colpitts oscillation circuit without the second feedback circuit H2 will be described. At the terminal of the capacitor CA, a voltage VA having a magnitude of (1 / ωCA) times Ix and a phase delayed by 90 degrees from Ix is generated. When the transconductance of the transistor is gm, a current of gm times VA flows through the control current source of the transistor. Since gm is a real number, the current generated by the mutual conductance is in phase with VA. Therefore, the current is delayed by 90 degrees from Ix. Further, the magnitude thereof is gm (Ix / ωCA) since the magnitude of VA is (Ix / ωCA). When this current flows through the capacitor CB, a voltage having a magnitude of gm (Ix / ωCA) (1 / ωCB) and a phase delayed by 90 degrees from the mutual conductance current is generated at the terminal.

  Since the mutual conductance current is 90 degrees behind the current Ix flowing from the coil, the voltage generated in the capacitor CB by the current flowing through the mutual conductance is 180 degrees behind the phase of Ix.

  That is, a voltage just opposite in phase to Ix is generated, and a negative voltage is generated with respect to Ix. Therefore, when viewed as a resistance, it becomes a negative resistance.

  On the other hand, not only current due to mutual conductance but also current Ix originally flowing from the base of the transistor flows in the capacitor. As in the case of the capacitor CA, the voltage generated at the terminal of the capacitor CB by this current is 90 degrees behind the phase Ix and has a magnitude of (1 / ωCB).

  The voltage generated in the capacitors CA and CB by the current Ix flowing from the coil is added, and the phase is delayed by 90 degrees from Ix, and the voltage is (Ix / ωCA) + (Ix / ωCB).

  The voltage Vx on the active circuit side as viewed from the coil terminal is a vector composition of the voltage due to the capacitors CA and CB and the voltage due to the negative resistance. Therefore, the phase of Vx is delayed with respect to Ix, and the delay is in the range of 0 degrees to -180 degrees.

  That is, the active circuit viewed from the coil terminal is capacitive, and can be represented as a series circuit of a negative resistance and a series capacitance of CA and CB.

  When the negative resistance is sufficiently large, the phase of Vx with respect to Ix is close to −180 degrees. The terminal voltage of the capacitor CB is a vector composition of the voltage generated by the control current of the transistor and the voltage by Ix. Therefore, the terminal voltages VB and Vx of the capacitor CB differ by the capacitive voltage of CA.

  Here, the feedback circuit H2 has a function of detecting, amplifying and phase-shifting the terminal voltage of the capacitor CB and applying it to the base terminal of the transistor in the same direction as Ix.

  Consider the case where the phase transition of H2 cancels the phase lag of V2 with respect to Ix. At this time, since the feedback current IF is originally in phase with the current Ix applied to the base terminal of the transistor, the current flowing into the base of the transistor is simply the sum of these and becomes Ix + IF.

The magnitude of IF is a value obtained by multiplying Ix by the magnitude of negative resistance (gm / (ω ² CA · CB)) and the degree of amplification A2 of H2, and is a value greater than 1. Therefore, Ib increases in magnitude at the same phase as Ix.

  Since the generation mechanism of the negative resistance is exactly the same as described above, it can be considered that Ix has increased, and therefore the negative resistance increases.

  In other words, by adding the feedback circuit H2, the current Ix flowing from the coil is increased and added by the second feedback circuit H2, and flows into the base of the transistor, thereby increasing the negative resistance.

  Thus, when the phase transition of the second feedback circuit H2 cancels the phase lag of V2 with respect to Ix, only an increase in negative resistance is realized.

  Next, consider a case where the phase transition of the second feedback circuit H2 is 0 or positive, that is, there is no phase transition or the phase is advanced.

  At this time, since the feedback current IF fed back by the second feedback circuit H2 is a current whose phase is advanced from the current Ix flowing from the coil, the current Ib flowing to the base of the transistor is a vector composition of Ix and IF, Although the phase angle is small, the current is advanced in phase from Ix. The magnitude of Ib is larger than Ix.

  Also in this case, since the negative resistance generation mechanism is exactly the same, the phase is advanced by increasing Ix in the vector diagram without the second feedback circuit H2. That is, on the vector diagram, it may be considered that the inclination is higher than Ix when there is no second feedback circuit H2.

  Therefore, the vector diagram not only increases in size but also rotates counterclockwise, so that the terminal voltage Vx of the active circuit finally becomes larger than that without the second feedback circuit H2, Since the rotation is counterclockwise, not only the negative resistance but also the equivalent reactance is increased and the equivalent capacitance value is also increased.

  On the other hand, let us consider a case where the phase transition of the second feedback circuit H2 is negative, that is, there is a phase delay. At this time, since the current IF fed back by the second feedback circuit H2 is a current delayed in phase from the current Ix flowing from the coil, the current Ib flowing to the base of the transistor is a vector composition of Ix and IF, and Ix The current is more delayed in phase. The magnitude of Ib is larger than Ix.

  Also in this case, since the negative resistance generation mechanism is exactly the same, in the vector diagram without the second feedback circuit H2, Ix is increased to delay the phase. That is, on the vector diagram, it may be considered that the inclination is lower than Ix when there is no second feedback circuit H2.

  Therefore, the vector diagram not only increases in size but also rotates in the clockwise direction, so that the terminal voltage Vx of the active circuit finally becomes larger than that without the second feedback circuit H2, and the clock Since it is rotated in the direction, it is above the line Ix and not only the negative resistance increases, but the equivalent reactance is inductive.

  In the case of feedback using the emitter-follower shown in Non-Patent Document 3, the voltage applied to both ends of the base-collector capacitor Cbc is adjusted by feeding back the voltage from the emitter terminal to the collector terminal of the oscillation stage transistor. Thus, the current flowing through the base-collector capacitance Cbc is suppressed, and the influence of the base-collector capacitance Cbc is reduced.

  On the other hand, the second feedback circuit of the present invention increases the current entering the transistor Q1 by performing current feedback to the base terminal of the oscillation stage transistor, as described in the vector diagram of FIG. Thus, by increasing the negative resistance and appropriately giving a phase shift to the feedback current, the equivalent reactance of the oscillation circuit can be changed.

  When the output impedance R0 of the second feedback circuit is a finite value, it is necessary to set the feedback capacitance so as to enhance the effect of current feedback to the oscillation stage transistor.

  Specifically, the feedback capacitance is determined based on the impedance relationship between the base-collector capacitance (mainly Cbc) of the oscillation stage transistor and the impedance of the feedback circuit of R0.

  The oscillation circuit of the present invention has the effect of selectively increasing the negative resistance on the frequency axis by the second feedback circuit and improving the equivalent reactance. Therefore, hereinafter, the frequency characteristic of the negative resistance will be described with reference to FIGS. 7 to 10, and the equivalent reactance will be described with reference to FIG.

  First, the frequency characteristic of the negative resistance of the Colpitts oscillation circuit will be described with reference to FIG.

  FIG. 7 shows the frequency characteristic of the negative resistance when the capacitance CA is changed in a normal Colpitts oscillation circuit. In FIG. 7, when the capacitors CA and CB are reduced, the negative resistance of the Colpitts oscillation circuit moves to the high frequency side at the peak frequency, but the peak value gradually decreases. In addition, the capacitance value is significantly reduced in the vicinity of 1 GHz.

  On the other hand, according to the oscillation circuit of the present invention, the negative resistance can be increased. Here, the frequency characteristics of the normal Colpitts oscillation circuit are compared using the base feedback type circuit configuration shown in FIG.

  FIG. 8 shows a configuration example of an oscillation circuit similar to that shown in FIG. 1A, and shows a case where the load resistance Rc is zero. FIG. 9 shows a simulation result designed so that the peak frequency of the negative resistance is 300 MHz in the configuration of the oscillation circuit. According to the simulation result of FIG. 9, the frequency characteristic of negative resistance by the oscillation circuit of the present invention (indicated by “proposed” in the figure) is the frequency characteristic of negative resistance by the normal Colpitts oscillation circuit (indicated by “colpitts” in the figure). ), It is confirmed that the negative resistance at the planned 300 MHz is greatly increased.

  In the oscillation circuit of the present invention, the relationship between the negative resistance and the mutual conductance gm (indicated by gm2 in the figure) of the second feedback circuit will be described with reference to FIG. In FIG. 10, the case where the circuit configuration is the base feedback type circuit configuration as in FIG. 8 and the load resistance Rc is 0 will be described. FIG. 10 shows the frequency characteristic of the negative resistance when the mutual conductance gm (gain) of the second feedback circuit H2 is changed. According to the simulation result of FIG. 10, it is confirmed that the negative resistance increases as the mutual conductance gm increases, and becomes maximum at a specific value of gm.

  In the oscillation circuit of the present invention, the relationship between the equivalent reactance and the mutual conductance gm (indicated by gm2 in the figure) of the second feedback circuit will be described with reference to FIG. The circuit configuration used for the simulation of FIG. 11 is the same as the base feedback type circuit configuration shown in FIG. 8, and the load resistance Rc is zero. FIG. 11 shows the frequency characteristic of the equivalent reactance when the mutual conductance gm (gain) of the second feedback circuit H2 is changed. According to the simulation result of FIG. 11, it is confirmed that the equivalent reactance is changed by the mutual inductance gm, and the equivalent reactance component of the oscillation circuit can be changed from capacitive (gm2 = 20 m) to inductive (gm2 = 60 m).

  In the above example, the configuration example in which the load resistance Rc is set to 0 has been described. However, the oscillation circuit of the present invention may be configured to connect the load resistance Rc as shown in FIG. The influence of this load resistance Rc will be described with reference to FIG. The circuit configuration used in the simulation of FIG. 12 is the same as that of the base feedback type circuit configuration shown in FIG. 8, and the load resistance Rc (indicated by Rc1 in the figure) is 0, 50, 100Ω. is there.

  According to the simulation result of FIG. 12, the negative resistance increases as the load resistance Rc decreases. This reduction in negative resistance is presumed to be due to the mirror effect.

  In the oscillation circuit, the load resistance Rc is not necessary for the generation of the negative resistance, so that Rc = 0 is desirable to obtain a large negative resistance.

  Note that the mutual conductance gm of the second feedback circuit can be determined by the bias condition and can be controlled by the bias resistance of the transistor. The amplitude and phase characteristics of the second feedback circuit can also be controlled by parameters such as capacitance in the second feedback circuit.

  The oscillation circuit of the present invention can be configured in a plurality of forms. Hereinafter, a first embodiment and a second embodiment in which the form of the feedback capacitor used for current feedback from the emitter terminal to the base terminal of the oscillation stage transistor is different, and a third form in which the second feedback circuit has resonance characteristics, A fourth embodiment in which the second feedback circuit is constituted by a differential feedback circuit will be described.

  First, a 1st form is demonstrated using FIG. 13, FIG. In the first embodiment, the voltage of the emitter terminal of the oscillation stage transistor of the Colpitts oscillation circuit is amplified and phase-shifted by the second feedback circuit, and current is fed back from the output terminal of the second feedback circuit to the base terminal via the feedback capacitor. This is a direct base feedback type.

  FIG. 13 shows a circuit configuration example of the first form, and FIG. 14 shows frequency characteristics of the negative resistance according to the circuit configuration example of the first form. The load resistance Rc is 0.

  In the circuit configuration example shown in FIG. 13, the second feedback circuit is configured by a two-stage grounded-emitter amplifier. The input terminal of the first-stage grounded-emitter amplifier is connected to the emitter terminal of the transistor Q1, and the output terminal of the second-stage grounded-emitter amplifier is connected to the base terminal of the transistor Q1 via a feedback capacitor CF (a sufficiently small capacity). The output current of the two-stage grounded-emitter amplifier is fed back to the base terminal of the oscillation stage transistor Q1.

  The feedback capacitance CF may be a small value of about 1/10 to 1/5 of the series connection capacitance of CA and CB. For example, when CA and CB are 2 to 3 pF, the feedback capacitance CF is preferably about 0.2 pF to 1 pF.

  The frequency characteristic of the negative resistance can be changed by changing the parameter of the second feedback circuit. For example, when the second feedback circuit is constituted by a two-stage grounded-emitter amplifier, resistors RD2 and RD3 connected to the collector terminal constituting the grounded-emitter amplifier, resistors RE2 and RE3 connected to the emitter terminal, and capacitors The frequency characteristics of the negative resistance can be changed by changing circuit constants such as CE2 and CE3.

  In changing this parameter, a plurality of resistors in which these resistors are connected in parallel can be connected and disconnected via the switch means, and the resistance value of the negative resistance is selected by selecting the resistance value to be connected by switching the switch means. Frequency characteristics (frequency and magnitude at which negative resistance peaks) can be selected.

  For example, the oscillation frequency of 900 MHz and 1.2 GHz used in the wireless LAN can be switched and output by one oscillation circuit.

  The frequency characteristics shown in FIG. 14 are simulation results when designed at 300 MHz, 600 MHz, 900 MHz, and 1.2 GHz.

  By setting circuit constants, it is possible to design a configuration that obtains sufficient negative resistance up to 2.0 GHz.

  Next, a 2nd form is demonstrated using FIG. 15, FIG. In the second embodiment, the collector-base capacitor of the oscillation stage transistor is used as the feedback capacitor CF ′ as the feedback capacitor, and the output terminal of the second feedback circuit is a capacitor different from the feedback capacitor CF ′ (a sufficiently large coupling capacitor). This is a mode in which the current is fed back from the collector terminal to the base terminal via the collector-base capacitance through the collector terminal of the oscillation stage transistor via C31.

  FIG. 15 shows a circuit configuration example of the second form, and FIG. 16 shows frequency characteristics of the negative resistance according to the circuit configuration example of the second form.

  The circuit configuration example of FIG. 15 is substantially the same as the circuit configuration example of FIG. 13 described above, and the second feedback circuit is configured by a two-stage grounded-emitter amplifier. The difference from the circuit configuration of the first embodiment shown in FIG. 13 is that the output current of the second-stage grounded-emitter amplifier is fed back to the collector terminal of the transistor Q1, which is an oscillation stage transistor, via the capacitor C31. Is a point where a resistor RD1 is connected as the load resistor Rc. Here, the capacitor C31 is an AC coupling capacitor that blocks direct current, and for example, a large capacitance of about 100 pF to 1000 pF is desirable. In this case, the resistor RD1 is essential.

  In the circuit configuration example shown in FIG. 15, the second feedback circuit is configured by a two-stage grounded-emitter amplifier as in FIG. The input terminal of the first-stage grounded-emitter amplifier is connected to the emitter terminal of the transistor Q1, and the output terminal of the second-stage grounded-emitter amplifier is connected to the collector terminal of the transistor Q1 via the capacitor C31. The output current of the two-stage grounded-emitter amplifier is fed back from the collector terminal of the oscillation stage transistor Q1 to the base terminal via the base-collector capacitance.

  Further, the frequency characteristic of the negative resistance can be changed by changing the parameter of the second feedback circuit, as in the above circuit configuration example. For example, the second feedback circuit is changed by a two-stage grounded emitter amplifier. When configured, the negative resistance is obtained by changing circuit constants such as resistors RD2 and RD3 connected to the collector terminal constituting the common-emitter amplifier, resistors RE2 and RE3 connected to the emitter terminal, and capacitors CE2 and CE3. The frequency characteristics can be changed.

  Also, in changing the parameters, a plurality of resistors connected in parallel with each other can be connected / disconnected via the switch means, and the resistance value to be connected is selected by switching the switch means, thereby reducing the frequency of the negative resistance. Properties can be selected.

  The frequency characteristics shown in FIG. 16 are simulation results when designed at 300 MHz, 600 MHz, 900 MHz, and 1.2 GHz.

  By setting circuit constants, it is possible to design a configuration that obtains sufficient negative resistance up to 2.0 GHz.

  Next, a 3rd form is demonstrated using FIGS. In the third form, the second feedback circuit has a resonance characteristic.

  17 shows a schematic diagram of the circuit of the third embodiment, FIG. 18 shows a circuit configuration example of the third embodiment, and FIG. 19 shows frequency characteristics of the negative resistance according to the circuit configuration example of the third embodiment. Yes.

  The schematic circuit configuration of FIG. 17 is obtained by giving resonance characteristics to the second feedback circuit in the circuit configuration example of FIG. 1, and FIG. 18 is an example of the circuit configuration. In this circuit configuration, resonance characteristics can be provided by adding a parallel circuit B of inductance and capacitance. Also in this circuit configuration, the output of the second feedback circuit composed of the two-stage grounded-emitter amplifier is fed back to the base terminal of the oscillation stage transistor.

  According to this circuit configuration, the frequency range where the peak of the negative resistance is obtained by the resonance characteristics can be narrowed.

  The circuit configuration example of FIG. 18 can also be applied to the circuit configuration shown in FIG. 15 of the second embodiment.

  In addition, the second feedback circuit can be provided with a plurality of resonance characteristics so as to be switchable, and can switch multi-frequency by switching the oscillation frequency.

  The frequency characteristics shown in FIG. 19 are simulation results when designed with Q = 20.

  Compared with the simulation results shown in FIG. 14 of the first embodiment and FIG. 16 of the second embodiment, it can be seen that the frequency band of the negative resistance peak can be narrowed.

  Next, a 4th form is demonstrated using FIG. The fourth form is a form in which the second feedback circuit is constituted by a differential feedback circuit. FIG. 20A shows a schematic diagram of the circuit of the fourth embodiment, and FIG. 20B shows an example of the circuit configuration of the fourth embodiment.

  The circuit configuration example of FIG. 20A is obtained by configuring the second feedback circuit with a differential feedback circuit in the circuit configuration example of FIG. 1, and FIG. 20B shows an example of the circuit configuration. It is.

  According to the differential feedback circuit, a differential output can be obtained and common-mode noise can be removed. Further, phase noise can be improved as compared with a circuit in which two stages of grounded emitters are connected.

  In FIG. 20B, the differential output can be taken out from the collector terminals of the two transistors of the differential pair or from the load indicated by the square in the drawing.

  Next, negative resistances in the first and second embodiments of the present invention, a normal Colpitts oscillation circuit, and a circuit example proposed in Non-Patent Document 3 will be compared using FIG. In order to clearly compare the effects of the second feedback circuit, the circuit element values of the oscillation stage (that is, the normal Colpit oscillation circuit portion) are equal in these circuits.

  When comparing the simulation results of the respective configuration examples shown in FIG. 21, the magnitude of the negative resistance is determined according to the first form (indicated by Base FB in the figure) and the second form (indicated by Collector FB in the figure). An example of a circuit using an emitter / follower proposed in Non-Patent Document 3 (indicated by “Follower” in the figure) and a normal Colpitts oscillation circuit (indicated by “colpitts” in the figure) are in this order. A large negative resistance can be obtained.

  Next, simulation results of the influence of the base-collector capacitance will be described with reference to FIGS.

  FIG. 22 is a simulation result of the influence of the base-collector capacitance in the first embodiment of the present invention. Here, in order to investigate the influence of the base-collector capacity, the case where the base-collector capacity is removed (Cjc = 0 in the figure) and the case where it exists (Cjc = 0.16p in the figure) ). Here, Rc = 0 and there is no mirror effect.

  According to the simulation results, due to the effect that the feedback signal is injected into the base terminal via the base-collector capacitance, a larger negative resistance is obtained when the base-collector capacitance is present.

  FIG. 23 is a simulation result of the influence of the base-collector capacitance in the second embodiment of the present invention. Again, in order to investigate the influence of the base-collector capacitance, the case where the base-collector capacitance is removed (Cjc = 0 in the figure) and the case where it exists (Cjc = 0.16p in the figure) ). Here, since the load resistance R is connected, there is a mirror effect, so that the negative resistance is smaller than that in the first embodiment shown in FIG. Further, according to the simulation result, due to the effect that the feedback signal is injected into the base terminal via the base-collector capacitance, a larger negative resistance can be obtained when the base-collector capacitance exists.

  FIG. 24 is a simulation result of the influence of the base-collector capacitance in the circuit example using the emitter-follower proposed in Non-Patent Document 3.

  According to this simulation result, there is little change due to the base-collector capacitance. That is, the emitter-follower amplifier sets the voltage at the emitter terminal and the collector terminal of the oscillation stage transistor to be substantially equal, and the magnitude of the negative resistance is sufficiently larger than the capacitive voltage of the capacitor CB. Therefore, the voltage at the base of the transistor Q1 is almost equal to the voltage at the collector of the transistor Q1, thereby reducing the negative resistance due to the influence of the base-collector capacitance. It is thought that.

  Next, application of the oscillation circuit of the present invention to a voltage controlled oscillation circuit will be described with reference to FIG. The voltage controlled oscillation circuit usually changes the frequency by inserting a variable capacitor Cs between the inductance L and the active circuit.

  The oscillation frequency is determined by the series capacitance of the variable capacitance Cs and the equivalent capacitance Ci of the active circuit and the inductance L. Therefore, when the equivalent capacitance Ci is small, the series capacitance of the variable capacitance Cs and the equivalent capacitance Ci is mainly determined by the equivalent capacitance Ci, so that the change in the oscillation frequency is small even if the variable capacitance Cs is changed. Therefore, the negative resistance can be compensated for by increasing CA and CB by increasing the negative resistance by the oscillation circuit of the present invention.

  This makes it possible to take a variable amount of frequency without difficulty in oscillation or stopping oscillation due to fluctuations.

  The variable capacitor Cs is usually configured using a varicap, but this varicap is not suitable for integration because of problems such as volume and characteristics, and is also unsuitable for frequency control from the viewpoint of linearity of oscillation frequency control. . On the other hand, according to the oscillation circuit of the present invention, the frequency can be changed without using a variable capacitor such as a varicap by directly changing the equivalent capacitance Ci by the second feedback circuit.

  FIG. 26 shows a configuration example of a voltage-controlled oscillation circuit using a resonator such as a piezoelectric vibrator, which is an example including an extension coil. FIG. 26B is an equivalent circuit of a resonator.

  In a voltage controlled oscillation circuit including a piezoelectric vibrator instead of an inductance, it is necessary to reduce the influence of Ci in order to increase the variable range of the oscillation frequency. Usually, in order to reduce the influence of Ci, an extension coil is inserted in series with a vibrator. However, by making the equivalent capacitance of the active circuit (second feedback circuit) of the present invention inductive, a voltage controlled oscillation circuit The extension coil can be replaced. The structure which does not require the extension coil or the extension coil can be reduced. As a result, the circuit configuration can be reduced in size.

  The present invention can be applied to an oscillator of a wireless system, an oscillator of an optical transmission system such as SONET / SDH, a reference frequency source such as a high-vision camera, and an oscillator such as a wireless LAN or UWB.

It is a figure which shows the basic circuit structure of the oscillation circuit of this invention. It is a figure which shows the circuit structure which paid its attention only about the alternating signal in the circuit of FIG. It is a figure which shows the circuit structure which rewritten the circuit structure of FIG. It is a figure which shows the simplified equivalent circuit of the basic circuit structure of the oscillation circuit of this invention. It is a figure which shows the 2 port type simplified equivalent circuit of the basic circuit structure of the oscillation circuit of this invention. It is a vector diagram for demonstrating the effect of the 2nd feedback circuit H2 of this invention. It is a figure which shows the frequency characteristic of a negative resistance when the capacity | capacitance CA is changed in a normal Colpitts oscillation circuit. It is a figure for demonstrating the circuit structure of the base feedback type | mold of this invention. It is a figure which shows the simulation result designed so that the peak frequency of a negative resistance might be 300 MHz in the structure of the oscillation circuit of this invention. In the oscillation circuit of this invention, it is a figure explaining the relationship between negative resistance and the mutual conductance gm of a 2nd feedback circuit. It is a figure which shows the frequency characteristic of an equivalent reactance when the mutual conductance gm of the 2nd feedback circuit H2 is changed. It is a figure for demonstrating the influence by load resistance Rc. It is a figure which shows the circuit structural example of a 1st form. It is a figure which shows the frequency characteristic of the negative resistance by the circuit structural example of a 1st form. It is a figure which shows the circuit structural example of a 2nd form. It is a figure which shows the frequency characteristic of the negative resistance by the circuit structural example of a 2nd form. It is a figure which shows the basic circuit structure of the circuit of a 3rd form. It is a figure which shows the circuit structural example of a 3rd form. It is a figure which shows the frequency characteristic of the negative resistance by the circuit structural example of a 3rd form. It is a figure for demonstrating a 4th form. It is a figure for comparing the negative resistance of the circuit example proposed in the 1st form, the 2nd form, a normal Colpitts oscillation circuit, and nonpatent literature 3. It is a simulation result of the influence by the capacity | capacitance between base-collectors in the 1st form of this invention. It is a simulation result of the influence by the capacity | capacitance between base-collectors in the 2nd form of this invention. It is a simulation result of the influence by the capacity | capacitance between bases and collectors in the circuit example by the emitter follower proposed by the nonpatent literature 3. It is a figure for demonstrating the application to the voltage control oscillation circuit of the oscillation circuit of this invention. 2 is a configuration example of a voltage controlled oscillation circuit using a resonator such as a piezoelectric vibrator. 10 is a diagram illustrating an oscillation circuit of Non-Patent Document 3. FIG.

Explanation of symbols

A ... Oscillation stage Q1 ... Oscillation stage transistor H1 ... First feedback circuit H2 ... Second feedback circuit CF ... Feedback capacitance

Claims (21)

In an oscillation circuit that oscillates by feedback amplification from output to input,
An oscillation circuit characterized by performing a first feedback for the oscillation and a double feedback of a second feedback that performs amplitude and phase transition.   A first feedback circuit for generating a negative resistance included in the Colpitts oscillation circuit and a second feedback circuit for current feedback from the emitter terminal of the oscillation stage transistor of the Colpitts oscillation circuit to the base terminal are provided. Oscillator circuit.   3. The oscillation circuit according to claim 2, wherein current is fed back from the emitter terminal to the base terminal via a second feedback circuit and a feedback capacitor.   4. The oscillation according to claim 2, wherein the second feedback circuit includes an active circuit, and the negative resistance and / or the equivalent reactance component is changed according to an amplification characteristic and a phase transition characteristic included in the active circuit. circuit.   5. The oscillation circuit according to claim 4, wherein a change in the negative resistance and / or equivalent reactance component by the second feedback circuit is independent of a capacitance that determines a frequency of the Colpitts oscillation circuit.   The feedback capacitor is a capacitive element included in a second feedback circuit, and the capacitive element is connected between an output terminal of the second feedback circuit and a base terminal of an oscillation stage transistor. The oscillation circuit according to any one of 3 to 5. The feedback capacitor is a collector-base capacitor of an oscillation stage transistor, and the emitter terminal is connected to a collector terminal of the oscillation stage transistor via a second feedback circuit. An oscillation circuit according to claim 1.   The oscillation circuit according to claim 6, wherein a collector terminal of the oscillation transistor and a power source are directly connected.   8. The oscillation circuit according to claim 6, wherein a load resistor is connected between the collector terminal of the oscillation transistor and a power source.   10. The oscillation circuit according to claim 6, wherein the second feedback circuit includes a two-stage grounded-emitter amplifier.   6. The oscillation circuit according to claim 2, wherein the second feedback circuit has a resonance characteristic.   The oscillation circuit according to claim 11, wherein the second feedback circuit includes a plurality of resonance characteristics that can be switched.   6. The oscillation circuit according to claim 2, wherein the second feedback circuit is a differential feedback circuit.   14. The oscillation circuit according to claim 2, wherein the oscillation stage transistor is a bipolar transistor or a MOS transistor.   The oscillation circuit according to claim 1, wherein an inverter oscillator is used for the oscillation stage that performs the oscillation.   16. The oscillation circuit according to claim 1, further comprising an inductance or a resonator as an oscillation element of the oscillation stage.   The resonator is a crystal resonator, a ceramic resonator, a surface acoustic wave resonator, an end surface reflection resonator, a piezoelectric resonator of a bulk resonator, or a dielectric resonator. The oscillation circuit described in 1. It has inductance, extension coil and capacity,
18. A voltage-controlled oscillation circuit, wherein the frequency is controlled by using the capacitance as an equivalent capacitance of an active circuit of the oscillation circuit according to claim 1 and changing the equivalent capacitance.   19. The voltage controlled oscillation according to claim 18, further comprising a resonator in place of the inductance, wherein the equivalent reactance of the active circuit is made inductive instead of a part or all of the expansion coil of the voltage controlled oscillation circuit. circuit.   2. The resonator according to claim 1, wherein the resonator is a crystal resonator, a ceramic resonator, a surface acoustic wave resonator, an end surface reflection resonator, a piezoelectric resonator of a bulk resonator, or a dielectric resonator. 20. The voltage controlled oscillation circuit according to 19.   The frequency temperature characteristic is compensated by making the equivalent capacitance or inductance of the active circuit of the oscillation circuit according to any one of claims 1 to 17 variable according to the frequency temperature characteristic of the oscillation circuit. Temperature compensated oscillation circuit.

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