JP2006017486A - Spectral analysis method and nuclear magnetic resonance apparatus using nuclear magnetic resonance - Google Patents

Abstract

In an NMR apparatus, since a mixer is provided for reception processing in order to perform frequency conversion, there is a risk that performance may deteriorate due to generation of noise or change in characteristics. The conventional phase correction method cannot cope with a fast change such as the phase change of the A / D sampling clock and cannot improve the S / N by the integration process.
An A / D converter clock frequency is made lower than the reception frequency of the received signal, and a received signal digitized by a signal having a frequency calculated from both is detected, thereby eliminating the need for a mixer. By detecting the phase change from the received signal, correcting the phase in the frequency domain, and integrating the received signals that have changed in the time domain again, it is possible to correct the phase change that has a fast time change and to improve performance.
[Selection] Figure 1

Description

  The present invention relates to a spectrum analysis method using nuclear magnetic resonance and a nuclear magnetic resonance apparatus for measuring a chemical bond state or the like of a sample to be measured using nuclear magnetic resonance (NMR).

  In a nuclear magnetic resonance apparatus, a sample to be measured is arranged in a static magnetic field, and a high-frequency magnetic field having a resonance frequency is transmitted in a pulse manner via a transmission / reception coil arranged around the sample. As a result, the resonance signal induced in the transmission / reception coil is received as a free induction decay signal (hereinafter abbreviated as FID signal). The NMR spectrum is obtained by Fourier-transforming this FID signal, and the chemical bonding state of the sample to be measured can be analyzed from the frequency and intensity of the measured spectrum. This technique is described in Patent Document 1 (Japanese Patent No. 3478924), Patent Document 2 (Japanese Patent Laid-Open No. 5-154128), and Patent Document 3 (Japanese Patent Laid-Open No. 10-99294).

  The high-frequency signal transmitted varies depending on the nuclide to be analyzed and the static magnetic field applied by the magnet, but is usually a frequency of several tens to several hundreds of MHz. The received signal has a plurality of spectra having frequencies in the vicinity of the transmission frequency, and is distributed in a range away from several kHz to several hundred kHz centering on the transmission frequency. Thus, although the transmission frequency which is the center frequency is high, the observation bandwidth is very narrow compared to the center frequency.

  Therefore, the received signal is input to the mixer, the center frequency is converted to a low frequency, and the center frequency is converted into a DC component using an IQ detector (In-Phase Quadrature detector). Therefore, the signal after IQ detection is converted into a signal having a maximum component of several kHz to several hundred kHz. When the signal is subjected to analog-digital conversion (A / D conversion) and Fourier transform (FFT) processing, the spectral distribution of the FID signal is detected.

  Further, the intensity of each spectrum is not constant and varies depending on the sample. For this reason, the transmission and reception processes described above are repeated a plurality of times, and the influence of noise is reduced by integrating the signals in the time domain. Also, analysis processing is performed using the absorption component of the spectral intensity of the FID signal. In this case, if the transmission and reception phase differences are different, the dispersion component is mixed into the absorption component and the measurement accuracy is reduced. It was necessary to correct.

Japanese Patent No. 3478924 JP-A-5-154128 Japanese Patent Laid-Open No. 10-99294

  As described above, the mixer and the detector are required for the reception process. Since these devices are composed of analog circuits, adjustment work at the time of circuit manufacture is necessary, and correction for characteristic changes due to aging of elements is also necessary. Furthermore, the mixer performs frequency conversion by multiplying local transmission signals with different frequencies with the FID signal. However, because the local transmission signal line and the FID signal line are close to each other on the circuit layout, interference occurs between the signals and the noise generation source. There is a risk of becoming.

  As described above, the phase difference between the transmission signal and the reception signal needs to be constant. The causes of the phase change are measurement delay, phase characteristics of circuit elements such as filters, amplifier gain, and phase change due to off-resonance, and several correction methods have been proposed. However, the method of correcting the phase change after IQ conversion and further integration processing cannot cope with a phase change for each transmission / reception. Further, no consideration is given to the phase change accompanying the frequency and phase change of the sampling clock when digitizing using the A / D converter. An object of the present invention is to provide a high-performance NMR apparatus by solving the above-described problems caused by a mixer of a receiving circuit and correcting a phase change.

According to one problem solving means of the present invention, the NMR apparatus does not provide a mixer on the receiving circuit side, performs A / D conversion without frequency conversion of the FID signal, and the frequency is lower than the frequency of the FID signal. The sampling frequency is IQ detected at a frequency calculated from the center frequency fo and sampling frequency fs of the FID signal. Thereby, since a mixer can be omitted, a high-performance NMR apparatus can be realized.
Furthermore, according to another problem solving means of the present invention, the NMR apparatus corrects the phase every time an FID reception signal is received. Thereby, since it can respond also to the phase change of the sampling clock with comparatively quick time fluctuation, a high-performance NMR apparatus can be realized.

  According to the present invention, the mixer can be deleted from the receiving circuit, so that the time for manufacturing and adjustment can be shortened, and the mixing of noise can be reduced. Moreover, since the phase change having a fast temporal variation can be corrected, the spread of the observation spectrum width can be suppressed. For this reason, it is possible to realize high-performance NMR transmission that can obtain a signal with little noise and a narrow spectrum width.

  The present invention will be described below with reference to FIGS. 1 and 2 showing an embodiment of the present invention. FIG. 1 is a block diagram showing the overall configuration of a nuclear magnetic resonance (NMR) apparatus. FIG. 2 is a graph showing the frequency of the free induction attenuation signal (FID signal) and the frequency change after digitization by a digital converter (A / D converter).

  In FIG. 1, a user of an NMR apparatus uses a personal computer 10 to set a pulse sequence such as a transmission frequency, a phase, an intensity, a transmission width, a transmission interval, the number of integration processes, a reception observation width, and the like. The set information is sent to the timing control circuit 12 and the reception processing circuit 14. The reception processing circuit 14 includes an A / D converter 16, an IQ detection processing unit 18, a phase correction unit 20, an integration processing unit 22, and the like.

  According to the signal information from the timing control circuit 12, the frequency / phase variable transmitter 24 generates a signal. The reference oscillator 26 having high stability oscillates at a frequency lower than the frequency transmitted to the transmission / reception coil 28 and supplies the output signal to the frequency / phase variable oscillator 24 and the two frequency synthesizers 30a and 30b. The frequency synthesizers 30a and 30b convert the output signal from the highly stable reference transmitter 26 to a high frequency at a frequency lower than the frequency transmitted to the transmission / reception coil 28.

  The mixer 32 converts the signal to the transmission frequency designated by the user in accordance with the signal from the frequency / phase variable oscillator 24 and the signal from the frequency synthesizer 30a. Further, a transmission signal having a predetermined intensity is transmitted to the transmission / reception coil 28 via the variable amplifier 34 a and the power amplifier 36. The transmission width of the transmission signal is several μsec to several hundred μsec, and is a short and pulse-like transmission signal.

  After the transmission is completed, a signal output from the transmission / reception coil 28 is sent to an analog-digital converter (hereinafter abbreviated as A / D converter) 16 of the reception processing circuit 14 via a variable amplifier 34b and a band pass filter (BPF) 38. input. The sampling frequency of the A / D converter 16 is determined by the frequency supplied from the frequency synthesizer 30b. The analog-digital converted signal is processed by the IQ detection processing unit 18, the phase correction unit 20, and the integration processing unit 22, and given to the personal computer 10.

  As shown in FIG. 3, the conventional NMR apparatus is configured such that the received signal from the transmitting / receiving coil 28 is input to the mixer 32b via the variable amplifier 34b and the bandpass filter (BPF) 38, and the frequency of the received signal is further increased. Convert to a lower frequency. The output signal of the mixer 32 b is supplied to the A / D converter 16 through the detection circuit 40.

  Since the received signal has a plurality of spectra having different frequencies and intensities, it is easily affected by noise. In the mixer 32b, in order to mix the signal from the frequency synthesizer 30 and the received signal, both signal lines are close to each other and noise due to interference is likely to occur. For this reason, it is necessary to take measures to prevent interference in circuit mounting.

  In addition, the analog circuit cannot avoid the aging of the element, and especially the phase change needs to be corrected because the spectrum width is widened as described above. However, in the embodiment as shown in FIG. 1, since the received FID signal is A / D converted without frequency conversion to the low frequency side, the above-described problem does not occur. 3 that are the same as those in FIG. 1 have the same functions, and their operations are omitted.

  Assuming that the frequency of the signal is f, in order to digitize this signal and detect it as its frequency component, it is necessary to sample at a frequency that is twice the frequency 2f or higher according to the sampling theorem. Since the frequency of the received signal reaches a maximum of close to 1 GHz, a sampling frequency fs of 2 GHz or more is required. Such a high-speed A / D converter is expensive, and the amplitude resolution is as low as about 8 bits. Furthermore, the data processing after A / D conversion also requires a high-speed processing circuit because the data is input at a rate of 2 GHz, and is very expensive.

  In the case of A / D conversion, a low-pass filter (hereinafter abbreviated as LPF) is inserted in the preceding stage in order to remove frequencies of fs / 2 or higher. Without a low-pass filter (LPF), the frequency of fs / 2 or higher is detected by being converted to a frequency of fs / 2 or lower due to an aerial effect. This will be described with reference to FIG. The frequencies of the received signal are f1 and f2, and the converted frequencies F1 and F2 differ as shown in the following formulas depending on whether the integer value n obtained by dividing the frequency by fN = fs / 2 is an even number or an odd number.

n: even number F2 = f2-n * fN
n: When odd number F1 = (n + 1) * fN−f1
For example, when the reception frequency f1 is 450 MHz and the sampling frequency fs = 125 MHz, n is an odd number, and the frequency of the signal after A / D conversion is 50 MHz. That is, since f1 / fN = f1 × (2 / fs) = 450 / (125/2) = 450 / 62.5≈7, F1 = (7 + 1) 62.5−450 = 50 MHZ. When the reception frequency is 150 MHz and fs = 125 MHz, n is an even number, and the frequency of the signal after A / D conversion is 25 MHz. That is, since f1 / fN = f1 × (2 / fs) = 150 / (125/2) = 150 / 62.5≈2, F2 = f2-2 × 62.5 = 150-125 = 25 MHZ .

  As described above, according to the present embodiment, frequency conversion is possible without using the conventional mixer 32b and only A / D conversion at a low sampling frequency, and relatively inexpensive A / D conversion. An NMR apparatus can be realized by using the device 16 and a signal processing circuit.

  As described above, if the transmission frequency is fo, the received signal has a frequency component in the vicinity of fo. When the received signal is frequency-converted in this embodiment, it is necessary to perform IQ detection at the converted frequency. In this way, the fo frequency becomes a DC component, and the spectrum in the vicinity of fo can be detected at a low sampling rate, and Fourier transform processing (hereinafter referred to as FFT processing) can be performed.

  FIG. 4 is a block diagram showing a detailed configuration of the IQ detection processing unit 18 and the phase correction unit 20 to which the output signal is given. The IQ detector (In-Quality detector) is also called a two-phase detector and detects a frequency difference and a phase difference between the detected signal y and the reference signal x. The signal y is input to two multipliers. One of the signals x is input to the multiplier, and the other is input to the multiplier whose phase is changed by 90 ° by the phase shifter. The output of each multiplier is input to a low-pass filter, and only the bandwidth to be detected is passed to detect the I and Q signals. If the frequencies of the signals x and y are the same, the detected signal does not change with time in coordinates on the IQ plane. If the frequency is different, it rotates on the IQ plane, and the rotation speed becomes the frequency difference.

  The A / D converted signal has a component whose frequency is converted to Fo as described above. The IQ detection processing unit 18 converts the frequency Fo component into DC, and generates a cos signal whose phase is shifted by 90 degrees with respect to the signals sin and sin of the phase θ at the frequency Fo from the numerical control oscillator (NCO) 42. To do.

  These signals are digital values. The center frequency Fo is frequency-converted into a DC component by multiplying the signal after A / D conversion by the sin and cos signals by the multipliers 44a and 44b, respectively. The outputs of the multipliers 44a and 44b are given to the inverse Fourier transformer 52 through the low-pass filters 46a and 46b, the Fourier transformer (FFT transformer) 48, and the phase change detector 50. In this way, the real part signal I and the imaginary part signal Q are obtained.

  In this way, compared to a method using two conventional analog mixers for digital processing, there is no possibility of noise being mixed, and therefore, detection can be performed with high accuracy. Furthermore, when IQ detection is performed by an analog circuit, it is necessary to generate sin and cos signals, which are generated by inputting one of the signals to a 90-degree transporter.

  Since the transfer device has frequency characteristics and the NMR apparatus has a resonance frequency different by several hundred MHz depending on the nuclide, it is necessary to provide a transfer device for each nuclide to be measured. However, since the numerically controlled oscillator (NCO) 42 does not distort the sin and cos signals due to the set frequency, the above problem does not occur. When the phase difference between the sin and cos signals deviates from 90 degrees, an image signal appears, which causes a problem in observing the spectrum. For example, when one spectrum of the received signal exists at the frequency Fo + f1, the spectrum is detected as if it also exists at the frequency Fo-f1.

  FIG. 5 is an example of transmission / reception timing in NMR transmission. The pulse is transmitted at a high frequency fo, and the FID signal is received from the transmission / reception coil 28 at a time indicated by a triangle after transmission. Some spectra of the FID signal have low intensity, and when the nuclide density in the sample is low, the signal is weak.

  For this reason, the S / N of the signal output from the transmission / reception coil 28 is often low. In order to increase the S / N, transmission / reception is repeated and IQ data is subjected to integration processing as shown in FIG. N is increased to improve the resolution of spectrum analysis by Fourier transform (FFT transform). The reception interval is 0.1 to several tens of seconds. However, if the transmission / reception phase difference changes every time each received signal is measured, the signal intensity S does not increase even if integration is performed because the phase is different. Cannot be improved. Furthermore, since the phase change causes the spread of the spectrum width along with the frequency fluctuation, the frequency resolution as the NMR apparatus is lowered.

  Factors that cause fluctuations in the transmission / reception phase difference can be classified into the following five. The first factor is the phase characteristics of the variable amplifier, transmission amplifier, analog and digital filter, the second factor is a measurement delay due to the time from the end of transmission to the start of reception, and the third factor is generated within the transmission pulse period. Phase shift due to the offense effect, the fourth factor is the phase difference between the phase of the detected signal and each spectrum of the received signal, and the fifth factor is the phase change of the A / D sampling clock. Phase changes due to these factors can be classified into three categories: zeroth order (constant with respect to frequency), first order, and higher order.

  Moreover, it is classified into those that change with time and those that change only depending on measurement conditions such as pulse width. A problem in the integration process is the phase change of each spectrum that occurs between the received signals. These factors are 2, 4, and 5 described above, and have zero-order and first-order phase characteristics with respect to frequency.

  In FIG. 5, the output signal from the detection processing unit 18 is subjected to Fourier transform processing (FFT processing) and converted into real part r and imaginary part i component data in the frequency domain. A phase change is detected using the data, and 0th-order and 1st-order phase correction is performed in the frequency domain, and this is converted into I and Q data in the time domain again by inverse Fourier transform (IFFT transform) processing. It is. The spectral distribution of the absorption component r after the inverse Fourier transform (IFFT transform) is shown in FIG. Figures (b) and (c) are enlarged versions of each spectrum. If there is no phase change, the peak waveform is as shown in (b), but if a phase difference occurs, the peak waveform is as shown in (c). Distorted.

  These are shown on the r and i axes as (d) and (e). The peak waveform is circular, and the data before and after the peak waveform are distributed on a straight line having an inclination of 90 degrees with respect to the phase difference. Accordingly, the phase difference can be detected from the r and i data at the maximum point of the peak waveform or the r and i data before and after the peak waveform. The spectrum width is 1 Hz or less, the phase difference θ (f) is detected over the entire observation frequency, and the complex number data S (f) consisting of r (f) and i (f) is on the frequency axis in the reverse direction by the following equation. It can be corrected by rotating with.

S (f) {cosθ (f) −j * sinθ (f)}
Since the change in the phase difference becomes a problem, the phase difference detected in the first received signal is stored, the phase difference is not corrected, the phase difference in the second received signal is detected, and the difference from the first time is detected. Only the phase difference may be corrected and integrated. Similarly, after the third time, only the difference from the previous received signal may be corrected and integrated. As a result, although the initial phase difference remains, it is a component that does not vary with time, and therefore it is sufficient to correct the integrated signal. As described above, according to the present embodiment, since the phase difference that changes for each received signal can be corrected, there is no spread of the spectrum width or S / N reduction due to the phase change of the integrated received signal, and high sensitivity. Measurement is possible.

  Next, another embodiment of the present invention will be described with reference to FIGS. FIG. 7 shows a configuration of NMR transmission. A variable frequency and phase transmitter 37 is provided, and its output signal (reference signal) is added to the FID signal by the adder 39 and digitized by the A / D converter 16. This is a difference from the embodiment of FIG.

  FIG. 8 shows the frequency relationship between the transmission frequency fo, the observation band W, and the reference signal. The frequency of the reference signal is set to be smaller than fo−W / 2 and larger than fo + W / 2. That is, it is made not to overlap with the spectrum to be detected existing in the observation band. Needless to say, since the frequency can be converted by the aerial effect, it is sufficient not to overlap the observation width after conversion. The reference signal is controlled by a signal from the timing control circuit 12 in order to match the initial phase although the frequency is different from the transmission frequency. Since the reference signal generated in this way has a higher S / N than the FID signal, it is possible to measure the phase change with high accuracy and to detect the phase with high accuracy.

  FIG. 9 is a diagram showing transmission / reception and the generation timing of the reference wave. As shown in FIG. 9C, the generation is performed simultaneously with the transmission, and the reference wave is output until the end of reception. In this case, as described above, it is necessary to set the reference signal in a frequency region outside the observation band. On the other hand, as shown in FIG. 9 (d), when a reference signal is output so as to be generated at the same time as transmission and to end at the start of reception, a frequency may be set within the observation band.

  Next, another embodiment of the present invention will be described with reference to FIG. The reference signal is output for the period of FIG. 9 (d), the signal output from the IQ detection processing unit 18 is FFT processed to detect the phase difference, and the phase difference is added to the phase for the NCO 42 instructed from the personal computer. , Which changes the phase of the NCO. When the reference signal has two or more types of frequency components, it is possible to detect 0th-order and first-order phase changes with respect to the frequency.

  In order to correct the primary change with respect to the frequency, in addition to rotating at a different θ (f) for each frequency in the frequency domain as described above, it is possible to correct by changing the frequency of 42 NCO. Become. Accordingly, as shown in FIG. 10, the phase and frequency change detected by the phase change detector 50 are added to the frequency and phase commanded from the personal computer by the adders 52a and 52, and the frequency and phase of the NCO 42 are changed. By doing so, it is possible to correct 0th and 1st order phase changes. Thus, according to the present embodiment, it is not necessary to perform inverse Fourier transform (IFFT) processing for phase correction, so that the amount of calculation processing can be reduced.

  In addition to the NMR apparatus, the present invention can be applied to fields that require stable detection over a long period of phase and frequency.

It is a block diagram of the nuclear magnetic resonance apparatus which is embodiment of this invention. It is the figure explaining conversion of the frequency of the received signal by undersampling which is an embodiment of the present invention. It is a figure which shows the structure of the conventional nuclear magnetic resonance apparatus. It is a figure which shows the structure of the IQ detection process part and phase correction part of embodiment of this invention. It is explanatory drawing of the transmission / reception timing in a nuclear magnetic resonance apparatus, and an integration process. It is explanatory drawing of the spectrum change when a phase change generate | occur | produces in a nuclear magnetic resonance apparatus. It is a figure which shows the structure of the nuclear magnetic resonance apparatus which is other embodiment of this invention. It is explanatory drawing of the spectrum of the reference signal of embodiment of this invention. It is explanatory drawing of the generation timing of the reference signal which is an Example of this invention. It is a block diagram of the IQ detection process part and phase correction part which show other embodiment of this invention.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 10 ... Personal computer, 12 ... Timing control circuit, 14 ... Reception processing circuit, 16 ... A / D converter, 18 ... IQ detection processing part, 20 ... Phase correction part, 22 ... Integration processing part, 24 ... Variable frequency / phase transmission 26 ... reference transmitter 28 ... transmission / reception coil 30 ... frequency synthesizer 32 ... mixer 34 ... variable amplifier 36 ... power amplifier 38 ... BPF 39 ... adder 40 ... detection circuit 42 ... NCO ( Numerical control oscillator), 44 ... multiplier, 46 ... LPF, 48 ... FFT, 50 ... phase change detector, 52 ... IFFT.

Claims (10)

  By transmitting a high-frequency signal to the coil, a high-frequency magnetic field is applied to the sample to be measured placed in the static magnetic field, the nuclear magnetic resonance spectrum radiated from the sample to be measured is received by the coil, and the received signal is converted from analog to digital. In a nuclear magnetic resonance apparatus for inputting to a detector and measuring a spectral distribution, the sampling frequency of the analog-digital converter is lower than the frequency of the nuclear magnetic resonance spectrum of the received signal. apparatus.   In the nuclear magnetic resonance apparatus according to claim 1, when the frequency of the high-frequency signal to be transmitted is f and the half of the sampling frequency fs is fN, when the integer value n of the calculation result of f / fN is an odd number, (N + 1) * fN−f calculated frequency, and when n is an even number, means for generating a signal having a frequency calculated by f−n * fN, and a received signal digitized by the signal is detected. A nuclear magnetic resonance apparatus comprising means.   3. The nuclear magnetic resonance apparatus according to claim 2, wherein the phase of a signal having a frequency calculated from the frequency of the high-frequency signal to be transmitted and the sampling frequency for detecting the received signal is compared with the phase of the high-frequency signal to be transmitted or A nuclear magnetic resonance apparatus comprising means for synchronizing the phase at the end of transmission.   A nuclear magnetic resonance apparatus that transmits a high-frequency signal to a coil, applies a high-frequency magnetic field to a sample to be measured placed in a static magnetic field, receives a nuclear magnetic resonance spectrum emitted from the sample to be measured by the coil, and measures the spectrum distribution The detector for detecting the received signal, means for detecting the phase change of the nuclear magnetic resonance spectrum from the detected signal, means for correcting the phase change amount, and means for integrating the corrected received signal. A nuclear magnetic resonance apparatus.   5. The nuclear magnetic resonance apparatus according to claim 4, wherein means for generating a signal whose phase is synchronized with the phase of a high-frequency signal to be transmitted and at the start or end of transmission, and the signal and the received signal are converted to the same analog-digital converter. A nuclear magnetic resonance apparatus comprising means for inputting.   5. The nuclear magnetic resonance apparatus according to claim 4, wherein means for Fourier-transforming the detected signal, means for detecting a phase change amount from the Fourier-transformed signal, and a signal Fourier-transformed from the phase change amount in the frequency domain. A nuclear magnetic resonance apparatus comprising: means for correcting; and means for converting the corrected signal into a signal in the time domain by inverse Fourier transform.   5. The nuclear magnetic resonance apparatus according to claim 4, wherein means for Fourier-transforming the detected signal, means for detecting a phase change amount from the Fourier-transformed signal, and a signal for detecting a received signal using the phase change amount And a means for changing the phase and frequency of the magnetic resonance apparatus.   5. The nuclear magnetic resonance apparatus according to claim 4, wherein means for generating a signal whose phase is synchronized with the phase of a high-frequency signal to be transmitted and at the start or end of transmission, and the signal and the received signal are converted to the same analog-digital converter. Means for inputting; means for Fourier transforming the detected signal; means for detecting a phase change amount from the Fourier transformed signal; means for correcting a signal obtained by Fourier transforming the phase change amount in the frequency domain; And a means for converting the processed signal into a signal in the time domain by inverse Fourier transform.   5. The nuclear magnetic resonance apparatus according to claim 4, wherein means for generating a signal whose phase is synchronized with the phase of a high-frequency signal to be transmitted and at the start or end of transmission, and the signal and the received signal are converted to the same analog-digital converter. Input means, means for Fourier transforming the detected signal, means for detecting the phase change amount from the Fourier transformed signal, and changing the phase and frequency of the signal for detecting the received signal using the phase change amount A nuclear magnetic resonance apparatus. By transmitting a high-frequency signal to the coil, a high-frequency magnetic field is applied to the sample to be measured placed in the static magnetic field, the nuclear magnetic resonance spectrum radiated from the sample to be measured is received by the coil, and the received signal is converted from analog to digital. A spectral analysis measurement method characterized in that the sampling frequency of the analog-digital converter is lower than the frequency of the nuclear magnetic resonance spectrum of the received signal. .

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