JP2004274922A - Drive device for stepping motor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a sensor-less pole position detecting method with a high resolution of a pole position detection in a multi-phase hybrid stepping motor. <P>SOLUTION: This driver is a drive device for feeding a motor winding current which changes at every application of an external position command. This drive includes a voltage detector which detects a motor applied voltage; a current detector which detects a phase current; an inverse function characteristics computing device of a motor winding constant, and a smoothing circuit which restrains an AC ripple. A condition estimate device is constituted so as to input a detection value which is an output of the voltage detector and a detection value which is an output of the current detector, and outputs a mean value of a difference between the value obtained after passing the phase current detection value through the inverse function characteristics computing device of the motor winding constant and the applied voltage detection value. It is provided with a position estimate function to estimate a motor rotational position from the output. <P>COPYRIGHT: (C)2004,JPO&NCIPI

Description

【数２】

また、モータの電圧方程式を数式３（α相）及び数式４（β相）で表す。
【数３】

【数４】

但し，数式で使用した記号は，ｅα，ｅβはモータの速度起電力，ｉα，ｉβはモータの相電流，θｒｅは電気角で表したモータ回転角度，ωｒｅは電気角で表したモータ軸回転角速度（モータの基本角周波数），Φはモータ磁束，ｖα，ｖβはモータの印加電圧，Ｒα，Ｒβは巻線抵抗，Ｌα，Ｌβは巻線インダクタンス，ｓはラプラス演算子である。なお，添え字のα，βはモータの相区分を示している。
ブラシレスモータの解析において、状態方程式をたてる場合速度起電力の１階微分を０とすることが一般的だが、ステッピングモータは過渡的な変化が繰り返されるため演算誤差が大きくなるという問題がある。
そこで、本発明では速度起電力の２階微分を０とするような状態方程式に基づき状態推定器を設計する。これにより状態推定器の推定精度の向上を図っている。速度起電力の２階微分が０となるモータの状態方程式を数式５（α相）及び数式６（β相）に示す。
【数５】

【数６】

数式５及び数式６を基にモータの速度起電力ｅα及びｅβを推定できれば数式７を用いてモータの磁極位置θｎｆを推定できる。
【数７】

【００１８】
速度起電力の推定は、ステッピングモータを永久磁石形同期電動機として取り扱った場合、相電流検出値をモータ巻線定数の逆関数特性演算器を通して得られる値と印加電圧検出値の差分を、該平滑回路にて平滑した値を出力とする状態推定器によって得られる。従って、状態推定器で得られた速度起電力推定値を数式７に従い演算しモータの磁極位置θｎｆを推定し、位置指令と比較して励磁タイミングを適正な条件に制御することで、脱調の回避や振動抑制が可能な駆動特性を実現できる。
状態推定器を構成する場合、平滑回路の特性決定が問題であるが、ステッピングモータはモータの基本周波数（速度起電力の基本周波数）が高いため、高速に磁極位置を推定する必要がある。このため、本発明では、状態推定器の推定演算を離散値系で取り扱い、且つ、収束を速くするために、有限整定特性とした。
数式６及び数式７を例えばオイラー法にて離散化した場合、平滑回路のゲインは、数式８とすればよい。
【数８】

但し、Ｔはサンプリング周期
Ｌはモータ巻線インダクタンス（Ｌ＝Ｌα＝Ｌβ）
【００１９】
この結果、推定遅れの少ないモータ磁極位置情報が得られ、しかも、連続的に変化する速度起電力を推定しているため、離散値系の演算精度に依存するものの、高分解能の磁極位置推定値が得られる。よって、マイクロステップ駆動の如く、ステッピングモータの基本分解能よりも微小な角度を取り扱う駆動方式においても、位置推定値を用いた閉ループ駆動が可能である。
また、有限時間制定特性の状態推定器は磁極位置推定の時間遅れを小さくできるため、正弦波状の位置変化を推定できるだけでなく変化の急峻なパルス電圧駆動においても推定誤差の小さい位置推定を行うことができるため、分解能の粗い通常のステップ運転にも適用可能である。
また、該位置指令と該状態推定器にて推定したモータ回転位置推定値を用いて、モータ励磁位置を制御するように構成することで、回転速度の上昇により生じる電流の遅れを補正することが可能となり、モータを高速まで回転させることができる。
また、本発明は２相ハイブリッド形ステッピングモータを例に記載しているが、周知の通り、座標変換を用いることで多相機を２相機に変換することが可能であることから、本発明は３相ステッピングモータを始めとする多相ステッピングモータに適用することができる。
【００２０】
【実施例】
図１は本発明の第１の実施例を示すブロック図である。
図１において、磁極位置推定器５０（詳細は後述する）は、ステッピングモータ７０のα相モータ通電電流を検出する第１の電流検出器６１の出力ｉαｆと、β相モータ通電電流を検出する第２の電流検出器６２の出力ｉβｆと、ステッピングモータ７０の第１相（α相）モータ印加電圧Ｖαと、第２相（β相）モータ印加電圧Ｖβを入力とし、磁極位置推定値θｎｆを発生する。位置指令入力端子１０に入力される位置指令θ＊と該磁極位置推定値θｎｆを位置制御器３０に入力する。位置制御器３０の出力と、電流振幅指令入力端子２０に加える電流振幅指令を電流制御器４０に加え、電流制御器４０（詳細は後述する）によりステッピングモータ７０の印加電圧を制御するように構成する。
【００２１】
ここで、磁極位置推定方式について図２に従い詳述する。図２はモータと磁極位置推定器５０の関係を示すブロック図である。但しモータ部は、２相ステッピングモータを例に、モータの一相分（α相）のみを示している。ここで、ステッピングモータは前述の通り永久磁石形同期電動機と類似した特性を有していることから、モータの挙動把握、解析に同期電動機の解析手法が適用できるものとしてブロック図を記述した。
ステッピングモータ７０は、電流制御器４０から与えられるα相印加電圧Ｖαを受け、モータ軸回転角速度ωｒｅを発生する。モータ内部では軸回転角速度ωｒｅに対して起電力定数７５（ｋｅ）に比例した速度度起電力Ｅｅｍｆが生成され加算器７１に帰還される。加算器７１ではα相印加電圧Ｖαと速度度起電力Ｅｅｍｆの差分が計算されモータ巻線定数７２に加えられ、α相巻線に相電流ｉαが流れる。相電流はトルク定数７３（ｋｔ）により軸トルクＴに変換され、更に軸トルクＴはモータ機械要素７４（図２では近似値１／Ｊｓとした）によってモータ軸回転角速度ωｒｅとなる。
【００２２】
以上のモータα相のブロック図において、モータ相電流ｉａはモータ巻線に流れる電流であるから図１に示した第１の電流検出器６１で検出することができる。
磁極位置推定器５０は、α相速度起電力推定部５１にてα相速度起電力推定値Ｅαｆを得る。つまり、第１の電流検出器６１の出力であるα相電流ｉａと、電流制御器４０の出力であるα相印加電圧Ｖαを入力とし、一方の入力であるα相電流ｉａをモータ巻線定数の逆関数特性演算器５４で係数倍した値と、他方の入力であるα相印加電圧Ｖαの差分を加算器５３で生成する。加算器５３の出力は平滑回路５５に入力されα相速度起電力推定値Ｅαｆを得る。図２には詳細を省略して表示しているが、モータの第２相（β相）についても、β相速度起電力推定部５２にてβ相速度起電力推定値Ｅβｆを得る。
更に、推定速度起電力Ｅαｆ及びＥβｆを位置演算器５６によって磁極位置推定値θｎｆに変換し、磁極位置推定値θｎｆを磁極位置推定器５０の出力として位置制御器３０に与える。
【００２３】
位置制御器３０について詳述する。
図３は位置制御器の実施例である。位置制御器３０は、位置指令入力端子１０に加えられる位置指令と、磁極位置推定器５０の出力である磁極位置推定値θｎｆを入力としている。加算器３１は位置指令θ＊と磁極位置推定値θｎｆの差（位置偏差）を求め、該位置偏差を位置補償器３２に与える。位置補償器３２は位置偏差を増幅し、加算器３３で位置補償器３２の出力と磁極位置推定値θｎｆを加えて位置制御信号λを出力する。ステッピングモータは位置指令に追従して回転するが、指令が印加されると過渡的に位置偏差を生じる。図３の構成で、位置偏差εが発生すると位置制御信号λは磁極位置推定値θｎｆに位置偏差を増幅した値が出力される。例えば、位置補償器３２がゲイン１倍の比例要素であれば、位置制御信号λは位置指令そのものとなる。通常、位置補償器３２にＰＩ補償器を用いる。この場合、位置偏差が継続的に発生すると、λは時間とともに増加する。
【００２４】
電流制御器４０について詳述する。
図４は電流制御器４０の実施例である。電流制御器４０は電流振幅指令入力端子２０から与えられる外部電流振幅指令ｉ＊と、位置制御器３０の出力である位置制御信号λと、該第１の電流検出器６１の出力である第１の電流検出値ｉαｆと、該第２の電流検出器６２の出力である第２の電流検出値ｉβｆを入力として、第１のモータ印加電圧Ｖαと、第２のモータ印加電圧Ｖβを出力するものである。
励磁信号発生器４１は位置制御信号λを位置信号としてモータ印加電圧の基本信号である２相交流ｃｏｓλ及びｓｉｎλびを発生する。
該２相交流は乗算器４２及び４３で外部電流振幅指令ｉ＊に比例した振幅に変換された電流指令（ｉ＊ｃｏｓλ及びｉ＊ｓｉｎλ）を生成する。加算器４４及び４５は該電流指令と、２相電流検出値ｉαｆ、ｉβｆとの電流偏差を演算する。該電流偏差は電流補償器４６及び４７で増幅した後、ＰＷＭ（パルス幅変調）変換器４８に加えられる。
ＰＷＭ変換器４８は、第１相の電流補償器４６の出力ＶａｃｒαをＰＷＭ変調しインバータ駆動信号Ｖｇαを生成する。同様にＰＷＭ変換器４８は、第２相の電流補償器４７の出力ＶａｃｒβをＰＷＭ変調しインバータ駆動信号Ｖｇβを生成する。インバータ４９は、第１相のインバータ駆動信号Ｖｇα及び第２相のインバータ駆動信号Ｖｇβ流により第１相（α相）モータ印加電圧Ｖαと、第２相（β相）モータ印加電圧Ｖβを出力する。
【００２５】
即ち、電流制御器４０は、電流指令とモータ電流との差が小さくなるような電流制御を行うとともに、モータの励磁位置は、位置制御信号λにより決定される。つまり、ステッピングモータとしての歩進位置及び速度上昇に伴うモータ励磁位置の補正が位置制御信号λを調整することで可能となっている。
【００２６】
図６は、本発明の第２の実施例である。
図６は、図１の第１の実施例に対して、電流制御器４０は同一構成で、位置制御器と磁極位置推定器の構成が異なる。
【００２７】
磁極位置推定器の構成については、第１の実施例では、図２に示した通り磁極位置推定器５０の入力はモータ端子電圧（Ｖα、Ｖβ）及びモータ相電流（ｉα、ｉβ）としたが、第２の実施例では、図６及び図８に示すように磁極位置推定器５０１の入力は、モータ端子電圧に替えて電流補償器４６の出力Ｖａｃｒα及び電流補償器４７の出力Ｖａｃｒβを用いている。モータ相電流（ｉα、ｉβ）は第１の実施例と同一のものである。電流補償器の出力は、モータ印加電圧と比例関係にあるため、図８でα相速度起電力推定部５１１内部に係数器５７を追加している。なお、β相速度起電力推定部５１２はα相速度起電力推定部５１１と同一構成であり、図８では簡略表現とした。
【００２８】
位置制御器の構成については、第１の実施例では、図３に示した通り位置制御信号λは磁極位置推定値θｎｆに位置偏差を増幅した値としたが、第２の実施例では、図７に示すように、位置制御信号λは磁極位置推定値θｎｆに位置偏差を増幅した値と、磁極位置推定値θｎｆ磁を微分し係数倍した値を加算している。これは、速度の変化に対応して、モータの励磁位置を調整するものである。
【００２９】
【発明の効果】
上記のごとく、ステッピングモータの駆動において、高速で高分解能のモータ磁極位置推定値が得られるため、通常のステッピングモータと同様に特別な検出器を設けることなく、位置検出器付のモータと同等の駆動特性を実現できる。よって、本発明によるステッピングモータ駆動装置は、開ループ制御方式に比べ同一モータを使いながら脱調の恐れが少なく信頼性の高い装置を構築できる。なお、離散値演算を実施する場合、マイクロプロセッサ等の集積回路が使用可能であるため、本発明による駆動装置は、小型、低価格で実現することが可能である。
【図面の簡単な説明】
【図１】本発明に係る第１の実施例を示すブロック図である。
【図２】本発明に係る第１の実施例の機能説明用部分ブロック図である。
【図３】本発明に係る第１の実施例における位置制御器の詳細ブロック図である。
【図４】本発明に係る第１の実施例における電流制御器の詳細ブロック図である。
【図５】従来のステッピングモータの駆動装置を示すブロック図である。
【図６】本発明に係る第２の実施例を示すブロック図である。
【図７】本発明に係る第２の実施例における位置制御器の詳細ブロック図である。
【図８】本発明に係る第２の実施例の機能説明用部分ブロック図である。
【００３１】
【符号の説明】
１０ 位置指令入力端子
２０ 電流振幅指令入力端子
３０、３００ 位置制御器
３１ 加算器
３２ 位置補償器
３３ 加算器
３４ 微分器
３５ 係数器
４０ 電流制御器
４１ 励磁信号発生器
４２、４３ 乗算器
４４、４５ 加算器
４６、４７ 電流補償器
４８ ＰＷＭ変換器
４９ インバータ
５０、５０１ 磁極位置推定器
５１、５１１ 速度起電力推定部
５２、５１２ 速度起電力推定部
５３ 加算器
５４ モータ巻線定数の逆関数特性演算器
５５ 平滑回路
５６ 位置演算器
５７ 係数器
６１、６２ 電流検出器
７０ ステッピングモータ
７１ 加算器
７２ モータ巻線定数
７３ トルク定数
７４ モータ機械要素
７５ 起電力定数
１０１ 位置指令入力端子
１１１ 偏差カウンタ
１２０ 加算器
１３０ 電圧指令制御回路
１４１ 同期運転用ＰＷＭ発生回路
１４２ ＰＷＭ制御回路
１５１、１５２ 切り替えスイッチ
１６０ ゲート信号発生回路
１７０ インバータ
１８０ ステッピングモータ
１９１ ロータ位置検出回路
１９２ 転流パルス発生回路[0001]
[Industrial applications]
The present invention relates to a driving device for a stepping motor for controlling position and speed.
[0002]
[Prior art]
A stepping motor is a control motor that generally performs open-loop position control, and reduces vibration and noise by devising a driving method. In order to avoid the step-out phenomenon, mounting a position detector on a stepping motor and performing closed-loop control is also being studied. A closed-loop drive system using a position detector is effective for avoiding step-out and achieving accurate position control. However, the range of application is limited due to an increase in wiring and cost.
Therefore, a sensorless driving technique for estimating the magnetic pole position, that is, the rotational position of the motor from the intrinsic constant of the motor and observable information during operation is being studied.
Hybrid type stepping motors have characteristics similar to those of permanent magnet type synchronous motors.Synchronous motor analysis methods can be used to grasp and analyze the behavior of the motors, and as a method to realize sensorless driving of hybrid type stepping motors. Application development of sensorless drive technology for synchronous motors is conceivable. However, since the applied voltage at the time of driving of the stepping motor is a square wave pulse waveform or a stepwise pseudo sine waveform, the voltage and current contain many high-order odd harmonics, for example, a resolution of 1.8 degrees / step. It is a great feature that a multi-pole machine has about 10 times as many motor poles as a synchronous motor, such as a phase machine having 100 poles. For this reason, there are technical problems such as a large influence of a time delay of the position estimation calculation and a large noise at the time of the differentiation calculation. Therefore, there are very few application examples of the sensorless drive to the synchronous motor.
[0003]
International Publication No. WO 00/04432 (hereinafter referred to as well-known document 1) is one of the few application examples related to a position sensorless control device for a stepping motor. Hereinafter, the outline of the known document 1 will be described.
FIG. 5 is an example of the known document 1. Known Document 1 targets a three-phase hybrid type stepping motor and forcibly switches the motor energization state according to a position command pulse externally applied to a position command input terminal 101 at zero speed (stop) and low speed rotation. To drive the motor. This is called synchronous operation.
[0004]
When the speed is increased and the speed electromotive force can be detected, the rotor position detection circuit 191 detects a change in polarity of the motor speed electromotive force from the motor terminal voltage, generates a rotor position signal, and generates a gate signal generation circuit 160. And outputs a commutation timing signal to a commutation pulse generation circuit. The commutation pulse generation circuit 192 generates a position detection pulse every time the rotational position changes according to the commutation timing signal generated by the rotor position detection circuit 191, and outputs the pulse to the deviation counter 111. The deviation counter 111 counts the position command pulse and the position detection pulse, respectively, and outputs a position deviation, which is a difference between them. The position deviation passes through a voltage command control circuit 130 and a PWM (pulse width modulation) control circuit 142 and is given to a gate signal generation circuit as a PWM modulation signal for adjusting a motor application voltage. That is, when the speed electromotive force can be detected, an operation for determining the excitation timing based on the speed electromotive force and adjusting the motor applied voltage to correct the positional deviation with respect to the command is performed. This is referred to as position sensorless closed loop operation.
[0005]
Here, the inputs of the voltage command control circuit are a position deviation, a PI compensator using the integrated value of the position deviation, and speed compensation using the frequency component of the position detection pulse detected by the commutation pulse generation circuit, that is, the rotation speed. It is described that the follow-up performance with respect to a position command pulse is improved by using a generator. Further, it is described that the rotor position detection compares the terminal voltage of each phase of the motor with the virtual neutral point voltage and detects and generates the polarity inversion position.
[0006]
The switching between the synchronous operation and the position sensorless closed-loop operation is realized by generating an operation switching signal based on predetermined switching conditions in the operation mode switching circuit 150 and switching the switches 151 and 152. The synchronous operation PWM generation circuit 141 and the bias control circuit 112 provide drive conditions during synchronous operation and appropriate conditions during switching.
[0007]
As described above, the known document 1 detects a terminal voltage of a motor, detects a magnetic pole position based on the speed electromotive force of the motor by comparing with a reference voltage, and, if the magnetic pole position can be detected, a position sensorless closed loop. The excitation of the motor is switched at the magnetic pole position based on the speed electromotive force, referred to as operation, and the motor applied voltage is adjusted based on the position deviation.
[0008]
However, in the case of the prior art according to the known document 1, since the polarity inversion timing of the motor speed electromotive force is detected, for example, when a minute change in the excitation position is required in the case of performing the micro-step driving, the detection is performed. There is a problem that the resolution is low.
In addition, since the stepping motor, which is a multi-pole machine, has a high fundamental frequency of the motor, the phase delay of the current due to the motor winding inductance increases as the speed increases. Excitation switching is performed according to the reversal position, and the applied voltage is adjusted according to the change in position deviation. Therefore, there is a problem that high-speed rotation after the region where the applied voltage is saturated is difficult.
Further, the prior art according to the known document 1 presupposes application to a three-phase stepping motor, and it is difficult to cope with stepping motors having different numbers of phases such as two-phase and five-phase.
[0009]
[Problems to be solved by the invention]
As described above, in the related art, when the magnetic pole position is detected from various motor constants while the stepping motor is driven, there is a problem that the resolution of the detected position is low. In addition, there is a problem that the high-speed rotation is difficult because the magnetic pole position and the excitation switching condition are fixed. Further, there is a problem that it is difficult to cope with stepping motors having different numbers of phases such as two phases and five phases.
[0010]
An object of the present invention is to solve the above problem and realize a sensorless magnetic pole position detection method in a polyphase hybrid type stepping motor with high magnetic pole position detection resolution.
It is another object of the present invention to realize a motor drive system capable of high-speed rotation in a multi-phase hybrid type stepping motor.
It is another object of the present invention to realize a sensorless magnetic pole position detection method and a motor driving method that are hardly affected by the number of motor phases in a hybrid type stepping motor.
[0011]
[Means to solve the problem]
In order to solve the above problem, in the present invention, a stepping motor driving device includes a voltage detector for detecting a motor applied voltage, a current detector for detecting a phase current, and an inverse function characteristic calculator for a motor winding constant. A smoothing circuit that suppresses AC ripple, receives an applied voltage detection value that is an output of a voltage detector, and a phase current detection value that is an output of a current detector, and uses the phase current detection value as a motor winding constant. A state estimator configured to output a value obtained by smoothing the difference between the value obtained through the inverse function characteristic calculator and the applied voltage detection value by the smoothing circuit is configured, and the motor rotational position is estimated from the output of the state estimator. The position estimation function configured as described above is provided.
[0012]
Further, the motor excitation position is controlled by the position command and the motor rotation position detection value using the position command and the motor rotation position estimation value estimated by the state estimator as a motor rotation position detection value. .
[0013]
Further, the state estimator is handled by a discrete value operation, and the state estimator is linearized by setting the second derivative of the speed electromotive force of the motor to 0.
[0014]
Further, the state estimator is handled by a discrete value operation, and the state estimator is configured to linearize the second-order derivative of the speed electromotive force of the motor as 0, so that the pole arrangement has a finite settling characteristic.
[0015]
In addition, a voltage detector that detects a control voltage proportional to the applied voltage with respect to the motor applied voltage, and a coefficient unit that amplifies the detection value of the voltage detector are provided, and are replaced with the motor applied voltage detector of the state estimator. It is constituted so that.
[0016]
[Action]
With the above configuration, it is possible to realize a sensorless magnetic pole position estimation method with high resolution, which is an object of the present invention, and to realize a motor driving method capable of high-speed rotation in a multi-phase hybrid type stepping motor. Can be. Further, in the hybrid stepping motor, it is possible to realize a sensorless magnetic pole position estimation method and a motor driving method that are hardly affected by the number of motor phases. The grounds are described below.
[0017]
The speed electromotive force of the motor can be expressed by Expression 1 (α phase) and Expression 2 (β phase).
(Equation 1)

(Equation 2)

Further, the voltage equation of the motor is expressed by Expression 3 (α phase) and Expression 4 (β phase).
[Equation 3]

(Equation 4)

Here, the symbols used in the formulas are eα, eβ: motor speed electromotive force, iα, iβ: motor phase current, θre: motor rotation angle in electrical angle, ωre: motor shaft rotation angular speed in electrical angle (Basic angular frequency of the motor), Φ is a motor magnetic flux, vα, vβ are applied voltages of the motor, Rα, Rβ are winding resistances, Lα, Lβ are winding inductances, and s is a Laplace operator. The subscripts α and β indicate the motor phase division.
In the analysis of the brushless motor, it is general to set the first derivative of the speed electromotive force to 0 when a state equation is established. However, a stepping motor has a problem that a calculation error increases because a transient change is repeated.
Therefore, in the present invention, a state estimator is designed based on a state equation that makes the second derivative of the speed electromotive force zero. This improves the estimation accuracy of the state estimator. Equations 5 (α phase) and 6 (β phase) of the state equation of the motor in which the second derivative of the speed electromotive force is 0 are shown.
(Equation 5)

(Equation 6)

If the speed electromotive forces eα and eβ of the motor can be estimated based on Expressions 5 and 6, the magnetic pole position θnf of the motor can be estimated using Expression 7.
(Equation 7)

[0018]
When the stepping motor is treated as a permanent magnet type synchronous motor, the speed electromotive force is estimated by calculating the difference between the value obtained from the phase current detection value through the inverse function characteristic calculator of the motor winding constant and the applied voltage detection value. It is obtained by a state estimator that outputs a value smoothed by the circuit. Therefore, the estimated value of the speed electromotive force obtained by the state estimator is calculated in accordance with Expression 7 to estimate the magnetic pole position θnf of the motor, and the excitation timing is controlled to an appropriate condition by comparing with the position command, thereby causing the step-out. Driving characteristics capable of avoiding and suppressing vibration can be realized.
When configuring a state estimator, determining the characteristics of the smoothing circuit is a problem. However, since the fundamental frequency of the stepping motor (the fundamental frequency of the speed electromotive force) is high, it is necessary to quickly estimate the magnetic pole position. Therefore, in the present invention, the estimation operation of the state estimator is handled in a discrete value system, and the finite settling characteristic is used in order to speed up convergence.
When Expressions 6 and 7 are discretized by, for example, Euler's method, the gain of the smoothing circuit may be Expression 8.
(Equation 8)

Here, T is the sampling period L is the motor winding inductance (L = Lα = Lβ)
[0019]
As a result, the motor magnetic pole position information with a small estimation delay is obtained, and since the continuously changing speed electromotive force is estimated, the magnetic pole position estimated value with high resolution depends on the calculation accuracy of the discrete value system. Is obtained. Therefore, even in a driving method that handles an angle smaller than the basic resolution of the stepping motor, such as microstep driving, closed-loop driving using the position estimation value is possible.
In addition, the state estimator with the finite time establishment characteristic can reduce the time delay of the magnetic pole position estimation, so it can not only estimate the sinusoidal position change, but also perform the position estimation with a small estimation error even in the pulse voltage driving with the steep change. Therefore, the present invention can be applied to a normal step operation having a coarse resolution.
Further, by configuring the motor excitation position using the position command and the motor rotation position estimation value estimated by the state estimator, it is possible to correct a current delay caused by an increase in rotation speed. It is possible to rotate the motor to a high speed.
Although the present invention describes a two-phase hybrid type stepping motor as an example, as is well known, it is possible to convert a multi-phase machine to a two-phase machine by using coordinate transformation. The present invention can be applied to a multi-phase stepping motor including a phase stepping motor.
[0020]
【Example】
FIG. 1 is a block diagram showing a first embodiment of the present invention.
In FIG. 1, a magnetic pole position estimator 50 (to be described in detail later) includes an output iαf of a first current detector 61 for detecting an α-phase motor energizing current of a stepping motor 70 and a second i-αf detecting a β-phase motor energizing current. The output iβf of the second current detector 62, the first-phase (α-phase) motor applied voltage Vα, and the second-phase (β-phase) motor applied voltage Vβ of the stepping motor 70 are input to generate the magnetic pole position estimated value θnf. I do. The position command θ * input to the position command input terminal 10 and the magnetic pole position estimated value θnf are input to the position controller 30. The output of the position controller 30 and the current amplitude command applied to the current amplitude command input terminal 20 are applied to the current controller 40, and the voltage applied to the stepping motor 70 is controlled by the current controller 40 (details will be described later). I do.
[0021]
Here, the magnetic pole position estimation method will be described in detail with reference to FIG. FIG. 2 is a block diagram showing the relationship between the motor and the magnetic pole position estimator 50. However, the motor section shows only one phase (α phase) of the motor, taking a two-phase stepping motor as an example. Here, since the stepping motor has characteristics similar to those of the permanent magnet type synchronous motor as described above, the block diagram is described assuming that the analysis method of the synchronous motor can be applied to grasp and analyze the behavior of the motor.
The stepping motor 70 receives the α-phase applied voltage Vα given from the current controller 40 and generates a motor shaft rotation angular velocity ωre. Inside the motor, a speed electromotive force Eemf proportional to the electromotive force constant 75 (ke) with respect to the shaft rotation angular speed ωre is generated and fed back to the adder 71. In the adder 71, the difference between the α-phase applied voltage Vα and the speed electromotive force Eemf is calculated and added to the motor winding constant 72, and the phase current iα flows through the α-phase winding. The phase current is converted into a shaft torque T by a torque constant 73 (kt), and the shaft torque T becomes a motor shaft rotation angular velocity ωre by a motor mechanical element 74 (approximate value 1 / Js in FIG. 2).
[0022]
In the above-described block diagram of the motor α phase, the motor phase current ia is a current flowing through the motor winding, and thus can be detected by the first current detector 61 shown in FIG.
The magnetic pole position estimator 50 obtains the α-phase speed electromotive force estimated value Eαf by the α-phase speed electromotive force estimation unit 51. That is, the α-phase current ia output from the first current detector 61 and the α-phase applied voltage Vα output from the current controller 40 are input, and one of the α-phase currents ia is input to the motor winding constant. The difference between the value obtained by multiplying the coefficient by the inverse function characteristic calculator 54 and the α-phase applied voltage Vα which is the other input is generated by the adder 53. The output of the adder 53 is input to the smoothing circuit 55 to obtain the estimated α-phase speed electromotive force Eαf. Although not shown in detail in FIG. 2, the β-phase speed electromotive force estimation unit 52 also obtains the β-phase speed electromotive force estimated value Eβf for the second phase (β phase) of the motor.
Further, the estimated speed electromotive forces Eαf and Eβf are converted into a magnetic pole position estimated value θnf by the position calculator 56, and the magnetic pole position estimated value θnf is given to the position controller 30 as an output of the magnetic pole position estimator 50.
[0023]
The position controller 30 will be described in detail.
FIG. 3 shows an embodiment of the position controller. The position controller 30 receives a position command applied to the position command input terminal 10 and a magnetic pole position estimated value θnf output from the magnetic pole position estimator 50 as inputs. The adder 31 obtains the difference (position deviation) between the position command θ * and the magnetic pole position estimated value θnf, and gives the position deviation to the position compensator 32. The position compensator 32 amplifies the position deviation, and the adder 33 adds the output of the position compensator 32 and the magnetic pole position estimated value θnf to output a position control signal λ. The stepping motor rotates following the position command, but when the command is applied, a position deviation occurs transiently. In the configuration of FIG. 3, when the position deviation ε occurs, the position control signal λ outputs a value obtained by amplifying the position deviation to the magnetic pole position estimated value θnf. For example, if the position compensator 32 is a proportional element with a gain of 1, the position control signal λ is the position command itself. Usually, a PI compensator is used as the position compensator 32. In this case, if the position deviation continuously occurs, λ increases with time.
[0024]
The current controller 40 will be described in detail.
FIG. 4 shows an embodiment of the current controller 40. The current controller 40 receives an external current amplitude command i * given from the current amplitude command input terminal 20, a position control signal λ output from the position controller 30, and a first current detector 61 output from the first current detector 61. And outputs a first motor applied voltage Vα and a second motor applied voltage Vβ by inputting the current detection value iαf of the second current detection value iαf and the second current detection value iβf output from the second current detector 62. It is.
The excitation signal generator 41 uses the position control signal λ as a position signal to generate two-phase AC cos λ and sin λ, which are basic signals of the motor applied voltage.
The two-phase alternating current generates current commands (i * cosλ and i * sinλ) which have been converted by the multipliers 42 and 43 into amplitudes proportional to the external current amplitude command i *. The adders 44 and 45 calculate a current deviation between the current command and the two-phase current detection values iαf and iβf. The current deviation is amplified by current compensators 46 and 47 and then applied to a PWM (pulse width modulation) converter 48.
The PWM converter 48 PWM-modulates the output Vacrα of the first phase current compensator 46 to generate an inverter drive signal Vgα. Similarly, the PWM converter 48 PWM-modulates the output Vacrβ of the second-phase current compensator 47 to generate an inverter drive signal Vgβ. The inverter 49 outputs a first-phase (α-phase) motor applied voltage Vα and a second-phase (β-phase) motor applied voltage Vβ by the first-phase inverter drive signal Vgα and the second-phase inverter drive signal Vgβ. .
[0025]
That is, the current controller 40 controls the current so as to reduce the difference between the current command and the motor current, and the excitation position of the motor is determined by the position control signal λ. That is, the stepping motor as a stepping motor and the motor excitation position accompanying the speed increase can be corrected by adjusting the position control signal λ.
[0026]
FIG. 6 shows a second embodiment of the present invention.
FIG. 6 is different from the first embodiment of FIG. 1 in that the current controller 40 has the same configuration, and the configurations of the position controller and the magnetic pole position estimator are different.
[0027]
Regarding the configuration of the magnetic pole position estimator, in the first embodiment, the inputs of the magnetic pole position estimator 50 are the motor terminal voltages (Vα, Vβ) and the motor phase currents (iα, iβ) as shown in FIG. In the second embodiment, as shown in FIGS. 6 and 8, the input of the magnetic pole position estimator 501 uses the output Vacrα of the current compensator 46 and the output Vacrβ of the current compensator 47 instead of the motor terminal voltage. I have. The motor phase currents (iα, iβ) are the same as in the first embodiment. Since the output of the current compensator is proportional to the motor applied voltage, a coefficient unit 57 is added inside the α-phase speed electromotive force estimation unit 511 in FIG. Note that the β-phase speed electromotive force estimating unit 512 has the same configuration as the α-phase speed electromotive force estimating unit 511, and is simplified in FIG.
[0028]
Regarding the configuration of the position controller, in the first embodiment, as shown in FIG. 3, the position control signal λ is a value obtained by amplifying the position deviation to the magnetic pole position estimated value θnf, but in the second embodiment, FIG. As shown in FIG. 7, the position control signal λ is obtained by adding a value obtained by amplifying the position deviation to the magnetic pole position estimated value θnf and a value obtained by differentiating the magnetic pole position estimated value θnf and multiplying the coefficient by a coefficient. This adjusts the excitation position of the motor according to the change in speed.
[0029]
【The invention's effect】
As described above, in driving the stepping motor, a high-speed and high-resolution motor magnetic pole position estimated value can be obtained, so that a special detector is not provided similarly to a normal stepping motor, and the same as a motor with a position detector is provided. Driving characteristics can be realized. Therefore, the stepping motor driving device according to the present invention can construct a highly reliable device that uses the same motor and is less likely to lose synchronism as compared with the open loop control system. When a discrete value operation is performed, an integrated circuit such as a microprocessor can be used. Therefore, the driving device according to the present invention can be realized at a small size and at a low price.
[Brief description of the drawings]
FIG. 1 is a block diagram showing a first embodiment according to the present invention.
FIG. 2 is a partial block diagram for explaining functions of the first embodiment according to the present invention.
FIG. 3 is a detailed block diagram of a position controller according to the first embodiment of the present invention.
FIG. 4 is a detailed block diagram of a current controller according to the first embodiment of the present invention.
FIG. 5 is a block diagram showing a conventional driving device for a stepping motor.
FIG. 6 is a block diagram showing a second embodiment according to the present invention.
FIG. 7 is a detailed block diagram of a position controller in a second embodiment according to the present invention.
FIG. 8 is a partial block diagram for explaining functions of a second embodiment according to the present invention.
[0031]
[Explanation of symbols]
10 Position Command Input Terminal 20 Current Amplitude Command Input Terminal 30, 300 Position Controller 31 Adder 32 Position Compensator 33 Adder 34 Differentiator 35 Coefficient Unit 40 Current Controller 41 Excitation Signal Generators 42, 43 Multipliers 44, 45 Adders 46, 47 Current compensator 48 PWM converter 49 Inverter 50, 501 Magnetic pole position estimator 51, 511 Speed electromotive force estimator 52, 512 Speed electromotive force estimator 53 Adder 54 Inverse function characteristic calculation of motor winding constant Unit 55 smoothing circuit 56 position calculator 57 coefficient units 61 and 62 current detector 70 stepping motor 71 adder 72 motor winding constant 73 torque constant 74 motor machine element 75 electromotive force constant 101 position command input terminal 111 deviation counter 120 adder 130 Voltage command control circuit 141 Synchronous operation PWM generation circuit 142 PWM control circuit 15 , 152 selector switch 160 gate signal generating circuit 170 inverter 180 stepper motor 191 rotor position detection circuit 192 commutation pulse generation circuit

Claims (8)

A voltage detector for detecting a motor applied voltage, a current detector for detecting a phase current, an inverse function characteristic calculator for a motor winding constant, and a smoothing circuit for suppressing an AC ripple are provided. A certain applied voltage detection value and a phase current detection value which is an output of the current detector are input, and the difference between the phase current detection value obtained through an inverse function characteristic calculator of the motor winding constant and the applied voltage detection value is calculated as A stepping motor control device having a position estimating function configured to constitute a state estimator that outputs a value smoothed by the smoothing circuit, and to estimate a motor rotational position from an output of the state estimator. A stepping motor control device having an input terminal for a position command and performing position control according to a position command given from the outside, wherein the position command and a motor rotation position estimation value estimated by the state estimator are used as a motor rotation position detection value. 2. The stepping motor drive device according to claim 1, wherein the motor excitation position is controlled by the position command and the motor rotation position detection value. 3. The stepping motor driving device according to claim 1, wherein the state estimator is handled by a discrete value operation, and the state estimator is linearized by setting a second derivative of a speed electromotive force of the motor to 0. Item 1 and item 2 wherein the state estimator is treated by a discrete value operation, and the state estimator is configured such that the second derivative of the speed electromotive force of the motor is linearized as 0 and the pole arrangement has a finite settling characteristic. Stepping motor drive. A voltage detector for detecting a control voltage proportional to a motor applied voltage, a current detector for detecting a phase current, an inverse function characteristic calculator for a motor winding constant, and a smoothing circuit for suppressing an AC ripple; Inputs the value obtained by multiplying the output of the detector by a coefficient and the phase current detection value that is the output of the current detector, and calculates the phase current detection value through the inverse function characteristic calculator of the motor winding constant and the applied voltage detection value. A stepping motor control device having a position estimating function configured to constitute a state estimator that outputs a value obtained by smoothing the difference of the above-mentioned difference by the smoothing circuit, and to estimate a motor rotational position from an output of the state estimator. A stepping motor control device having an input terminal for a position command and performing position control according to a position command given from the outside, wherein the position command and a motor rotation position estimation value estimated by the state estimator are used as a motor rotation position detection value. 6. The stepping motor driving device according to claim 5, wherein the motor excitation position is controlled by the position command and the motor rotation position detection value. 3. The stepping motor driving device according to claim 1, wherein the state estimator is handled by a discrete value operation, and the state estimator is linearized by setting a second derivative of a speed electromotive force of the motor to 0. Item 5 and Item 6 in which the state estimator is treated by a discrete value operation, and the state estimator is configured such that the second derivative of the speed electromotive force of the motor is linearized as 0 and the pole arrangement has a finite settling characteristic. Stepping motor drive.

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