JP2003319685A - Motor-driven power steering controlier and motor current operating method therefor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a motor-driven power steering control device capable of canceling an offset current generated by the switching of a driving method of a booster circuit for boosting a DC power voltage, and preventing the motor from being malfunctioned in switching the driving method for the booster circuit, and to provide a motor current operating method therefor. <P>SOLUTION: A CPU saves an A/D value equivalent to the current detected by S110 in a RAM buffer. S120 determines whether the booster circuit is synchronously rectified. In the case of synchronous rectification, S130 performs the arithmetic operation of a motor current with a current calibration formula corresponding to the synchronous rectification. I=G0×A/D value +p0 G0 is a constant in the case of the synchronous rectification, and p0 is an offset value in the case of the synchronous rectification. In the case of non-synchronous rectification, S140 performs the arithmetical operation of the motor current with the following current calibration formula corresponding to the non- synchronous rectification. I=G1×A/D value+p1 G1 is a constant for the non- synchronous rectification, and p1 is an offset value for the non-synchronous rectification. <P>COPYRIGHT: (C)2004,JPO

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an electric power steering control device for applying an assisting force by a motor to a steering system of an automobile or a vehicle, and more particularly, it can adjust a current supplied from an on-vehicle battery to the motor. The present invention relates to an electric power steering control device having a booster circuit and a method for calculating a motor current of the electric power steering control device.

[0002]

2. Description of the Related Art Conventionally, the rotational force of a motor is used to
An electric power steering device that assists the operation of a steering wheel is used.

In such a control device for an electric power steering system, when a driver rotates a steering wheel to perform steering, an assist command value corresponding to a steering torque by the assist control is calculated, and the assist command value is calculated. A steering assist force based on the steering force is applied from the motor to the steering mechanism.

By the way, the control device of the electric power steering device (electric power steering control device) as described above is a system which requires a large current in order to obtain a large torque.

Conventionally, an on-vehicle battery (DC12V) is directly applied, a motor of DC12V specification is used, and a large current is supplied to the motor.
Increasing the size of the motor and increasing the capacity of the wiring used (thick line) cannot be avoided.

Further, if the power consumed by the control device is large, the current is large, so that the control device generates a large amount of heat (loss), and even if the control device is mounted in the engine room, In some cases, it cannot be mounted because it is restricted by temperature conditions.

In order to solve this problem, the present applicant has proposed an electric power steering control device capable of adjusting the current supplied from the vehicle battery. In this electric power steering control device, a booster circuit 100 and a central processing unit (CPU 21) as shown in FIG. 4 are provided in a circuit that supplies a current to a motor. Note that FIG.
FIG. 4 is a circuit diagram for explaining an embodiment of the present invention.
Since the hardware configuration is the same as the configuration proposed by the present applicant, for convenience of description, reference numerals in the embodiments will be used for description.

The booster circuit 100 is provided between the application point P1 of the battery voltage VPIG (DC12V) from the on-vehicle battery and the application point P2 of the voltage to the motor. The booster circuit 100 includes capacitors C1 and C2, a coil L, a first transistor Q1 for switching, and a second transistor Q2.

The CPU 21 calculates the duty ratio of the first transistor Q1 of the booster circuit 100 by PWM calculation for boosting. Then, the CPU 21 uses the duty ratio to drive the duty ratio drive signal (PWM
Drive signal) and the duty ratio drive signal causes the first transistor Q to operate regardless of the load state of the motor.
The first and second transistors Q2 are alternately turned on / off, that is, controlled by the synchronous rectification method (see FIG. 6A).

As a result, energy accumulation and discharge are repeated in the coil L, and a high voltage at the time of discharge appears on the capacitor C2 side of the second transistor Q2. When the first transistor Q1 is turned on, a current flows through the coil L, and when the first transistor Q1 is turned off, the current flowing through the coil L is cut off.

When the current flowing through the coil L is cut off, a high voltage is generated on the capacitor C2 side of the second transistor Q2 so as to prevent the change of the magnetic flux due to the cutoff of the current. By repeating this, a high voltage is repeatedly generated on the side of the capacitor C2 of the second transistor Q2, smoothed (charged) by the capacitor C2, and generated as the output voltage VBPIG at the voltage application point P2.

By the booster circuit 100, the current supplied to the motor can be reduced and heat generation (loss) can be suppressed. on the other hand,
A shunt resistor is provided in the circuit for supplying a current to the motor so that the motor current flows, and after converting the potential difference between both ends of the shunt resistor into an A / D value, using a current calibration formula,
The motor current is calculated and used as the current detection value. The current calibration formula is obtained in advance by driving the booster circuit 100 by synchronous rectification by a test or the like.
The current calibration formula is stored in the ROM or the like and is read and used when the motor current is calculated. The detected current value is used in a feedback system for motor control of the control device.

In the proposed construction, the first transistor Q1 and the second transistor Q2 are driven by the synchronous rectification method in case of high load according to the load state of the motor during power running, and in case of low load, First transistor Q
An asynchronous rectification method is conceivable in which only 1 is PWM-driven and the second transistor Q2 is all off.

This is because if the motor is constantly under low load condition and the synchronous rectification is always performed, one of the transistors is constantly turned on, so that current always flows through the booster circuit 100 to generate heat and the efficiency is remarkably lowered. This is to prevent this.

For example, when the motor rotation speed is high, synchronous rectification is performed as a high load, and when the motor rotation speed is low (including a case where the motor rotation speed is 0), the load is low. Use asynchronous rectification.

[0016]

However, when the driving method of the booster circuit 100 is changed to a method other than the synchronous rectification when the motor has a low load, the actual motor current flowing through the shunt resistor and the shunt resistor are changed. A corresponding to the potential difference between both ends of
Based on the / D value, an offset occurs in the current detection value calculated by using the current calibration formula previously obtained by synchronous rectification.

As a result, there is a problem that the control device malfunctions and the motor operates due to the current (offset current) caused by the offset at the moment when the driving method of the booster circuit 100 is switched.

It is an object of the present invention to cancel an offset current generated by switching the drive system of a booster circuit for boosting a DC power supply voltage, and prevent an erroneous operation of a motor when switching the drive system of the booster circuit. An object of the present invention is to provide a motor current calculation method for a device and an electric power steering control device.

[0019]

In order to solve the above problems, the invention according to claim 1 is a boosting means for boosting the voltage of a DC power source to supply electric power to the motor driving means of an electric motor. A booster having a coil connected to a DC power source, a first switching element and a second switching element connected together to an output terminal of the coil, and a capacitor connected to an output terminal of the second switching element Means and
Depending on the load state of the electric motor, a step-up control unit that changes the drive method of both the switching elements to perform step-up control, and an electric current of the electric motor that flows through the electric motor drive unit are set to an A / D value corresponding to a potential difference between both ends of a shunt resistance. In the electric power steering control device provided with a motor current calculation means for calculating with a current calibration formula, the motor current calculation means is a current calibration formula according to a load state of the motor, and a potential difference between both ends of the shunt resistor is calculated. The gist of an electric power steering control device is characterized in that a motor current is calculated based on a considerable A / D value.

According to a second aspect of the present invention, in the first aspect, there is provided storage means for storing a current calibration formula corresponding to a load state of the electric motor, and the electric motor current calculation means is a current calibration formula stored in the storage means. Among these, the current calibration formula corresponding to the load state of the electric motor is read to calculate the electric motor current.

The invention of claim 3 is the invention of claim 1 or claim 2.
In the above, there is provided load state determination means for determining the load state of the electric motor, and the electric motor current calculation means selects a current calibration formula based on the determination result of the load state determination means.

The invention of claim 4 is the first to third aspects of the invention.
In any one of the above items, the boost control means is characterized in that the switching element synchronously rectifies both switching elements when the electric motor has a high load and asynchronous rectification when the electric motor has a low load.

According to a fifth aspect of the present invention, there is provided boosting means for boosting the voltage of the DC power source to supply electric power to the motor driving means of the electric motor, wherein the coil connected to the DC power source and the output terminal of the coil are connected to the coil. Boosting means including a first switching element and a second switching element, which are connected together, and a capacitor connected to the output terminal of the second switching element,
Booster control means for performing boosting control by changing the driving method of both of the switching elements according to the load state of the electric motor, and the electric motor current flowing through the electric motor drive means is A / A corresponding to the potential difference between both ends of the shunt resistor. In a motor current calculation method of an electric power steering controller that calculates a current value based on a D value, an A / D value corresponding to a potential difference between both ends of the shunt resistance is calculated by a current calibration expression according to a load state of the motor. Based on the above, the present invention is directed to a motor current calculation method for an electric power steering control device characterized by calculating a motor current.

[0024]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS (First Embodiment) An embodiment of an electric power steering controller embodying the present invention will be described below with reference to FIGS.

FIG. 1 shows an outline of an electric power steering controller. A torsion bar 3 is provided on a steering shaft 2 connected to a steering wheel 1 (handle). A torque sensor 4 is attached to the torsion bar 3. Then, when the steering shaft 2 rotates and a force is applied to the torsion bar 3, the torsion bar 3 is twisted according to the applied force,
The twist, that is, the steering torque τ applied to the steering wheel 1 is detected by the torque sensor 4.

The torque sensor 4 constitutes steering torque detecting means. In addition, a reduction gear 5 is attached to the steering shaft 2.
Is stuck. A gear 7 attached to a rotating shaft of an electric motor (hereinafter referred to as a motor 6) as an electric motor is meshed with the speed reducer 5. The motor 6 is a brushless motor composed of a three-phase synchronous permanent magnet motor.

A rotation angle sensor 30 composed of a rotary encoder for detecting a rotation angle of the motor 6 is attached to the motor 6 (see FIG. 2). The rotation angle sensor 30 is π depending on the rotation of the rotor of the motor 6.
It outputs a two-phase pulse train signal having a phase difference of / 2 and a zero-phase pulse train signal indicating the reference rotation position.

Further, a pinion shaft 8 is fixed to the speed reducer 5. A pinion 9 is fixed to the tip of the pinion shaft 8, and the pinion 9 meshes with the rack 10. Tie rods 12 are fixedly provided at both ends of the rack 10, and a knuckle 13 is rotatably connected to a tip portion of the tie rod 12. A front wheel 14 as a tire is fixed to the knuckle 13. Further, one end of the knuckle 13 is rotatably connected to the cross member 15.

Therefore, when the motor 6 rotates, the number of rotations is reduced by the speed reducer 5 and transmitted to the pinion shaft 8 and then to the rack 10 via the pinion and rack mechanism 11. The rack 10 can change the traveling direction of the vehicle by changing the direction of the front wheels 14 provided on the knuckle 13 via the tie rods 12.

A vehicle speed sensor 16 is provided on the front wheel 14. Next, the electrical configuration of this electric power steering control device (hereinafter referred to as control device 20) will be described.

The torque sensor 4 outputs a voltage according to the steering torque τ of the steering wheel 1. The vehicle speed sensor 16 outputs the vehicle speed at that time as a pulse signal having a cycle corresponding to the rotation speed of the front wheels 14.

The control unit 20 includes a central processing unit (CPU2).
1), read only memory (ROM 22) and read and write only memory (RAM) for temporarily storing data
23). This ROM 22 has a CPU 21
A control program for performing the arithmetic processing by is stored. The RAM 23 temporarily stores the calculation processing result and the like when the CPU 21 performs the calculation processing.

The ROM 22 stores a basic assist map (not shown). The basic assist map is for obtaining the basic assist current corresponding to the steering torque τ (turning torque) and corresponding to the vehicle speed.
The basic assist current for is stored.

The function of the control device 20 to drive and control the three-phase synchronous permanent magnet motor is a known structure, and will be briefly described. FIG. 3 is a control block diagram showing functions executed by a program inside the CPU 21. Each part illustrated in the control block diagram, that is, each part indicated by reference numerals 51 to 65, 73, and 76 does not indicate independent hardware but indicates a function executed by the CPU 21.

The CPU 21 is provided with a basic assisting force calculating section 51, a returning force calculating section 52 and an adding section 53 for calculating the command torque τ *. The basic assist force calculation unit 51
The steering torque τ from the torque sensor 4 and the vehicle speed sensor 16
The vehicle speed V detected by is input, and an assist torque that increases as the steering torque τ increases and decreases as the vehicle speed V increases is calculated.

The return force calculation unit 52 inputs the vehicle speed V as well as the electrical angle θ (corresponding to the rotation angle) of the rotor of the motor 6 and the angular velocity ω, and based on these input values, the steering shaft 2 is moved to the basic position. Return force and steering shaft 2
The return torque corresponding to the resistance to rotation of is calculated. The addition unit 53 calculates the command torque τ * by adding the assist torque and the return torque, and outputs the command torque τ * to the command current setting unit 54.

The command current setting unit 54, based on the command torque τ *, has a two-phase d-axis command current Id * and a q-axis command current Iq *.
To calculate. Both command currents respectively correspond to the d axis in the same direction as the direction of the magnetic flux of the permanent magnet and the q axis orthogonal thereto in the rotating coordinate system synchronized with the rotating magnetic flux generated by the permanent magnet on the rotor of the motor 6.

The d-axis command current Id * and the q-axis command current Iq * are supplied to the subtractors 55 and 56. The subtracters 55 and 56 are d
Axis command current Id *, q axis command current Iq *, and d axis detection current I
A difference value ΔId between the d and q axis detection currents Iq,
ΔIq is calculated, and the result is supplied to PI control units (proportional and integral control units) 57 and 58.

The PI control units 57 and 58 are provided with a difference value ΔId,
Based on ΔIq, the d-axis command voltage Vd * and the q-axis command voltage Vq * are calculated so that the d-axis detection current Id and the q-axis detection current Iq follow the d-axis command current Id * and the q-axis command current Iq *, respectively. .

D-axis command voltage Vd * and q-axis command voltage Vq *
Is a non-interference control correction value calculation unit 63 and subtractors 59 and 60.
Thus, the d-axis correction command voltage Vd ** and the q-axis correction command voltage Vq ** are corrected and supplied to the two-phase / 3-phase coordinate conversion unit 61.

The non-interference control correction value calculation unit 63 calculates the d-axis command voltage Vd * and the q-axis command voltage Vq * based on the d-axis detection current Id and the q-axis detection current Iq and the angular velocity ω of the rotor of the motor 6. Non-interference control correction value for ω ・ La ・ Iq, −ω ・
Calculate (φa + La · Id). Note that the inductance L
a and the magnetic flux φa are predetermined constants.

Subtractors 59 and 60 subtract the non-interference control correction values from the d-axis command voltage Vd * and the q-axis command voltage Vq *, respectively, to obtain d-axis correction command voltages Vd ** and q.
The axis correction command voltage Vq ** is calculated and output to the 2-phase / 3-phase coordinate conversion unit 61. The two-phase / three-phase coordinate conversion unit 61 converts the d-axis correction command voltage Vd ** and the q-axis correction command voltage Vq ** into three-phase command voltages Vu *, Vv *, Vw *, and the same conversion is performed. The phase command voltages Vu *, Vv *, Vw * are output to the PWM control unit 62.

The PWM control section 62 uses the three-phase command voltage V
PWM control signals UU, VU, WU corresponding to u *, Vv *, Vw *
(Including a PWM wave signal and a signal indicating the rotation direction of the motor 6), and the motor drive device 3 that is an inverter circuit
Output to 5.

The q-axis command current Iq * corresponds to the motor control value. The motor drive device 35 corresponds to an electric motor drive means. As shown in FIG. 2, the motor driving device 35 uses the FET (F
Field-Effect Transistor) 81U, 82U series circuit, FET 81V, 82V series circuit, FET 81
It is configured by connecting a series circuit of W and 82W in parallel. A voltage boosted higher than the voltage of the battery mounted on the vehicle is applied to each series circuit. And FE
The connection point 83U between T81U and 82U is connected to the U-phase winding of the motor 6, and the connection point 83V between the FETs 81V and 82V.
Is connected to the V phase winding of the motor 6, and FETs 81W and 82
The connection point 83W between W is connected to the W-phase winding of the motor 6.

FET 81U, 82U, FET 81V, 8
The PWM control signals UU, VU, WU (PWM of each phase) are supplied to the 2V and the FETs 81W, 82W from the PWM control unit 62, respectively.
A PWM wave signal and a signal indicating the rotation direction of the motor 6 are input to the control signal.

The motor drive unit 35 uses the PWM control signal U
Three-phase exciting currents corresponding to U, VU, and WU are generated and supplied to the motor 6 via the three-phase exciting current paths.
Also, FETs 82U, 82V, 82W are shunt resistors R
It is grounded to the ground PGND of a circuit board (not shown) via U, RV, and RW.

The input terminals of the shunt resistors RU, RV, RW are connected to the non-inverting input terminals of the amplifier circuits 74U, 74V, 74W, and the output terminals of the shunt resistors RU, RV, RW are the inverting input terminals, respectively. It is connected. The amplifier circuits 74U, 74V, 74W amplify the potential difference across the shunt resistors RU, RV, RW by a predetermined gain of the amplifier circuits, and A / D converters 75U, 75V,
Input to 75W. The input value is the A / D converter 7
The A / D value, that is, the digital value is converted by 5U, 75V, and 75W, and the converted values are respectively the U-phase current calculation unit 76U, the V-phase current calculation unit 76V, and the W-phase current calculation unit 76.
Input to W.

A / D converters 75U, 75V, 75
W performs A / D conversion based on a command signal from an A / D control unit (not shown) of the CPU 21. The U-phase current calculation unit 76U calculates the U-phase current Iu, which is the U-phase motor current (motor current), from the input value using a predetermined current calibration formula, and outputs the calculated U-phase current Iu. Is output to the 3-phase / 2-phase coordinate conversion unit 73.

The V-phase current calculator 76V calculates the V-phase current Iv, which is the V-phase motor current (motor current), from the input value using a predetermined current calibration formula, and the calculated result V The phase current Iv is output to the 3-phase / 2-phase coordinate conversion unit 73.

The W-phase current calculator 76W calculates the W-phase current Iw, which is the W-phase motor current (motor current), from the input value using a predetermined current calibration formula, and the calculated result is obtained. A certain W-phase current Iw is output to the 3-phase / 2-phase coordinate conversion unit 73. The current calibration formula will be described later.

The three-phase / two-phase coordinate conversion unit 73 converts the motor current of each of these phases into a two-phase d-axis detection current Id and a q-axis detection current Iq, and subtractors 55, 56 and non-interference control correction. Input to the value calculation unit 63.

The two-phase pulse train signal and the zero-phase pulse train signal from the rotation angle sensor 30 are continuously supplied to the electrical angle converter 64 at a predetermined sampling cycle. The electrical angle conversion unit 64 uses the motor 6 based on the pulse train signals.
The electrical angle θ of the rotor with respect to the stator is calculated, and the calculated electrical angle θ is input to the angular velocity conversion unit 65. The angular velocity converter 65 differentiates the electrical angle θ to calculate the angular velocity ω of the rotor with respect to the stator. The positive angular velocity ω represents rotation of the rotor in the positive direction, and the negative angular velocity ω represents rotation of the rotor in the negative direction.

(Regarding Current Calibration Formula) Here, the current calculation unit (U-phase current calculation unit 76U, V-phase current calculation unit 76 for each phase).
The current calibration formula used in the V and W phase current calculator 76W) will be described.

Since the resistance values of the shunt resistors RU, RV, RW and the amplification degree of the amplifier circuits 74U, 74V, 74W vary, A corresponding to the potential difference across the shunt resistors is present.
/ D value based on the motor current of each phase (U phase current Iu, V
It is necessary to obtain in advance a current calibration formula for calculating the phase current Iv, W-phase current Iw).

The current calibration formula is a linear approximation formula of I = G × A / D value + offset value. Note that G is a constant. I is the motor current.

When obtaining the current calibration formula, a pseudo load (the pseudo load may be used instead of the motor 6) may be connected to the motor drive unit 35 connected to the booster circuit 100. A current sensor (reference device) for detecting the current flowing in each phase is provided in each phase. Then, the PWM control signals UU, VU, WU are appropriately output to the motor drive device 35 to drive the motor drive device 35, and a current is supplied to each phase.

The currents flowing in the respective phases at this time are respectively detected by the respective current sensors, the detected values (current values) are output to the CPU 21 of the control device 20, and the shunt resistance across the shunt resistor is output for each phase. At least two combinations of the A / D value corresponding to the potential difference and the current value detected by the current sensor (see FIG. 6).
(See points a to c in (c)). The current calibration formula is obtained for each phase by linear approximation that passes through or is close to each of these two points.

In order to obtain this current calibration formula, in the present embodiment, in the booster circuit 100 to be described later, each time it is driven by synchronous rectification and asynchronous rectification, that is, the booster circuit 100.
In different driving methods, the current calibration formula is required for each phase. In this embodiment, the current calibration formula obtained in advance as described above is such that the booster circuit 100 performs synchronous rectification,
Or, R can be read for each phase according to asynchronous rectification.
It is stored in a predetermined storage area of the OM 22. The current calibration formula may be stored in advance in an external storage device (not shown) so that it can be read during control.

The ROM 22 corresponds to storage means.
When the current calibration formula is stored in the external storage device,
The external storage device corresponds to the storage means. (Booster Circuit Controller) Next, a booster circuit 100 for boosting the battery voltage VPIG and a booster circuit controller for controlling the booster circuit 100 will be described. In this embodiment, the booster circuit control device is also used by the CPU 21. The booster circuit 100 corresponds to booster means.

The booster circuit 100 is provided in a current supply circuit between a vehicle-mounted battery (hereinafter referred to as battery B) as a DC power source and the motor drive device 35. In the booster circuit 100 of the present embodiment, the application point P1 and the voltage application point P
A boosting coil (coil L) and a second transistor Q2 are connected between the two. The second transistor Q2
Has a source connected to the output terminal of the coil L and a drain connected to the voltage application point P2. The gate of the second transistor Q2 is connected to the CPU 21. D2
Is a parasitic diode of the second transistor Q2.

The application point P1 is a rectifying capacitor C1.
It is grounded to the ground PGND of the circuit board (not shown) via. The circuit board of the booster circuit 100 is common to the circuit board of the motor drive device 35, and each circuit mounted on the circuit board or an element forming the circuit is grounded to the same ground PGND.

The application point P1 corresponds to the output terminal of the DC power supply. The voltage application point P2 is grounded to the ground PGND of the circuit board (not shown) via the boosting capacitor C2.

The capacitor C2 is the second transistor Q.
It is connected to the drain that serves as the output terminal of No.2. The capacitor C2 corresponds to a boosting capacitor that charges the boosted voltage generated by the boosting coil.

The first transistor Q1 has a drain connected to the connection point between the output terminal of the coil L and the second transistor Q2, and a source connected to the ground PGND of a circuit board (not shown).
Grounded to. The gate of the first transistor Q1 is connected to the CPU 21 of the booster circuit control device 101. D1 is a parasitic diode of the first transistor Q1. To detect the voltage at the voltage application point P2, the voltage application point P
Reference numeral 2 is connected to a voltage input port (not shown) of the CPU 21 so that the output voltage VBPIG can be detected as a measured value.

The first transistor Q1 and the second transistor Q2 are n-channel type MOSFETs.
The first transistor Q1 constitutes a first switching element, and the second transistor Q2 corresponds to a second switching element.

Next, a booster circuit control device for controlling both the transistors will be described. FIG. 5 shows a functional block diagram of a booster circuit control device for controlling both the transistors. That is, it is a control block diagram showing functions executed by a program inside the CPU 21.

Each unit shown in the control block diagram does not represent independent hardware, but a CPU.
21 shows the functions performed. The CPU 21 constitutes boost control means, electric motor current calculation means, and load state determination means.

The CPU 21 comprises a calculator 110, a PID controller 120, a PWM calculator 130, and an A / D converter 150. The arithmetic unit 110 uses the target output voltage VBPIG * (in the present embodiment, 20 V) stored in advance in the ROM 22.
V) and VBPIG input via the A / D converter 150 are calculated, and the deviation is supplied to the PID controller 120.

The PID control unit 120 performs proportional (P) / integral (I) / derivative (D) processing to reduce the deviation, that is, in order to perform feedback control, and then the first
This is a circuit for calculating the control amount of the transistor Q1 and the second transistor Q2. The control amount calculated by the PID control unit 120 is further converted into a duty ratio drive signal (PWM drive signal) by calculating a duty ratio α corresponding to the control amount by the PWM calculation unit 130, and the converted duty ratio. The drive signal is applied to each transistor of the booster circuit 100.

In the present embodiment, the calculated duty ratio drive signal is applied to the first transistor Q1 and the second transistor Q2 to perform on / off control alternately (see FIG. 6A). ), Or an asynchronous rectification method in which only the first transistor Q1 is applied and PWM driving is performed (see FIG. 6B).

The synchronous rectification method is performed when the motor 6 is under high load during power running and during regeneration, and the asynchronous rectification method is performed when the motor 6 is under low load during power running. Note that the low load is meant to include the case of no load to which no load is applied in the present specification.

FIG. 6A shows a pulse signal (duty ratio drive signal) applied to the first transistor Q1, where Tα is the on-time, T is the pulse period, and α is the duty ratio of the first transistor Q1 ( On-duty). The duty ratio of the second transistor Q2 is (1- | α |).

When the duty ratio α is "+", it is in the power running state, and when it is "-", it is in the regenerative state. In the present embodiment, the duty ratio α in the power running state is 0 ≦ α ≦ α0 <1.
I am trying. α0 is a limit value, and the PWM calculation unit 130
When the result of calculating the duty ratio α at exceeds α0, α0 is determined as the duty ratio α.

The duty ratio α in the regenerative state is 0 ≦ | α
| ≦ 1. In addition, in another embodiment including this embodiment, when the second transistor Q2 is turned on and off alternately with the first transistor Q1, the duty ratio of the second transistor Q2 is calculated by (1- | α |). Since it is possible, the description is omitted.

Further, to the second transistor Q2, a pulse signal (duty ratio drive signal) is applied which turns off when the first transistor Q1 is on and turns on when the first transistor Q1 is off. .

(Operation of this Embodiment) FIG. 7 shows CP
It is a flowchart of the booster circuit drive change control program executed at the time of power running executed by U21, and is executed at a predetermined control cycle when the duty ratio α is “+”.

At S10, it is determined whether the motor 6 is in a low load state or a high load state. In S10, based on the steering torque τ and the threshold value τ0 stored in the ROM 22 in advance, the motor 6
Determine the load state of.

That is, when the steering torque τ is less than or equal to the threshold value τ0, it is determined that the motor 6 has a low load, and the process proceeds to S30. When the steering torque τ exceeds the threshold value τ0, the motor 6 is high. Assuming that the load is present, the process proceeds to S20.

In S20, the CPU 21 is operated by the synchronous rectification method.
PW the first transistor Q1 and the second transistor Q2
M drive. That is, according to the duty ratio drive signal of the drive pattern shown in FIG.
The first and second transistors Q2 are alternately turned on and off.

More specifically, at the time of high load during power running, in the booster circuit 100, the first transistor Q1 performs the switching operation by the duty control by the duty ratio drive signal. As a result, energy accumulation and discharge are repeated in the coil L, and the second transistor Q2
A high voltage appears on the drain side of the. That is,
The first transistor Q1 is turned on and the second transistor Q1
When 2 is turned off, a current flows to the ground PGND via the first transistor Q1. Next, the first transistor Q1
When is turned off, the current flowing through the coil L is cut off.
When the current flowing through the coil L1 is cut off, a high voltage is generated on the drain side of the second transistor Q2 which is turned on so as to prevent the change in the magnetic flux due to the cutoff of the current. By repeating this, a high voltage is repeatedly generated on the drain side of the second transistor Q2, smoothed (charged) by the capacitor C2, and generated at the voltage application point P2 as the output voltage VBPIG.

At this time, the voltage boosted by the booster circuit 100 is related to the duty ratio α of the duty ratio drive signal output from the CPU 21. When the duty ratio α is large, the output voltage VBPIG is high, and when the duty ratio α is small, the output voltage VBPIG is low.

At the end of the processing of S20, the drive discrimination flag F is set to 1, and then this flow chart is terminated. In S30, the CPU 21 PWM-drives only the first transistor Q1 by asynchronous rectification (see FIG. 6).
(See (b)), the drive determination flag F is reset to 0, and this flowchart is ended.

The asynchronous rectification will be described. In the asynchronous rectification, the second transistor Q2 is always turned off, and only the first transistor Q1 is PW by the duty ratio drive signal.
M drive. In the case of this asynchronous rectification, the capacitor C2 is charged, and the voltage at the voltage application point P2 changes to the target output voltage VBP.
When IG * is reached, the first transistor Q1 is actually
The duty ratio (on-duty) of is close to zero.

The reason for this is that since the motor 6 has a low load, the discharge current supplied from the capacitor C2 to the motor 6 is small, and particularly when there is no load, the discharge current does not flow.

In such a state, as shown in FIG. 5, since feedback control is performed, once the voltage at the voltage application point P2 reaches the target output voltage VBPIG *, boosting is performed. This is because the duty ratio (on-duty) for is close to zero. The duty ratio (on-duty) approaching 0 means that the capacitor C
This is because a leakage current is generated in 2 and charges are actually discharged little by little, and feedback control corresponding to the leakage current is actually performed so that the duty ratio does not become 0 completely.

If the first transistor Q1 and the second transistor Q2 are driven on and off only by the synchronous rectification method during the power running of the motor 6, when the motor 6 has a low load, the motor 6 is not driven and the capacitor C2 is not driven. No discharge current is consumed. That is, in this state, the capacitor C
The charge charged in 2 will be returned to the battery B via the coil L when the second transistor Q2 is turned on. At this time, switching loss due to on / off of the second transistor Q2 and heat generation of the coil L occur.
As described above, due to the occurrence of the switching loss of the second transistor Q2 and the heat generation of the coil L, the booster circuit 100 heats up, resulting in a great decrease in efficiency.

However, as in the present embodiment, when the motor 6 is under a low load and the second transistor Q2 is completely turned off by the asynchronous rectification and the first transistor Q1 is PWM driven, the second transistor Q2 is turned on / off. Switching loss is eliminated, and electric charges do not flow from the capacitor C2 to the coil L. Therefore, the switching loss of the second transistor Q2 and the heat generation of the coil L are eliminated, and the temperature rise of the booster circuit 100 can be suppressed.

When only the first transistor Q1 is turned on and off by PWM driving only by asynchronous rectification during power running of the motor 6, a current is applied to the voltage application point P2 side via the parasitic diode D2 of the second transistor Q2. It will be in the form of supply. In this case, when the motor 6 has a high load, the current value to the voltage application point P2 side increases, and the loss (heat generation) in the second transistor Q2 (parasitic diode D2) increases, which is not preferable.

When the motor 6 enters the regenerative state, the CPU 21 causes the first transistor Q to operate by the synchronous rectification method.
The first and second transistors Q2 are PWM-driven. At this time, the output voltage VBPIG increases due to the regenerative current from the motor 6, but the second transistor Q2 is turned on by the duty control. Therefore, the second transistor Q2
A regenerative current flows to the battery B via the and is absorbed.

Next, the arithmetic control of the motor currents (U-phase current Iu, V-phase current Iv, W-phase current Iw) will be described with reference to FIG. The A / D converters 75U, 75V, 75
W performs A / D conversion for each phase based on a command signal from an A / D control unit (not shown) of the CPU 21.

At the end of A / D conversion for each phase, the CPU
21 performs interrupt processing of the current detection program of FIG. The CPU 21 executing this process realizes the functions of the U-phase current calculation unit 76U, the V-phase current calculation unit 76V, and the W-phase current calculation unit 76W.

The current detection program of FIG. 8 processes each phase, but in the following description, calculation of the U-phase current Iu corresponding to the function of the U-phase current calculation unit 76U will be described.

In this process, in S110, the CPU 21 saves (stores) the A / D value corresponding to the detected current in the buffer of the RAM 23. In S120, the booster circuit 10
Based on the drive determination flag F, it is determined whether 0 is driven by synchronous rectification. That is, whether the A / D value saved in the buffer of the RAM 23 is the value obtained by the synchronous rectification,
It is determined whether the value is obtained by asynchronous rectification.

If the drive determination flag F is set to 1, it is determined to be "YES" and the process proceeds to S130, and if it is reset to 0, it is determined to be "NO" and the process proceeds to S140.

In step S130, since the booster circuit 100 is driven by the synchronous rectification method, the CPU 21 stores the current calibration expression (1) of the following current of the U phase corresponding to the synchronous rectification in the ROM.
It is read from 22 and the motor current is calculated.

I = G0 × A / D value + p0 (1) G0 is a constant during synchronous rectification, and p0 is an offset value during synchronous rectification. When the process of S130 ends, this program ends.

On the other hand, in step S140, since the booster circuit 100 is driven by the asynchronous rectification method, the CPU 21 reads the following U-phase current calibration equation (2) corresponding to the asynchronous rectification from the ROM 22 and calculates the motor current. To do.

I = G1 × A / D value + p1 (2) G1 is a constant during asynchronous rectification, and p1 is an offset value during asynchronous rectification. The current calibration formula (1) above,
Since the equation (2) is also used to explain the calculation of the U-phase current Iu, I is the motor current corresponding to the U-phase current Iu.

When the processing of S140 is completed, this program is once terminated. Since the same processing is performed for the motor currents of the other phases, the description can be made by replacing the characters related to the U phase with the characters related to the other phase in the above description, and a duplicate description will be omitted.

The first embodiment has the following features. (1) The control device 20 of the present embodiment boosts the voltage of the battery B (DC power supply) and supplies electric power to the motor drive device 35 (electric motor drive means) of the motor 6 (electric motor). ). The booster circuit 100 is
The coil L, the first transistor Q1 (first switching element), the second transistor Q2 (second switching element), and the capacitor C2 are provided.

Then, the CPU 21 (step-up control means) of the controller 20 synchronously rectifies the first transistor Q1 and the second transistor Q2 when the motor 6 has a high load, and turns the first transistor Q1 on and off when the motor 6 has a low load, and the second transistor Q1. Asynchronous rectification is used to control Q2 off. That is, the driving method of both transistors is changed according to the load state of the motor 6.

Further, the CPU 21 (motor current calculation means) calculates the motor current (motor current) flowing through the motor drive device 35 based on the A / D value corresponding to the potential difference between both ends of the shunt resistors RU, RV, RW. The calculation was performed using a calibration formula.

Further, the CPU 21 uses the current calibration formula ((1) formula or (2) formula) according to the load state of the motor 6,
A / corresponding to the potential difference across the shunt resistors RU, RV, RW
The motor current is calculated based on the D value.

In this case, in the synchronous rectification, when the first transistor Q1 is turned on, a current flows through the ground PGND, so that the ground potential of the ground PGND rises.
In the case of synchronous rectification, the detected motor current is amplified by the amplifier circuit 7 under the increased ground potential of the ground PGND.
Amplification is performed with 4 U, 74 V, and 74 W, and the A / D value corresponding to the potential difference across the shunt resistors RU, RV, and RW is the A / D value affected by this. Then, the equation (1) of the current calibration equation is previously obtained by a test or the like as being affected by this.

Therefore, the A / D value corresponding to the potential difference across the shunt resistors RU, RV, RW during synchronous rectification is correctly calibrated by the equation (1). On the other hand, in asynchronous rectification, the second transistor Q2 is always turned off, and only the first transistor Q1 is PWMed by the duty ratio drive signal.
To drive. In the case of this asynchronous rectification, the capacitor C2 is charged, and the voltage at the voltage application point P2 changes to the target output voltage VBPIG.
When it reaches *, the duty ratio (on-duty) of the first transistor Q1 actually becomes close to zero. That is, even when the first transistor Q1 is turned on, the on-time is extremely short compared with the time of synchronous rectification.

Therefore, when the first transistor Q1 is turned on, even if a current flows through the ground PGND, the increase in the ground potential of the ground PGND is smaller than that in the case of synchronous rectification.

In the case of the asynchronous rectification, the detected motor current is amplified by the amplifier circuits 74U, 74V, 7 under the ground potential in which the rise of the ground PGND is smaller than that in the synchronous rectification.
Amplified at 4W, shunt resistors RU, RV, R
The A / D value corresponding to the potential difference between both ends of W is
/ D value. Then, the equation (2) of the current calibration equation is previously obtained by a test or the like as being affected by this.

Therefore, the A / D value corresponding to the potential difference across the shunt resistors RU, RV, RW during asynchronous rectification is
It is correctly calibrated by the equation (2). Here, if the motor 6 has a low load and the drive system of the booster circuit 100 is changed from synchronous rectification to asynchronous rectification, the shunt resistor RU is used.
Etc., the actual motor current flowing, etc., and the shunt resistance RU
It is assumed that the calculation is performed using the current calibration equation (Equation (1)) previously obtained by synchronous rectification based on the A / D value corresponding to the potential difference between both ends.

In this case, the value of the motor current flowing during the asynchronous rectification is calculated using the current calibration equation (Equation (1)) obtained under the rise of the ground potential during the synchronous rectification. . However, the motor current value detected at the time of asynchronous rectification is amplified by the amplifier circuit 74U or the like under the ground potential at which the rise of the ground PGND is smaller than that at the time of synchronous rectification, and corresponds to the potential difference between both ends of the shunt resistor RU or the like. The A / D value of is the A / D value affected by this.

For this reason, as explained in the prior art, based on the actual motor current flowing through the shunt resistor RU and the A / D value corresponding to the potential difference across the shunt resistor RU and the like, synchronous rectification is performed beforehand. There is a problem that an offset occurs with the current detection value calculated using the obtained current calibration formula (equation (1)). Further, there is a problem that the control device malfunctions and the motor operates due to the current (offset current) caused by the offset at the moment when the driving method of the booster circuit 100 is switched.

On the other hand, in the present embodiment, the booster circuit 100 for boosting the battery voltage VPIG (DC power supply voltage).
Since there is no offset current generated by the switching of the driving method of (No), it is possible to prevent the malfunction of the motor 6 when the driving method of the booster circuit 100 is switched.

(2) The control device 20 of the present embodiment is provided with the ROM 22 (storage means) for storing the current calibration formula according to the load state of the motor 6 (electric motor). And CPU
21 (motor current calculation means) reads the current calibration formula corresponding to the load state of the motor 6 from the current calibration formulas stored in the ROM 22 and calculates the motor current (motor current).

As a result, by reading the current calibration formula corresponding to the load of the motor 6 from the ROM 22, the motor current of each phase can be calculated according to the load state of the motor 6.

(3) In the control device 20 of this embodiment,
The CPU 21 (load state determination means) determines the load state of the motor 6. Further, the CPU 21 (electric motor current calculation means) selects the current calibration formula based on the determination result of the load state of the motor 6.

As a result, the CPU 21 determines whether the load state of the motor 6 is low load or high load.
A current calibration formula corresponding to synchronous rectification or a current calibration formula corresponding to asynchronous rectification can be selected.

(4) In the control device 20 of this embodiment,
The CPU 21 (step-up control means) synchronously rectifies the first transistor Q1 and the second transistor Q2 when the motor 6 has a high load and asynchronous rectifies when the motor 6 has a low load.

As a result, depending on the load state of the motor 6,
The driving method of the booster circuit 100 can be changed. (5) In the motor current calculation method of the control device 20 of the present embodiment, the current calibration formulas (Formulas (1) and (2)) corresponding to the load state of the motor 6 are used to calculate the potential difference across the shunt resistor RU. The motor current (motor current) is calculated based on the A / D value.

As a result, the same effect as the above (1) can be obtained by the above motor current calculation method. (6) In the present embodiment, the torque sensor 4 (steering torque detecting means) for detecting the steering torque τ is provided. And
When the steering torque τ detected by the torque sensor 4 is less than or equal to the threshold value τ0, the CPU 21 (load state determination means) determines that the load state of the motor 6 is low, and when the steering torque τ exceeds the threshold value τ0. Determines that the load state of the motor 6 is high.

As a result, the steering torque τ and the threshold value τ0 make it easy to determine whether the motor 6 has a low load or a high load. (Second Embodiment) Next, a second embodiment will be described with reference to FIG.

In the first embodiment, the load state detection parameter of the motor 6 is the steering torque τ, but in the second embodiment, the motor rotation speed n of the motor 6 is used as the load state detection parameter of the motor 6. The location is different.

That is, in this embodiment, the rotation angle sensor 30 also serves as a rotation position sensor for detecting the rotation position of the motor 6, and the CPU 21 corresponds to the electric motor rotation speed estimating means.

Then, in the flowchart of FIG. 7, S10
Then, the CPU 21 calculates the motor rotation speed n using a known arithmetic expression based on the detection signal from the rotation angle sensor 30.

Then, the motor speed n and the ROM 22 are stored in advance.
The magnitude relationship with the rotation speed threshold value n0 stored in, that is, whether the motor 6 is in a low load state or a high load state is determined. When the motor rotation speed n is less than or equal to the rotation speed threshold value n0,
It is determined that the motor 6 has a low load, and the process proceeds to S30. If the motor rotation speed n is greater than the rotation speed threshold value n0, the motor 6 is considered to be a high load and the process proceeds to S20.

In the second embodiment, (1) to (1) of the first embodiment
In addition to (4), there are the following features. (1) In the second embodiment, the CPU 21 is used as an electric motor rotation speed estimation means for estimating the rotation speed of the motor 6. When the estimated rotation speed of the motor 6 (motor rotation speed n) is equal to or lower than the rotation speed threshold value n0, the CPU 21 (load state determination means) determines that the load state of the motor 6 is low, and the motor rotation speed is low. When the number n exceeds the rotation speed threshold value n0,
The load state of the motor 6 is determined to be high.

As a result, the motor speed n and the speed threshold n
With 0, it is possible to easily determine whether the motor 6 has a low load or a high load. You may change the structure of 2nd Embodiment as follows.

(A) Although the brushless motor is used in the second embodiment, a brushed motor is used instead of the brushless motor (hereinafter, simply referred to as a motor in this section).
You may change to. Even in this case, the flowchart of FIG. 7 is executed.

In S10, the motor rotation speed n is calculated (estimated) as follows. In order to detect the motor current of the motor, the motor is provided with a motor current detection circuit (not shown) and a motor terminal voltage detection circuit (not shown) for detecting the voltage between the motor terminals.

In order to calculate the motor rotation speed n of the motor 6, the CPU 21 first calculates the average value of the motor current (motor current average value Ia) detected by the motor current detection circuit (not shown) and the motor terminal. The average value of the terminal voltage detected by the voltage detection circuit (the terminal voltage average value Va) is calculated. From the obtained motor current average value Ia and inter-terminal voltage average value Va, the instantaneous value of the motor internal resistance (motor internal resistance instantaneous value R) is calculated according to the equation (3).

R = Va / Ia (3) Next, the motor internal resistance instantaneous value R is integrated over time to obtain the motor internal resistance value Ri, and the internal resistance value Ri, the motor current average value Ia and the inter-terminal voltage are obtained. The back electromotive force Vc of the motor is calculated using the equation (4) based on the average value Va.

Vc = Va-IaRi (4) Then, using the equation (5), the counter electromotive voltage Vc is multiplied by the motor power generation constant K which is the ratio of the rotation speed to the counter electromotive voltage Vc. , The motor rotation speed n is calculated.

The motor speed n is the counter electromotive voltage Vc of the motor.
It has a code corresponding to the code of. The motor speed n
Has a positive value for the right rotation of the motor and a negative value for the left rotation of the motor. That is, the motor rotation speed n is the rotation speed including the rotation direction component of the motor.

N = KVc (5) Therefore, in this modification, in S10, | n |> n0 is used to determine the magnitude relationship.

The other structure is similar to that of the second embodiment.
The CPU 21 corresponds to a motor rotation speed estimation means. (B) Further, a steering wheel rotation speed sensor for detecting the rotation speed of the steering wheel 1 (steering wheel) is provided, and the CPU 21 calculates (estimates) the motor rotation speed n based on the steering wheel rotation speed detected by the steering wheel rotation speed sensor. You may do it. This is also acceptable because the handle rotation speed and the motor rotation speed n are in a proportional relationship.

Also in this case, the CPU 21 corresponds to the electric motor rotation speed estimating means. (Third Embodiment) Next, a third embodiment will be described with reference to FIG.

In the first embodiment, the steering torque τ is used as the load state detection parameter of the motor 6, but in the third embodiment, the assist command current, that is, the q-axis command current Iq * is used as the load state detection parameter of the motor 6. The difference is that (motor control value) is used.

In S10 of FIG. 7, the CPU 21 reads the q-axis command current Iq * (motor control value), and the q-axis command current Iq.
The magnitude relationship between q * and the command value threshold Iq * s stored in the ROM 22 in advance, that is, whether the motor 6 is in a low load state or a high load state is determined.

If the q-axis command current Iq * is less than or equal to the command value threshold value Iq * s, it is determined that the motor 6 has a low load and S30
If the q-axis command current Iq * is larger than the command value threshold value Iq * s, it is determined that the motor 6 has a high load and S
Move to 20.

In the third embodiment, (1) to (1) of the first embodiment
In addition to (4), there are the following features. (1) In the third embodiment, the CPU 21 (load state determination means) determines that the load state of the motor 6 is low load or high load based on the q-axis command current Iq * (motor control value). I chose Specifically, the magnitude relation between the q-axis command current Iq * and the command value threshold Iq * s stored in advance, that is, whether the motor 6 is in a low load state or a high load state is determined.

As a result, the q-axis command current Iq * and RO
With the command value threshold Iq * s stored in M22, the motor 6
Can easily determine whether the load is low or high. The embodiment of the present invention may be modified as follows.

In each of the above embodiments, the steering torque τ
Instead of the embodiment using the vehicle speed V and the vehicle speed V, the electric motor control value may be determined only by the steering torque τ. In each of the above-described embodiments, the booster circuit 100 is driven by an asynchronous rectification method in which the second transistor Q2 is always off and only the first transistor Q1 is PWM-driven by the duty ratio drive signal when the motor 6 has a low load. I went there.

Alternatively, when the motor 6 has a low load, the first transistor Q1 and the second transistor Q2 may all be turned off, that is, the booster circuit 100 may stop boosting.
Therefore, in the case where the current calibration formula is obtained in advance in this modified example, in the booster circuit 100, synchronous rectification and the first transistor Q1 and the second transistor Q2 are all turned off, so that different phases are used for different phases. Calculate the current calibration formula.

Even in this case, the effects of each embodiment can be obtained. The motor 6 may be synchronously rectified when the load is high, and when the load is low, only the first transistor Q1 may be completely turned off and the second transistor Q2 may be PWM-driven.

Therefore, when the current calibration formula is obtained in advance in this modified example, in the booster circuit 100, synchronous rectification and only the first transistor Q1 are all turned off, and the second transistor Q2 is PWM-driven, so that they are different from each other. In the drive method, a current calibration formula is obtained for each phase.

Even in this case, the effects of each embodiment can be obtained. Next, technical ideas other than the invention described in the claims that can be understood from the above-described embodiment will be described below.

(1) In claim 3, steering torque detecting means for detecting a steering torque is provided, and the load state determining means determines the load state of the electric motor when the steering torque detected by the steering torque detecting means is small. Is determined to be a low load, and when the steering torque is large, it is determined that the load state of the electric motor is a high load.

(2) In claim 3, an electric motor revolution speed estimating means for estimating the revolution speed of the electric motor is provided, and the load condition judging means is provided when the revolution speed estimated by the electric motor revolution speed estimating means is small. The electric power steering control device is characterized in that the load state of the electric motor is determined to be low load, and when the rotational speed is high, the load state of the electric motor is determined to be high load.

(3) The electric power steering control device according to claim 3, wherein the load condition judging means judges the load condition of the electric motor based on the electric motor control value.

[0148]

As described in detail above, the inventions of claims 1 to 5 can cancel the offset current generated by switching the driving method of the booster circuit for boosting the DC power supply voltage. Then, when the driving method of the booster circuit is switched, it is possible to prevent the malfunction of the motor.

[Brief description of drawings]

FIG. 1 is a schematic diagram of an electric power steering control device embodied in the present embodiment of the invention.

FIG. 2 is a control block diagram of the electric power steering control device.

FIG. 3 is a control block diagram of CPU 21.

FIG. 4 is an electric circuit diagram of the booster circuit.

FIG. 5 is a control block diagram of the control device for boosting the same.

6A is a waveform diagram of PWM drive signals of both transistors in the case of a synchronous rectification system, FIG. 6B is a waveform diagram of PWM drive signals of both transistors in the case of an asynchronous rectification system, and FIG. , Explanatory drawing of a calibration formula.

FIG. 7 is a flowchart of a control program executed by the CPU 21 of this embodiment.

FIG. 8 is a flowchart of a control program executed by the CPU 21 of this embodiment.

[Explanation of symbols]

4 ... Torque sensor (steering torque detection means) 6 ... Motor (electric motor) 20 ... Control device 21 ... CPU (boost control means, electric motor current calculation means, load state determination means) 35 ... Motor drive device (electric motor drive means) 100 ... Step-up circuit (step-up means) B ... Battery (DC power supply) L ... Coil (step-up coil) C2 ... Capacitor (step-up capacitor) Q1 ... First transistor (first switching element) Q2 ... Second transistor (second switching element) ) RU, RV, RW ... Shunt resistance

─────────────────────────────────────────────────── ─── Continuation of front page (51) Int.Cl. ⁷ Identification code FI theme code (reference) // B62D 101: 00 H02P 6/02 321N 119: 00 321P F term (reference) 3D032 DA15 DA23 DA64 DA65 DC01 DD10 DD17 GG01 3D033 CA03 CA13 CA16 CA20 CA21 5H560 AA10 BB04 BB12 DA01 DB01 EB01 JJ01 SS02 TT12 TT15 UA05 XA12 XA13 XB04 5H570 AA21 BB09 DD04 GG01 GG04 HA09 HB03 JJ03 JJ16 JJ24 LL12 MM02

Claims (5)

[Claims] 1. A step-up means for stepping up the voltage of a DC power supply to supply electric power to a motor driving means of an electric motor, the coil being connected to the DC power supply and connected to an output terminal of the coil. A step-up device including a first switching element, a second switching element, and a capacitor connected to an output terminal of the second switching element, and a driving method for the both switching elements is changed according to a load state of the electric motor. The electric power is provided with a step-up control means for performing step-up control by electric current, and a motor current calculation means for calculating a motor current flowing through the motor drive means by a current calibration formula based on an A / D value corresponding to the potential difference across the shunt resistor. In the steering control device, the electric motor current calculation means is a current calibration formula according to a load state of the electric motor, and an A / D corresponding to a potential difference between both ends of the shunt resistor. Based on, electric power steering control apparatus characterized by computing the motor current. 2. A storage unit for storing a current calibration formula corresponding to a load state of the electric motor, wherein the electric motor current calculating unit is a current calibration formula stored in the storage unit and corresponds to a load state of the electric motor. The electric power steering control device according to claim 1, wherein the electric current steering formula is read to calculate the electric motor current. 3. A load state determination means for determining a load state of the electric motor, wherein the electric motor current calculation means selects a current calibration formula based on a determination result of the load state determination means. The electric power steering control device according to claim 1 or 2. 4. The step-up control means performs synchronous rectification of both switching elements when the electric motor has a high load, and asynchronous rectification when the electric motor has a low load. 2. The electric power steering control device according to item 1. 5. A step-up means for stepping up the voltage of a DC power supply to supply electric power to a motor driving means of an electric motor, the coil being connected to the DC power supply and being connected to an output terminal of the coil. A step-up device including a first switching element, a second switching element, and a capacitor connected to an output terminal of the second switching element, and a driving method for the both switching elements is changed according to a load state of the electric motor. And a step-up control means for performing step-up control by means of an electric power steering control device for calculating a motor current flowing through the motor drive means by a current calibration formula based on an A / D value corresponding to a potential difference across the shunt resistor. In the current calculation method, the motor current is calculated based on the A / D value corresponding to the potential difference between both ends of the shunt resistance, using a current calibration formula according to the load state of the motor. Motor current calculation method of an electric power steering control apparatus according to claim Rukoto.