JP2003319652A - Switching power supply circuit - Google Patents

Abstract

(57) [Problem] To improve the cross-regulation characteristics of a composite resonance type converter having a current resonance type converter on the primary side and a voltage resonance circuit on the secondary side, and having a zero converter transformer gap. SOLUTION: A secondary winding for obtaining a second DC output voltage is wound so as to be sandwiched by first secondary windings divided so as to have the same number of turns. Then, at this time, the first secondary winding is wound separately in half. Alternatively, for the entire secondary winding, a first tap output terminal and a second tap output terminal provided so that the number of turns from the winding start end and the number of turns from the winding end end are equal. By partitioning, a secondary winding for obtaining a second DC output voltage is formed. Thus, in the secondary winding for obtaining the second DC output voltage, the alternating voltage is evenly excited at the center tap.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit provided as a power supply in various electronic devices.

[0002]

2. Description of the Related Art As a switching power supply circuit, a switching power supply circuit that employs a switching converter such as a flyback converter or a forward converter is widely known. Since the switching operation waveforms of these switching converters are rectangular waves, there is a limit in suppressing switching noise. Also, due to its operating characteristics,
It is known that there is a limit to the improvement of power conversion efficiency. On the other hand, the present applicant has recently proposed switching power supply circuits using various resonant converters. This resonant converter can easily obtain high power conversion efficiency and realize low noise because the switching operation waveform is sinusoidal. It also has the advantage that it can be configured with a relatively small number of parts.

[0003]

However, in such a resonance type converter, in order to prevent magnetic saturation, a gap is formed with respect to the magnetic leg of the core of the converter transformer. There was also a limit to improving the power conversion efficiency. Therefore, the present applicant previously invented a resonance type converter in which the gap of the converter transformer is zero, and further improves the power conversion efficiency in the switching power supply circuit.

The circuit diagram of FIG. 8 shows an example of a switching power supply circuit as a prior art in which the gap of the converter transformer is zero, which can be constructed based on the invention previously made by the present applicant. As the basic configuration of the power supply circuit shown in this figure, a self-exciting current resonance type converter is provided as a primary side switching converter.

In the power supply circuit shown in this figure, first,
A full-wave rectifying / smoothing circuit including a bridge rectifying circuit Di and one smoothing capacitor Ci is provided for the commercial AC power supply AC. The full-wave rectification operation of the bridge rectification circuit Di and the smoothing capacitor Ci results in the rectification smoothing voltage Ei (DC input voltage) across the smoothing capacitor Ci. This DC input voltage Ei is
The level corresponds to the same size as the AC input voltage VAC.

The switching converter of the power supply circuit shown in this figure is of a current resonance type, and two switching elements Q1 and Q2 are connected by a half bridge coupling as shown in the figure. In this case, bipolar transistors are selected as the switching elements Q1 and Q2. The base of the switching element Q1 is connected to a self-excited oscillation drive circuit formed by series-connecting a base current limiting resistor RB1-resonance capacitor CB1-drive winding NB1. A damper diode DD1 is connected between the base and emitter of the switching element Q1 in the direction shown. A starting resistor Rs1 is connected between the collector and the base of the switching element Q1 to allow a current at the time of starting to flow to the base. Similarly, for the base of the switching element Q2, the base current limiting resistor RB2-resonance capacitor CB2
-A self-excited oscillation drive circuit formed by connecting drive windings NB2 in series is connected. A damper diode DD2 is connected between the base and the emitter, and a starting resistor Rs2 is connected between the collector and the base.

Here, a series resonance circuit is formed by the capacitance of the resonance capacitor CB1 forming the self-excited oscillation drive circuit on the switching element Q1 side and the inductance of the drive winding NB1. Similarly, a series resonance circuit is formed by the capacitance of the resonance capacitor CB2 forming the self-excited oscillation drive circuit on the switching element Q2 side and the inductance of the drive winding NB2. Then, the switching elements Q1 and Q2 are switching-driven by the self-exciting method at the switching frequency determined by the resonance frequency of these series resonant circuits. Further, as will be described later, in the drive transformer PRT, since the alternating voltages of which the drive windings NB1 and NB2 have opposite polarities are excited, the switching element Q1 and NB2 are excited.
Q2 performs switching operation by alternately turning on / off.

Further, the collector of the switching element Q2
A partial resonance capacitor Cp is connected in parallel between the emitters.
Are connected. Depending on the capacitance of the partial resonance capacitor Cp and the leakage inductance component L1 of the primary winding N1, a parallel resonance circuit (partial voltage resonance circuit)
To form. Then, a partial voltage resonance operation is obtained in which the voltage resonates only when the switching elements Q1 and Q2 are turned off.

Drive transformer PRT (Power Regulati
The ng transformer) is provided for switching-driving the switching elements Q1 and Q2 and for variably controlling the switching frequency for constant voltage control. The drive transformer PRT winds the drive windings NB1 and NB2 and the resonance current detection winding NA, and further, saturable by winding the control winding Nc in a direction orthogonal to each of these windings. It is said to be a reactor. The drive winding N
B1 and drive winding NB2 are adapted to excite voltages having opposite polarities.

Isolation Converter Transformer PIT (Power Is
(Olation Transformer) transmits the switching outputs of the switching elements Q1 and Q2 to the secondary side. The winding end of the primary winding N1 of the insulating transformer PIT is connected to the contact (switching output point) between the emitter of the switching element Q1 and the collector of the switching element Q2 via the resonance current detection winding NA. , A switching output is obtained.

In this case, the winding start end of the primary winding N1 is connected to the primary side ground via the series resonance capacitor C1. The series resonance capacitor C
The capacitance of 1 and the inductance component of the insulating converter transformer PIT including the primary winding N1 form a primary side series resonance circuit for making the operation of the primary side switching converter a current resonance type. In this way, as the primary side switching converter shown in this figure, the current resonance type operation and the partial voltage resonance operation described above are obtained in a composite manner.

A secondary winding N2 and a secondary winding N2 are provided on the secondary side of the insulating converter transformer PIT in this case.
A tertiary winding N3 having a smaller number of turns (number of turns) than that is wound. The tertiary winding N3 is provided with a center tap as shown in the drawing, and the tertiary winding N3 is divided at the center tap. Here, the portions divided by the center tap will be referred to as "tertiary winding N3a" and "tertiary winding N3b" for the sake of explanation. The number of turns of each winding in this insulating converter transformer PIT is as follows: primary winding N1 = 30T (turn), secondary winding N2 = 55T, tertiary winding N3 = tertiary winding N3a + tertiary winding N3b = 6T + 6T. It

The insulating converter transformer PIT has a structure shown in FIG. 9, for example. Insulation converter transformer PIT consists of E-shaped cores CR1 and CR2 made of ferrite material.
An EE type core in which the magnetic legs are opposed to each other is provided. Then, for the bobbin B which is divided so that the winding areas on the primary side and the secondary side are independent of each other and is integrated, the winding area on the primary side has the primary winding N1 and the winding area on the secondary side. A secondary winding N2 and a tertiary winding N3 are wound around the winding region as shown in the figure. That is, the tertiary winding N3a and N3b are wound in this order from the inside, and the secondary winding N2 is wound outside the tertiary winding N3. The gap G is not formed in the central magnetic leg as shown in the figure. Accordingly, the coupling coefficient k is, for example, k≈
A tightly coupled state of about 0.95 is obtained.

In the circuit of FIG. 8, the insulating converter transformer PIT is not magnetically saturated even if the gap is not formed in this way, because the primary winding N1 and the secondary winding N2 are respectively as described above. The number of windings,
This is because the number of windings of the secondary winding N2 with respect to the primary winding N1 is increased more than in the conventional case, and the magnetic flux density of the ferrite magnetic core is reduced.

On the secondary side of the insulating converter transformer PIT, a secondary side partial voltage resonance capacitor C2 is connected in parallel with the secondary winding N2 as shown in the figure. Then, the capacitance of the secondary partial voltage resonance capacitor C2 and the leakage inductance L2 of the secondary winding N2.
Depending on and, a secondary side partial voltage resonance circuit is formed.
Therefore, the secondary winding N of the isolation converter transformer PIT
When the alternating voltage is excited in 2, the voltage (partial voltage) resonance operation can be obtained even on the secondary side. That is, the power supply circuit shown in FIG. 1 is configured as a current resonance type converter so that the current resonance operation and the partial voltage resonance operation are obtained on the primary side and the partial voltage resonance operation is obtained on the secondary side. It will be.

Since the resonance operation can be obtained on the secondary side as described above, the circuit shown in this figure prevents the abnormal oscillation during the intermediate load when the gap is zero as described above. Is what you are doing.

For the secondary winding N2, a bridge rectifier circuit DBR and a smoothing capacitor C are provided as shown in the figure.
By connecting O1, the secondary side DC output voltage EO1 is obtained across the smoothing capacitor CO1 by the full-wave rectification operation. For the tertiary winding N3,
The rectifying diode DO1 is connected to the end on the side of the tertiary winding N3a, and the rectifying diode D02 is connected to the end on the side of the tertiary winding N3b. Then, these rectifying diodes DO1 and DO2
Are connected to the positive terminal of the smoothing capacitor CO2, respectively, thereby forming a double-wave rectification circuit, and a secondary side DC output voltage lower than the DC output voltage EO1 is applied across the smoothing capacitor CO2. EO2 can be obtained. These secondary side DC output voltages EO1 and EO2 are supplied to loads not shown. The secondary side DC output voltage EO1 is also branched and input as a detection voltage for the control circuit 1.

In the control circuit 1, the secondary side DC output voltage EO1
By varying the level of the control current (DC current) flowing through the control winding NC in accordance with the level change of the control winding NC, the inductance LB of the drive winding NB wound around the orthogonal control transformer PRT is variably controlled. As a result, the resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the main switching element Q1 formed including the inductance LB of the drive winding NB changes. This is an operation to change the switching frequency of the main switching element Q1,
By this operation, the secondary side DC output voltage (EO1, EO2)
Stabilize.

FIG. 11 is a waveform diagram showing the operation of the main part of the power supply circuit shown in FIG. 8 by the switching cycle. The operation waveform shown in this figure is the AC input voltage VAC
= 100V system, the measurement result is shown under the condition of load power Po = 125W. In this figure, the switching element Q2 is turned on during the period TON and the period TO
Switching operation is performed so that it is turned off in FF.
As shown in the figure, the switching current IQ2 flowing through the switching element Q2 is 0 level in the period TOFF, and in the period TON, first, at the start, from the damper diode DD2 through the base → collector of the switching element Q2. A damper current flows in the positive direction of the negative electrode, and then has a waveform flowing through the collector-emitter.

Further, the collector of the switching element Q2
The emitter-to-emitter voltage VQ2 becomes a pulse clamped at the level of the rectified and smoothed voltage Ei (DC input voltage) in the period TOFF, and has a waveform of 0 level in the period TON. The switching element Q1 is switched with respect to the switching element Q2 at the timing of being alternately turned on / off. Therefore, the switching element Q
The switching current and the collector-emitter voltage of 1 have substantially the same waveform shapes as the switching current IQ2 and the collector-emitter voltage VQ2.
The phase is shifted by 80 °.

The primary winding current I1 flowing through the primary winding N1 as a switching output is shown in this figure. The primary winding current I1 is a resonance current obtained by the resonance operation of the series resonance circuit of the primary side series resonance circuit (C1-N1) according to the switching operation of the switching elements Q1 and Q2. Then, this primary winding current I1
The switching element Q1 is off and the switching element Q2 is
During the period TON when the switching element is turned on, the switching element Q
The second switching current IQ2 flows through the switching element Q2. Further, during the period TOFF in which the switching element Q2 is off and the switching element Q12 is on, the switching current of the switching element Q1 flows to the switching element Q1.

As the primary winding current I1 thus flows, as the voltage V3 generated in the tertiary winding N3, a waveform having a cycle corresponding to the primary winding current I1 is obtained as shown in the figure. Then, as the voltage V3, a waveform whose absolute value level is clamped at the level of the secondary side DC output voltage EO2 is obtained. In addition, the rectifying diode DO1
The waveforms of the rectifying current ID1 flowing through the rectifying current ID1 and the rectifying current ID2 flowing through the rectifying diode DO2 are as shown in the figure. From these waveforms, it is understood that the rectifying diodes DO1 and DO2 are alternately turned on / off. .

In order to obtain the operation shown in FIG. 11 as the power supply circuit shown in FIG. 8, each part is selected as follows. Primary winding N1 = 30T Secondary winding N2 = 55T Tertiary winding N3 = Tertiary winding N3a + Tertiary winding N3b = 6T +
6T Primary side series resonance capacitor C1 = 0.15 μF Primary side partial resonance capacitor Cp = 680 pF Secondary side partial voltage resonance capacitor = 4700 pF

Here, in the power supply circuit shown in FIG. 8, in addition to the secondary side DC output voltage EO1 to the load side, the secondary side DC output voltage EO2 lower than this is separately output. Has been done. As the secondary side DC output voltage EO2, the tertiary winding N3 (tertiary winding N3a, tertiary winding N3b) is placed inside the secondary winding N2 that obtains the secondary side DC output voltage EO1. It is designed to be obtained by winding it.

However, since the tertiary winding N3 is wound inside the secondary winding N2 in this way, in the circuit shown in FIG. 8, the tertiary winding N3a is wound more than the innermost tertiary winding N3a. The tertiary winding N3b wound on the outer side has a larger transformer coupling with the secondary winding N2. Along with this, the rectifying diode DO1 (the tertiary winding N
The rectification current ID1 flowing in 3a) is the rectification diode DO2.
It becomes larger than the rectified current ID2 flowing through (the tertiary winding N3b). According to the experiment, the peak level of the rectified current ID1 becomes 4.1 Ap as shown in FIG.
The rectified current ID2 is 2.5 Ap.

Thus, the tertiary winding N3a and the tertiary winding N3b
Due to the difference in the magnitude of the currents flowing between and, the power loss in the rectifying diode DO1 and the rectifying diode DO2 becomes uneven. Therefore, in the circuit of FIG.
Heat generation on the side of these rectifying diodes DO1 will increase.

Further, since the tertiary winding N3 is wound inside the secondary winding N2 as described above,
The waveforms of the rectified current ID1 and the rectified current ID2 are shown in FIG.
As shown in FIG. 4, the rectified current ID1 and the rectified current ID2 have noises during the turn-off period of the rectified diode DO1 and the rectified diode DO2, respectively.

Further, as described above, since the levels of the rectified current ID1 and the rectified current ID2 are biased, the difference between the levels shown in FIG.
In the circuit of No. 2, since the change in the secondary side DC output voltage EO2 with respect to the load variation on the secondary side DC output voltage EO1 side becomes large, the cross regulation characteristic cannot be improved as shown in FIG. The problem arises.

FIG. 10 is a diagram for explaining the cross regulation characteristic in the circuit of FIG. In this figure, the load current IO2 on the secondary side DC output voltage EO2 side is shown.
(A) and load power Po on the secondary side DC output voltage EO1 side
Secondary side DC output voltage EO2 against level fluctuation of (W)
The change characteristic of (V) is shown. As shown in this figure,
In the circuit of FIG. 8, when the load power Po is 0 W and the load current IO2 is 2 A, the level of the secondary side DC output voltage EO2 is EO2 = 7 V, which is the lowest level. On the other hand, the secondary side DC output voltage EO2 is highest at 14.5 V when the load power Po is 100 W and the load current IO2 is 0 A. Therefore,
In the circuit shown in FIG. 8, the secondary side DC output voltage EO2 is 7V to 1 due to the fluctuations of the load power Po and the load current IO2.
It changes to 4.5V.

Here, as for the output which is not detected and input to the control circuit 1 like the secondary side DC output voltage EO2 in the circuit of FIG. Local regulators such as so-called three-terminal series regulators and step-down converters have been constructed. Considering the case where such a local regulator is used in the circuit of FIG. 8, since the level fluctuation of the secondary side DC output voltage EO2 is large as described above, the control range of these local regulators is also large. Will be. That is, when the local regulator as described above is used in the circuit of FIG. 8, there is a problem that the loss in the local regulator cannot be reduced.

[0031]

In view of the above problems, the present invention has a switching power supply circuit configured as follows. That is, first, in order to transmit the switching output of the switching means from the primary side to the secondary side, the current resonance type switching means for performing the switching operation by intermittently connecting the input DC input voltage is provided. A primary winding, a first secondary winding, and a second secondary winding having a smaller number of windings than the first secondary winding, and forming a gap in the magnetic leg. For EE type cores or U-U type cores that are not
The primary winding is wound on the primary side, 1/2 of the first secondary winding is wound on the secondary side, and the second secondary winding is wound on the outer side. Further, 1/2 of the first secondary winding is wound on the outer side, and the primary winding and the secondary winding are tightly coupled with each other by a coupling coefficient higher than necessary. An insulating converter transformer configured as follows is provided. Then, at least the leakage inductance component of the primary winding of the insulating converter transformer and the capacitance of the primary side series resonance capacitor connected in series to the primary winding are formed, and the operation of the switching means is a current resonance type. The primary side series resonance circuit and the capacitance of a partial resonance capacitor connected in parallel to a predetermined switching element among a plurality of switching elements forming the switching means, and the leakage inductance of the primary winding of the insulating converter transformer. A primary side partial voltage resonance circuit that performs a voltage resonance operation during a turn-off period of a plurality of switching elements that form the switching means, and that further includes a first side of the isolation converter transformer on the secondary side. Secondary winding leakage inductor And vinegar component,
And a secondary side partial voltage resonance circuit formed by the capacitance of a secondary side partial voltage resonance capacitor connected in parallel to the first secondary winding.
Then, the alternating voltage obtained in the first secondary winding is input to perform a rectifying operation to generate a first DC output voltage, and a first DC output voltage generating means configured to generate a first DC output voltage. , The second secondary winding is center tapped and grounded, and then a pair of rectifying diodes is provided to input the alternating voltage obtained in the second secondary winding, A second direct current output voltage generating means configured to generate a second direct current output voltage by performing a rectifying operation, and further, according to the level of the first direct current output voltage, The constant frequency control means is configured to perform constant voltage control on the first DC output voltage and the second DC output voltage by changing the switching frequency of the switching means.

Further, in the present invention, the switching power supply is configured as follows. That is, a current resonance type switching means for performing a switching operation by interrupting an input DC input voltage, and a means for transmitting the switching output of the switching means from the primary side to the secondary side, A primary winding is wound around the primary side of the EE type core or the U-U type core having no gap formed in the legs, and the first tap output terminal and the second tap are provided on the secondary side. And the number of windings from the winding start end to the first tap output terminal is the same as the number of windings from the second tap output terminal to the winding end. With the secondary winding installed,
An insulating converter transformer configured such that the primary winding and the secondary winding are in a tightly coupled state with a coupling coefficient higher than a required value is provided. Then, at least the leakage inductance component of the primary winding of the insulating converter transformer and the capacitance of the primary side series resonance capacitor connected in series to the primary winding are formed, and the operation of the switching means is a current resonance type. The primary side series resonance circuit and the capacitance of a partial resonance capacitor connected in parallel to a predetermined switching element among a plurality of switching elements forming the switching means, and the leakage inductance of the primary winding of the insulating converter transformer. A partial voltage resonance circuit on the primary side that performs a voltage resonance operation during the turn-off period of a plurality of switching elements that form the switching means, and a leakage on the secondary winding of the insulation converter transformer on the secondary side. Inductance component and this secondary winding So as comprising a secondary side partial voltage resonant circuit formed by the capacitance of the secondary side partial voltage resonant capacitor connected in parallel to the. Then, a first DC output voltage generating means configured to input an alternating voltage obtained in the secondary winding of the insulating converter transformer to perform a rectifying operation to generate a first DC output voltage. A center tap is further provided between the first tap output terminal and the second tap output terminal in the secondary winding for grounding, and a pair of rectifying diodes is provided. A second direct current output voltage generating means configured to generate a second direct current output voltage by inputting an alternating voltage obtained between the second tap output terminals and performing a double wave rectification operation, Further, by varying the switching frequency of the switching means according to the level of the first DC output voltage, constant voltage control for the first DC output voltage and the second DC output voltage is performed. And to include a constant voltage control means constituted.

According to each of the above configurations, the secondary winding for obtaining the second DC output voltage is sandwiched by the first secondary windings which are divided so that the number of turns is the same. The first tap output terminal and the second tap which are wound or provided so that the number of turns from the winding start end to the number of turns to the winding end end are the same for the secondary winding. The tap output terminal is formed in a partitioned form. As a result, in the secondary winding for obtaining the second DC output voltage, the alternating voltage from the first secondary winding or another secondary winding portion is uniformly excited with the center tap as a boundary. As a result, the levels of the currents flowing through the respective rectifying diodes provided in the second DC output voltage generating means are made equal.

[0034]

1 shows an example of the configuration of a switching power supply circuit as an embodiment of the present invention. The power supply circuit shown in this figure is provided with a self-excited current resonance type converter as a primary side switching converter.

In the power supply circuit shown in this figure, first,
A full-wave rectifying / smoothing circuit including a bridge rectifying circuit Di and one smoothing capacitor Ci is provided for the commercial AC power supply AC. The full-wave rectification operation of the bridge rectification circuit Di and the smoothing capacitor Ci results in the rectification smoothing voltage Ei (DC input voltage) across the smoothing capacitor Ci. This DC input voltage Ei is
The level corresponds to the same size as the AC input voltage VAC.

The switching converter of the power supply circuit shown in this figure is of a current resonance type, and as shown in the figure, two switching elements Q1 and Q2 are connected by half bridge coupling. In this case, bipolar transistors are selected as the switching elements Q1 and Q2. The base of the switching element Q1 is connected to a self-excited oscillation drive circuit formed by series-connecting a base current limiting resistor RB1-resonance capacitor CB1-drive winding NB1. A damper diode DD1 is connected between the base and emitter of the switching element Q1 in the direction shown. A starting resistor Rs1 is connected between the collector and the base of the switching element Q1 to allow a current at the time of starting to flow to the base. Similarly, for the base of the switching element Q2, the base current limiting resistor RB2-resonance capacitor CB2
-A self-excited oscillation drive circuit formed by connecting drive windings NB2 in series is connected. A damper diode DD2 is connected between the base and the emitter, and a starting resistor Rs2 is connected between the collector and the base.

Here, a series resonance circuit is formed by the capacitance of the resonance capacitor CB1 forming the self-excited oscillation drive circuit on the side of the switching element Q1 and the inductance of the drive winding NB1. Similarly, a series resonance circuit is formed by the capacitance of the resonance capacitor CB2 forming the self-excited oscillation drive circuit on the switching element Q2 side and the inductance of the drive winding NB2. Then, the switching elements Q1 and Q2 are switching-driven by the self-exciting method at the switching frequency determined by the resonance frequency of these series resonant circuits. Further, as will be described later, in the drive transformer PRT, since the alternating voltages of which the drive windings NB1 and NB2 have opposite polarities are excited, the switching element Q1 and NB2 are excited.
Q2 performs switching operation by alternately turning on / off.

Further, the collector of the switching element Q2
A partial resonance capacitor Cp is connected in parallel between the emitters.
Are connected. Depending on the capacitance of the partial resonance capacitor Cp and the leakage inductance component L1 of the primary winding N1, a parallel resonance circuit (partial voltage resonance circuit)
To form. Then, a partial voltage resonance operation is obtained in which the voltage resonates only when the switching elements Q1 and Q2 are turned off.

Drive Transformer PRT (Power Regulati
The ng transformer) is provided for switching-driving the switching elements Q1 and Q2 and for variably controlling the switching frequency for constant voltage control. The drive transformer PRT winds the drive windings NB1 and NB2 and the resonance current detection winding NA, and further, saturable by winding the control winding Nc in a direction orthogonal to each of these windings. It is said to be a reactor. The drive winding N
B1 and drive winding NB2 are adapted to excite voltages having opposite polarities.

Insulation Converter Transformer PIT (Power Is
(Olation Transformer) transmits the switching outputs of the switching elements Q1 and Q2 to the secondary side. The winding end of the primary winding N1 of the insulating transformer PIT is connected to the contact (switching output point) between the emitter of the switching element Q1 and the collector of the switching element Q2 via the resonance current detection winding NA. , A switching output is obtained.

The winding start end of the primary winding N1 is
It is connected to the primary side ground via a series resonance capacitor C1. The capacitance of the series resonance capacitor C1 and the inductance component of the insulating converter transformer PIT including the primary winding N1 form a primary side series resonance circuit for making the operation of the primary side switching converter a current resonance type. . In this way, as the primary side switching converter shown in this figure, the current resonance type operation and the partial voltage resonance operation described above are obtained in a composite manner.

A secondary winding N2 and a tertiary winding N3 having a smaller number of turns (number of turns) than the secondary winding N2 are wound around the secondary side of the insulating converter transformer PIT. In this case, the secondary winding N2 is wound so as to sandwich the tertiary winding N3 as shown in the drawing. The tertiary winding N3 is provided with a center tap, and this center tap is grounded to the secondary side ground. Here, for convenience of explanation, the winding portion on the winding start end side of the secondary winding N2 wound as described above is referred to as "secondary winding N2a", and the winding portion on the winding end end side is referred to as "secondary winding N2a". The secondary winding N2b "is called, and the winding portion on the winding start end side of the portion of the tertiary winding N3 divided by the center tap is" tertiary winding N3a "and the winding end portion side winding portion. "The tertiary winding N3
b ”. In this insulating converter transformer PIT, the number of turns of each winding formed as described above is as follows: primary winding N1 = 30T (turn), secondary winding N2 = 55T, tertiary winding N3 = tertiary winding N3a + The tertiary winding N3b = 6T + 6T.

In the present embodiment, the insulating converter transformer PIT has the structure shown in FIGS. 2 and 3, for example. FIG. 2 is a structural example using a pair of E-shaped cores. The core of the insulating converter transformer PIT in this case is an EE type core in which E type cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs face each other. Then, for the bobbin B which is divided so that the winding areas on the primary side and the secondary side are independent of each other and is integrated, the winding area on the primary side has the primary winding N1 and the winding area on the secondary side. A secondary winding N2 and a tertiary winding N3 are wound around the winding region. In the case of the present embodiment, with respect to the winding region on the secondary side, first, the secondary winding N2 is wound as the secondary winding N2b on the innermost side by half the number of windings. The tertiary winding N3a is wound on the outer side of the secondary winding N3b in this order, and the other half of the secondary winding N2 is wound on the outer side of the tertiary winding N3 (secondary winding N2a). Will be wrapped around. The gap G is not formed in the central magnetic leg of the core as shown in the figure. As a result, the coupling coefficient k is set to a tightly coupled state of, for example, k≈0.95.

FIG. 3 shows an example of a structure using a pair of U-shaped cores. In this case, in the insulating converter transformer PIT, as its core, as shown in FIG. 3, U-shaped cores CR11 and CR12 each having two magnetic legs are combined,
It is adapted to form a U-U type core. Further, with respect to one of the magnetic legs of the U-U type core thus formed, the primary winding N1 is provided on the primary side, and the secondary winding N2 and the tertiary winding N3 are provided on the secondary side. The wound bobbin B is attached in the same manner as in the above case. Also in this case, a gap is not formed in the central magnetic leg of the U-U type core formed as described above, and for example, a tight coupling state with a coupling coefficient k≈0.95 is obtained. ing.

In the power supply circuit of the present embodiment, the insulating converter transformer PIT is not magnetically saturated even if the gap is not formed in this manner because the primary winding N
The magnetic flux density of the ferrite core is increased by increasing the number of windings of the secondary winding N2 with respect to the primary winding N1 by setting the number of windings as described above for the primary winding N2 and the secondary winding N2, respectively. Because it lowered.

On the secondary side of this power supply circuit, a secondary side partial voltage resonance capacitor C2 is connected in parallel with the secondary winding N2 as shown in the figure. A secondary side partial voltage resonance circuit is formed by the capacitance of the secondary partial voltage resonance capacitor C2 and the leakage inductance L2 of the secondary winding N2. Therefore, when an alternating voltage is excited in the secondary winding N2 of the insulating converter transformer PIT, a voltage (partial voltage) resonance operation is obtained on the secondary side as well. That is, the power supply circuit shown in FIG. 1 is configured as a current resonance type converter so that the current resonance operation and the partial voltage resonance operation are obtained on the primary side and the partial voltage resonance operation is obtained on the secondary side. It will be.

Since the resonance operation is also obtained on the secondary side as described above, in the circuit shown in this figure, at the time of an intermediate load, which occurs when the gap is zero as described above. It is possible to prevent abnormal oscillation in.

For the secondary winding N2, a bridge rectifier circuit DBR and a smoothing capacitor C are provided as shown in the figure.
By connecting O1, the secondary side DC output voltage EO1 is obtained across the smoothing capacitor CO1 by the full-wave rectification operation. For the tertiary winding N3,
The anode of the rectifying diode D01 is connected to the end of the tertiary winding N3a side, and the anode of the rectifying diode D02 is connected to the end of the tertiary winding N3b side. The cathode terminals of these rectifying diodes DO1 and DO2 are connected to the positive terminal of the smoothing capacitor CO2, thereby forming a double-wave rectifying circuit, and the DC output voltage EO1 is applied across the smoothing capacitor CO2. Also, a low-voltage secondary side DC output voltage EO2 can be obtained. These secondary side DC output voltages EO1 and EO2 are supplied to loads not shown. In addition, the secondary side DC output voltage EO1 is
Is also branched and input as a detection voltage for.

In the control circuit 1, the secondary side DC output voltage EO1
By varying the level of the control current (DC current) flowing through the control winding NC in accordance with the level change of the control winding NC, the inductance LB of the drive winding NB wound around the orthogonal control transformer PRT is variably controlled. As a result, the resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the main switching element Q1 formed including the inductance LB of the drive winding NB changes. This is an operation to change the switching frequency of the main switching element Q1,
By this operation, the secondary side DC output voltage (EO1, EO2)
Stabilize.

FIG. 6 is a waveform diagram showing the operation of the main part in the power supply circuit of the present embodiment by the switching cycle. The operation waveform shown in this figure is the AC input voltage VAC
= 100V system, the measurement result is shown under the condition of load power Po = 125W. In this figure, the switching element Q2 is turned on during the period TON and the period TO
Switching operation is performed so that it is turned off in FF.
As shown in the figure, the switching current IQ2 flowing through the switching element Q2 is 0 level in the period TOFF, and in the period TON, first, at the start, from the damper diode DD2 through the base → collector of the switching element Q2. A damper current flows in the positive direction of the negative electrode, and then has a waveform flowing through the collector-emitter.

Further, the collector of the switching element Q2
The emitter-to-emitter voltage VQ2 becomes a pulse clamped at the level of the rectified and smoothed voltage Ei (DC input voltage) in the period TOFF, and has a waveform of 0 level in the period TON. The switching element Q1 is switched with respect to the switching element Q2 at the timing of being alternately turned on / off. Therefore, the switching element Q
The switching current and the collector-emitter voltage of 1 have substantially the same waveform shapes as the switching current IQ2 and the collector-emitter voltage VQ2.
The phase is shifted by 80 °.

The primary winding current I1 flowing through the primary winding N1 as a switching output is shown in this figure. The primary winding current I1 is a resonance current obtained by the resonance operation of the series resonance circuit of the primary side series resonance circuit (C1-N1) according to the switching operation of the switching elements Q1 and Q2. Then, this primary winding current I1
The switching element Q1 is off and the switching element Q2 is
During the period TON when the switching element is turned on, the switching element Q
The second switching current IQ2 flows through the switching element Q2. Further, during the period TOFF in which the switching element Q2 is off and the switching element Q1 is on, the switching current of the switching element Q1 flows to the switching element Q1.

As the primary winding current I1 thus flows, the voltage V3 generated in the tertiary winding N3 on the secondary side of the power supply circuit corresponds to the primary winding current I1 as shown in the figure. A periodic waveform can be obtained. Then, as the voltage V3, a waveform whose absolute value level is clamped at the level of the secondary side DC output voltage EO2 is obtained. Also, on the secondary side of this power circuit,
The waveforms of the rectifying current ID1 flowing through the rectifying diode DO1 and the rectifying current ID2 flowing through the rectifying diode DO2 are as shown in the figure. From these waveforms, the rectifying diodes DO1, D
It can be seen that O2 is turned on / off alternately.

In order to obtain the operation shown in FIG. 6 as the power supply circuit shown in FIG. 1, each part is selected as follows. Primary winding N1 = 30T Secondary winding N2 = 55T Tertiary winding N3 = Tertiary winding N3a + Tertiary winding N3b = 6T +
6T Primary side series resonance capacitor C1 = 0.15 μF Primary side partial resonance capacitor Cp = 680 pF Secondary side partial voltage resonance capacitor = 4700 pF

Here, as described above, in the power supply circuit of the present embodiment, the secondary winding N is first on the innermost side of the secondary winding region of the insulating converter transformer PIT.
As the secondary winding 2b, the secondary winding N2 is wound by half the number of windings, and the tertiary windings N3a and N3b are wound on the outer side of the secondary winding N3 in this order. Then, the remaining 1/2 of the secondary winding N2 is wound on the outer side thereof as the secondary winding N2a. As a result, in the power supply circuit of the present embodiment, the transformer coupling between the tertiary winding N3a and the secondary winding N2a and the transformer coupling between the tertiary winding N3b and the secondary winding N2b are made equal. ing.

As a result, the rectifying diodes Do connected to the tertiary winding N3a and the tertiary winding N3b, respectively.
1. Currents of uniform level flow through the rectifier diode D02. That is, the rectified currents ID1, I2 flowing through the rectifier diodes DO1, DO2,
As shown in FIG. 6, the level of D2 is 3.3 Ap each.
Is equal to. Then, as the rectified currents IID1 and ID2 have the same level of current, the cross regulation characteristic for the secondary side DC output voltage EO2 is improved as shown in FIG. become.

FIG. 7 is a diagram for explaining the cross regulation characteristic in the power supply circuit of the present embodiment. In this figure, the secondary side DC output voltage EO2 with respect to the level fluctuations of the load current IO2 (A) on the secondary side DC output voltage EO2 side and the load power Po (W) on the secondary side DC output voltage EO1 side. The change characteristic of (V) is shown. As shown in this figure, in the power supply circuit of the present embodiment, when the load power Po is 0 W and the load current IO2 is 2 A, the level of the secondary side DC output voltage EO2 is 10 V, which is the lowest level. On the other hand, this secondary side DC output voltage EO2 has a load power Po of 100
When the load current IO2 is W and the load current IO2 is 0A, the maximum value is 14.5V. Therefore, in the power supply circuit of the present embodiment, the fluctuation range of the secondary side DC output voltage EO2 with respect to the fluctuations of the load power Po and the load current IO2 is 10V to 14.5V.

On the other hand, as shown in FIG.
In the case of the circuit shown in the prior art, the fluctuation range of the secondary side DC output voltage EO2 under the same condition is from 7.0V to 14.5.
It was V. That is, in the power supply circuit of the present embodiment, the fluctuation range of the secondary side DC output voltage EO2 is reduced by about 3.0 V as compared with the case of the circuit of FIG. 8, and the cross regulation characteristic is reduced accordingly. Improvements are being made.

Here, for the output which is not detected and input to the control circuit 1 such as the secondary side DC output voltage EO2 in the power supply circuit of the present embodiment, this is conventionally made to be a constant voltage. As a means, a local regulator such as a so-called three-terminal series regulator or a step-down converter has been configured. Considering the case where such a local regulator is used for the secondary side DC output voltage EO2, in the power supply circuit of the present embodiment, the fluctuation width of the DC output voltage EO2 is reduced as described above. As a result, the control range of these local regulators is also reduced as compared with the case of the circuit of FIG. That is, in the power supply circuit of the present embodiment,
As described above, when the local regulator is used for the secondary side DC output voltage EO2, the loss in the local regulator can be reduced.

Further, as described above, the rectifying diode D
As the rectified currents ID1 and ID2 flowing through O1 and DO2, respectively, the currents of equal level do not flow, so that the power loss in both the rectifier diodes DO1 and DO2 becomes equal, and the heat generation amounts of both are also equal. Becomes

Further, as described above, the transformer coupling between the tertiary winding N3a and the secondary winding N2a and the transformer coupling between the tertiary winding N3b and the secondary winding N2b become equal, so that the rectified current ID1 , And the rectified current ID2, a sinusoidal waveform is obtained as shown in FIG. Further, as can be seen by comparing FIG. 6 with FIG. 11, which is a waveform diagram of the circuit of FIG. 8, in the power supply circuit of the present embodiment,
At the rectified currents ID1 and ID2, the rectified diode D0
The noise level generated during the turn-off period of 1 and D02 is suppressed. This is because the rectified current ID1 and the rectified current ID2 have sinusoidal waveforms as described above, so that the power loss due to the forward voltage drop and the reverse recovery time becomes equal as shown in the figure. is there.

Here, as a modification of the circuit configuration of the power supply circuit shown in FIG. 1, the configuration on the secondary side is shown in FIG. In FIG. 4, the parts already described in FIG. 1 are omitted. In the configuration shown in FIG. 4, for each winding provided on the secondary side of the insulating converter transformer PIT, in the configuration of FIG. 1, the secondary winding N2 and the tertiary winding N3 are configured as separate windings. The secondary winding N2 is constituted by only one winding.
That is, first, the secondary winding N2 is provided with two tap output terminals as shown in the figure, and the winding part partitioned by these tap output terminals is further center-tapped. Then, the rectifying diodes DO1 and DO are connected to the winding part partitioned by the tap output terminal in this way.
2. A double-wave rectifying / smoothing circuit using a smoothing capacitor CO2 is provided to obtain the secondary side DC output voltage EO2. Further, regarding the secondary side DC output voltage EO1, the rectifier diode DO is applied to the entire secondary winding N2.
3, a double-wave rectification circuit including a rectification diode D04 and a smoothing capacitor C01 is provided.

In the structure of FIG. 4, the secondary winding N
The winding part from the winding start end of 2 to the tap output terminal provided near this winding start end is the "secondary winding N2a", and from this tap to the center tap is the "secondary winding N2a".
3a ", when the center tap to the tap output terminal provided near the end of winding is" secondary winding N3b ", and from this tap to the end of winding" secondary winding N2b ", The number of windings is made equal to that in the case of FIG.

Further, FIG. 5 shows a structural sectional view of the insulating converter transformer PIT in this case. In this modified example also, as shown in the drawing, no gap is formed in the magnetic leg of the core, so that the primary The tight coupling state of about k = 0.95 is obtained for the coupling coefficient between the side and the secondary side. Further, in the insulating converter transformer PIT in this case, as in the case of FIG. 1, with respect to the winding region on the secondary side, from the inside to the secondary winding N2a as shown in the figure.
The secondary winding N2 is wound in the order of → N3a → N3b → N2b. Although only the case of using the EE type core is shown in this figure, the configuration of FIG.
It is also possible to use a UU type core as shown in FIG.

By adopting the configuration shown in FIG. 4 as the switching power supply circuit of the present invention,
It is possible to equalize the transformer coupling between the secondary winding N3a and the secondary winding N2a and the transformer coupling between the secondary winding N3b and the secondary winding N2b. That is, also in the configuration of FIG. 4, the same operation and effect as in FIG. 1 can be obtained.

The switching power supply circuit according to the present invention is not limited to the configuration of each of the embodiments described above, and, for example, the constants of the component elements of the main part can be appropriately set under various conditions. It may be changed to an appropriate value depending on the situation. For example, as the switching element used in the primary side switching converter, in addition to the bipolar transistor shown in FIG.
T or the like may be adopted. Further, when a MOS-FET, an IGBT or the like is adopted, it may be configured to be driven by a separately excited type.

Further, in the present embodiment, the configuration in which only the secondary side DC output voltage EO2 is output as the secondary side low voltage DC output voltage is taken as an example. The output voltage of the voltage may be configured to output DC voltages of a plurality of systems.

[0068]

As described above, according to the present invention, as a basic configuration, a partial voltage resonance circuit is combined with a current resonance type converter on the primary side, and a secondary voltage is used on the secondary side. It constitutes a side partial voltage resonance circuit. And for the isolated converter transformer,
After winding the primary winding and the secondary winding by the required number of turns,
A gap is not formed in the magnetic leg of the core so that the primary side and the secondary side are tightly coupled. Then, with such a basic configuration, the power conversion efficiency is significantly improved as compared with the case of configuring a resonant converter that forms a gap.

Further, in the present invention, the secondary windings (N3a, N3b) for obtaining the second DC output voltage are divided into the first secondary winding (N3a, N3b) having the same number of turns. N2a, N2
Wind it so that it is sandwiched by b). Alternatively, with respect to the secondary winding (N2), a first tap output terminal and a second tap output terminal provided so that the number of turns from the winding start end is equal to the number of turns up to the winding end end. It is formed by partitioning with. As a result, a transformer coupling between the winding N3a in the secondary winding for obtaining the second DC output voltage and the winding N2a in the first secondary winding, and a winding for obtaining the second DC output voltage are provided. The transformer coupling between the winding N3b in the next winding and the winding N2b in the first secondary winding can be made equal.

As described above, since the transformer coupling between the winding N3a and the winding N2a and the transformer coupling between the winding N3b and the winding N2b are equal to each other, the secondary winding for obtaining the second DC output voltage is obtained. In the line, the alternating voltage is evenly excited with the center tap as a boundary. Along with this, the level of the current flowing through each rectifying diode provided in the secondary winding for obtaining the second DC output voltage becomes equal, and as a result, the cross regulation characteristic of the second DC output voltage is obtained. Will be able to improve.

Further, the improvement of the cross regulation characteristic of the second DC output voltage can be achieved in this way. For example, when a local regulator is provided for the second DC output voltage, the control of the local regulator is performed. It becomes possible to reduce the range, and it is possible to reduce the loss in the local regulator thus provided.

Further, by making the current levels flowing through the respective rectifying diodes uniform as described above, it becomes possible to suppress the heat generation which is unevenly generated in one of the rectifying diodes.

Further, as described above, the transformer coupling between the winding N3a and the winding N2a, and the winding N3b and the winding N2b.
When the transformer coupling with and becomes equal, the waveform of the current flowing through each rectifying diode becomes sinusoidal. As a result, in the currents flowing through these rectifying diodes, it is possible to reduce the noise that has occurred during the period in which the respective rectifying diodes are turned off.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing a configuration example of a switching power supply circuit as an embodiment of the present invention.

FIG. 2 is a diagram illustrating a structural example of an insulation converter transformer provided in the power supply circuit of the embodiment.

FIG. 3 is a diagram illustrating a structural example of an insulation converter transformer provided in the power supply circuit of the embodiment.

FIG. 4 is a circuit diagram showing a modified example of the configuration on the secondary side of the switching power supply circuit as the embodiment.

FIG. 5 is a diagram illustrating a structural example of an insulating converter transformer provided in the configuration of the above modification.

FIG. 6 is a waveform diagram showing the operation of the power supply circuit of the embodiment.

FIG. 7 is a diagram illustrating cross regulation characteristics in the power supply circuit according to the embodiment.

FIG. 8 is a circuit diagram showing a configuration example of a switching power supply circuit as a prior art.

9 is a cross-sectional view showing a structural example of an insulating converter transformer used in the power supply circuit shown in FIG.

10 is a diagram illustrating cross regulation characteristics in the power supply circuit shown in FIG.

11 is a waveform chart showing an operation of the power supply circuit shown in FIG.

[Explanation of symbols]

1 Control circuit, Di, DBR bridge rectifier circuit, DO
1, DO2 rectifier diode, CO1, CO2 smoothing capacitor, Q1, Q2 switching element, PIT insulation converter transformer, N1 primary winding, N2 secondary winding, N3
Tertiary winding, C1 primary side series resonance capacitor, Cp primary side partial resonance capacitor, C2 secondary side partial voltage resonance capacitor

Claims (2)

[Claims] 1. A current resonance type switching means for performing a switching operation by interrupting an input DC input voltage; and a switching output of the switching means, the switching output being provided from a primary side to a secondary side. Primary winding, and first
Secondary winding, and a second secondary winding having a smaller number of windings than the first secondary winding, and having no magnetic leg gap formed in the EE type core or U-core. The primary winding is wound on the primary side of the U-shaped core, and the primary winding is mounted on the secondary side.
After winding 1/2 of the secondary winding, the second secondary winding is wound on the outer side, and 1/2 of the first secondary winding is further wound on the outer side. And an insulating converter transformer configured so that the primary side winding and the secondary side winding are in a tightly coupled state with a coupling coefficient higher than a required value, and at least the insulating converter transformer described above. A primary side series resonant circuit formed by a leakage inductance component of the primary winding and a capacitance of a primary side series resonant capacitor connected in series to the primary winding, and wherein the operation of the switching means is a current resonant type; Of the plurality of switching elements forming the switching means, a capacitance of a partial resonance capacitor connected in parallel to a predetermined switching element and one of the insulating converter transformers. A primary side partial voltage resonance circuit formed by a leakage inductance component of the winding and performing a voltage resonance operation during a turn-off period of a plurality of switching elements forming the switching means; and a first secondary winding of the insulation converter transformer. A secondary side partial voltage resonance circuit formed by a leakage inductance component and a capacitance of a secondary side partial voltage resonance capacitor connected in parallel to the first secondary winding; and the first secondary side. A first DC output voltage generating unit configured to generate a first DC output voltage by inputting an alternating voltage obtained in the winding and performing a rectifying operation, and the second secondary winding. On the other hand, by providing a center tap and grounding it and then providing a pair of rectifier diodes, the alternating voltage obtained in this second secondary winding is input to perform both-wave rectification operation, A second direct current output voltage generating means configured to generate a second direct current output voltage; and a variable switching frequency of the switching means according to the level of the first direct current output voltage A constant voltage control means configured to perform constant voltage control on the first DC output voltage and the second DC output voltage, and a switching power supply circuit. 2. A current resonance type switching means for performing a switching operation by interrupting an input DC input voltage, and for transmitting a switching output of the switching means from a primary side to a secondary side. The primary winding is wound on the primary side of the EE type core or the UU type core having no magnetic leg gap, and the first tap output terminal is provided on the secondary side. And a second tap output terminal, the number of turns from the winding start end to the first tap output terminal is the same as the number of turns from the second tap output terminal to the end end. And an insulating converter transformer configured such that the primary winding and the secondary winding are tightly coupled with each other by a coupling coefficient higher than a required value, and at least, Insulation converter A leakage inductance component of the primary winding of the impedance and a capacitance of a primary side series resonant capacitor connected in series to the primary winding, and a primary side series resonant circuit in which the operation of the switching means is a current resonance type, Of the plurality of switching elements forming the switching means, a capacitance of a partial resonance capacitor connected in parallel to a predetermined switching element and a leakage inductance component of the primary winding of the insulating converter transformer, The primary side partial voltage resonance circuit that performs voltage resonance operation during the turn-off period of the plurality of switching elements that form the means, the leakage inductance component of the secondary winding of the insulation converter transformer, and the secondary winding in parallel with the secondary winding. Key of connected secondary side partial voltage resonant capacitor And a secondary side partial voltage resonance circuit formed by a capacitance and an alternating voltage obtained in the secondary winding of the insulation converter transformer are input to perform a rectification operation to generate a first DC output voltage. A configured first direct current output voltage generating means, the first tap output terminal in the secondary winding, and a second
A center tap is further provided between the tap output terminals of, and a pair of rectifying diodes is provided, so that the alternating voltage obtained between the first and second tap output terminals is input and both waves are input. A second DC output voltage generating unit configured to generate a second DC output voltage by performing a rectifying operation, and a switching frequency of the switching unit according to a level of the first DC output voltage. A switching power supply circuit comprising: a constant voltage control unit configured to perform constant voltage control on the first DC output voltage and the second DC output voltage by varying the constant voltage control unit.