JP2002034280A - Brushless DC motor control method and device - Google Patents

Abstract

(57) [Problem] To drive a brushless DC motor without detecting a magnetic pole position angle. SOLUTION: An output waveform from an inverter unit 1 is supplied to a brushless DC motor 2, a voltage, a current or a magnetic flux of the brushless DC motor 2 is detected, and a detection signal is supplied to a phase reference generating unit 3 to determine a phase reference. A waveform control signal is output from the waveform control unit 4 based on the generated phase reference and a phase command given from the outside, and the inverter unit 1 is controlled to set the output waveform phase to a predetermined value.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a brushless DC motor control method and an apparatus for supplying an output of an inverter to a brushless DC motor.

[0002]

2. Description of the Related Art A brushless DC motor in which a permanent magnet is mounted on a rotor has higher efficiency than a synchronous motor and is widely used in applications where energy-saving operation is first desired. As the control circuit configuration, as shown in FIG. 20, an encoder 73 for detecting a rotor position (magnetic pole position) of the brushless DC motor 72 and a magnetic pole position are detected based on an output signal from the encoder 73. A general configuration includes a magnetic pole position detection unit 74 and a waveform control unit 75 that controls the inverter unit 71 to output an inverter output waveform synchronized with the rotor position angle (magnetic pole position angle) based on the magnetic pole position detection signal. Has been adopted.

In applications where it is difficult to incorporate an encoder in terms of reliability and cost (for example, applications as a drive source for a compressor used in an air conditioner, a refrigerator, etc.), a magnet voltage is derived from the terminal voltage of a brushless DC motor. A method of detecting a back electromotive voltage and indirectly detecting a rotor position angle from the detected back electromotive voltage is employed. Note that this method
(1) A method of detecting a zero cross of a fundamental wave of a back electromotive voltage to generate a magnetic pole position angle signal (see Japanese Patent Application Laid-Open No. Hei 5-276782); (See Japanese Patent Application Laid-Open No. 7-828328).

[0004]

When the method (1) is adopted, a predetermined energization OFF period (a period during which no current flows) is provided for each phase, and the terminal voltage of the brushless DC motor must be detected. Therefore, the energization waveform is restricted, and it becomes impossible to drive the brushless DC motor with a continuous energization waveform (for example, a sine wave).

When the method (2) is adopted, the magnetic pole position can be detected without depending on the current, so that the restriction on the current waveform as in the method (1) is eliminated. It becomes impossible to apply to a motor that does not include the third harmonic.

That is, in order to reduce the harmonic loss and noise of the motor, it is preferable to drive the motor with a sine wave or trapezoidal wave in which the current changes smoothly. However, in the method (1), it is not possible to detect the magnetic pole position. This makes it impossible to drive a brushless DC motor, and the method (2) limits the motors that can be used, making it impossible to use a desired motor.

[0007]

SUMMARY OF THE INVENTION The present invention has been made in view of the above-mentioned problems, and has been made in consideration of the above-described problems, and is intended to provide a brushless DC motor capable of driving a brushless DC motor without detecting a magnetic pole position angle.
An object of the present invention is to provide a motor control method and a device thereof.

[0008]

A brushless D according to claim 1
The C motor control method is a method of generating a reference signal that is not restricted by the rotor position angle and controlling the inverter to set the inverter output waveform phase based on the generated reference signal.

A brushless DC motor control method according to a second aspect of the present invention generates a reference signal not restricted by the rotor position angle, detects a load state, and outputs an inverter output based on the generated reference signal and the detected load state. This is a method of controlling the inverter to set the waveform phase.

A brushless DC motor control method according to a third aspect is a method of detecting a primary magnetic flux and generating a reference signal that is not restricted by a rotational position angle. Here and in this specification, the “primary magnetic flux” means the one obtained by integrating the inverter terminal voltage.

A brushless DC motor control method according to a fourth aspect is a method for detecting a load state based on the number of revolutions of the brushless DC motor.

A brushless DC motor control method according to a fifth aspect is a method of detecting a primary magnetic flux and directly controlling an inverter with the detected primary magnetic flux. Here and in the following, “directly control the inverter” means controlling the inverter without performing a conversion process from a primary magnetic flux to a position or a position calculation process.

According to a sixth aspect of the present invention, there is provided a brushless DC motor control device comprising: a reference signal generating means for generating a reference signal not restricted by a rotor position angle; and an inverter for setting an inverter output waveform phase based on the generated reference signal. And an inverter control means for controlling.

According to a seventh aspect of the present invention, there is provided a brushless DC motor control device, wherein a reference signal generating means for generating a reference signal not restricted by the rotor position angle, a load state detecting means for detecting a load state; And inverter control means for controlling the inverter so as to set the inverter output waveform phase based on the load state thus set.

According to an eighth aspect of the present invention, the brushless DC motor control device employs, as the reference signal generating means, a device which detects a primary magnetic flux and generates a reference signal which is not restricted by a rotational position angle.

According to a ninth aspect of the present invention, in the brushless DC motor control device, the load state detecting means detects a load state based on a rotation speed of the brushless DC motor.

According to a tenth aspect of the present invention, there is provided a brushless DC motor control device including primary magnetic flux detecting means for detecting a primary magnetic flux, and inverter controlling means for directly controlling an inverter based on the detected primary magnetic flux.

[0018]

According to the brushless DC motor control method of the present invention, when the output of the inverter is supplied to control the brushless DC motor, a reference signal not restricted by the rotor position angle is generated, and the generated reference signal is generated. Since the inverter is controlled so as to set the inverter output waveform phase on the basis of this, it is possible to control the brushless DC motor without providing a power-off period, thereby improving the efficiency and reducing the noise, and furthermore, the back electromotive voltage. The present invention can also be applied to a brushless DC motor that does not include the third harmonic.

According to the brushless DC motor control method of the second aspect, the output of the inverter is supplied and the brushless DC motor is controlled.
In controlling the motor, a reference signal not restricted by the rotor position angle is generated, and a load state is detected.
Since the inverter is controlled to set the inverter output waveform phase based on the generated reference signal and the detected load state, the brushless DC motor is controlled in consideration of the load state without providing a power-off period. Improvement in efficiency and reduction in noise can be achieved, and the present invention can be applied to a brushless DC motor in which the back electromotive force does not include the third harmonic.

According to the brushless DC motor control method of the third aspect, since the primary magnetic flux is detected and the reference signal not generated by the rotation position angle is generated, the reference signal can be easily generated. The same operation as the first or second aspect can be achieved.

According to the brushless DC motor control method of claim 4, since the load state is detected based on the rotation speed of the brushless DC motor, the load state can be easily and accurately detected. The same operation as that of the item 2 can be achieved.

According to the brushless DC motor control method of the fifth aspect, the output of the inverter is supplied and the brushless DC motor is controlled.
In controlling the motor, the primary magnetic flux is detected, and the inverter is directly controlled by the detected primary magnetic flux. Therefore, the brushless DC motor is controlled without providing a power-off period to improve efficiency and reduce noise. The present invention can be applied to a brushless DC motor that does not include the third harmonic in the back electromotive voltage.

According to the brushless DC motor control device of the sixth aspect, the output of the inverter is supplied to the brushless DC motor control device.
In controlling the motor, a reference signal generation unit generates a reference signal that is not restricted by the rotor position angle, and the inverter control unit controls the inverter to set an inverter output waveform phase based on the generated reference signal. Can be.

Therefore, the efficiency can be improved and the noise can be reduced by controlling the brushless DC motor without providing the power-off period, and the brushless DC motor which does not include the third harmonic in the back electromotive voltage can be realized. Can be applied.

According to the brushless DC motor control device of the present invention, the output of the inverter is supplied and the brushless DC motor is controlled.
In controlling the motor, the reference signal generating means generates a reference signal not restricted by the rotor position angle, the load state detecting means detects the load state, and the inverter control means generates the generated reference signal and the detected load. The inverter can be controlled to set the inverter output waveform phase based on the state.

Therefore, the efficiency can be improved and the noise can be reduced by controlling the brushless DC motor in consideration of the load state without providing the power-off period, and the third harmonic is included in the back electromotive voltage. It can also be applied to brushless DC motors without.

According to the brushless DC motor control device of the present invention, the reference signal generating means which detects the primary magnetic flux and generates a reference signal which is not restricted by the rotational position angle is adopted. The signal can be easily generated, and the same operation as that of claim 6 or 7 can be achieved.

In the brushless DC motor control device according to the ninth aspect, a brushless DC motor may be used as the load state detecting means.
Since the load state is detected based on the number of rotations of the C motor, the load state can be easily and accurately detected, and the same operation as in claim 7 can be achieved.

According to the brushless DC motor control apparatus of the tenth aspect, the output of the inverter is supplied to the brushless DC motor control apparatus.
In controlling the C motor, a primary magnetic flux is detected by a primary magnetic flux detecting means, and the primary magnetic flux is detected by an inverter controlling means.
The inverter can be directly controlled by the detected primary magnetic flux.

Therefore, the efficiency can be improved and the noise can be reduced by controlling the brushless DC motor without providing the power-off period, and the brushless DC motor which does not include the third harmonic in the back electromotive voltage can be achieved. Can be applied.

Further description will be given.

Brushless D generally captured in dq coordinate system
The voltage equation of the C motor can be expressed by Equation 1 when the d-axis is set to match the direction of the magnetic flux of the permanent magnet.

[0033]

(Equation 1)

Here, R [Ω] is the winding resistance and λa [w
b] is the number of permanent magnet magnetic fluxes linked to the winding, ω [rad /
s] is the rotational angular velocity, n is the number of motor pole pairs, Ld [H]
Represents a d-axis inductance, and Lq [H] represents a q-axis inductance. Here, p is a differential operator.

The voltages in the dq coordinate system are represented by vd, vq,
If the current is represented by id and iq, these quantities are 3
It can be converted to a phase coordinate system (uvw).

[0036]

(Equation 2)

Note that θe represents the rotor position angle [rad] as an electrical angle (electrical angle = number of pole pairs × mechanical angle).

Assuming that the amplitude of the three-phase alternating current flowing through the brushless DC motor is i and the current phase with respect to the q axis is β, iq = i · cosβ id = −i · sinβ. Then, the instantaneous torque τ of the brushless DC motor
Is τ = n · λa · i · cos β− (1/2) · Lv · i ²
・ Sin2β Here, Lv = Ld-Lq.

Further, the steady state where the differential term is 0 is considered, ignoring the resistance in Equation 1, and the relationship of Equation 3 is obtained from this and the above-described current equation and instantaneous torque equation.

[0040]

[Equation 3]

Focusing on the instantaneous torque equation and the equation (3), the motor torque is represented by the current amplitude i and the phase β,
Alternatively, it can be seen that it is determined by the voltage amplitude v and the phase δ.

That is, in order to control the motor torque, the conventional brushless DC motor drive system controls the current or the voltage phase based on the rotor position angle signal.

In addition, in servo applications such as FA (factory automation), the servo applied voltage is accurately calculated based on the equation (1) so that a transient phenomenon does not occur in the current and a high-speed response can be obtained. For this purpose, the rotor position angle information has been used.

FIGS. 1 and 2 schematically show the phase relationship between the magnetic flux, the voltage and the current, and the force acting between the stator magnetic flux and the rotor generated by energization.

According to the present invention, there is provided means for determining the phase of the inverter output waveform based on a reference signal not restricted by the rotor position angle, and based on this phase command or responding this phase command to the motor load torque. By changing it, the brushless DC motor is driven. Therefore, it is not necessary to directly or indirectly detect the rotor position angle.

Further description will be given.

In a brushless DC motor in which a permanent magnet is mounted on the surface of a rotor (hereinafter referred to as a brushless DC motor having a surface magnet structure), Lv = 0 because of its magnetic structure. Does not generate the torque of the second term (hereinafter, referred to as reluctance torque), but generates only the torque of the first term (hereinafter, referred to as magnet torque).

On the other hand, a brushless DC motor having a permanent magnet mounted inside a rotor (hereinafter referred to as a brushless DC motor having an embedded magnet structure) generates magnet torque and reluctance torque. As a high-efficiency motor, it is widely used in devices that require energy saving. Therefore, the operation of the brushless DC motor having an embedded magnet structure will be described as an example. Table 1 shows the electrical parameters of the brushless DC motor used.

[0049]

[Table 1]

A brushless DC motor having various electric quantities shown in Table 1 was employed, and the output current effective value of the inverter was 5
[A], the motor rotation speed was set to 30 [rps], and the torque when the current phase was changed was calculated from the equation representing the instantaneous torque. As a result, the result shown in FIG. 3 was obtained.

Here, the inverter connected to drive the brushless DC motor does not feed back the rotor position angle information, that is, does not perform phase control based on the position angle information, and has a predetermined frequency. Assuming that an AC current having an amplitude is output, the brushless DC motor operates at the operating point A in synchronization with the output waveform of the inverter.
(See FIG. 3).

In this state, when the load torque increases, the rotation speed becomes lower than the inverter output frequency. In this case, the dq coordinate axis changes in the direction shown by the broken line in FIG. 1, and the inverter current phase β with respect to the back electromotive voltage increases, and the torque increases. Then, it moves to the operating point B (see FIG. 3) equal to the load torque, and continues to rotate at the synchronous speed. That is, the state where the inverter output frequency is equal to the frequency obtained by multiplying the motor rotation speed by the number of pole pairs is maintained.

Conversely, when the load torque decreases, the reverse trajectory may be considered.

This is because the brushless DC motor itself has a function of absorbing the torque mismatch and adjusting the phase if the load torque changes within a certain range. The output waveform and the rotor position angle are controlled on the inverter control side. This indicates that even if the phase is not precisely controlled, the rotation of the brushless DC motor does not become unstable or stop immediately, but can be continued.

This can be easily and qualitatively understood by assuming a "spring" effect on the electromagnetic force acting between the magnetic flux of the stator and the rotor magnet as shown in FIG. .

Therefore, the brushless DC motor can be controlled by appropriately changing and optimizing the phase command that is not restricted by the rotor position in response to the load state of the brushless DC motor, that is, the torque or the number of revolutions. It turns out that it is.

Further, a brushless DC motor having various electric quantities shown in Table 1 is employed, the output voltage amplitude of the inverter is fixed to be equal to the back electromotive voltage e, and the motor speed is set to 30 [r].
ps], and the torque when the voltage phase was changed was calculated from Equation 3, and the result shown in FIG. 4 was obtained.

Consider the case where the brushless DC motor is rotating at the operating point A (see FIG. 4) as described above.

In this state, when the load torque increases, the rotation speed becomes lower than the inverter output frequency. In this case, the dq coordinate axis changes in the direction indicated by the broken line in FIG. 1, and the inverter current phase δ with respect to the back electromotive voltage increases, and the torque increases. Then, it moves to the operating point B (see FIG. 4) equal to the load torque, and continues to rotate.

Conversely, when the load torque decreases, the reverse trajectory may be considered.

That is, similarly to the case where the current is controlled,
It can be seen that stable rotation control can be realized if the load torque changes within a certain range.

FIG. 5 is a block diagram showing the configuration of the brushless DC motor control device according to the present invention.

This brushless DC motor control device converts the output waveform from the inverter unit 1 into a brushless DC motor 2
To detect the voltage, current, or magnetic flux of the brushless DC motor 2, and supply a detection signal to the phase reference generator 3 to generate a phase reference, based on the generated phase reference and an externally applied phase command. Thus, a waveform control signal is output from the waveform control unit 4 to control the inverter unit 1 so that the output waveform phase becomes a predetermined value.

Therefore, even after the voltage amplitude is saturated (fixed) to the maximum value, the weak magnetic flux control of the motor by the phase control (the number of magnet magnetic fluxes linked to the winding is equivalently reduced by the d-axis current). Control) can be performed, and the rotation speed range (constant output range) can be expanded. In particular, brushless DC motors with an embedded magnet structure have a greater magnetic flux weakening effect than brushless DC motors with a surface magnet structure, and can further expand the motor operating range (Takeda et al., “PM motor equipment constants and output ranges”, Academic Theory D, Vol. 110, No. 11, 1990).

[0065]

BRIEF DESCRIPTION OF THE DRAWINGS FIG. 1 is a block diagram of a brushless DC motor control method according to an embodiment of the present invention;

For convenience of description, unless otherwise noted, the “primary magnetic flux” will be simply referred to as “magnetic flux” in the following description.

FIG. 6 is an electric circuit diagram showing one embodiment of the brushless DC motor control device of the present invention, and FIG. 7 is a block diagram for explaining internal processing of the control microcomputer of FIG.

In this brushless DC motor control device, the switching command output from the control microcomputer 5 is supplied through the base drive circuit 11 to the voltage source inverter 1
2, a resistor circuit 13 Y-connected to output terminals for three phases of the voltage-source inverter unit 12, an integrator 14 that performs an integration operation by using output voltages of each phase of the voltage-source inverter unit 12 as inputs, A comparator 15 for comparing the magnitudes of the integrated outputs from two integrators 14, a photocoupler 16 to which the comparison output from each comparator 15 is input, and a magnetic flux zero crossing to which the outputs from all the photocouplers 16 are input. And a magnetic flux zero-cross signal generation unit 17 that generates a signal and supplies the signal to an external interrupt terminal of the control microcomputer 5.

Each of the integrators 14 has a resistor connected in series between the output terminal of each phase of the voltage source inverter 12 and the inverting input terminal of the operational amplifier, and has a resistor and a capacitor between the inverting input terminal and the output terminal. Are connected in parallel with each other, and a non-inverting input terminal is connected to a neutral point of the resistance circuit 13. However, in the frequency band for driving the brushless DC motor 18, by setting the values of the resistor and the capacitor so as to operate as a complete integrator, it can be considered as a complete integrator.

The magnetic flux zero-cross signal generation unit 17 includes three first NAND gates, each of which receives two of the outputs from all the photocouplers 16, and all the first NAs.
A second NAND gate which receives an output from the ND gate as an input. Then, the logic truth table when the outputs from all the photocouplers 16 are Su, Sv, and Sw and the output from the second NAND gate is Sφ is set as shown in Table 2.

[0071]

[Table 2]

The operation of the brushless DC motor control device shown in FIG. 6 is as follows.

If the output potentials of the respective phases of the voltage source inverter unit 12 are expressed as vu ′, vv ′, vw ′, the potential vn ′ at the neutral point of the resistance circuit 13 is vn ′ = (vu ′ + vv ′ + vw ′). ) / 3, and the output waveform of the inverter 12 fluctuates with the 3n-order harmonic component in which each phase becomes the same phase. Since one input terminal (non-inverting input) of the operational amplifier constituted by the differential input is connected to this neutral point, the output of the operational amplifier connected to each phase output is a main low-order harmonic component, which is 3 It is obtained as the integral value of the phase voltage from which the subharmonic has been removed. In other words, the time integral of the fundamental wave voltage (the number of winding flux linkages) can be detected without distortion.

Then, the detected magnetic flux waveform of each phase is input to the comparator 15. For example, the u-phase magnetic flux is supplied to the non-inverting input terminal of the comparator Quw, and the w-phase magnetic flux is supplied to the inverting input terminal of the comparator Quw. Therefore, this comparator Quw
Outputs a rectangular wave that is inverted at the intersection of each phase magnetic flux waveform (equivalent to the zero cross of the integrated waveform of the uw line voltage).
Similarly, other comparators output rectangular waves that are inverted at the intersections of the magnetic flux waveforms of the respective phases. By performing these comparison processes, the in-phase noise component superimposed on the detected magnetic flux waveform can be removed, and a stable comparator output waveform can be output.

The three comparator outputs obtained in this manner are supplied to a magnetic flux zero cross signal generation unit 17 and perform a logical operation shown in a logic truth table of Table 2 to generate a 1-bit magnetic flux zero cross signal Sφ. It is supplied to the external interrupt terminal of the control microcomputer 5.

The control microcomputer 5 reads the timer values of the capture unit 51, the index calculation unit 52, the carrier interrupt timer 53, the free-run timer 54, and the free-run timer 54, which operate using the magnetic flux zero cross signal Sφ as an external interrupt signal. A previous timer value storage unit 55 that reads through the capture unit 51 and stores it as a previous timer value, and a period calculation unit that performs a period calculation using the timer value of the free-run timer 54 and the previous timer value read through the capture unit 51 as inputs. 56, a speed calculating unit 57 for performing a speed calculation from the cycle calculation result, and a speed for performing a speed control calculation (for example, PI calculation) using the calculated speed and a speed command given from the outside to output an amplitude command. Control unit 58, calculated speed, externally applied phase command α *, free The timer value of the run timer 54 and the index value I output from the index calculator 52
x and the previous timer value as input, perform phase calculation,
A phase calculator 59 for outputting a voltage phase, and a speed controller 58
Pulse width calculation unit 50 that performs a pulse width calculation using the amplitude command output from the controller and the phase output from the phase calculation unit 59 as an input, and a three-phase PWM (pulse width modulation) that outputs a switching command using the pulse width calculation result as an input ) Control timer 60;

The processing in the phase calculation section 59 and the pulse width calculation section 50 is performed in response to an interrupt signal from the carrier interrupt timer 53.

Next, the processing in the control microcomputer 5 will be described with reference to the flowcharts of FIGS.

The interrupt processing of FIG. 8 is started with the magnetic flux zero cross signal as a trigger signal, and in step SP1,
Timer value T of free-run timer 54 at the start of processing
0 is read into the capture unit 51, and in step SP2, the cycle T (= T1−T0) is calculated by the cycle calculation unit 56 based on the read timer value T0 and the previous timer value T1 stored in the previous timer value storage unit 55. ) Is calculated, and in step SP3, the timer value T0 is stored in the previous timer value storage unit 55 as a new previous timer value T1. Then, in step SP4, the actual speed ω is calculated from the calculated period T, and in step SP5, the index value Ix for the phase calculation is incremented by one. Then, in step SP6, it is determined whether or not the index value Ix is greater than 6, and if it is determined that the index value Ix is greater than 6, the index value Ix is set to 0 in step SP7.
Set to.

If it is determined in step SP6 that the index value Ix is not larger than 6, or if the processing in step SP7 is performed, in step SP8, the speed control calculation is performed, and the original processing is performed. Return to

FIG. 9 is a flowchart for explaining the processing in step SP8 of FIG. 8 in detail.

At step SP1, the actual speed ω and the speed command ω * are input. At step SP2, the speed deviation Δω (= ω * −ω) is calculated. At step SP3, the speed deviation Δω is calculated for each speed control calculation. , The result is multiplied by an integral gain Ki to obtain an integral term, and a proportional term obtained by multiplying a speed gain Δω by a proportional gain Kp and an integral term are added (by performing PI calculation. ), Is supplied to the pulse width calculation unit as the voltage amplitude command V *, and returns to the original processing as it is.

FIG. 10 is a flowchart for explaining a carrier interrupt process for controlling the inverter waveform. The processing of this flowchart starts with a trigger signal output from the carrier interrupt timer at a predetermined cycle.

At step SP1, a phase command α * given from the outside, an index value Ix updated by an external interrupt process, a previous timer value T1, and a voltage amplitude command V * are input. The timer value t is read, and in step SP3, the power supply phase φ ｛= 60 ° · Ix + ω · (t−T1)
+ Α *｝ is calculated, and in step SP4, the pulse width of each phase (on / off timing of the transistor of the inverter unit) is calculated from the power supply phase φ (details will be described later). In step SP5, the switching pattern for each phase transistor is calculated. Is initialized, the respective pulse widths are set in the PWM timer corresponding to each phase, and the processing returns to the original processing.

FIG. 11 is a flowchart for explaining the PWM timer process per phase, which is executed in response to the time set in the PWM timer.

At step SP1, the transistor signal is inverted and the process returns to the original processing.

Next, the pulse width calculation will be described.

For all three-phase AC motors, including brushless DC motors, the terminal voltage to be applied to the motor winding is ideally a sine wave AC voltage.

Here, the three-phase AC voltage is represented by Equation 4.

[0090]

(Equation 4)

Then, this is projected on a complex plane by Expression 5 for convenience.

[0092]

(Equation 5)

Here, exp (jθ) = cos θ + j ·
sin θ, j is an imaginary number.

Then, λ = ∫vp · dt = j (V / ω) · exp (−j · ω ·
t), and converted to a time product (magnetic flux) locus, and drawn on a complex plane, becomes a circular locus having a radius of V / ω and a tangential velocity of V (FIG. 1).
2).

On the other hand, when the inverter terminal voltage is based on the negative side of the inverter DC section, there are seven states as shown in Table 3 depending on the ON state of the transistor.

[0096]

[Table 3]

A transistor whose collector terminal is connected to the + side of the inverter DC unit (hereinafter referred to as an upper arm transistor) is “1”, and a transistor whose emitter terminal is connected to the − side of the inverter DC unit (hereinafter referred to as “upper arm transistor”). The state where the lower arm transistor is on)
0 ". Since the function of the voltage source inverter exclusively controls the upper arm transistor and the lower arm transistor, it is not necessary to consider switch states other than those in Table 3. For convenience, seven types of switches are used. Switching states are shown in correspondence with symbols of voltage vectors V0 to V7.

Here, when each voltage vector is drawn on a complex plane according to Equation 5, as shown in FIG. 13, six types of vectors of length Vdc (= voltage of the inverter DC section) arranged at every 60 ° are provided. And two kinds of zero vectors having no length.

Here, consider the output time of each voltage vector such that the magnetic flux locus based on the output waveform of the inverter approaches the circular locus based on the sinusoidal voltage.

FIG. 14 is an enlarged view of the sine wave voltage phase ω · t in the range of 0 ° to 60 °.

Focusing on the triangle P0qP1, the side P0P
1 can be obtained as the product of the tangential velocity V of the circular locus generated by the sine wave voltage and the carrier period Tc, provided that the PWM carrier period Tc is sufficiently short. Also, ∠qP
Since 0P1 = ω · t and ∠P0qP1 = 120 °, Equation 6
Get the relationship.

[0102]

(Equation 6)

A voltage phase of 0 ° to 60 ° is defined as a region I, and one cycle of the voltage phase is divided into six regions as shown in FIG. 15, and voltage vectors used in each region are selected as shown in Table 4, thereby obtaining a circle. A polygonal locus close to the locus can be obtained.

[0104]

[Table 4]

For example, V4 in region I is called a main vector and V6 is called a subvector. Table 4 shows voltage vectors to be used as a main vector and a subvector for each region.

Then, the output time of each vector is expressed as: zero vector output time τ0 / T = 1−Ks · sin (π
/ 3 + ψ0) Main vector output time τmain / T = Ks · sin
(Π / 3−ψ0) Subvector output time τsub / T = Ks · sinψ0 where Ks = (21/2 · V) / Vdc (0 ≦ Ks ≦
1), ψ0 = MOD (φ / 60 °), MOD is a function that takes a remainder. It becomes.

FIG. 16 shows a case of starting from V0 {FIG.
6 (a)} and starting from V7 {FIG.
For each of the middle (b)｝, the voltage vector for each of the regions I and II is defined in order to clarify the initial value of the transistor signal of each phase and the period to be set in the PWM timer of each phase. The example is expanded to the on / off state of the transistor.

In the region I starting from V0, the transistor signal of each phase is initialized to “0” for each carrier cycle, τ0 for the u-phase timer, and τ0 + τma for the v-phase timer.
In may be set, and the transistor signal may be inverted at each time. Tc is set in the w-phase timer so that switching is not performed.

In the region II starting from V7, the transistor signal of each phase is set to "1" every carrier cycle.
Τ0 for the w-phase timer and τ0 for the u-phase timer.
+ Τmain may be set, and the transistor signal may be inverted at each time, and Tc may be set in the v-phase timer to prevent switching.

By expanding the other regions in the same manner, the timer value to be set and the initial value of the transistor signal to be set for each carrier can be known.

Table 5 shows the initial value (indicated by a voltage vector) of the transistor signal at the start time of the carrier and the timer value set in the PWM timer for each area.

[0112]

[Table 5]

FIG. 17 is a diagram showing a part of operation waveforms of the brushless DC motor control device of FIG. In the brushless DC motor control device of FIG. 6, since the difference of the magnetic flux is obtained at the comparator input unit, the difference waveforms λuv, λvw, λwu of each phase magnetic flux detection are shown for convenience.

Here, the case where the inverter output voltage V is constant is indicated by a thick solid line, and the case where the inverter output voltage changes is indicated by a thin solid line.

Hereinafter, how the motor speed, that is, the inverter output frequency is controlled will be described with reference to FIG.

In the brushless DC motor control device of FIG. 6, the speed information is obtained by measuring the zero-cross period of the magnetic flux waveform of FIG.

When the inverter output voltage V is constant, the zero-cross points are point A, point B, point C, and point D, and the cycle information in this case is constant.

Here, when a speed increase command is given, the inverter output voltage V increases by speed control calculation. As a result,
The magnetic flux waveform, which is an integral waveform of the voltage, follows a locus of a thin solid line that changes faster than at a constant voltage, and crosses zero at point E. Then, the speed calculated from the zero-cross period also increases, and this is reflected in the calculation of the voltage phase φ, so that the output frequency of the inverter increases.

On the other hand, in the case of the deceleration command, the trajectory of the magnetic flux waveform becomes a trajectory of a thin solid line that crosses zero at point F, and the detected speed is reduced. The output frequency decreases.

In both the acceleration and deceleration, a deviation between the speed detected from the zero cross of the magnetic flux waveform and the speed command is obtained, and the PI is calculated by feedback control, so that steady operation is performed at a point of a predetermined voltage amplitude V and frequency ω. (Equal to the speed command ω *).

FIG. 18 is a view showing a result of experimentally determining a phase command value at which the motor can be operated (no step-out or the like) by changing the rotation speed in the brushless DC motor control device of FIG.

Therefore, the relationship between the rotation speed and the phase command can be tabulated in advance.

FIG. 19 is a block diagram showing an example of a configuration for controlling a brushless DC motor by incorporating a table showing the relationship between the rotation speed and the phase command.

This brushless DC motor control device converts the output voltage from the inverter 1 into a brushless DC motor 2
To detect the voltage of the brushless DC motor 2 and supply it to the magnetic flux zero-cross generator 3 ′, and supply the magnetic flux zero-cross detection signal output from the magnetic flux zero-cross generator 3 ′ to the waveform controller 4 and the speed detector 6. are doing. Then, the speed (rotation speed) detected by the speed detection unit 6 is supplied to the phase table unit 7, a phase command corresponding to the rotation speed is read, and supplied to the waveform control unit 4.

The waveform controller 4 performs a waveform control operation based on the magnetic flux zero cross detection signal and the phase command, outputs a switching command, and supplies the switching command to the inverter 1.

Therefore, in this case, the brushless DC motor can be controlled without externally giving a phase command.

In the above embodiment, the zero cross point of the magnetic flux waveform is used to generate the phase reference. For example, when a voltage type inverter is used, the zero cross point of the inverter output current is detected and the phase is detected. A reference can be generated, and when a current-source inverter is used, a phase reference can be generated by detecting a zero-cross point of the inverter output voltage. It is also possible to generate a phase reference by detecting a point (for example, a peak point) other than the zero cross point.

Further, the inverter output terminal voltage is integrated, and
Although the configuration is such that the magnetic flux waveform is obtained, in order to accurately detect the magnetic flux generated in the stator, instead of integrating the voltage obtained by subtracting the voltage drop at the winding resistance from the output voltage of each phase of the inverter, Is also good.

As described above, the brushless D according to the present invention
If the C motor control method and its device are adopted, it is not necessary to mount a sensor (for example, a rotary encoder or the like) for directly detecting the magnetic pole position on the rotor, and furthermore, the magnetic pole position is indirectly detected. For the inverter output waveform (for example, to provide a non-energized (off) period when the output terminal of the voltage source inverter is open) is unnecessary, and a preferable waveform from the viewpoint of reducing noise and improving efficiency of the brushless DC motor. Can be adopted without detecting the magnetic pole position.

[0130]

According to the first aspect of the present invention, the efficiency can be improved and the noise can be reduced by controlling the brushless DC motor without providing the power-off period. It has a unique effect that it can be applied to a brushless DC motor that does not include it.

According to the second aspect of the present invention, it is possible to improve the efficiency and reduce the noise by controlling the brushless DC motor in consideration of the load state without providing the power-off period. Brushless D without subharmonic
This has a specific effect that it can be applied to the C motor.

According to the third aspect of the invention, the reference signal can be easily generated, and the same effects as those of the first or second aspect can be obtained.

According to the fourth aspect of the present invention, the load state can be easily and accurately detected, and the same effects as those of the second aspect can be obtained.

According to the fifth aspect of the present invention, the brushless DC motor can be controlled without providing a power-off period to improve the efficiency and reduce the noise, and the back electromotive voltage does not include the third harmonic. It has a unique effect that it can be applied to a brushless DC motor.

According to the sixth aspect of the present invention, it is possible to control the brushless DC motor without providing a power-off period, thereby improving the efficiency and reducing the noise, and furthermore, the back electromotive voltage does not include the third harmonic. It has a unique effect that it can be applied to a brushless DC motor.

According to the seventh aspect of the present invention, it is possible to control the brushless DC motor in consideration of the load state without providing a power-off period, thereby improving the efficiency and reducing the noise. Brushless D without subharmonic
This has a specific effect that it can be applied to the C motor.

According to the eighth aspect of the invention, the reference signal can be easily generated, and the same effect as that of the sixth or seventh aspect can be obtained.

According to the ninth aspect of the present invention, the load state can be easily and accurately detected, and the same effects as those of the seventh aspect can be obtained.

According to the tenth aspect of the present invention, it is possible to control the brushless DC motor without providing a power-off period, thereby improving the efficiency and reducing the noise, and furthermore, the back electromotive voltage does not include the third harmonic. It has a unique effect that it can be applied to a brushless DC motor.

[Brief description of the drawings]

FIG. 1 is a diagram showing a phase relationship among a magnetic flux, a voltage, and a current.

FIG. 2 is a diagram schematically showing a force acting between a stator magnetic flux and a rotor generated by energization.

FIG. 3 is a diagram showing a current phase-torque characteristic.

FIG. 4 is a diagram showing a voltage phase-torque characteristic.

FIG. 5 is a block diagram showing a configuration of a brushless DC motor control device according to the present invention.

FIG. 6 is an electric circuit diagram showing one embodiment of the brushless DC motor control device of the present invention.

FIG. 7 is a block diagram illustrating internal processing of the control microcomputer of FIG. 6;

FIG. 8 is a flowchart illustrating an interrupt process using a magnetic flux zero cross signal as a trigger signal.

FIG. 9 is a flowchart illustrating the process of step SP8 in FIG. 8 in detail.

FIG. 10 is a flowchart illustrating a carrier interrupt process for controlling an inverter waveform.

FIG. 11 is a flowchart illustrating a PWM timer process per phase.

FIG. 12 is a diagram illustrating a magnetic flux trajectory based on a sine wave voltage.

FIG. 13 is a diagram illustrating a voltage vector on a complex plane.

FIG. 14 is a diagram illustrating a locus of a desired voltage vector in a sine wave voltage phase of 0 ° to 60 °.

FIG. 15 is a diagram illustrating the definition of an area.

FIG. 16 is a diagram showing a voltage vector developed in an on / off state of each phase transistor.

FIG. 17 is a diagram showing a magnetic flux waveform.

FIG. 18 is a diagram illustrating a relationship between a rotation speed and a phase command.

FIG. 19 is a block diagram schematically showing another embodiment of the brushless DC motor control device of the present invention.

FIG. 20 is a block diagram schematically showing a conventional brushless DC motor control device.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 Inverter part 2 Brushless DC motor 3 Phase reference generation part 3 'Magnetic flux zero cross generation part 4 Waveform control part 5 Control microcomputer 6 Speed detection part

 ──────────────────────────────────────────────────続 き Continued on the front page F term (reference) 5H560 BB04 BB12 DA15 DC12 DC13 DC14 EB01 GG03 SS02 TT07 TT15 UA02 XA06 XA12

Claims (10)

[Claims] An output of an inverter (1) is a brushless D.
A method for controlling a brushless DC motor to be supplied to a C motor (2), the method comprising: generating a reference signal not restricted by a rotor position angle; and setting an inverter output waveform phase based on the generated reference signal. Controlling the brushless DC motor. 2. The output of the inverter (1) is a brushless D
A brushless DC motor control method for supplying to a C motor (2), wherein a reference signal not restricted by a rotor position angle is generated, a load state is detected, and a load state is detected based on the generated reference signal and the detected load state. And controlling the inverter (1) to set the inverter output waveform phase. 3. The brushless DC motor control method according to claim 1, wherein the primary magnetic flux is detected to generate a reference signal that is not restricted by the rotational position angle. 4. A brushless DC motor control method according to claim 2, wherein a load state is detected based on a rotation speed of the brushless DC motor. 5. The output of the inverter (1) is a brushless D
A brushless DC motor control method for supplying a C motor (2), wherein a primary magnetic flux is detected and an inverter is directly controlled by the detected primary magnetic flux. 6. The output of the inverter (1) is a brushless D
A brushless DC motor control device for supplying to the C motor (2), a reference signal generating means (3) (3 ') for generating a reference signal not restricted by a rotor position angle; A brushless DC motor control device comprising: inverter control means (4) and (5) for controlling the inverter (1) to set the inverter output waveform phase. 7. An output of the inverter (1) is a brushless D
A brushless DC motor control device for supplying a C motor (2), wherein reference signal generating means (3) and (3 ') for generating a reference signal not restricted by a rotor position angle; and load state detection for detecting a load state. Means (6); and inverter control means (4) (5) for controlling the inverter (1) to set the inverter output waveform phase based on the generated reference signal and the detected load condition. Brushless DC motor control device. 8. The brushless DC motor according to claim 6, wherein said reference signal generating means (3 ') detects a primary magnetic flux and generates a reference signal not restricted by a rotational position angle. Control device. 9. The brushless DC motor control device according to claim 6, wherein said load state detection means (6) detects a load state based on a rotation speed of the brushless DC motor. 10. A brushless DC motor control device for supplying an output of an inverter (1) to a brushless DC motor (2), comprising: a primary magnetic flux detecting means for detecting a primary magnetic flux; Brushless D comprising inverter control means for controlling an inverter
C motor control device.