JP2001110647A - High frequency power transformer and power conversion device using the same - Google Patents

Abstract

[PROBLEMS] To provide a high-frequency power transformer having a secondary winding with a center tap for full-wave rectification output, and a highly efficient and highly reliable power conversion device using the same. thing. SOLUTION: Fine crystal grains having a peak value of magnetizing force of 0.05 A / m and a magnetic permeability at a frequency of 20 kHz of not less than 10,000 and not more than 30,000 and having a crystal grain size of not more than 50 nm are formed in the structure. A primary winding without a center tap and at least one set of secondary windings for full-wave rectification output are provided on a non-cut nanocrystalline soft magnetic alloy ribbon wound core occupying 50% or more of the entire volume. In the high-frequency power transformer, the two coupling coefficients at 20 kHz of the primary winding and the secondary windings for full-wave rectification output, which are the main outputs, are both 0.9999 or more, and the main output is the full-wave A high-frequency power transformer in which each of the output secondary windings is short-circuited and a value obtained by dividing a large one of leakage inductances measured from the primary winding side by a small one is 1.1 or less.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an uncut nanocrystalline soft magnetic alloy ribbon core comprising Fe as a main component, wherein fine grains having a grain size of 50 nm or less occupy 50% or more of the entire volume of the structure. And a high-frequency power transformer having at least one set of secondary windings for full-wave rectification output and a power converter using the same.

[0002]

2. Description of the Related Art A full-bridge type DC-DC converter shown in FIG. 5 is used as one of insulated power converters using a high-frequency power transformer. In FIG. 5, 1 is an input DC power supply, 2, 3, 4 and 5 are main switches, 6, 7, 8 and 9 are feedback diodes, 10 is a capacitor for blocking DC current, and 20 has no center tap. A high-frequency power transformer having a primary winding and a secondary winding for full-wave rectification output, 21 is a primary winding of the high-frequency power transformer 20, and 22 and 23 are for full-wave rectification output of the high-frequency power transformer 20. Secondary windings, 31 and 32 are output rectifier diodes, 33 is an output smoothing choke coil, 34 is an output smoothing capacitor, 35 and 36
Is an output terminal, and 37 is a load.

In the full-bridge type DC-DC converter shown in FIG. 5, main switches 2 and 3 and 4 and 5 each have 1
A pair of switches, these two sets of switches are alternately switched, so that the high-frequency power transformer 2
0 through the capacitor 10 to the primary winding 21 of FIG.
A high-frequency voltage such as 21 is applied, and from the secondary windings 22 and 23 for full-wave rectification output of the high-frequency power transformer 20, loads are output via output rectifier diodes 31 and 32, a smoothing choke coil 33, and a capacitor smoothing 34. Power is supplied to 37. In FIG. 6, the period in which the main switches 2 and 3 are on is Ton23, and the period in which the main switches 4 and 5 are on is Tn.
on45, and Tp is the cycle. (Ton23 + Ton45) /
Tp is an on-duty ratio Don, and PWM control (pulse width) for controlling the output voltage Vo to be constant with respect to the fluctuation of the voltage E of the input DC power supply 1 and the fluctuation of the load 37 by keeping Tp constant and changing Don. Modulation control) is generally used. Note that the driving frequency f of the high-frequency power transformer 20 is
Is given by 1 / Tp.

FIG. 7 is a conceptual diagram of the operation BH loop of the high-frequency power transformer 20 in the present converter. The direction of the magnetic field generated in the high-frequency power transformer 20 when a current flows from the black circle side of the primary winding 21 of the high-frequency power transformer 20 shown in FIG. 5 is set to the positive side of the H-axis in FIG. Therefore, while the main switches 2 and 3 are on, Ton23
7, the magnetic flux density of the high-frequency power transformer 20 changes by 2 Bm from the point a to the point b in FIG.
During the period Ton45 during which the power is on, the magnetic flux density of the high-frequency power transformer 20 changes by -2 Bm from the point b to the point a in FIG. That is, the high-frequency power transformer 20 in the present converter operates to draw a symmetric minor loop with respect to the origin of the BH loop.

In the high-frequency power transformer 20 of the present converter, downsizing and low loss are important issues. As a general method of reducing the size of the high-frequency power transformer 20, increasing the driving frequency is performed. But,
Extremely high frequency without considering the high frequency characteristics of elements such as the magnetic core used in the high frequency power transformer 20, the main switches 2, 3, 4, and 5, the feedback diodes 6, 7, 8, and 9, or the output rectifier diodes 31 and 32 This not only increases the loss of these elements, but also increases the loss of the high-frequency power transformer 20, which causes a decrease in converter efficiency and a decrease in reliability due to an excessive rise in temperature.

The high-frequency power transformer 20 of the present converter generally has a driving frequency selected in consideration of the high-frequency characteristics of the main switches 2, 3, 4 and 5.
It is necessary to select the most compact and low loss core.

For example, when the output power is relatively small up to about several kW, a power MOS-FET is usually selected for the main switches 4 and 5, and the driving frequency is 50 kHz.
It is selected more than degree. In this case, the magnetic core of the high-frequency power transformer 20 is conventionally mainly provided with a saturation magnetic flux density B at room temperature.
Although the s is as small as about 0.5T, a Mn-Zn ferrite core having a small core loss at several hundred kHz or more has been used.

On the other hand, in the region where the output power exceeds several kW, the main switches 2, 3, 4 and 5 are generally connected to the IGBT.
Is selected, and the driving frequency is selected from several kHz to about 20 kHz. In this case, Hagiwara, Saito, Kamo, Toyota, Yamauchi, Yoshizawa: "Inverter transformer using microcrystalline alloy for iron core", IEEJ Technical Report, MAG-90-19
4. As described on December 6, 1990 (hereinafter referred to as Document 1), the magnetic core of the high-frequency power transformer 20 has a saturation magnetic flux density at room temperature exceeding twice that of the Mn-Zn ferrite core. At 1T or more, the core loss at 20 kHz is small, and fine grains having Fe as a main component and having a grain size of 50 nm or less as described in JP-A-63-302504 account for 50% of the entire volume of the structure. Fe accounts for more than
It is known that a base nanocrystalline soft magnetic alloy ribbon wound core is excellent.

In the above-mentioned Fe-based nanocrystalline soft magnetic alloy ribbon wound core, the above-mentioned reference 1 or Fukunaga, Koga, Eguchi,
Ota, Kakehashi: "Magnetic properties of cut core with gap using iron-based magnetic ribbon, IEEJ Technical Report, MAG-89-
203, December 1, 1989 (hereinafter referred to as Reference 2)
The resin or varnish penetrates between the layers of the nanocrystalline soft magnetic alloy ribbon that composes the core by subjecting the core to resin impregnation or surface fixation, as described in It is known that when a stress is applied to a magnetic alloy ribbon, its magnetic core loss is significantly increased.

By the way, in the magnetic core for the high frequency power transformer 20, a cut magnetic core is widely used in order to facilitate the winding operation. In order to cut the nanocrystalline soft magnetic alloy ribbon wound core, the wound core is impregnated with an impregnating material such as an epoxy adhesive, and the layers of the nanocrystalline soft magnetic alloy ribbon constituting the wound core are separated. After fixing with the impregnating material, it is necessary to cut with a rotating grindstone or the like. Also, from the viewpoint of reducing magnetic core loss, after cutting, end faces are also mirror-polished. However, a core obtained by cutting a nanocrystalline soft magnetic alloy ribbon wound core using such a technique is:
As described above, the manufacturing process becomes complicated, and as described in the above-mentioned documents 1 and 2, there is a problem that the magnetic core loss is significantly increased.

[0011] Therefore, in order to reduce the core loss of the core cut using a nanocrystalline soft magnetic alloy ribbon wound core, Yoshizawa,
Mori, Arakawa, Yamauchi: "High-frequency magnetic properties of Fe-Cu-Nb-Si-B nanocrystalline alloys", IEICE Technical Meeting, M
AG-94-202, November 22, 1994 (hereinafter referred to as
As described in Literature 3, a nanocrystalline soft magnetic alloy ribbon whose saturation magnetostriction constant λs is as small as 10 ⁻⁶ or less is used, and the surface of the ribbon is covered with ceramics to perform an interlayer insulation treatment. Is valid. However, in the case of a magnetic core coated with the above-mentioned ceramic and subjected to interlayer insulation treatment, the hardness of the insulating layer of this ceramic is high, so that the number of cutting steps is significantly increased. It reaches about 1.4 times the core loss of the alloy ribbon wound core.

Therefore, in order to effectively utilize the low core loss characteristic of the nanocrystalline soft magnetic alloy ribbon core, stress should not be applied to the nanocrystalline soft magnetic alloy ribbon constituting the core as much as possible. Must be configured as follows. As a nanocrystalline soft magnetic alloy thin-band wound core with such a configuration, the uncut wound core is housed in an insulating case made of plastic or ceramic, etc., using silicon grease or gel-like silicon rubber as a buffer, and externally. The one in which the stress is hardly applied directly to the wound core is widely used.

[0013]

As in the full-bridge DC-DC converter of FIG. 5, the magnetic flux density of the high-frequency power transformer 20 is symmetric with respect to the origin of the BH loop as shown in FIG. In the high-frequency power transformer 20 of the power conversion device that performs an operation of drawing a minor loop, the BH minor loop during the operation performs an asymmetric operation with respect to the origin of the BH loop as shown in FIG. The high frequency power transformer 20 suppresses the magnetic saturation and the exciting current from increasing remarkably, and the main switches 2, 3, 4 and 5
It is extremely important to ensure safe operation of the vehicle.

As is well known, the magnetic polarization of the high-frequency power transformer 20 is mainly caused by variations in the electrical characteristics of the main switches 2, 3, 4 and 5.
The exciting current is balanced at a certain value by the circuit impedance. However, when the magnetic bias of the high frequency power transformer 20 is large, the BH minor loop during operation reaches one saturation region as shown in FIG. 8 and the exciting current is significantly increased. An excessive current flowed between the main electrodes of the main switches 5 and 5, and the main switches 2, 3, 4 and 5 could be broken. In particular, when the voltage of the input DC power supply 1 suddenly changes or the load 27 suddenly changes, the amount of change ΔB in the magnetic flux density during the operation of the high-frequency power transformer 20 increases transiently, so that the amount of increase in the exciting current due to the magnetization increases. , The main switches 2, 3, 4 and 5 were at high risk of breaking.

When an overcurrent protection circuit having a fast response speed for suppressing an overcurrent flowing between the main electrodes of the main switches 2, 3, 4 and 5 is provided, the main switch 2 is remarkably demagnetized. An excessive current can be prevented from flowing through 3, 4, and 5, and the main switches can be prevented from being broken. However, when the overcurrent protection circuit operates, there is a problem in that sufficient power cannot be supplied to the output side, and thus the accuracy of constant output voltage cannot be ensured.

As the most general method for preventing the high-frequency power transformer 20 from being demagnetized, a magnetic core having a small magnetic permeability is adopted for the high-frequency power transformer 20, and the operating magnetic flux density peak value Bm of the magnetic core is determined. The selection is made to have a sufficiently small value with respect to the saturation magnetic flux density Bs of the magnetic core. The simplest method for obtaining a magnetic core with a small magnetic permeability is to provide a gap in the cut magnetic core and reduce the effective relative magnetic permeability. According to this method, there is also an advantage that the effective relative permeability of the magnetic core can be arbitrarily selected by adjusting the gap width.

However, a method of reducing the effective relative permeability by providing a gap as described above in the case of a nanocrystalline soft magnetic alloy ribbon wound core is also described in the above-mentioned documents 1 to 3. As described above, since the core loss is greatly increased by providing the cut core and providing the gap, the feature of low core loss, which is the greatest advantage when used in the high-frequency power transformer 20, is impaired, and the gap portion is reduced. There is a problem that the copper loss increases due to the influence of the leakage magnetic flux generated in the above.

In the nanocrystalline soft magnetic alloy ribbon wound core,
As a method of reducing the relative magnetic permeability without providing a gap, the direction of the ribbon width direction of the same wound core (the height direction of the wound core)
And a method of applying stress to the nanocrystalline soft magnetic alloy ribbon constituting the wound core. However, in a composition system where the core loss is relatively small,
The level of relative permeability achievable using the former method is 20
It reaches about 10,000 or more at kHz, and Mn-
The value was one digit larger than about several thousands of Zn ferrite cores, and did not reach a level sufficient for countermeasures against magnetic polarization. On the other hand, the latter method has a problem that the core loss is significantly increased as described in the above-mentioned documents 1 and 2.

For this reason, the full bridge type DC-D shown in FIG.
Like a C converter, the magnetic flux density of the high-frequency power transformer 20 has a high frequency power transformer 20 that operates to draw a BH minor loop symmetric with respect to the origin of the BH loop as shown in FIG. In order to use the nanocrystalline soft magnetic alloy ribbon wound core and exhibit the characteristic of low core loss, the high frequency power transformer is saturated by extreme magnetic polarization so that the main switch is not destroyed. 20. A demagnetization suppression circuit capable of detecting the amount of magnetization of 20 and correcting the correction by independently controlling the ON period of each of two sets of switches formed by the main switches 2, 3, 4 and 5. It was necessary to take countermeasures such as adding.

In the above description, a full-bridge DC
The problems of the high-frequency power transformer 20 using the nanocrystalline soft magnetic alloy ribbon wound core and the power converter using the same have been described by taking the -DC converter as an example. A high-frequency power transformer 20 of another power converter such as a half-bridge type converter having a primary winding having no tap and at least one set of secondary windings for full-wave rectification output, and using the same. However, the same problem has been encountered in the conventional power converter.

An object of the present invention is to provide a small-sized uncut nanocrystalline soft magnetic alloy thin film which can prevent the occurrence of a level of demagnetization at a level which is difficult to realize in the prior art, and which can be a practical obstacle. A high-frequency power transformer having a band-wound core having a primary winding having no center tap and at least one set of secondary windings having a center tap for full-wave rectification output, and high efficiency and reliability using the same. It is to provide a high power conversion device.

[0022]

According to the present invention, the peak value of the magnetizing force is 0.05 A / m, and the permeability at a frequency of 20 kHz is 10,000.
A center tap is applied to the uncut nanocrystalline soft magnetic alloy ribbon wound core in which fine crystal grains having Fe as a main component of not less than 00 and not more than 30,000 and having a grain size of not more than 50 nm account for 50% or more of the entire volume of the structure. In a high-frequency power transformer provided with a primary winding having no primary winding and at least one set of secondary windings for full-wave rectification output, the coupling coefficient at 20 kHz is 0.9999 or more, and the full-wave output is the main output. And a value obtained by short-circuiting each of the secondary windings and dividing a larger one of the leakage inductances measured from the primary winding side by a smaller one is 1.1 or less. High-frequency power transformer.

With this configuration, a low-loss coreless uncut nanocrystalline soft magnetic alloy thin-film wound core has a primary winding having no center tap and at least one set of a secondary winding for full-wave rectification output. Magnetic saturation due to magnetic demagnetization, which had become a problem when using high-frequency power transformers with secondary windings in power converters such as full-bridge converters and half-bridge converters, without adding complicated demagnetization suppression circuits It is preferable because it can be prevented.

In the high-frequency power transformer, the peak value of the magnetizing force of the magnetic core of the nanocrystalline soft magnetic alloy ribbon is set to 80.
Squareness ratio Br / B, which is the ratio of the residual magnetic flux density Br and the saturation magnetic flux density Bs in the DC magnetic characteristics measured as 0 A / m
When s is set to 0.2 or less, the magnetic saturation due to the magnetic bias is less likely to occur, so that the change amount of the operating magnetic flux density can be set to a larger value, and the number of turns of the winding can be reduced.
This is preferable because the high-frequency power transformer can be downsized.

In the high-frequency power transformer, at least one set of secondary windings for full-wave rectification output, which is a main output of the transformer, has each of its windings connected to the nanocrystalline soft magnetic alloy ribbon core. Bifilar wound almost evenly,
When the secondary winding is sandwich-wound by a primary winding wound substantially evenly around the nanocrystalline soft magnetic alloy ribbon core, the primary winding and its main output, full-wave, Since the leakage inductance between the secondary windings for the rectified output can be reduced and the coupling coefficient can be easily increased, the increase in the excitation current due to the demagnetization can be further reduced, and the copper loss due to the influence of the leakage magnetic flux can be reduced. Since the increase can be suppressed, downsizing and high efficiency of the high frequency power transformer can be achieved, which is preferable.

When the driving frequency of the high-frequency power transformer is selected in the range of 5 kHz to 150 kHz, the size and efficiency of the selected high-frequency power transformer can be further reduced as compared with the conventional high-frequency power transformer. .

The power converter using the high-frequency power transformer according to the present invention is different from the conventional power converter.
The size and efficiency can be reduced, and an increase in the exciting current due to the demagnetization of the high-frequency power transformer can be suppressed with a simple circuit configuration.

[0028]

Embodiments of the present invention will be described below in detail. (Embodiment) A full-bridge type DC having a switching frequency f of 20 kHz with a circuit configuration shown in FIG.
The high-frequency power transformer 20 of the DC converter and the performance of the full-bridge DC-DC converter were examined.

In FIG. 5, reference numeral 1 denotes an input DC power supply;
3, 4 and 5 are main switches, 6, 7, 8 and 9 are feedback diodes, 10 is a capacitor for blocking DC current, 20 is a primary winding having no center tap and 2 for a full-wave rectified output. High frequency power transformer with secondary winding,
21 is a primary winding of the high-frequency power transformer 20;
And 23 are secondary windings for full-wave rectified output of the high-frequency power transformer 20, 31 and 32 are output rectifier diodes, 33 is an output smoothing choke coil, 34 is an output smoothing capacitor, 35 and 36 are output terminals, and 37 is an output terminal. It is a load.

[0030]

[Table 1]

In this embodiment, in order to suppress variations in the ON periods of the main switches 2, 3, 4, and 5, which are one of the causes of the magnetic bias of the high-frequency power transformer 20, these four main switches are provided with power. Using a MOS-FET,
In order to suppress variations in the turn-off time, the gate current peak value at the time of the turn-off is increased to shorten the turn-off time as much as possible.

In the circuit of FIG. 5, the feedback diodes 6, 7, 8 and 9 regenerate the excitation energy of the high-frequency power transformer 20 to the input DC power supply 1 to improve the efficiency of the converter and to increase the efficiency of the converter. 20 has the function of suppressing the magnetization. Further, the capacitor 10 for blocking the DC current can also prevent the DC current component from flowing into the primary winding 21 of the high-frequency power transformer 20, and thus has the function of suppressing the magnetization of the high-frequency power transformer 20.

The magnetic core shown in Table 2 was used for the high frequency power transformer 20. In Table 2, from core A to core, uncut nanocrystalline soft magnetic alloy ribbons each containing Fe as a main component and fine crystal grains having a crystal grain size of 50 nm or less occupying 50% or more of the entire volume of the structure. It is a magnetic core.

[0034]

[Table 2]

Each of the magnetic cores in Table 2 has an outer diameter of 4
4 mm, inner diameter 24 mm, height 20 mm in toroidal shape with dimensions of outer diameter 49 mm, inner diameter 21 mm, height 25
mm in a plastic insulating case. The effective sectional area of each core is 142.5 mm ² , and the average magnetic path length is 1
It is the same at 06.8 mm.

Table 3 shows the winding specifications of the high-frequency power transformer 20. In Table 3, the configurations of the high-frequency power transformers 20 of the present invention A to the present invention D and the comparative examples s and t are shown in FIG. FIG. 1 is a radial cross-sectional view of a high-frequency power transformer 20 of the present invention A to the present invention D, and comparative examples s and t. 5. In FIG. 5, reference numeral 50 denotes a magnetic core including a case, black circles 51 and white circles 52 represent primary windings 21 in FIG. 5, shaded circles 53 and hatched circles 54 in FIG. They correspond to the secondary windings 22 and 23.

The secondary windings 53 and 54 are each 0.23φ.
The litz wire composed of 105 pieces of the polyurethane insulated wire was wound around the magnetic core 50 substantially bifilar. Meanwhile, 1
The next winding is a 2-layer, 1.0φ three-layer insulated wire.
A winding 51 wound substantially evenly around a magnetic core 50 including a case and a three-layer, 1.0φ three-layer insulated wire 20
The secondary windings 53 and 54 are sandwiched by windings 52 in which the turns are wound substantially evenly around the magnetic core 50, and at the same time, the windings 51 and 52 are connected in parallel to form four parallel windings.
The primary winding 21 of the turn is configured, and is a so-called sandwich winding configuration.

In Table 3, the configurations of the high-frequency power transformers 20 of Comparative Examples a to f are shown in FIG. FIG.
FIG. 9 is a radial cross-sectional view of the high-frequency power transformer 20 of Comparative Example a to Comparative Example f. In the same figure, 50 is a magnetic core including a case, black circle 61 is the primary winding 21 in FIG. 5, shaded circle 62 and hatched circle 63 are the secondary windings 22 and 23 in FIG. 5, respectively. I do.

The primary winding 61 is formed by winding three turns of a 1.0-φ three-layer insulated wire around the magnetic core 50 substantially four times in four paras. On the other hand, for the secondary windings 62 and 63, litz wires composed of 105 0.23φ polyurethane insulated wires were substantially uniformly aligned and wound around the magnetic core 50, respectively.

In Table 3, the configurations of the high-frequency power transformers 20 of Comparative Examples g to l are shown in FIG. FIG.
It is a radial cross-sectional view of the high frequency power transformer 20 of Comparative Example g to Comparative Example l. In the same figure, 50 is a magnetic core including a case, black circle 61 is the primary winding 21 in FIG. 5, shaded circle 62 and hatched circle 63 are the secondary windings 22 and 23 in FIG. 5, respectively. I do.

[0041]

[Table 3]

The primary winding 61 is formed by winding three turns of a 1.0-diameter three-layer insulated wire around the magnetic core 50 substantially four times in four paras. On the other hand, the secondary windings 62 and 63
A litz wire composed of 105 polyurethane-insulated wires each having a diameter of 0.23φ was bifilar wound around the magnetic core 50 substantially uniformly.

In Table 3, the configurations of the high frequency power transformers 20 of Comparative Examples m to r are shown in FIG. FIG.
It is a radial cross section of the high frequency power transformer 20 of the comparative example m to the comparative example r. 5. In FIG. 5, reference numeral 50 denotes a magnetic core including a case, black circles 51 and white circles 52 represent primary windings 21 in FIG. 5, shaded circles 53 and hatched circles 54 in FIG. They correspond to the secondary windings 22 and 23.

For the secondary windings 62 and 63, litz wires composed of 105 0.23φ polyurethane insulated wires were wound around the magnetic core 50 almost uniformly. Meanwhile, 1
The next winding is a 2-layer, 1.0φ three-layer insulated wire.
A winding 51 wound substantially evenly around a magnetic core 50 including a case and a three-layer, 1.0φ three-layer insulated wire 20
At the same time, the secondary windings 53 and 54 are sandwiched between the windings 52 in which the turns are wound substantially evenly around the magnetic core 50, and at the same time, the windings 51 and 52 are connected in parallel to form four parallel windings. The primary winding 21 has 21 turns.

Table 4 shows that among the coupling coefficients between the primary winding and one of the secondary windings at a frequency of 20 kHz of the 24 types of high-frequency power transformers 20 shown in Table 3, k1 is the smaller one.
The larger one was k2, and only one of each of the secondary windings was short-circuited, and the larger one of the leakage inductances at the frequency 20 kHz measured from the primary winding side was divided by the smaller one Ll2. The ratio of leakage inductance Ll1 / Ll2, the circuit configuration is shown in FIG. 5, the degree of demagnetization of the high-frequency power transformer 20 when mounted on the full-bridge type DC-DC converter shown in Table 1 and the magnetic flux density during operation. The change amount ΔB and the temperature rise ΔT are shown. Note that the coupling coefficients k1 and k2 correspond to the inductance of the primary winding L.
p and the leakage inductances Ll1 and Ll2 can be obtained by the following equations. k1 = (1- (L11 / Lp)) ^0.5 (1) k2 = (1- (L12 / Lp)) ^0.5 (2)

Regarding the magnetic polarization, in the range of the specifications in Table 1, the case where the saturation of the high frequency power transformer does not occur due to the magnetic polarization even when the load is suddenly changed from 2 to 25 A, and the load is 2
The case where the high frequency power transformer is saturated due to the magnetic polarization only when the voltage is suddenly changed from 25 to 25A is indicated by Δ.
And Further, the change amount ΔB of the magnetic flux density and the temperature rise ΔT during the operation are the same as those of the input voltage 200 at an ambient temperature of 25 ° C.
It is a result measured at the time when the value is saturated by continuous energization under the input / output conditions of V, output voltage 40V, and load current 25A.

As can be seen from Table 4, the magnetic permeability was 10,000 at 20 kHz and the peak value of the magnetizing force was 0.05 A / m.
An uncut nanocrystalline soft magnetic alloy ribbon wound core of not more than 30,000 or less, a coupling core k1 and k2 of 0.9999 or more, and a leakage inductance ratio Ll1 / Ll2 of 1.1 or less. According to the high-frequency power transformers of the inventions A to D, the magnetic polarization can be suppressed to a level that does not cause any practical problem, and the temperature rise ΔT is also suppressed to 60 ° C. or less, which is an allowable value that does not hinder practical use. Can be. Here, the allowable value of the temperature rise ΔT is 60 ° C. which is obtained by adding 20 ° C. to 40 ° C. which is the upper limit of the ambient temperature during the operation in Table 1 and the assumed upper limit of the temperature rise inside the DC-DC converter case during the operation. Was subtracted from the allowable temperature of class E insulation, 120 ° C., to obtain 60 ° C. or less.

On the other hand, the leakage inductance ratio Ll
In the high-frequency power transformers 20 of Comparative Examples a to e and 1 to Lq in which 1 / L12 exceeds 1.1, the maximum output cannot be taken out stably due to the influence of the magnetic polarization, and the temperature rise ΔT is measured. Was impossible.

Also, at 20 kHz, the peak value of the magnetizing force is 0.0.
When a magnetic core having a magnetic permeability of less than 10,000 measured at 5 A / m was used, as shown in Comparative Examples f and r, the ratio L11 / L12 of leakage inductance was 1.1.
Even when the load exceeds the limit, there was no influence of the magnetic polarization in the steady state, but there was a problem that the magnetic polarization was generated when the load suddenly changed, and the temperature rose to 80 ° C. or more.

[0050]

[Table 4]

Leakage inductance ratio L1 / L1
2 is 1.1 or less, but the coupling coefficients k1 and k2 are 0.9.
In Comparative Examples g to j of less than 998, the influence of the magnetic declination was not observed in the steady state, but there was a problem that the magnetic demagnetization occurred when the load suddenly changed. In Comparative Examples g to j,
It can be seen that the temperature rise ΔT is high even though the same magnetic cores a to d are used as in the present inventions A to D.

The comparative examples k and t show the leakage
The inductance ratio L11 / L12 is 1.1 or less, the coupling coefficients k1 and k2 are 0.9999 or more, and the temperature rise ΔT
Is less than the allowable value, but since the magnetic permeability measured at 20 kHz and the peak value of the magnetizing force exceeds 30,000, the high-frequency power transformer is saturated due to the influence of the magnetic polarization at the time of a sudden change in load. There was a problem to do.

20 kHz, peak value of magnetizing force 0.05 A /
In Comparative Examples 1 and t using magnetic cores having a magnetic permeability of less than 10,000 measured in m, the ratio of leakage inductance Ll1 / Ll2 is 1.1 or less and the coupling coefficients k1 and k2 are 0.9998. Even if the load is less than 1, even if the load is suddenly changed, the high frequency power transformer does not saturate due to the influence of the magnetic polarization, but the temperature rise ΔT reaches 70 ° C. or more.

As described above, 20 kHz
z, the magnetic permeability is 1 when the peak value of the magnetizing force is 0.05 A / m.
Using uncut nanocrystalline soft magnetic alloy ribbon wound cores of not less than 0000 and not more than 30,000, coupling coefficients k1 and k2
In any case, according to the high frequency power transformers of the present invention A to D of the present invention in which the leakage inductance ratio Ll1 / Ll2 is 1.1 or less and the leakage inductance ratio is 0.9999 or more, the magnetic demagnetization can be suppressed to a practically acceptable level. While being able to
The temperature rise ΔT is also an allowable value which does not hinder practical use.
It can be seen that a high-efficiency, high-efficiency, high-frequency power transformer can be obtained that can be suppressed to below ℃, and a highly efficient and highly reliable power converter can be realized.

In the present invention A to D according to the present embodiment, even when the load was suddenly changed from 2 A to 25 A as described above, no saturation of the high frequency power transformer due to the magnetization was recognized at all. However, the peak value of the magnetizing force was 0.05.
A / m, permeability at a frequency of 20 kHz is 10,000
0 to 30,000 and the peak value of the magnetizing force is 8
The squareness ratio Br /, which is the ratio of the residual magnetic flux density Br and the saturation magnetic flux density Bs in the DC magnetic characteristics measured as 00 A / m.
Fe having a Bs of more than 0.20 as a main component and a crystal grain of 50 n
m, a high-frequency power transformer having a winding structure similar to that of the present invention A to D using an uncut nanocrystalline soft magnetic alloy ribbon wound core in which fine crystals of 50% or more occupy 50% or more of the structure. Saturation of the high-frequency power transformer due to magnetization was slightly observed when the load suddenly changed. Therefore,
In order to perform a more stable operation, the squareness ratio Br / Bs is set to 0.2 like the magnetic core used in the present invention A to the present invention D.
It was also found that a value of 0 or less was more preferable.

Further, for a DC-DC converter whose circuit configuration is shown in FIG. 5 and whose specifications are given in Table 1, when the driving frequency was changed, the uncut nanocrystalline soft magnetic alloy ribbon wound core according to the present invention was used. High frequency power transformer,
For a high-frequency power transformer using a Mn-Zn ferrite core and a high-frequency power transformer using a Fe-based amorphous soft magnetic alloy thin-film wound core, the conditions under which the temperature rise ΔT is 60 ° C or less and no abnormal operation due to magnetic demagnetization will occur. Satisfactory product sizes were compared and studied. As a result, the high-frequency power transformer using the Fe-based amorphous soft magnetic alloy thin-film wound core can be most miniaturized when the driving frequency is less than 5 kHz, and the high-frequency power transformer using the Mn-Zn ferrite core can be used when the driving frequency exceeds 150 kHz. Most compact. On the other hand, it was also found that the high-frequency power transformer using the uncut nanocrystalline soft magnetic alloy thin-film wound core according to the present invention can be most miniaturized when the driving frequency is in the range of 5 kHz to 150 kHz.

[0057]

As described above, according to the present invention,
In particular, it is possible to suppress the occurrence of demagnetization, which is a practical obstacle, without providing a complicated demagnetization suppression circuit, and to use a small, uncut nanocrystalline soft magnetic alloy ribbon wound core with low core loss and a small core to increase the temperature. A high-efficiency high-frequency power transformer 20 having a small size and a highly efficient and highly reliable power converter using the same can be obtained. In the above embodiment, an application example to a full-bridge DC-DC converter as a typical power converter using a high-frequency power transformer has been described in detail. Applied to all high-frequency power transformers 20 provided with a primary winding having no tap and at least one set or more of secondary windings having a center tap, and to all power converters using the high-frequency power transformer 20;
Similarly, it exerts an effective effect, and the effect is extremely large.

[Brief description of the drawings]

FIG. 1 shows a high-frequency power transformer 1 according to the present invention.
The conceptual diagram of the winding structure cross section of an Example.

FIG. 2 is a conceptual diagram of a winding structure cross section of a high-frequency power transformer as a comparative example.

FIG. 3 is a conceptual diagram of a winding structure cross section of a high-frequency power transformer as a comparative example.

FIG. 4 is a conceptual diagram of a winding structure cross section of a high-frequency power transformer as a comparative example.

FIG. 5 is a circuit configuration block diagram of a full-bridge DC-DC converter.

6 is a conceptual diagram of a terminal voltage of a primary winding 21 of the high-frequency power transformer 10 of the full-bridge DC-DC converter of FIG.

FIG. 7 shows a full-bridge type DC-
Operation B- of high frequency power transformer 20 of DC converter
H loop conceptual diagram.

FIG. 8 is a conceptual diagram of an operation BH loop of the high-frequency power transformer 20 of the full-bridge DC-DC converter of FIG. 8 when the magnetic core is saturated due to the magnetization.

[Explanation of symbols]

 1: Input DC power supply 2, 3, 4, 5: Main switch element 7, 8, 9: Feedback diode 10: Capacitor for blocking DC current 20: High frequency power transformer 21: Primary winding of high frequency power transformer 20 23: Secondary winding 31, 32 of the high-frequency power transformer 20, 32: Output rectifier diode 33: Output smoothing choke coil 34: Output smoothing capacitor 35, 36: Output terminal 37: Load

──────────────────────────────────────────────────の Continued on the front page (51) Int.Cl. ⁷ Identification symbol FI Theme coat ゛ (Reference) H02M 3/335 H01F 31/00 M (72) Inventor Katsuhiro Ogura 33-12 Minamisakaemachi, Tottori City, Tottori Prefecture Hitachi F-term in the Tottori Plant of Metal Corporation (reference) 5E043 BA01 5H730 AA14 AA19 BB27 EE03 EE72 ZZ15

Claims (5)

[Claims] 1. Microstructure having a peak value of magnetizing force of 0.05 A / m, Fe having a magnetic permeability of 10,000 to 30,000 at a frequency of 20 kHz and a crystal grain size of 50 nm or less as a main component. A primary winding without a center tap and at least one set of secondary windings for full-wave rectification output are provided on a non-cut nanocrystalline soft magnetic alloy ribbon wound core occupying 50% or more of the whole volume In the high-frequency power transformer, the two coupling coefficients at 20 kHz of the primary winding and each of the secondary windings for full-wave rectification output, which are the main outputs, are all 0.9999 or more, and the primary output is the total output. The value obtained by short-circuiting each of the secondary windings for wave output and dividing the larger one of the leakage inductances measured from the primary winding side by the smaller one is 1.1 or less. Characteristic high circumference Power transformer. 2. The high-frequency power transformer according to claim 1, wherein a residual magnetic flux density Br and a saturation in a DC magnetic characteristic measured by setting a peak value of a magnetizing force of the nanocrystalline soft magnetic alloy ribbon core to 800 A / m. A high-frequency power transformer, wherein a squareness ratio Br / Bs, which is a ratio of magnetic flux density Bs, is 0.2 or less. 3. The high-frequency power transformer according to claim 1, wherein the secondary winding for full-wave rectification output, which is the main output, has each of the windings constituting the nanocrystalline soft magnetic alloy ribbon. The secondary winding is sandwiched by a primary winding that is wound substantially evenly around the nanocrystalline soft magnetic alloy ribbon winding core. High frequency power transformer. 4. The high-frequency power transformer according to claim 1, wherein the driving frequency is 5 kHz.
A high-frequency power transformer characterized by being in the range from 150 kHz to 150 kHz. 5. A power converter using the high-frequency power transformer according to any one of claims 1 to 4.