JP2000125558A - Voltage resonance type switching power source circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To prolong the turning-off period of the rectifier diode for boosting of a boosting circuit and to increase the maximum load power or to reduce the size of an orthogonal transformer and, at the same time, to reduce unnecessary radiation noise by connecting a parallel resonance capacitor in parallel with the rectifier diode for boosting. SOLUTION: A second parallel resonance capacitor C2A is connected in parallel with a diode DB for boosting. Then the capacitance of the capacitor C2A is selected so that the resonance current IC3 flowing during the turning-off period of the diode DB may become a 1-cycle parallel resonance current by means of a parallel resonance circuit in a boosting circuit. Hereby, the current of the boosting current is made to increase by prolonging the turning-off period of the diode DB and the maximum load current is increased by raising the boosting voltage of the boosting circuit. In addition, noise can be made lower by shaping the waveforms of the current and voltage in the boosting circuit to smooth sine waves.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a voltage resonance type switching power supply circuit provided as a power supply for various electronic devices, and more particularly to a so-called wide range voltage resonance type switching power supply circuit.

[0002]

2. Description of the Related Art In recent years, switching power supply circuits have become mainstream as power supply circuits provided in various electronic devices. Further, as a switching power supply circuit type, a so-called hard switching power supply circuit in which the switching output has a rectangular wave, and a so-called switching output current flowing into the primary winding of the insulating converter transformer having a sine wave shape, and a so-called hard switching power supply circuit. It can be divided into soft switching power supply circuits. The soft switching power supply circuit has lower noise and higher power conversion efficiency than the hard switching power supply circuit. In addition, since it can be configured with a small number of parts, it is advantageous in terms of cost reduction and reduction in size and weight of the circuit board.

[0003]

The present applicant has proposed various types of current-resonant switching power supply circuits and voltage-resonant switching power supply circuits as soft switching power supply circuits. , Further downsizing and lightening of circuit scale, low noise,
In addition, various characteristics should be improved, such as handling high load power.

[0004]

In view of the above-mentioned problems, the present invention provides a rectifying / smoothing means for generating a rectified / smoothed voltage having a level corresponding to an input commercial AC power supply level by full-wave rectification, A gap is formed so as to obtain a required coupling coefficient that is loosely coupled, and an insulating converter transformer provided for transmitting the primary output to the secondary side and a rectified smoothing voltage output from the rectifying and smoothing means are intermittently provided. Switching means for outputting to the primary winding of the insulated converter transformer and at least an inductance component including the primary winding of the insulated converter transformer and the capacitance of the resonance capacitor, so that the operation of the switching means is a voltage resonance type. And the inductance component of the secondary winding of the isolated converter transformer A secondary side resonance circuit formed on the secondary side by the capacitance of a resonance capacitor, and an alternating voltage obtained in a secondary winding of an insulating converter transformer, and a DC that obtains a secondary side DC output voltage by full-wave rectification. An output voltage generating means, a controlled winding provided so as to function as an inductance component of the primary resonance circuit, and a control winding whose winding direction is orthogonal to the controlled winding. Constant-voltage control for the secondary-side DC output voltage by providing a variable control current to the control winding according to the level of the DC output voltage and changing the inductance of the controlled winding And at least an alternating voltage that is wound around the insulating converter transformer and that is in accordance with the switching output of the switching means. Boost voltage is provided by providing a boost winding, a boost rectifier diode for rectifying an alternating voltage obtained in the boost winding, and a boost smoothing capacitor for smoothing a rectified output by the boost rectifier diode. Generate
This boost voltage is superimposed on the rectified smoothed voltage to obtain a boost rectified smoothed voltage and includes a controlled winding, so that the boost rectified smoothed voltage becomes constant by a change in inductance of the controlled winding. And a parallel resonance capacitor connected in parallel with the boost rectifier diode and forming a parallel resonance circuit in the boost circuit by an inductance component forming the boost circuit. A resonant switching power supply circuit is configured.

[0005] According to the above configuration, the insulation converter transformer is loosely coupled, and on the secondary side, a secondary-side DC output voltage is generated by the full-wave rectifier circuit to supply power to the load. In other words, a required load condition is handled by providing a full-wave rectifier circuit on the secondary side, and accordingly, the primary side is not a voltage doubler rectifier circuit, but one time of the AC input voltage level. Is provided with a full-wave rectifier circuit that generates a rectified smoothed voltage corresponding to. A boost circuit is provided on the primary side to obtain a boosted smoothed voltage by superimposing the boosted voltage on the rectified smoothed voltage, thereby increasing the apparent DC input voltage level to the switching converter. The boost circuit is configured to stabilize the boost smoothed voltage in accordance with constant voltage control having an orthogonal control transformer. Further, in the present invention, a parallel resonance capacitor is connected in parallel to the boost rectifier diode forming the boost circuit to form a parallel resonance circuit in the boost circuit. This increases the off period of the boost rectifier diode,
A parasitic oscillation component generated in the boost circuit is absorbed.

[0006]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Prior to describing a power supply circuit as an embodiment of the present invention, a power supply circuit based on the invention proposed by the present applicant and serving as a base of the present invention will be described. One configuration example will be described with reference to FIG. The switching power supply circuit shown in FIG. 1 includes a voltage resonance type converter. Also, this power supply circuit is, for example, a so-called AC1
It is assumed to be a 00V system and to meet the condition that the maximum load power is 150 W to 160 W or more.

In the power supply circuit shown in FIG. 5, a bridge rectifier circuit D is provided as a rectifier circuit for receiving a commercial AC power supply AC (AC input voltage VAC) and obtaining a rectified smoothed voltage Ei.
i and a full-wave rectifier circuit including one smoothing capacitor Ci. In this case, for example, an inrush current limiting resistor Ri for limiting an inrush current flowing from the commercial AC power supply AC to the smoothing capacitor Ci when the power is turned on is inserted in series with the commercial AC power supply AC.

The voltage resonance type converter shown in FIG.
It employs a self-excited configuration with a stone switching element Q1. In this case, a high withstand voltage bipolar transistor (BJT; junction transistor) is employed as the switching element Q1. The base of the switching element Q1 is connected to the positive side of the smoothing capacitor Ci1 (rectified smoothed voltage Ei) via the starting resistor RS so that the base current at the time of starting can be obtained from the rectified smoothing line. Further, an inductor LB, a driving winding NB, a resonance capacitor CB, between the base of the switching element Q1 and the temporary ground.
A resonance circuit for self-excited oscillation composed of a damping resistor RB is connected in series. In this case, the drive winding NB is provided with an insulation converter transformer PIT (Power Isolation Transformer).
So that a required inductance for setting the switching frequency can be obtained together with the inductor LB. A clamp diode DD inserted between the base of the switching element Q1 and the negative primary side (earth) of the smoothing capacitor Ci forms a path for a damper current flowing when the switching element Q1 is off. , The collector of switching element Q1 is connected to one end of primary winding N1 of insulating converter transformer PIT, and the emitter is grounded.

A parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. This parallel resonance capacitor Cr
Is a voltage based on its own capacitance and a combined inductance (L1 + LR) obtained by connecting a primary winding N1 of an isolated converter transformer PIT described later and a controlled winding NR of a quadrature control transformer PRT (Power Regulating Transformer) in series. A parallel resonant circuit of the resonant converter is formed. When the switching element Q1 is turned off, the operation of the parallel resonance circuit causes the voltage VCP across the resonance capacitor Cr to be actually a sinusoidal pulse waveform to obtain a voltage resonance type operation.

[0010] The insulating converter transformer PIT is provided with a winding N3 so as to wind up the primary winding N1 in addition to the primary winding N1, the secondary winding N2 and the driving winding NB. The end of the winding N3 is connected to a positive electrode of a smoothing capacitor CiB for generating a boost voltage described later.
The negative electrode of the smoothing capacitor CiB is connected to the positive electrode (Ei line) of the smoothing capacitor Ci. Further, in the power supply circuit shown in this figure, a boost diode DB is provided. The boost diode DB has an anode connected to a connection point (Ei line) between the negative electrode of the smoothing capacitor CiB and the positive electrode of the smoothing capacitor Ci, and has a cathode connected in series with the controlled winding NR of the orthogonal control transformer PRT. Therefore, it is connected to a connection point between the primary winding N1 and the winding N3. According to such a connection configuration, the switching output voltage obtained in the winding N3 is rectified by the boost diode DB and smoothed by the smoothing capacitor CiB, thereby generating a boost voltage VB across the smoothing capacitor CiB. A circuit will be formed. However, as described above, the controlled winding NR is inserted in series in this boost circuit.

By providing the boost circuit, the voltage resonance type switching converter including the switching element Q1 is capable of operating the boost smoothed voltage EB obtained by superimposing the boost voltage VB on the rectified smoothed voltage Ei. The switching is performed as follows. That is, by operating the boost circuit, the apparent DC input voltage level to be supplied to the voltage resonance type switching converter is increased.

In the configuration having the boost circuit as described above, the boost smoothing voltage EB obtained by superimposing the boost voltage VB on the rectified smoothing voltage Ei is applied across the smoothing capacitor CiB and the smoothing capacitor Ci connected in series. As a result, the boost smoothing voltage EB is

(Equation 1) Can be represented by Then, the relationship of L3 = L1 is obtained as the inductance of the winding N3 and the primary winding N1, and the rectified smoothing voltage Ei, the drop voltage VF of the boost diode DB, and the saturation voltage V (SAT) of the switching element Q1 are obtained. Assuming that the relationship of Ei≫VF, V (SAT) holds, the boost smoothed voltage EB is calculated by the above (Equation 1)
On the basis of the,

(Equation 2) Will be indicated by In this case, in the power supply circuit shown in FIG. 5, for example, the inductance LR of the controlled winding NR of the orthogonal transformer PRT is set to 0.1 × L1 to 1.2 × L1.
, The boost smoothing voltage EB is substantially changed from Ei to 2Ei (Ei is the smoothing capacitor Ci).
(Corresponding to the level of the rectified and smoothed voltage obtained at both ends). In this way, the boost DC voltage is apparently increased by obtaining the boost smoothed voltage EB by the boost circuit. For example, as the maximum load power, the DC input voltage is obtained by the voltage doubler rectifier circuit. It is possible to increase it to a level that is almost equivalent to the configuration described above.

On the secondary side of the power supply circuit shown in FIG. 5, a center tap is provided for the secondary winding N2, and rectifier diodes DO1, DO2, DO3, DO4 and smoothing capacitors CO1, CO2 are connected. By connecting as shown in the figure, a set of [rectifier diodes DO1, DO2, smoothing capacitor CO1] and [rectifier diodes DO3, DO4, smoothing capacitor CO1]
2], two sets of full-wave rectifier circuits are provided.
A full-wave rectifier circuit composed of [rectifier diodes DO1, DO2, smoothing capacitor CO1] generates a DC output voltage EO1, and a full-wave rectifier circuit composed of [rectifier diodes DO3, DO4, smoothing capacitor CO2] generates a DC output voltage EO2. I do. That is, a full-wave rectifier circuit is provided to obtain a DC output voltage on the secondary side. In this case, the DC output voltage EO1 and the DC output voltage EO2 are also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO1 is used as a detection voltage, and the DC output voltage EO2 is used as an operation power supply of the control circuit 1.

As shown in FIG. 6, the insulating converter transformer PIT has an E-type core CR made of, for example, a ferrite material.
1. An EE-type core in which CR2 and CR2 are combined such that their magnetic legs face each other is provided. A primary winding N1 (and N3, N3) is attached to a central magnetic leg of this EE-type core using a bobbin B.
B) and the secondary winding N2 is wound separately. In this case, a gap G is formed for the center magnetic leg as shown in the figure. As a result, loose coupling with a required coupling coefficient can be obtained.

In the insulation converter transformer PIT, the inductance L1 of the primary winding N1 and the inductance L1 of the primary winding N1 are determined by the relationship between the polarity (winding direction) of the primary winding N1 and the secondary winding N2 and the connection of the rectifier diode DO (DO1, DO2). Secondary winding N2
In some cases, the mutual inductance M with the inductance L2 is + M or -M. For example, when the connection configuration shown in FIG. 7A is employed, the mutual inductance is + M, and when the connection configuration shown in FIG. 7B is employed, the mutual inductance is -M. If this is made to correspond to the configuration on the secondary side described above, for example, when the alternating voltage obtained in the secondary winding N2 has a positive polarity, the operation in which the rectified current flows through the rectifier diodes DO1 and DO3 is the + M operation mode. On the contrary, when the alternating voltage obtained in the secondary winding N2 has a negative polarity, the rectification current flowing through the rectification diodes DO2 and DO4 is in the -M operation mode (flyback method). It can be seen that there is. That is, each time the alternating voltage obtained in the secondary winding becomes positive / negative, the mutual inductance operates in the mode of + M / -M.

Further, in the case of the power supply circuit shown in FIG. 1, a secondary side parallel resonance capacitor C2 is provided for the secondary winding N2, and the capacitance of the secondary side parallel resonance capacitor C2 and the secondary winding A parallel resonance circuit is formed by the inductance of N2.

Here, assuming a configuration in which a secondary parallel resonance capacitor C2 is provided to form a secondary parallel resonance circuit, for example, as in the power supply circuit shown in this figure, the operation of this secondary parallel resonance circuit Since power is supplied to the load side,
Compared to the case where the secondary side parallel resonance capacitor C2 is not provided,
Further, the maximum load power increases. In addition to this, when a full-wave rectifier circuit is connected to the secondary-side parallel resonance circuit as in the circuit shown in this figure, as described above, both operation modes having a mutual inductance of + M / -M And the rectified current flows alternately. In other words, since the alternating voltage allows the rectified output to be obtained during both the positive and negative periods, the power supplied to the load increases accordingly.
The rate of increase of the maximum load power is also improved.

In this case, the control circuit 1 supplies a control current of a level corresponding to the fluctuation of the DC output voltage EO1 to the control winding NC of the orthogonal transformer PRT so that the DC output voltage EO1 is constant. Thus, the operation is performed to vary the inductance LR of the controlled winding NR. That is, the parallel resonance circuit on the primary side can be regarded as formed by the combined inductance (L1 + LR) of the primary winding N1 and the controlled winding NR and the capacitance of the parallel resonance capacitor Cr. When the inductance LR of the controlled winding NR changes due to the operation, the resonance condition of the primary side parallel resonance circuit changes and the operation of controlling the pulse width of the resonance voltage Vcr obtained at both ends of the switching element Q1 and the parallel resonance capacitor Cr is performed. can get. Here, the resonance voltage V
The pulse width of cr is the off period of the switching element Q1, in other words, controlling the pulse width of the resonance voltage Vcr controls the on period of the switching element Q1 to reduce the DC output voltage on the secondary side. There is nothing to control. That is, in the present embodiment, in the case of the AC 200 V system, the operation is performed so as to stabilize the secondary-side output voltage by controlling the ON period of the switching element Q1 (here, referred to as an inductance control method). I will). Further, the inductance LR of the controlled winding NR has a circuit configuration inserted in series with the boost circuit, and changes the impedance characteristic in the boost circuit by changing the inductance LR. As a result, the boost smoothing voltage EB represented by (Equation 1) is controlled to be constant. That is, in the power supply circuit shown in this figure, by controlling the secondary side DC output voltage to be constant, the primary side boost smoothing voltage EB is also controlled to be constant.

For example, in the configuration of the soft switching power supply circuit previously proposed by the present applicant, for example, the AC input voltage A
C100V system and maximum load power 150W ~ 160W
In the case of adopting the configuration corresponding to the above, a rectified smoothed voltage corresponding to almost twice the level of the AC input voltage is obtained by the voltage doubler rectifier circuit. Thereby, the DC voltage level input to the switching converter is increased to cope with the condition of relatively high load.

On the other hand, in the power supply circuit shown in FIG. 5, by increasing the maximum load power as described above, the rectifying and smoothing circuit for generating the DC input voltage (rectified and smoothed voltage) is a voltage doubler rectifier. It is no longer necessary to cover the load power by taking As a result, as shown in FIG. 5, it is possible to adopt a configuration of a normal equal-voltage rectifier circuit using, for example, a bridge rectifier circuit. Thus, for example, in the power supply circuit shown in FIG. 5, the rectified smoothed voltage Ei at the time of the AC input voltage VAC = 144 V becomes about 200 V. The resonance voltage Vcr is generated when the switching element Q1 is turned off by the action of the parallel resonance circuit on the primary side on the rectified smoothed voltage Ei. In the circuit of FIG. 5, the rectified smoothed voltage Ei is doubled as described above. That is, about 約 of the voltage rectification. However, in the configuration shown in FIG.
Since the boosted voltage VB is superimposed on i to generate the boosted smoothed voltage EB, the resonance voltage Vcr depends on the level of the boosted smoothed voltage EB, but the resonance voltage Vcr is still suppressed to about 1200 V. For example, in a configuration provided with a voltage doubler rectifier circuit, the resonance voltage V
cr rises to about 1800V. Therefore, in the circuit shown in FIG. 5, the switching element Q1 and the parallel resonance capacitor Cr need only be selected with a withstand voltage of 1200V.

Further, as described above, the full-wave rectifier circuit is provided on the secondary side so that the rectified current flows during both the positive and negative periods of the alternating voltage of the secondary winding N2A, so that the resonance on the secondary side is achieved. The voltage V2 is suppressed to the same level as the rectified smoothed voltage Ei in both the positive and negative periods.
As a result, as the rectifier diodes (DO1 to D04) forming the secondary-side full-wave rectifier circuit, a withstand voltage product substantially corresponding to the level of the rectified smoothed voltage Ei can be selected.

As described above, in the circuit shown in FIG. 5, since a low withstand voltage product can be used for the switching element Q1, the parallel resonance capacitor Cr, and the rectifier diode forming the full-wave rectifier circuit on the secondary side, Will be cheaper. Therefore, for example, the switching element Q1 and a rectifying diode forming a full-wave rectifier circuit on the secondary side have improved characteristics (for the switching element Q1, the saturation voltage VCE (SAT ), Good characteristics such as accumulation time tSTG, fall time tf, and current amplification factor hFE, and rectifier diodes having good characteristics such as forward voltage drop VF and reverse recovery time trr). Therefore, reduction of power loss is promoted accordingly. That is, power conversion efficiency can be improved at a lower cost or at substantially the same cost as compared with a configuration in which a DC input voltage is generated by a voltage doubler rectifier circuit. Further, by improving the power conversion efficiency, for example, a radiator plate or the like which is necessary for radiating heat of the rectifier diode of the voltage doubler rectifier circuit becomes unnecessary.

If a high switching frequency of, for example, about 100 KHz is set as the configuration of the voltage resonance type converter, the various components can be reduced in size and weight. Here, if the winding N3 is selected such that the boost voltage VB is optimized according to the maximum load power to be actually handled, further reduction in size and weight of various components can be realized. Furthermore, from the viewpoint of reducing the size and weight of the power supply circuit, in the configuration including the voltage doubler rectifier circuit for generating the DC input voltage, two sets of rectifier diodes and a smoothing capacitor were required, respectively. In the circuit shown in FIG. 5, for example, a full-wave rectifier circuit using a normal bridge rectifier circuit can be used, so that a set of a block-type smoothing capacitor and a bridge rectifier diode can be employed. In addition, miniaturization of parts can be achieved. That is, in the circuit shown in FIG. 5, various components including the insulating converter transformer PIT and the orthogonal control transformer PRT can be reduced in size. Experiments have also shown that the power conversion efficiency can be improved.

The operation of the main part of the power supply circuit shown in FIG. 5 will now be described with reference to the waveform diagram of FIG. FIG. 8A shows a resonance voltage Vcr generated at both ends of a parallel connection of the switching element Q1 // parallel resonance capacitor Cr. The resonance voltage Vcr has a waveform in which a sine-wave shaped resonance pulse appears as shown in the figure during the period TOFF when the switching element Q1 is turned on and the period TON when the switching element Q1 is turned off. That is, the operation is a voltage resonance type operation. On this occasion,
The rectified current IDB flowing through the boost diode DB and the current I1 flowing into the primary winding N1 via the boost diode DB and the controlled winding NR correspond to the switching operation as shown in FIG. Obtained as a waveform. The voltage VDB across the boost diode DB shown in FIG. 8C, the voltage VNR across the controlled winding NR shown in FIG. 8D, and the boost diode DB shown in FIG. 8E.
The voltage V1 between the cathode and the primary side ground is
In a period TR in which the boost diode DB is almost off, the waveform has a waveform in which a high-frequency pulse is generated.

As can be understood from the above description, the power supply circuit shown in FIG. 5 has been improved in characteristics such as an increase in load power that can be handled and an improvement in power conversion efficiency, and has a reduced number of parts. The reduction in size, the size of component elements, and the like have promoted the reduction in size, weight, and cost of circuit boards.

However, in a configuration in which the boost voltage VB is variably controlled to control the boost smoothed voltage EB to be constant, the variable control range of the boost smoothed voltage EB is expressed by the following equation. As can be seen from Expression 1), the inductance L1 of the primary winding N1 and the controlled winding N
It is determined by the ratio of R to the inductance LR. For this reason, in the circuit shown in FIG. 5, when the boost smoothing voltage EB is actually controlled to be constant with respect to the fluctuations of the load power and the AC input voltage, for example, the quadrature control transformer PRT is provided without providing the boost circuit. In comparison with the case where the constant voltage control is performed by the inductance control of (1), the variable range of the inductance LR is determined as follows.
1.2) As with L1, a 12-fold dynamic range is required. In order to secure this dynamic range, for example, it is necessary to set an appropriate number of turns for the controlled winding NR, which hinders miniaturization and cost reduction of the orthogonal control transformer PRT.

In the power supply circuit shown in FIG.
As shown in (b), (c), (d), and (e), a high-frequency pulse voltage or current is generated in the boost circuit during the period TR in which the boost diode DB is turned off. This high-frequency pulse is composed of a depletion layer capacitance (about several tens of pF) when the boost diode DB is off and an inductance component (L3,
LR) appears in response to the parasitic vibration generated by the action, and for example, reaches several MHz. For this reason, in the boost diode DB and the quadrature control transformer PRT, radiation noise due to the parasitic vibration is generated. Such unnecessary radiation noise must be reduced as much as possible.

In view of the above, the present invention is based on the voltage-resonant switching power supply circuit having the configuration shown in FIG. 5 and utilizes the merits thereof to further reduce the size and weight of the circuit and reduce the cost. The aim is to reduce noise and reduce noise.

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to a first embodiment of the present invention.
In this figure, the same parts as those in FIG.

In the power supply circuit shown in FIG. 1, a second parallel resonance capacitor C2A is connected in parallel with the boost diode DB. This second parallel resonance capacitor C
The capacitance of 2A is selected so that a parallel resonance circuit (hereinafter, also referred to as a "parallel resonance circuit in the boost circuit") is formed together with the inductance (L3 + LR) in the boost circuit. Further, the second parallel resonance capacitor C
As 2A, for example, a film capacitor is used.

FIG. 2 is a waveform diagram showing the operation of the main part of the power supply circuit shown in FIG.
2A shows a resonance voltage Vcr, FIG. 2B shows a rectified current IDB flowing through the boost diode DB, FIG. 2C shows a voltage VDB across the boost diode DB, and FIG. 2D shows a controlled winding NR. 2 (e) is a voltage V1 between the cathode of the boost diode DB and the primary side ground, FIG. 2 (f) is a resonance current IC3 flowing through the second parallel resonance capacitor C2A, and FIG. The current I1 flowing into the primary winding N1 from the boost circuit is shown.

Here, the resonance current IC3 flowing during the period TR during which the boost diode DB is turned off (FIG. 2 (f))
Is a second parallel resonance capacitor C2A such that the parallel resonance current of one cycle by the parallel resonance circuit in the boost circuit is obtained.
Is assumed to be selected. With such a configuration, as can be seen by comparing with the operation waveform of the power supply circuit of FIG. 5 shown in FIG. 8, the boost diode D
The period TR in which B is off is about twice as large as the period TR shown in FIG. 8, and by this operation, the rectified current IDB (FIG. 2B) flowing through the boost diode DB and the primary winding The current I1 (FIG. 2 (g)) flowing into N1 increases. That is, the current level flowing through the boost circuit is increased. In the configuration shown in FIG. 1, the boost diode D
During the period TR in which B is off, the second parallel resonance capacitor C2A → the controlled winding NR → the winding N3 → the smoothing capacitor Ci
Although the parallel resonance current IC3 flows through B, the current level flowing through the boost circuit is increased as described above, so that the boost voltage VB is higher than that of the power supply circuit having the configuration shown in FIG. And as a result, the maximum load power that can be handled further increases. Along with this, the waveforms of FIGS. 2B to 2G and FIGS.
As can be seen from comparison with the waveform (e), during the period TR in which the boost diode DB is turned off, the voltage and current in the boost circuit have no high-frequency pulse waveform component corresponding to the parasitic oscillation. As a result, a smooth sinusoidal waveform is obtained. As a result, radiation noise from the boost circuit (the boost diode DB, the controlled winding NR) is not generated, and the noise is further reduced.

For example, in the configuration shown in FIG. 1, the insulation converter transformer PR is assumed to correspond to the condition when the maximum load power is 150 W and the AC input voltage VAC is 100 V.
For T, using an EE-45 type ferrite core, for each number of turns of the primary winding N1, the secondary winding N2, and the winding N3,
N1 = N2 = N3, switching frequency fs = 100KH
z, the second parallel resonance capacitor C2A = 2200 pF, the level of the boost voltage VB is 5 compared to the circuit shown in FIG. 5 in which the second parallel resonance capacitor C2A is not provided.
%, And the maximum load power that can be supported increases to 165 W. Therefore, if the variable amount of the inductance LR of the control winding NR of the orthogonal control transformer PRT is redesigned so that the maximum load power that can be handled is 150 W, the variable range of the inductance LR decreases by about 10%. can get. Thereby, the number of turns of the control winding NR is reduced,
Accordingly, it is possible to reduce the size of the orthogonal control transformer PRT.

FIG. 3 is a circuit diagram showing a configuration of a power supply circuit according to a second embodiment of the present invention. The same parts as those in FIG. 1 are denoted by the same reference numerals and description thereof is omitted. In the power supply circuit shown in FIG. 3, the winding N3 is provided independently of the primary winding N1. That is, it is not formed by winding up the primary winding N1 as in the case of FIG.

Further, by providing a center tap for the controlled winding NR, the center tap divides the controlled winding NR1 as the inductance LR1 and the controlled winding NR2 as the inductance LR2. One end of the controlled winding NR1 is inserted in series between the cathode of the boost diode DB and one end of the winding N3. The other end of the winding N3 is connected to the positive electrode of the smoothing capacitor CiB. Therefore, in this case, a boost circuit is formed by the boost diode DB, the controlled winding NR2, the winding N3, and the smoothing capacitor CiB. Further, the end of the controlled winding NR2 is connected to the end of the primary winding N1, thereby being connected to the primary-side parallel resonance circuit (N1, Cr).

In such a configuration, substantially the same operation as described above with reference to FIG. 2 can be obtained, and the corresponding maximum load power can be increased, or the orthogonal control transformer PRT can be reduced in size (controlled winding). (Reduction in the number of turns of NR), and elimination of radiation noise.

The control circuit 1 according to the present embodiment supplies a control current of a level corresponding to the fluctuation of the DC output voltage EO1 to the control winding NC of the orthogonal control transformer PRT, thereby Inductance LR of lines NR1, NR2
1, LR2 is varied. In the connection form of the present embodiment, since the inductance LR2 forms a primary side parallel resonance circuit, the resonance condition of the primary side parallel resonance circuit is changed by the variable inductance LR2. That is, the secondary side DC output voltage is operated to be constant by the above-described inductance control method. In addition, in the present embodiment, the boost circuit including the inductance LR1 is formed, so that the inductance LR1 is varied, thereby stabilizing the boost smoothing voltage EB. It has the function of stabilizing the side output voltage.

FIG. 4 is a circuit diagram showing the configuration of a power supply circuit according to a third embodiment of the present invention. The same parts as those in FIG. 1 are denoted by the same reference numerals, and the same parts are described. Is omitted.

In the power supply circuit shown in FIG. 4, the controlled winding NR of the orthogonal control transformer PRT is inserted in series between the primary winding N3 and the positive electrode of the smoothing capacitor CiB.
In this configuration, the primary-side parallel resonance circuit is formed by a combined inductance (L1 + L3 + LR) obtained by connecting the primary winding N1, the winding N3, and the controlled winding NR in series, and the capacitance of the parallel resonance capacitor Cr. It can be seen as something. Even with the configuration shown in FIG. 4, for example, as in the first embodiment, the maximum load power that can be handled is increased, or the orthogonal control transformer PRT is downsized (the number of turns of the controlled winding NR is reduced). , And radiation noise.

The control circuit 1 shown in FIG. 4 also applies a DC output voltage EO to the control winding NC of the orthogonal transformer PRT.
A control current of a level corresponding to the fluctuation of 1 is passed to operate to vary the inductance LR of the controlled winding NR.
Also in this case, since the inductance LR forms a primary-side parallel resonance circuit, the variable inductance LR operates to keep the secondary-side DC output voltage constant by the inductance control method described above. In addition, in the present embodiment, by providing the inductance LR in the boost circuit, the boost smoothing voltage EB is controlled to be constant by the change in the inductance LR, as shown in FIG. Further, when the inductance LR of the controlled winding NR is varied, even in this configuration,
The pulse width of the resonance voltage Vcr (that is, the TOFF period) is controlled, and this operation also operates to keep the secondary-side DC output voltage constant.

The present invention is not limited to the configurations shown in FIGS. 1, 3 and 4, and can be modified. For example, the voltage resonance type converter in each of the above-described embodiments employs a self-excited configuration in which a switching element is driven by a self-excited oscillation circuit. It is also possible. As for the switching elements, not only bipolar transistors but also other types of switching elements such as MOS-FET transistors may be employed.

[0042]

As described above, according to the present invention, an insulation converter transformer is loosely coupled as a voltage resonance type switching power supply circuit corresponding to a relatively high load condition of 150 W or more with an AC input voltage of 100 V AC, for example. so,
An operation mode (+ M / -M) in which the mutual inductances of the primary winding and the secondary winding have opposite polarities is obtained. In addition, by providing the full-wave rectifier circuit on the secondary side, the secondary-side DC output voltage is obtained from the alternating voltage obtained in the secondary winding. That is, as a result of supplying power to the load by the full-wave rectifier circuit on the secondary side, in the present invention, it is possible to improve the maximum load power that can be handled, compared to the case where the secondary-side DC output voltage is obtained by the half-wave rectifier circuit. Will be possible. Along with this, the primary side is not a voltage doubler rectifier circuit, but a full-wave rectifier circuit is used to input a rectified smoothed voltage at a level corresponding to the AC input voltage level. It will be able to respond.

For example, conventionally, when the above condition is satisfied, it is necessary to obtain a rectified smoothed voltage corresponding to twice the AC input voltage level by a voltage doubler rectifier circuit. For the resonance capacitor, it is necessary to select a withstand voltage product corresponding to the switching voltage generated according to the rectified smoothing voltage level. In addition, since the DC output voltage is generated by the half-wave rectifier circuit on the secondary side, a voltage of about 2.5 to 3.5 times the rectified smoothed voltage is applied during the non-conductive period of the rectifier diode. Therefore, a withstand voltage product according to this voltage level has been selected.

On the other hand, in the present invention, since the switching voltage depending on the rectified smoothing voltage level is １ ／ of the conventional one, the withstand voltage of the switching element and the resonance capacitor on the primary side is much lower than the conventional one. Component elements can be used. On the secondary side, an alternating voltage (resonance voltage) obtained in a secondary winding forming a parallel resonance circuit
Is rectified by a full-wave rectifier circuit to obtain a DC output voltage. As a result, after increasing the maximum load power that can be handled, the voltage applied to the rectifier diode is suppressed to approximately the same level as the rectification smoothed voltage level. Things can be selected. As a result, firstly, costs for the switching element, the primary-side parallel resonance capacitor, the secondary-side rectifier diode, and the like can be reduced. In addition, if the switching element and the secondary rectifier diode having improved characteristics are selected, the power conversion efficiency can be improved. Further, if the characteristics of the switching element are improved, it becomes easy to increase the switching frequency. Therefore, if the switching frequency is increased, it is possible to reduce the size and weight of circuit components around the switching element. . Further, since a circuit for obtaining a rectified smoothed voltage from a commercial AC power supply is a normal full-wave rectifier circuit, for example, a normal set of a block-type smoothing capacitor and a bridge rectifier diode can be employed. However, cost and circuit scale can be reduced.

Further, according to the present invention, a boost circuit (boost means) for obtaining a boost smoothed voltage by superimposing a boost voltage on a rectified smoothed voltage is provided. This also provides the increase in the maximum load power described above.
In addition, this boost circuit is formed including the controlled winding of the orthogonal control transformer, so that the boost smoothing voltage is controlled to be constant with the constant voltage control operation using the orthogonal control transformer. Is done. With this configuration, it is possible to reduce the current to be passed through the controlled winding. This makes it possible to reduce the size of the orthogonal control transformer. Further, the boost circuit includes a boost winding, a controlled winding, a rectifying diode for boost, and a smoothing capacitor for boost for obtaining a boost voltage by charging a rectified output of the rectifying diode for boost. It is formed by a simple circuit. Further, as for the boosting smoothing capacitor, a rectified smoothed voltage (DC input voltage) is obtained by a normal full-wave rectification method instead of a voltage doubler rectification method.
An inexpensive and small-sized product can be adopted as long as a pressure-resistant product of a sufficient level can be selected.

In the present invention, a parallel resonance capacitor is connected in parallel with the boost rectifier diode of the boost circuit in the above configuration, thereby forming a parallel resonance circuit in the boost circuit. By the operation of the parallel resonance circuit in the boost circuit, first, the off period of the rectifier diode for boost is extended, the current flowing in the boost circuit is increased, and the boost voltage can be increased. This promotes downsizing of the orthogonal transformer due to an increase in the maximum load power or a reduction in the number of turns of the controlled winding. Furthermore, since the operation is performed so that the high-frequency pulse component due to the parasitic vibration generated when the boost rectifier diode is turned off is eliminated, unnecessary radiation noise is eliminated, and the noise reduction is promoted accordingly.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a configuration example of a power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a waveform chart showing an operation of a main part of the power supply circuit of the embodiment.

FIG. 3 is a circuit diagram illustrating a configuration example of a power supply circuit according to a second embodiment of the present invention.

FIG. 4 is a circuit diagram illustrating a configuration example of a power supply circuit according to a third embodiment of the present invention.

FIG. 5 is a circuit diagram showing a configuration example of a power supply circuit that can be configured based on the invention previously proposed by the present applicant.

FIG. 6 is a cross-sectional view illustrating a configuration of an insulating converter transformer.

FIG. 7 is an explanatory diagram showing each operation when the mutual inductance is + M / −M.

8 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIG.

[Explanation of symbols]

1 control circuit, 2 oscillation circuit, 3 drive circuit, C
i, CiB smoothing capacitor, Cr parallel resonance capacitor, C2 (secondary side) parallel resonance capacitor, Di, DO
Bridge rectifier circuit, DO1, DO2, DO3, DO4 rectifier diode, PIT isolation converter transformer, N1, primary winding, N2 secondary winding, PRT orthogonal control transformer,
NC control winding, NR controlled winding, Q1 switching element, DB boost diode, N3 (for boost)
Winding, C2A 2nd parallel resonance capacitor

Claims (1)

[Claims] 1. A rectifying / smoothing means for generating a rectified / smoothed voltage having a level corresponding to an input commercial AC power supply level by full-wave rectification, and a gap formed so as to obtain a required coupling coefficient which is loosely coupled. An insulating converter transformer provided for transmitting a primary output to a secondary side; and a rectifying and smoothing voltage output from the rectifying and smoothing means being intermittently output to a primary winding of the insulating converter transformer. Switching means, at least a primary-side resonance circuit formed by an inductance component including a primary winding of the insulated converter transformer and a capacitance of a resonance capacitor to make the operation of the switching means a voltage resonance type; and Of the secondary winding and the capacitance of the secondary side parallel resonant capacitor And a secondary-side resonance circuit formed on the secondary side, and an alternating voltage obtained in a secondary winding of the insulating converter transformer, and a DC output voltage generation for obtaining a secondary-side DC output voltage by full-wave rectification. Means, a controlled winding provided so as to function as an inductance component of the primary side resonance circuit, and a control winding whose winding direction is orthogonal to the controlled winding. An orthogonal control transformer having a variable control current flowing through the control winding according to the level of the DC output voltage to change the inductance of the controlled winding.
Constant voltage control means configured to perform constant voltage control on the secondary side DC output voltage; and at least a boost winding wound around the insulating converter transformer and exciting an alternating voltage according to the switching output of the switching means. Line, a boost rectifier diode for rectifying the alternating voltage obtained in the boost winding, and a boost smoothing capacitor for smoothing the rectified output by the boost rectifier diode to generate a boost voltage. A voltage is superimposed on the rectified and smoothed voltage to obtain a boost rectified and smoothed voltage. In addition to including the controlled winding, the boost rectified and smoothed voltage is kept constant by a change in inductance of the controlled winding. Boost means having a controllable configuration so that It is connected in parallel for, the boost circuit voltage resonance type switching power supply circuit, characterized in that it comprises a parallel resonant capacitor, a forming a parallel resonant circuit boosting circuit by the inductance component for forming a.